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RT8811A

Dual-Phase COT Buck PWM Controller with Dynamic Voltage Control

General Description

The RT8811A is a dual-phase synchronous Buck PWM

controller which is optimized for high performance graphic

microprocessor and computer applications. The IC

integrates a Constant-On-Time (COT) PWM controller, two

MOSFET drivers with internal bootstrap diodes, as well

as channel current balance and protection functions

including Over-Voltage Protection (OVP), Under-Voltage

Protection (UVP), current limit and thermal shutdown into

the WQFN-24L 4x4 package.

The RT8811A adopts R

Current limit is accomplished through continuous inductor-

current-sense, while R

accurate channel current balance. Using the method of

current sampling utilizes the best advantages of each

technique.

The RT8811A features external reference input and PWM-

VID dynamic output voltage control, in which the feedback

voltage is regulated and tracks external input reference

voltage. Other features include adjustable switching

frequency, dynamic phase number control, internal/external

soft-start, power good indicator, and enable functions.

current sensing technique.

DS(ON)

current sensing is used for

DS(ON)

Features



 Dual-Phase PWM Controller



 Two Embedded MOSFET Drivers and Embedded



Switching Boot Diode



 External Reference Input Control



 PWM-VID Dynamic Voltage Control



 Dynamic Phase Number Control



 Lossless R



 Internal Fixed and External Adjustable Soft-Start



 Built-In 220mA 5V LDO



 Adjustable Current Limit Threshold



 Adjustable Switching Frequency



 UVP/OVP Protection



 Shoot-Through Protection and Short Pulse Free



Technology



 Single IC Supply Voltage : 4.5V to 13.2V



 Support an Ultra-Low Output Voltage as Standby



Voltage



 Thermal Shutdown



 Thermal Alert Indicator



 Power Good Indicator



 RoHS Compliant and Halogen Free



Current Sensing for Current Balance

DS(ON)

Simplified Application Circuit

RT8811A

PVCC

BOOT1

UGATE1

PHASE1

PGOOD

TON

PGOOD

PSI

VID

GND

UGATE2

PHASE2

RGND

DS8811A-02 March 2014 www.richtek.com

OUT

GND_SNS

RT8811A

Applications

 Motherboard to High End GPU Core Power

 High End Desktop PC Memory Core Power

 Low Voltage. High Current DC/DC Converter

 Voltage Regulator Modules

Ordering Information

RT8811A

Package Type QW : WQFN-24L 4x4 (W-Type) (Exposed Pad-Option 1)

Lead Plating System G : Green (Halogen Free and Pb Free)

Note :

Richtek products are :

 RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

 Suitable for use in SnPb or Pb-free soldering processes.

Marking Information

0T= : Product Code

0T=YM

YMDNN : Date Code

DNN

Pin Configurations

(TOP VIEW)

LGATE1

GND

VREF

GND

21 20 1924 2223

TON

PVCC

RGND

BOOT1

UGATE1

EN PSI VID

REFADJ

PHASE1

78910 1211

REFIN

WQFN-24L 4x4

PHASE2

LGATE2

BOOT2

UGATE2

PGOOD

VCC

TALERT

TSNS

VSNS

Function Pin Description

Pin No. Pin Name Pin Function

1 BOOT1 Bootstrap Supply for PWM 1. This pin powers the high-side MOSFET driver.

High-Side Driver of PWM 1. This pin provides the gate drive for the

2 UGATE1

3 EN

4 PSI

5 VID

6 REFADJ Reference Adjustment Output. Refer to PWM-VID Dynamic Voltage Control.

7 REFIN External Reference Input.

8 VREF

9 TON ON-Time/Switching Frequency Adjustment Input.

10 RGND Negative Remote Sense Input. Connect this pin to the ground of output load.

11 VSNS

converter's high-side MOSFET. Connect this pin to the Gate of high-side MOSFET.

Enable Control Input. Drive EN higher than 1.6V to turn on the controller, lower than 0.8V to turn it off. If the EN pin is open, it will be pulled to high by internal circuit.

Power Saving Interface. When the voltage is pulled below 0.8V, the device will operate into 1 phase DEM. When the voltage is between 2.4V to 5.5V, the device will operate into 2 phases force CCM.

Programming Output Voltage Control Input. Refer to PWM-VID Dynamic Voltage Control.

Reference Voltage Output. This is a high precision voltage reference (2V) from the VREF pin to the RGND pin.

Positive Remote Sense Input. Connect this pin to the positive terminal of output load.

DS8811A-02 March 2014www.richtek.com

Pin No. Pin Name Pin Function

Soft-Start Time Setting. Connect an external capacitor to adjust soft-start

12 SS

13 TSNS Temperature Sensing Input.

time. When the external capacitor is removed, the internal soft-start function will be chose.

RT8811A

15 VCC

16 PGOOD Power Good Indicator Output. Active high open-drain output.

17 UGATE2

18 BOOT2 Bootstrap Supply for PWM 2. This pin powers the high-side MOSFET driver.

19 PHASE2

20 LGATE2

21 PVCC

22,

25 (Exposed Pad)

23 LGATE1

24 PHASE1

TALERT

GND

Thermal Alert. Active low open-drain output. LDO Regulator Output. Connect a minimum 4.7F ceramic capacitor between

this pin and ground.

High-Side Driver of PWM 2. This pin provides the gate drive for the converter's high-side MOSFET. Connect this pin to the high-side MOSFET.

Switch Node for PWM2. Connect this pin to the Source of high-side MOSFET together with the Drain of low-side MOSFET and the inductor.

Low-Side Driver of PWM 2. This pin provides the gate drive for the converter's low-side MOSFET. Connect this pin to the low-side MOSFET.

Supply Voltage Input. Place a high quality bypass capacitor from this pin to GND.

Ground. Must be connected to GND on PCB. The Exposed pad should be soldered to a large PCB and connected to GND for maximum thermal dissipation.

Low-Side Driver of PWM 1. This pin provides the gate drive for the converter's low-side MOSFET. Connect this pin to the Gate of low-side MOSFET.

Switch Node for PWM1. Connect this pin to the Source of high-side MOSFET together with the Drain of low-side MOSFET and the inductor.

DS8811A-02 March 2014 www.richtek.com

RT8811A

Function Block Diagram

VID

REFADJ

PSI

REFIN

VSNS

TON

TALERT

TSNS

VREF

RGND

150% REFIN

40% REFIN

Soft-Start

& Slew Rate

Control

2µA

Enable

Logic

VIN

Detection

Internal

Die Temperature

Sense

RGND

Reference Output

Gen. & BG

Mode Select

PWM

CMP

To Driver Logic To Power On Reset

To Power On Reset

Power On Reset

& Central Logic

Control & Protection Logic

TON

Gen 1

TON

Gen 2

VCC

Current

Balance

Current

Limit

VCC

Internal

Regulator

Boot-Phase Detection 1

Boot-Phase Detection 2

PWM1

Driver

Logic

PWM2

S/H

PVCC

PGOOD

BOOT1 UGATE1 PHASE1

LGATE1

BOOT2 UGATE2

PHASE2

LGATE2

DS8811A-02 March 2014www.richtek.com

Operation

RT8811A

The RT8811A integrates a Constant-On-Time (COT) PWM

controller, the controller provides the PWM signal which

relies on the output ripple voltage comparing with internal

reference voltage. Referring to the function block diagram

of TON Genx, the synchronous UGATE driver is turned on

at the beginning of each cycle. After the internal one-shot

timer expires, the UGATE driver will be turned off. The

pulse width of this one-shot is determined by the

converter's input voltage and the output voltage to keep

the frequency fairly constant over the input voltage and

output voltage range. Another one-shot sets a minimum

off-time.

The RT8811A also features a PWM-VID dynamic voltage

control circuit driven by the pulse width modulation

method. This circuit reduces the device pin count and

enables a wide dynamic voltage range.

Current Balance

The RT8811A implements the internal current balance

mechanism in the current loop. The RT8811A senses per

phase current and compares it with the average current. If

the sensed current of any particular phase is higher than

average current, the on-time of this phase will be adjusted

to be shorter.

PGOOD

Current Limit

The current limit circuit employs a unique “valley” current

sensing algorithm. If the magnitude of the current sense

signal at PHASE is above the current limit threshold, the

PWM is not allowed to initiate a new cycle. Thus, the

current to the load exceeds the average output inductor

current, the output voltage falls and eventually crosses

the under-voltage protection threshold, inducing IC

shutdown.

Over-Voltage Protection (OVP)

The output voltage can be continuously monitored for over-

voltage protection. When the output voltage exceeds its

set voltage threshold (If V

> 1.33V, OV = 1.5 x V

REFIN

≤ 1.33V, OV = 2V, or V

REFIN

), UGATE goes low and LGATE

is forced high. The controller is latched until VCC is re-

supplied and exceeds the POR rising threshold voltage.

Under-Voltage Protection (UVP)

The output voltage is continuously monitored for under-

voltage protection. When the output voltage is less than

40% of its set voltage, under-voltage protection is triggered

and then both UGATE and LGATE gate drivers are forced

low. The controller is latched until VCC is re-supplied and

exceeds the POR rising threshold voltage.

The power good output is an open-drain architecture.

When the soft-start is finished, the PGOOD open-drain

output will be high impedance.

Soft-Start (SS)

For internal soft-start function, an internal current source

charges an internal capacitor to build the soft-start ramp

voltage. The output voltage will track the internal ramp

voltage during soft-start interval.

For external soft-start function, an additional capacitor

connected from SS to GND will be charged by a current

source and determines the soft-start time.

DS8811A-02 March 2014 www.richtek.com

RT8811A

Absolute Maximum Ratings (Note 1)

 TON to GND ------------------------------------------------------------------------------------------------------- −0.3 to 32V

 PVCC to GND ---------------------------------------------------------------------------------------------------- −0.3 to 15V

 RGND to GND ---------------------------------------------------------------------------------------------------- −0.7V to 0.7V

 PHASEx to GND

DC------------------------------------------------------------------------------------------------------------------- −0.3V to 26V

<20ns -------------------------------------------------------------------------------------------------------------- −8V to 38V

 BOOTx to PHASEx --------------------------------------------------------------------------------------------- 15V

 UGATEx to GND

DC------------------------------------------------------------------------------------------------------------------- 0.3V to (V

<20ns -------------------------------------------------------------------------------------------------------------- −5V to (V

 LGATEx to GND

DC------------------------------------------------------------------------------------------------------------------- −0.3V to (V

<20ns -------------------------------------------------------------------------------------------------------------- −5V to (V

 Other Pins--------------------------------------------------------------------------------------------------------- −0.3V to 6V

 Power Dissipation, P

@ T

= 25°C

WQFN-24L 4x4 -------------------------------------------------------------------------------------------------- 3.57W

 Package Thermal Resistance (Note 2)

WQFN-24L 4x4, θJA--------------------------------------------------------------------------------------------- 28°C/W

WQFN-24L 4x4, θJC-------------------------------------------------------------------------------------------- 7°C/W

 Lead Temperature (Soldering, 10 sec.) --------------------------------------------------------------------- 260°C

 Junction Temperature ------------------------------------------------------------------------------------------- 150°C

 Storage Temperature Range ---------------------------------------------------------------------------------- −65°C to 150°C

 ESD Susceptibility (Note 3)

HBM (Human Body Model)------------------------------------------------------------------------------------ 2kV

BOOT

PVCC

+ 0.3V)

+ 5V)

+ 0.3V)

+ 5V)

Recommended Operating Conditions (Note 4)

 Supply Input Voltage, V

 Control Voltage, V

 Junction Temperature Range---------------------------------------------------------------------------------- −40°C to 125°C

 Ambient Temperature Range---------------------------------------------------------------------------------- −40°C to 85°C

PVCC

-------------------------------------------------------------------------------------- 7V to 20V

----------------------------------------------------------------------------------------- 4.5V to 13.2V

Electrical Characteristics

= 25°C unless otherwise specified)

Parameter Symbol Test Conditions Min Typ Max Unit

PWM Controller

PVCC Supply Voltage V

PVCC Supply Current I

PVCC Shutdown Current I

VCC POR Threshold 3.7 4 4.3 V

POR Hysteresis -- 0.3 -- V

VCC (LDO Output) PVCC > 10.8V; I

4.5 -- 13.2 V

PVCC

SUPPLY

SHDN

EN = 3.3V, Not Switching -- 2 4 mA

EN = 0V, VCC Remains Active -- -- 220 A

< 220mA 4.75 5 5.25 V

VCC

DS8811A-02 March 2014www.richtek.com

RT8811A

Parameter Symbol Test Conditions Min Typ Max Unit

VCC Output Current VCC = 4.5V 220 -- 400 mA

Switching Frequency R

Minimum TON T

Minimum Off-Time T

Zero Current Crossing Threshold

ON(MIN)

OFF(MIN)

8 -- 8 mV

-- 70 -- ns

-- 300 -- ns

= 500k (Note 5) 270 300 330 kHz

TON

EN Threshold

EN Input Voltage

Logic-High V

Logic-Low V

1.6 -- --

ENH

-- -- 0.8

ENL

Mode Decision

PSI High Threshold V

PSI Low Threshold V

Logic-High V

VID Input Voltage

Logic-Low V

Enable Two Phases with FCCM 2.4 -- -- V

PSIH

Enable One Phase with DEM -- -- 0.8 V

PSIL

2 -- --

VIDH

-- -- 1

VIDL

Protection Function

Current Limit Setting Current I

Current Limit Setting Current Temperature Coefficient

9 10 11 A

OCSET

OCSET_TC

On the basis of 25C -- 6300 -- ppm/C

Current Limit Threshold 20 -- 20 mV

Current Limit Threshold Setting Range

Absolute Over-V oltage Protection Threshold

Relative Over-Voltage Protection Threshold

OVP Delay

Relative Under-Voltage Protection Threshold

UVP Delay

Internal Die Temperature Sense Threshold

50 -- 300 mV

OVP, Absolute VREFIN

OVP, Relative VREFIN

D_OVP

UVP 35 40 45 %

UVP

D_UVP

-- 5 -- s

-- 3 -- s

-- 95 -- C

 1.33V 1.9 2 2.1 V

> 1.33V 145 150 155 %

Thermal Shutdown Threshold TSD -- 140 -- C

TSNS Threshold V

PGOOD Blanking Time

Internal Soft-Start Time TSS

OUT

(No Shutting Down) 0.98 1 1.02 V

TSEN

From EN = High to PGOOD = High with

within Regulation Point

OUT

From First UGATE to V Point, V

REFIN

= 1V and V

Regulation

OUT

Initial = 0V

-- 3.7 -- ms

-- 0.7 -- ms

Soft-Start Current Source ISS -- 5 -- A

DS8811A-02 March 2014 www.richtek.com

RT8811A

Parameter Symbol Test Conditions Min Typ Max Unit

Er ror Am p l if ier

VSNS Error Comparator Threshold (Valley)

Reference

= 1V 5 -- 5 mV

REFIN

Reference Voltage V

VREF

Sourcing Current = 1mA, VID No Switching

1.98 2 2.02 V

Driver On-Resistance

UGATE Driver Source R

UGATE Driver Sink R

LGATE Driver Source R

LGATE Driver Sink R

UGATEsr

UGATEsk

LGATEsr

LGATEsk

Dead-Time

UGATEx

LGATEx

= 150mA -- 1.5 3 

UGATEx

 V

PHASEx

= 0.1V -- 2 4 

= 150mA -- 1.5 3 

= 0. 1V -- 0.7 1.4 

LGATEx

From LGATE falling to UGATE rising -- 30 --

From UGATE falling to LGATE rising -- 30 --

Boost Switch Ron R

Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θ

Note 3. Devices are ESD sensitive. Handling precaution is recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Note 5. Not production tested. Test condition is V

is measured at T

measured at the exposed pad of the package.

= 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC is

PVCC to BOOTx, I

BOOT

= 8V, V

OUT

= 1V, I

= 10mA -- 40 80 

BOOT

= 20A using application circuit.

OUT

DS8811A-02 March 2014www.richtek.com

Typical Application Circuit

RT8811A

V_STANDBY

RGND

PVCC

4.75k

STANDBY

1.07k

VREF

PGOOD

REF1

BOOT

2.2

PSI

VID

Enable

0.1µF

RGND

REFADJ

0.01µF

RGND

REF2

4.22k

RGND RGND

OTSET

10k

2.2µF

TON

500k

1µF

100k

REFADJ

6.34k

REFIN

NTC

10k

PVCC

RT8811A

TON

PGOOD

PSI

VID

VREF

REFADJ

REFIN

TSNS

VCC

VSNS

RGND

TALERT

BOOT1

UGATE1

PHASE1

LGATE1

BOOT2

UGATE2

PHASE2

LGATE2

GND

22, 25 (Exposed pad)

4.7µF

30k

0.1µF

R 10k

47pF

0.1µF

OCSET

10µF x 6

0.36µH/1.05m

470µF 50V x 2



22µF x 15

OUT

820µF

2.5V x 4

0.36µH/1.05m



1010

OUT_SNS

GND_SNS

Figure 1. 2 Active Phase Configuration

DS8811A-02 March 2014 www.richtek.com

RT8811A

V_STANDBY

RGND

PVCC

STANDBY

1.07k

VREF

PGOOD

REF1

4.75k

BOOT

2.2

PSI

VID

Enable

RGND

REFADJ

6.34k

REFADJ

0.01µF

RGND

REF2

4.22k

RGND RGND

OTSET

10k

2.2µF

TON

500k

1µF

100k

C NC

R 10k

REFIN

NTC

PVCC

RT8811A

TON

PGOOD

PSI

VID

VREF

REFADJ

REFIN

TSNS

VCC

BOOT2

VSNS

RGND

TALERT

BOOT1

UGATE1

PHASE1

LGATE1

UGATE2

PHASE2

LGATE2

GND

22, 25 (Exposed pad)

4.7µF

30k

47pF

0.1µF

10k

OCSET

0.36µH/1.05m

10µF x 6



22µF x 15

470µF 50V x 2

OUT

820µF

2.5V x 4

Floating

10 10

OUT_SNS

GND_SNS

Figure 2. 1 Active Phase Configuration

DS8811A-02 March 2014www.richtek.com

Typical Operating Characteristics

RT8811A

285.0

282.5

280.0

277.5

(ns)

275.0

272.5

Efficiency vs. Load Current

1.0

100

0.9

0.8

0.7

0.6

0.5

0.4

Efficiency (%)

0.3

0.2

0.1

0.0

0 102030405060

= 0.94V, 2 Phases Operation

OUT

VIN = V

PVCC

= 12V,

Load Current (A)

TON vs. Temperature

Efficiency vs. Load Current

1.0

100

0.9

0.8

0.7

0.6

0.5

0.4

Efficiency (%)

0.3

0.2

0.1

0.0

0.01 0.1 1 10

= 0.94V, 1 Phase with DEM Operation

OUT

Load Current (A)

vs. Temperature

2.04

2.03

2.02

2.01

(V)

2.00

REF

1.99

REF

VIN = V

PVCC

= 12V,

270.0

267.5

265.0

-50-250 255075100125

VIN = V

= 12V, No Load

PVCC

Temperature (° C)

Inductor Current vs. Output Current

Inductor Current (A)

0 102030405060

Phase 1

Phase 2

VIN = V

Output Current (A)

PVCC

= 12V

1.98

1.97

1.96

(5V/Div)

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

VIN = V

-50-250 255075100125

= 12V, No Load

PVCC

Temperature (° C)

Power On from EN

VIN = V

Time (1ms/Div)

PVCC

= 12V, I

OUT

= 50A

DS8811A-02 March 2014 www.richtek.com

RT8811A

(5V/Div)

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

PVCC

(10V/Div)

Power Off from EN

VIN = V

Time (1ms/Div)

PVCC

= 12V, I

Power Off from PVCC

OUT

= 50A

PVCC

(10V/Div)

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

DVID

(2V/Div)

Power On from PVCC

VIN = V

Time (5ms/Div)

PVCC

= 12V, I

OUT

Dynamic Output Voltage Control

VIN = V

PVCC

= 12V

= 50A

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

DVID

(2V/Div)

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

VIN = V

PVCC

= 12V, I

OUT

= 50A

Time (5ms/Div)

Dynamic Output Voltage Control

OUT

= 50A, V

VIN = V

= 1.22V to 0.7V

REFIN

PVCC

= 12V

OUT

(1V/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

OUT

(50mV/Div)

OUT

(50A/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

= 50A, V

OUT

= 0.7V to 1.22V

REFIN

Time (50μs/Div)

Load Transient Response

VIN = V

PVCC

= 12V

Time (50μs/Div)

Time (20μs/Div)

DS8811A-02 March 2014www.richtek.com

RT8811A

OUT

(50mV/Div)

OUT

(50A/Div)

UGATE1

(40V/Div)

UGATE2

(40V/Div)

VSNS

(1V/Div)

UGATE1

(20V/Div)

LGATE1

(10V/Div)

Load Transient Response

VIN = V

PVCC

Time (20μs/Div)

UVP

VIN = V

PVCC

= 12V, I

OUT

= 12V

= 40A

VSNS

(1V/Div)

UGATE1

(20V/Div)

LGATE1

(10V/Div)

(10A/Div)

UGATE1

(30V/Div)

LGATE1

(20V/Div)

OVP

VIN = V

PVCC

Time (100μs/Div)

OCP

VIN = V

= 12V, No Load

= 12V

PVCC

Time (50μs/Div)

Time (20μs/Div)

DS8811A-02 March 2014 www.richtek.com

RT8811A

Application Information

The RT8811A is a dual-phase synchronous Buck PWM

controller with integrated drivers which is optimized for

high performance graphic microprocessor and computer

applications. A COT (Constant-On-Time) PWM controller

and two MOSFET drivers with internal bootstrap diodes

are integrated so that the external circuit can be easily

designed and the number of component is reduced.

The topology solves the poor load transient response timing

problems of fixed frequency current mode PWM and avoids

the problems caused by widely varying switching

frequencies in conventional constant on-time and constant

off-time PWM schemes.

The RT8811A supports dynamic mode transition function

with various operating states, which include dual-phase

with CCM operation, single phase with diode emulation

mode. These different operating states make the system

efficiency as high as possible.

The RT8811A provides a PWM-VID dynamic control

operation in which the feedback voltage is regulated and

tracks external input reference voltage. It also features

complete fault protection functions including over-voltage,

under-voltage and current limit.

PWM Operation

The RT8811A integrates a Constant-On-Time (COT) PWM

controller, and the controller provides the PWM signal

which relies on the output ripple voltage comparing with

internal reference voltage as shown in Figure 3. Referring

to the function block diagram of TON Genx, the

synchronous UGATE driver is turned on at the beginning

of each cycle. After the internal one-shot timer expires,

the UGATE driver will be turned off. The pulse width of

this one-shot is determined by the converter's input voltage

and the output voltage to keep the frequency fairly constant

over the input voltage and output voltage range. Another

one-shot sets a minimum off-time.

OUT

PEAK

OUT

VALLEY

REF

Figure 3. Constant On-Time PWM Control

Remote Sense

The RT8811A uses the remote sense path (VSNS and

RGND) to overcome voltage drops in the power lines by

sensing the voltage directly at the end of GPU to make

sure the voltage drops on PCB has no impact on the load

transient response. Normally, to protect remote sense path

disconnecting, there are two resistors (R

) connecting

Local

between local sense path and remote sense path. That

is, in application with remote sense, the R

Local

recommended to be 10Ω to 100Ω. If no need of remote

sense, the R

BOOT

UGATE

PHASE

LGATE

RGND

VSNS

is recommended to be 0Ω.

Local

Local Sense Path

OUT

Local

GPU

Remote Sense Path

Figure 4. Output Voltage Sensing

On-Time Control

The on-time one-shot comparator has two inputs. One

input monitors the output voltage, the other input samples

the input voltage and converts it to a current. This input

voltage proportional current is used to charge an internal

on-time capacitor. The on-time is the time required for

the voltage on this capacitor to charge from zero volts to

, thereby making the on-time of the high-side switch

OUT

directly proportional to output voltage and inversely

proportional to input voltage. The implementation results

in a nearly constant switching frequency without the need

for a clock generator.

DS8811A-02 March 2014www.richtek.com

RT8811A



)

2 V 3.2p



T = R

ON TON

OUT

V0.5





and then the switching frequency FS is :

F= V V T

SOUT INON

TON

The value of R



is the resistor connected from the VIN to TON pin.

can be selected according to Figure 5.

TON

The recommended operation frequency range is 150kHz

to 600kHz.

Frequency vs. R

900

800

700

600

500

400

Frequency (kHz

300

200

150 250 350 450 550 650 750

(kΩ)

(kohm)

TON

Figure 5. Frequency vs. R

TON

Active Phase Circuit Setting

The RT8811A can operate in 2/1 phase. For one phase

operation, leave the UGATE2, BOOT2, PHASE2, and

LGATE 2 as floating, and keep the voltage on the PSI pin

under 2.4V before POR.

Diode-Emulation Mode

In diode-emulation mode, the RT8811A automatically

reduces switching frequency at light-load conditions to

maintain high efficiency. As the output current decreases

from heavy-load condition, the inductor current is also

reduced, and eventually comes to the point that its valley

touches zero current, which is the boundary between

continuous conduction and discontinuous conduction

modes. By emulating the behavior of diodes, the low-side

MOSFET allows only partial of negative current when the

inductor freewheeling current reaches negative level. As

the load current is further decreased, it takes longer and

longer to discharge the output capacitor to the level that

requires the next “ON” cycle. In reverse, when the output

current increases from light load to heavy load, the

switching frequency increases to the preset value as the

inductor current reaches the continuous condition.

The switching waveforms may appear noisy and

asynchronous when light loading causes diode-emulation

operation, but this is a normal operating condition that

results in high light load efficiency. Trade-off in DEM noise

vs. light load efficiency is made by varying the inductor

value. Generally, low inductor values produce a broader

efficiency vs. load curve, while higher values result in higher

full-load efficiency (assuming that the coil resistance

remains fixed) and less output voltage ripple. The

disadvantages for using higher inductor values include

larger physical size and degrade load-transient response

(especially at low input voltage levels).

Mode Selection

The RT8811A can operate in 2 phases with force CCM

and 1 phase with DEM according to PSI voltage setting. If

PSI voltage is pulled below 0.8V, the controller will operate

into 1 phase with DEM. In DEM operation, the RT8811A

automatically reduces the operation frequency at light load

conditions for saving power loss. If PSI voltage is pulled

between 2.4V to 5.5V, the controller will switch operation

into 2 phases with force CCM. Moreover, the PSI pin is

Forced-CCM Mode

The low noise, forced-CCM mode disables the zero-

crossing comparator, which controls the low-side switch

on-time. This causes the low-side gate drive waveform to

become the complement of the high-side gate drive

waveform. This in turn causes the inductor current to

reverse at light loads as the PWM loop to maintain a duty

ratio V

/ VIN. The benefit of forced-CCM mode is to

OUT

keep the switching frequency fairly constant.

valid after POR of VR.

DS8811A-02 March 2014 www.richtek.com

RT8811A

Enable and Disable

The EN pin allows power sequencing between the

controller bias voltage and another voltage rail. The RT8811A

remains in shutdown if the EN pin is lower than 0.8V.

When the EN pin rises above 1.6V, the RT8811A will begin

a new initialization and soft-start cycle. If the EN pin is

open it will be pulled to high by internal circuit.

PVCC

OUT

Internal SS

External SS

Power On Reset (POR), UVLO

Power On Reset (POR) occurs when VCC rises above to

approximately 4V (typ.), the RT8811A will reset the fault

latch and preparing the PWM for operation. Below 3.7V

(typ.), the VCC Under-Voltage Lockout (UVLO) circuitry

inhibits switching by keeping UGATE and LGATE low.

Soft-Start

The RT8811A provides an internal soft-start function and

an external soft-start function. The soft-start function is

used to prevent large inrush current and output voltage

overshoot while the converter is being powered up. The

soft-start function automatically begins once the chip is

enabled.

If external capacitor from SS to GND is removed, the

internal soft-start function will be chosen. An internal

current source charges the internal soft-start capacitor

such that the internal soft-start voltage ramps up uniformly.

The output voltage will track the internal soft-start voltage

during the soft-start interval. After the internal soft-start

voltage exceeds the REFIN voltage, the output voltage no

longer tracks the internal soft-start voltage but follows the

REFIN voltage. Therefore, the duty cycle of the UGATE

signal as well as the input current at power up are limited.

The soft-start process is finished until both the single

internal SSOK and external SSOK go high and protection

is not triggered. Figure 6 shows the internal soft-start

sequence.

Internal SSOK

External SSOK

LGATE

UGATE

PGOOD

Soft

Discharged

Current Limit

Programming

Soft-Start

Normal

Figure 6. Internal Soft-Start Sequence

The RT8811A also provides a proximate external soft-start

function, and the external soft-start sequence is shown in

Figure 7, an additional capacitor can be connected from

SS to GND. The external capacitor will be charged by

5μA current source to build soft-start voltage ramp. If

external soft-start function is chosen, the external soft-

start time should be set longer than internal soft-start time

to avoid output voltage tracking the internal soft-start ramp,

the external soft-start time setting is shown in Figure 8,

the recommend external soft-start slew rate is 0.1V/ms

to 0.4V/ms.

DS8811A-02 March 2014www.richtek.com

RT8811A

PWM VID and Dynamic Output Voltage Control

PVCC

OUT

Internal SS

External SS

Internal SSOK

External SSOK

LGATE

UGATE

PGOOD

Current Limit

Programming

Soft-Start

Normal

Soft

Discharged

Figure 7. External Soft-Start Sequence

VCC

OUT

REFIN

Figure 8. External Soft-Start Time Setting

The soft-start time can be calculated as :

(C V )



t =

where ISS = 5μA (typ.), V

SS REFIN

is the voltage of REFIN pin,

REFIN

and CSS is the external capacitor placed from SS to GND.

Power Good Output (PGOOD)

The power good output is an open-drain architecture, and

it requires a pull-up resistor. During soft-start, PGOOD is

actively held low and is allowed to be pulled high after

achieved over UVP threshold, under OVP threshold,

OUT

and soft start is completed. In addition if any protection is

triggered during operation, PGOOD will be pulled low

immediately.

The RT8811A features a PWM VID control as shown in

Figure 9, which reduces the number of device pin and

enables a wide dynamic voltage range. The output voltage

is determined by the applied voltage on the REFIN pin.

After VCC POR, the buffer output is available, the VID

PWM duty cycle determines the variable output voltage

at REFIN.

STANDBY

Standby

Control

RGND

PWM IN

REF1

BOOT

REF2

RGND

REFADJ

REFIN

VID

VREF

REFADJ

REFIN

Buffer

RGND

Figure 9. PWM VID Analog Circuit Diagram

According to the PWM VID and external circuit control,

the controller can be set three modes which is shown in

Figure 10.

VREF

REFIN

PWM VID

STANDBY

CONTROL

BOOT MODE

NORMAL

MODE

BOOT MODE

STANDBY

MODE

Figure 10. PWM VID Time Diagram

Boot Mode

When VID is not driven, and the buffer output is tri-state.

At this time, turn off the switch Q1 and connect a resistor

divider as shown in Figure 9 that can set the REFIN voltage

into V

V = V

BOOT VREF

where V

Choose R

and R

according to below calculation :

BOOT







RRR

REF1 REF2 BOOT



= 2V (typ.)

VREF

to be approximately 10kΩ, and the R

REF2

can be calculated by the following equations :

BOOT

REF2



REF1

DS8811A-02 March 2014 www.richtek.com

RT8811A

RR =



REF1 BOOT

RVV

R = R

REF1 BOOT

R = R

BOOT REF1

REF2 VREF BOOT

RVV







REF2 VREF BOOT





REF2 VREF BOOT

BOOT



BOOT



Standby mode. An external control can provide a very low

voltage to meet V

going to standby mode, If the VID

OUT

pin is not driven and switch Q1 is enabled, the REFIN pin

can be set for standby voltage according to the calculation

below :

V = V

STANDBY VREF

R// R

By choosing R



R R (R // R )

REF1 BOOT REF2 STANDBY

, R

REF1

REF2 STANDBY



REF2

, and R

BOOT

, the R

STANDBY

can

be calculated by the following equation :

R =

STANDBY

RRR V

 



REF2 REF1 BOOT STANDBY

RVV RRR

   

REF2 REF STANDBY REF1 REF2 BOOT



REF1



Normal Mode

V = V NV

OUT min STEP

where V

V =

STEP

is total available voltage step numbers and N is the

max

number of steps at a specific V

VID period (T

is the resolution of each voltage step :

STEP



(V V )

max min

max

. The dynamic voltage

OUT

= Tu x N

vid

) is determined by the unit

max

pulse width (Tu) and the available step number (N

The recommend Tu is 27ns.

min

N = 1

N = 2

REFIN

N = 1

N = 2

0.5

= N

max

x T

vid

N = N

max

VID Duty

VID Input

Figure 11. PWM VID Analog Output

max

If the VID pin is driven and switch Q1 is disabled, the

can be adjusted from V

REFIN

the zero percent duty cycle voltage value and V

one hundred percent duty cycle voltage value. V

can be set according to below calculation :

max

V = V

min VREF

V = V



R R // (R R )

REF1 REFADJ BOOT REF2

max VREF

By choosing R



R // (R R )



REF1

REF2

REFADJ BOOT REF2





REF2 BOOT



(R // R ) R R

REF1 REFADJ BOOT REF2

, R

, and R

REF2

to V

min

max



REF2



, the R

BOOT

, where V

max

min

can be

REFADJ

min

is the

and

calculated by the following equation :



R =

REFADJ

The relationship between VID duty and V

Figure 11, and V

REF1 min



max min

REFIN

can be set according to the calculation

OUT

is shown in

below :

VID Slew Rate Control

In RT8811A, the V

slew rate is proportional to PWM

REFIN

VID duty. The rising time and falling time are the same

because the voltage of REFIN pin traveling is the same. In

normal mode, the V

by C

When choose C

SR =

R = (R // R ) // (R +R )

SR REF1 REFADJ BOOT REF2

When choose C

SR =

R = R // R R // R

SR REF1 REFADJ BOOT REF2

or C

REFADJ

(V V ) 80%

REFIN_Final REFIN_initial

REFIN

REFADJ

2.2R C



REFIN

(V V ) 80%

REFIN_Final REFIN_initial

2.2R C







The recommend SR is estimated by C

slew rate SR can be estimated

REFIN

as the following equation :



SR REFADJ



SR REFIN



REFADJ

DS8811A-02 March 2014www.richtek.com

RT8811A



Current Balance

The RT8811A implements internal current balance

mechanism in the current loop. The RT8811A senses per

phase current signal and compares it with the average

current. If the sensed current of any particular phase is

higher than average current, the on-time of this phase will

be adjusted to be shorter.

Current Limit

The RT8811A provides cycle-by-cycle current limit control

by detecting the PHASE voltage drop across the low-side

MOSFET when it is turned on. The current limit circuit

employs a unique “valley” current sensing algorithm. If

the magnitude of the current sense signal at PHASE is

above the current limit threshold, the PWM is not allowed

to initiate a new cycle.

In order to provide both good accuracy and a cost effective

solution, the RT8811A supports temperature compensated

MOSFET R

DS(ON)

sensing.

In an over-current condition, the current to the load exceeds

the average output inductor current. Thus, the output

voltage falls and eventually crosses the under-voltage

protection threshold, inducing IC shutdown.

Current Limit Setting

Current limit threshold can be set by a resistor (R

OCSET

between LGATE1 and GND. Once PVCC exceeds the

POR threshold and chip is enabled, an internal current

source I

OCSET

R =

OCSET

where I

(valley inductor current) and I

flows through R

OCSET

. The voltage across

OCSET

is stored as the current limit protection threshold

. After that, the current source is switched off.

can be determined using the following equation :

IR 40mV



VALLEY LGDS(ON)

represents the desired inductor limit current

VALLEY



OCSET

is current limit setting

OCSET

current which has a temperature coefficient to compensate

the temperature dependency of the R

If R

is not present, there is no current path for I

OCSET

DS(ON)

OCSET

to build the current limit threshold. In this situation, the

current limit threshold is internally preset to 300mV (typ.).

Negative Current Limit

The RT8811A supports cycle-by-cycle negative current

limit. The value of negative current limit is set as the

positive current limit. If negative inductor current is rising

to trigger negative current limit, the low-side MOSFET

will be turned off and the current will flow to input side

through the body diode of the high-side MOSFET. At this

time, output voltage tends to rise because this protection

limits current to discharge the output capacitor. In order

to prevent shutdown because of over-voltage protection,

the low-side MOSFET is turned on again 400ns after it is

turned off. If the device hits the negative current limit

threshold again before output voltage is discharged to the

target level, the low-side MOSFET is turned off and process

repeats. It ensures maximum allowable discharge

capability when output voltage continues to rise. On the

other hand, if the output is discharged to the target level

before negative current limit threshold is reached, the low-

side MOSFET is turned off, the high-side MOSFET is

then turned on, and the device resumes normal operation.

Output Over-V oltage Prote ction (OVP)

The output voltage can be continuously monitored for over-

voltage protection. If REFIN voltage is lower than 1.33V,

the output voltage threshold follows to absolute over-voltage

2V. If REFIN voltage is higher than 1.33V, the output voltage

)

threshold follows relative over-voltage 1.5 x V

OVP is triggered, UGATE goes low and LGATE is forced

high. The RT8811A is latched once OVP is triggered and

can only be released by PVCC or EN power on reset. A

5μs delay is used in OVP detection circuit to prevent false

trigger.

Output Under-Voltage Protection (UVP)

The output voltage can be continuously monitored for under-

voltage protection. When the output voltage is less than

40% of its set voltage, under-voltage protection is triggered

and then all UGATE and LGATE gate drivers are forced

low. There is a 3μs delay built into the UVP circuit to

prevent false transitions. During soft-start, the UVP blanking

time is equal to PGOOD blanking time.

REFIN

. When

DS8811A-02 March 2014 www.richtek.com

RT8811A



Thermal Monitoring and Temperature Reporting

The RT8811A provides thermal monitoring function via

sensing the TSNS pin voltage, and which can indicate

ambient temperature through the voltage divider R

and R

shown in Figure 12. The voltage of V

NTC

OTSET

TSNS

typically set to be higher than 1V, when ambient

temperature rises, V

will fall, the TALERT signal will

TSNS

be pulled to low level if TSNS voltage drops below 1V or

internal die temperature high than 95°C (typ.).

TSNS

OTSET

TSNS

NTC

Die Temperature

Internal

Sense

CMP

TALERT

Figure 12. External OTP Setting

where R

can be determined using the following equation :

OTSET

R = R V1

OTSET NTC,T C X

is the thermistor's resistance at OTP trigger

NTC,T°C







temperature.

The standard formula for the resistance of the NTC

thermistor as a function of temperature is given by :













R=Re

where R



NTC,T C 25 C

is the thermistor's nominal resistance at room

25°C









T 273 298

temperature 25°C, β (beta) is the thermistor's material

constant in Kelvins, and T is the thermistor's actual

temperature in Celsius.

MOSFET Gate Driver

The RT8811A integrates high current gate drivers for the

MOSFETs to obtain high efficiency power conversion in

synchronous Buck topology. A dead-time is used to prevent

the crossover conduction for high-side and low-side

MOSFETs. Because both the two gate signals are off

during the dead-time, the inductor current freewheels

through the body diode of the low-side MOSFET. The

freewheeling current and the forward voltage of the body

diode contribute to the power loss. The RT8811A employs

adaptive dead-time control scheme to ensure safe operation

without sacrificing efficiency. Furthermore, elaborate logic

circuit is implemented to prevent cross conduction. For

high output current applications, two power MOSFETs are

usually paralleled to reduce R

. The gate driver needs

DS(ON)

to provide more current to switch on/off these paralleled

MOSFETs. Gate driver with lower source/sink current

capability result in longer rising/falling time in gate signals,

and therefore higher switching loss. The RT8811A embeds

high current gate drivers to obtain high efficiency power

conversion.

Inductor Selection

Inductor plays an importance role in step-down converters

because the energy from the input power rail is stored in

it and then released to the load. From the viewpoint of

efficiency, the DC Resistance (DCR) of inductor should

be as small as possible to minimize the copper loss. In

addition, because inductor occupies most of the board

space, the size of it is also important. Low profile inductors

can save board space especially when the height has

limitation. However, low DCR and low profile inductors are

usually not cost effective.

Additionally, larger inductance results in lower ripple

current, which means the lower power loss. However, the

inductor current rising time increases with inductance value.

This means the transient response will be slower. Therefore,

the inductor design is a trade-off between performance,

size and cost.

In general, inductance is designed to let the ripple current

ranges between 20% to 40% of full load current. The

inductance can be calculated using the following equation :

VV V



L =

min

IN OUT OUT

FkI V



SW OUT_rated IN



where k is the ratio between inductor ripple current and

rated output current.

Input Capacitor Selection

Voltage rating and current rating are the key parameters

in selecting input capacitor. Generally, input capacitor has

a voltage rating 1.5 times greater than the maximum input

voltage is a conservatively safe design.

The input capacitor is used to supply the input RMS

current, which can be approximately calculated using the

following equation :

DS8811A-02 March 2014www.richtek.com

RT8811A

I = I 1

RMS OUT





OUT OUT



IN IN



The next step is to select proper capacitor for RMS current

rating. Use more than one capacitor with low Equivalent

Series Resistance (ESR) in parallel to form a capacitor

bank is a good design. Besides, placing ceramic capacitor

close to the Drain of the high-side MOSFET is helpful in

reducing the input voltage ripple at heavy load.

Output Capacitor Selection

The output filter capacitor must have ESR low enough to

meet output ripple and load transient requirement, yet have

high enough ESR to satisfy stability requirements. Also,

the capacitance must be high enough to absorb the inductor

energy going from a full load to no load condition without

tripping the OVP circuit. Organic semiconductor

capacitor(s) or special polymer capacitor(s) are

recommended.

MOSFET Selection

The majority of power loss in the step-down power

conversion is due to the loss in the power MOSFETs. For

low voltage high current applications, the duty cycle of

the high-side MOSFET is small. Therefore, the switching

loss of the high-side MOSFET is of concern. Power

MOSFETs with lower total gate charge are preferred in

such kind of application.

However, the small duty cycle means the low-side

MOSFET is on for most of the switching cycle. Therefore,

the conduction loss tends to dominate the total power

loss of the converter. To improve the overall efficiency, the

MOSFETs with low R

are preferred in the circuit

DS(ON)

design. In some cases, more than one MOSFET are

connected in parallel to further decrease the on-state

resistance. However, this depends on the low-side

MOSFET driver capability and the budget.

Thermal Considerations

For continuous operation, do not exceed absolute

maximum junction temperature. The maximum power

dissipation depends on the thermal resistance of the IC

package, PCB layout, rate of surrounding airflow, and

difference between junction and ambient temperature. The

maximum power dissipation can be calculated by the

following formula :

where T

the ambient temperature, and θ

D(MAX)

= (T

J(MAX)

− TA) / θ

J(MAX)

is the maximum junction temperature, TA is

is the junction to ambient

thermal resistance.

For recommended operating condition specifications, the

maximum junction temperature is 125°C. The junction to

ambient thermal resistance, θJA, is layout dependent. For

WQFN-24L 4x4 package, the thermal resistance, θJA, is

28°C/W on a standard JEDEC 51-7 four-layer thermal test

board. The maximum power dissipation at TA = 25°C can

be calculated by the following formula :

= (125°C − 25°C) / (28°C/W) = 3.57W for

D(MAX)

WQFN-24L 4x4 package

The maximum power dissipation depends on the operating

ambient temperature for fixed T

and thermal

J(MAX)

resistance, θJA. The derating curve in Figure 13 allows

the designer to see the effect of rising ambient temperature

on the maximum power dissipation.

4.0

3.5

3.0

2.5

2.0

1.5

1.0

Four-Layer PCB

0.5

Maximum Power Dissipation (W) 1

0.0 0 25 50 75 100 125

Ambient Temperature (°C)

Figure 13. Derating Curve of Maximum Power

Dissipation

DS8811A-02 March 2014 www.richtek.com

RT8811A

Layout Considerations

Layout is very important in high frequency switching

converter design. If designed improperly, the PCB could

radiate excessive noise and contribute to the converter

instability. Certain points must be considered before

starting a layout for the RT8811A.

 Place the RC filter as close as possible to the PVCC

pin.

 Keep current limit setting network as close as possible

to the IC. Routing of the network should avoid coupling

to high voltage switching node.

 Connections from the drivers to the respective gate of

the high-side or the low-side MOSFET should be as

short as possible to reduce stray inductance.

 All sensitive analog traces and components such as

VSNS, RGND, EN, PSI, VID, PGOOD, VREF,

VREFADJ, VREFIN and TSNS should be placed away

from high voltage switching nodes such as PHASE,

LGATE, UGATE, or BOOT nodes to avoid coupling. Use

internal layer(s) as ground plane(s) and shield the

feedback trace from power traces and components.

 Power sections should connect directly to ground

plane(s) using multiple vias as required for current

handling (including the chip power ground connections).

Power components should be placed to minimize loops

and reduce losses.

DS8811A-02 March 2014www.richtek.com

Outline Dimension

RT8811A

SEE DETAIL A

1 2

DETAIL A

Pin #1 ID and Tie Bar Mark Options

Note : The configuration of the Pin #1 identifier is optional,

1 2

but must be located within the zone indicated.

Dimensions In Millimeters Dimensions In Inches

Symbol

Min Max Min Max

A 0.700 0.800 0.028 0.031

A1 0.000 0.050 0.000 0.002

A3 0.175 0.250 0.007 0.010

b 0.180 0.300 0.007 0.012

D 3.950 4.050 0.156 0.159

Option 1 2.400 2.500 0.094 0.098

Option 2 2.650 2.750 0.104 0.108

E 3.950 4.050 0.156 0.159

Option 1 2.400 2.500 0.094 0.098

Option 2 2.650 2.750 0.104 0.108

e 0.500 0.020

L 0.350 0.450 0.014 0.018

Richtek Technology Corporation

14F, No. 8, Tai Yuen 1st Street, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

W-Type 24L QFN 4x4 Package

Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should

obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot

assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be

accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third

parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

DS8811A-02 March 2014 www.richtek.com

Richtek RT8811AGQW Schematic [ru]

Specifications and Main Features

Frequently Asked Questions

User Manual