Rainbow Electronics MAX863 User Manual

_______________General Description

The MAX863 dual-output DC-DC converter contains
two independent step-up controllers in a single com­pact package. This monolithic Bi-CMOS design draws
only 85µA when both controllers are on. The input
range extends down to 1.5V, permitting use in organiz­ers, translators, and other low-power hand-held prod­ucts. The MAX863 provides 90% efficiency at output
loads from 20mA to over 1A. This space-saving device
is supplied in a 16-pin QSOP package that fits in the
same area as an 8-pin SOIC.

The device uses a current-limited, pulse-frequency­modulated (PFM) control architecture that reduces start­up surge currents and maintains low quiescent currents
for excellent low-current efficiency. Each controller
drives a low-cost, external, N-channel MOSFET switch,
whose size can be optimized for any output current or
voltage.

In larger systems, two MAX863s can be used to gener­ate 5V, 3.3V, 12V, and 28V from just two or three bat­tery cells. An evaluation kit (MAX863EVKIT) is available
to speed designs. For a single-output controller, refer to
the MAX608 and MAX1771 data sheets.

________________________Applications

2- and 3-Cell Portable Equipment

Organizers

Translators

Hand-Held Instruments

Palmtop Computers

Personal Digital Assistants (PDAs)

Dual Supply (Logic and LCD)

____________________________Features

♦ Smallest Dual Step-Up Converter: 16-Pin QSOP

♦ 90% Efficiency

♦ 1.5V Start-Up Voltage

♦ 85µA Max Total Quiescent Supply Current

♦ 1µA Shutdown Mode

♦ Independent Shutdown Inputs

♦ Drives Surface-Mount, Dual N-Channel MOSFETs

♦ Low-Battery Input/Output Comparator

♦ Step-Up/Down Configurable

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

________________________________________________________________ Maxim Integrated Products 1

__________________Pin Configuration

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

SENSE1 REF

SHDN2

LBI

LBO

FB2

SHDN1

CS2

EXT2

TOP VIEW

MAX863

QSOP

V

DD

FB1

EXT1

BOOT

CS1

GND

PGND

__________Typical Operating Circuit

19-1218; Rev 2; 2/98

PART

MAX863C/D

MAX863EEE -40°C to +85°C

0°C to +70°C

TEMP. RANGE PIN-PACKAGE

Dice*

16 QSOP

EVALUATION KIT MANUAL

AVAILABLE

______________Ordering Information

*Dice are tested at TA= +25°C.

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

V

IN

OUT1

SENSE1 V

N

EXT1

CS1

BOOT

DD

EXT2

CS2

OUT2

N

MAX863

PGND

FB2

SHDN1

SHDN2

REF

GND

ON/OFF

LOW-BATTERY
DETECTOR OUTPUT

LBO

LBI

FB1

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VDD= +5V, I

LOAD

= 0mA, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

VDDto GND............................................................-0.3V to +12V

PGND to GND .......................................................-0.3V to +0.3V

SHDN1, SHDN2, SENSE1, LBO to GND ................-0.3V to +12V

EXT1, EXT2 to PGND..................................-0.3V to (V

DD

+ 0.3V)

FB1, FB2, CS1, CS2, SEL,

LBI, BOOT to GND.................................-0.3V to (V

DD

+ 0.3V)

LBO Continuous Output Current.........................................15mA

EXT1, EXT2 Continuous Output Current .............................50mA

Continuous Power Dissipation (T

A

= +70°C)

QSOP (derate 8.30mW/°C above +70°C) ...................667mW

Operating Temperature Range

MAX863EEE ....................................................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +160°C

Lead Temperature (soldering, 10sec) .............................+300°C

V

DD

= OUT1 = BOOT (Note 1)

CONDITIONS

1.5 11

UNITSMIN TYP MAXSYMBOLPARAMETER

(Note 2)

V

2.7 11

V

DD

VDDInput Voltage

SHDN1 = VDD, SHDN2 = GND,
measured from V

DD

µA

35 60

I

DD

Quiescent Current

SHDN1 = SHDN2 = VDD, measured from V

DD

50 85

VIN= 2.7V to 5V, V

OUT1

= 5V,

I

LOAD

= 300mA, Figure 2

mV/V8Line Regulation

VIN= 3.3V, V

OUT1

= 5V,

I

LOAD

= 0mA to 500mA, Figure 2

mV/A40Load Regulation

nA210IFB, I

LBI

FB1, FB2, LBI Input Current

VDD= 1.5V

V

0.7 x V

DD

V

IH

2.7V < VDD< 11V 1.6

SHDN1, SHDN2, SEL, BOOT
Input High Voltage

mV85 100 115V

CS

CS1, CS2 Threshold Voltage

µs14 17.5 22

Logic input = VDDor GND

t

ON

Maximum Switch On-Time

µA125CS1, CS2 Input Current

µA1I

I

SHDN1, SHDN2, SEL, BOOT
Input Current

V1.225 1.25 1.275VFB, V

LBI

FB1, FB2, LBI
Threshold Voltage (Note 4)

C

LOAD

= 1nF, 10% to 90% ns50EXT Rise/Fall Time (Note 5)

µs1.6 2 2.4t

OFF

Minimum Switch Off-Time

FB1 = GND

V

4.85 5 5.15

V

OUT1

OUT1 Output Voltage
(Note 3)

FB1 = V

DD

3.2 3.3 3.4

VDD= 1.5V

V

0.2 x V

DD

V

IL

2.7V < VDD< 11V 0.4

SHDN1, SHDN2, SEL, BOOT
Input Low Voltage

SHDN1 = SHDN2 = GND

µA1I

DD, SHDN

Shutdown Current

Ω

5EXT On-Resistance

V

LBO

= 11V, V

LBI

> 1.275V µA1I

LBO

LBO Leakage Current

I

LBO,SINK

= 1mA, V

LBI

< 1.225V V0.1 0.4V

LBO,L

LBO Low Level

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS

(VDD= +5V, I

LOAD

= 0mA, TA= -40°C to +85°C, unless otherwise noted.) (Note 6)

Note 1: When bootstrapped, an internal low-voltage oscillator drives the EXT1 pin rail-to-rail for low supply voltages.
Note 2: For non-bootstrapped operation, V

DD

> 2.7V is required to allow valid operation of all internal circuitry.

Note 3: For adjustable output voltages, see the Set the Output Voltage section.
Note 4: Measured with LBI falling. Typical hysteresis is 15mV.
Note 5: EXT1 and EXT2 swing from V

DD

to GND.

Note 6: Specifications to -40°C are guaranteed by design and not production tested.

VDD= OUT1 (Note 1) 1.6 11

CONDITIONS

VDDInput Voltage

(Note 2)

V

2.8 11

V

DD

V1.21 1.285V

FB

FB1, FB2 Threshold Voltage

60

mV85 115V

CS

CS1, CS2 Threshold Voltage

UNITSMIN TYP MAXSYMBOLPARAMETER

FB1 = V

DD

3.15 3.45

OUT1 Output Voltage
(Note 3)

FB1 = GND

V

4.8 5.2

V

OUT1

SHDN1 = SHDN2 = VDD, measured from V

DD

85

Quiescent Current

SHDN1 = VDD, SHDN2 = GND,
measured from V

DD

µAI

DD

SHDN1 = SHDN2 = GND

µA1I

DD, SHDN

Shutdown Current

__________________________________________Typical Operating Characteristics

(TA = +25°C, unless otherwise noted.)

0.01 0.1 1 10 100 1000

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT1

= 3.3V, BOOTSTRAPPED)

MAX863 toc01

OUTPUT CURRENT (mA)

EFFICIENCY (%)

B

C

A

10

0

30

20

50

40

70

60

90

80

100

V

OUT1

= 3.3V

A: V

IN

= 1.5V

B: V

IN

= 2.4V

C: V

IN

= 2.7V

0.01 0.1

10

0

30

20

50

40

70

60

90

80

100

1 10 100 1000

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT1

= 5.0V, BOOTSTRAPPED)

MAX863 toc02

OUTPUT CURRENT (mA)

EFFICIENCY (%)

V

OUT1

= 5.0V

A: V

IN

= 1.5V

B: V

IN

= 2.4V

C: V

IN

= 2.7V

D: V

IN

= 3.3V

E: V

IN

= 3.6V

F: V

IN

= 4.0V

B

C

A

D

E

F

0.01 0.1

0

10

20

30

40

50

60

70

80

90

100

1 10 100 1000

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT1

= 5.0V, NON-BOOTSTRAPPED)

MAX863 toc03

OUTPUT CURRENT (mA)

EFFICIENCY (%)

V

OUT1

= 5.0V

A: V

IN

= 2.7V

B: V

IN

= 3.3V

C: V

IN

= 3.6V

D: V

IN

= 4.0V

A

B

C

D

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

4 _______________________________________________________________________________________

____________________________Typical Operating Characteristics (continued)

(TA = +25°C, unless otherwise noted.)

0.01 0.1

10

20

30

40

50

60

70

80

90

100

0

1 10 100 1000

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT1

= 12V, NON-BOOTSTRAPPED)

MAX863 toc04

OUTPUT CURRENT (mA)

EFFICIENCY (%)

V

OUT1

= 5.0V

A: V

IN

= 2.7V

B: V

IN

= 3.3V

C: V

IN

= 3.6V

D: V

IN

= 4.0V

E: V

IN

= 6.0V

A

B

C

D

E

3.5

1

1 10 1000

BOOTSTRAPPED-MODE MINIMUM

START-UP INPUT VOLTAGE

vs. OUTPUT CURRENT

0.5

1.0

1.5

2.0

2.5

3.0

MAX863toc05

OUTPUT CURRENT (mA)

START-UP INPUT VOLTAGE (V)

100

V

OUT1

= 3.3V

V

OUT1

= 5V

0

012

VDD CURRENT

vs. V

DD

VOLTAGE

10

20

60

MAX863 toc15

VDD VOLTAGE (V)

V

DD

CURRENT (µA)

40

30

8

50

10

24 6

Cond: Single +5V

BOTH ON

CONVERTER 1 ON

CONVERTER 2 ON

LOAD-TRANSIENT RESPONSE

A

MAX863 toc08

B

100µs/div

V

OUT1

= 3.3V, I

OUT1

= 100mA TO 600mA

A: V

OUT1

, 100mV/div, 3.3V DC OFFSET

B: I

OUT1

, 200mA/div

RESPONSE ENTERING/

EXITING SHUTDOWN (BOOTSTRAPPED)

B

A

MAX863 toc09

C 3.3V

200µs/div

V

OUT1

= 3.3V, I

OUT1

= 100mA, VIN = 2.4V
A: SHDN1, 5V/div
B: INDUCTOR CURRENT, 2A/div
C: V

OUT1

, 3.3V OFFSET, 500mV/div

LINE-TRANSIENT RESPONSE

B

A

MAX863 toc10

C0A

500µs/div

V

OUT1

= 5V, I

OUT1

= 800mA

A: V

IN

= 2.7V TO 3.7V, 500mV/div

B: V

OUT1

, AC COUPLED, 50mV/div

C: INDUCTOR CURRENT, 2A/div

0

012

EXT RISE AND FALL TIMES vs.

SUPPLY VOLTAGE AND MOSFET CAPACITANCE

20

140

MAX863 toc07

SUPPLY VOLTAGE (V)

RISE/FALL TIME (ns)

6

60

80

40

24 8

120

100

C,1

C,2

B,1

B,2

A,1

A,2

10

A: 470pF
B: 1.0nF
C: 2.2nF
1: RISE
2: FALL

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

_______________________________________________________________________________________ 5

_______________Detailed Description

The MAX863 dual, bi-CMOS, step-up, switch-mode
power-supply controller provides preset 3.3V, 5V, or
adjustable outputs. Its pulse-frequency-modulated
(PFM) control scheme combines the advantages of low
supply current at light loads and high efficiency with
heavy loads. These attributes make the MAX863 ideal
for use in portable battery-powered systems where
small size and low cost are extremely important, and
where low quiescent current and high efficiency are
needed to maximize operational battery life. Use of
external current-sense resistors and MOSFETs allows
the designer to tailor the output current and voltage
capability for a diverse range of applications.

PFM Control Scheme

Each DC-DC controller in the MAX863 uses a one-shot­sequenced, current-limited PFM design, as shown in
Figure 1. Referring to the Typical Operating Circuit
(Figure 2) and the switching waveforms (Figures 3a–3f),
the circuit works as follows. Output voltage is sensed
by the error comparator using either an internal voltage
divider connected to SENSE1 or an external voltage
divider connected to FB1. When the output voltage
drops, the error comparator sets an internal flip-flop.
The flip-flop turns on an external MOSFET, which allows
inductor current to ramp-up, storing energy in a mag­netic field.

______________________________________________________________Pin Description

PIN

Feedback Input for DC-DC Controller 1 in Fixed-Output ModeSENSE11

FUNCTIONNAME

IC Power-Supply InputV

DD

2

Bootstrap Low-Voltage-Oscillator Enable Input. BOOT is an active-high, logic-level input. It enables the
low-voltage oscillator to allow start-up from input voltages down to 1.5V while in a bootstrapped circuit
configuration. Connect BOOT to GND when in a non-bootstrapped configuration. If BOOT is high, V

DD

must be connected to OUT1.

BOOT4

Adjustable Feedback and Preset Output Voltage Selection Input for DC-DC Controller 1. Connect to V

DD

for 3.3V preset output or to GND for 5V output. Connect a resistor voltage divider to adjust the output volt­age. See the section Set the Output Voltage.

FB13

Gate-Drive Output of DC-DC Controller 1. Drives an external N-channel power MOSFET.EXT16

High-Current Ground Return for Internal MOSFET DriversPGND8

Analog Ground for Internal Reference, Feedback, and Control CircuitsGND7

Input to the Current-Sense Comparator of DC-DC Controller 1CS15

Input to the Current-Sense Amplifier of DC-DC Controller 2CS210

Adjustable Feedback Input for DC-DC Controller 2. Connect a resistor voltage divider to adjust the output
voltage. See the section Set the Output Voltage.

FB212

Active-Low Shutdown Input for DC-DC Controller 1. Connect to VDDfor normal operation.

SHDN1

11

Low-Battery Comparator Input. When the voltage on LBI drops below 1.25V, LBO sinks current. If unused,
connect to GND.

LBI14

Reference Bypass Input. Connect a 0.1µF ceramic capacitor from REF to GND.REF16

Active-Low Shutdown Input for DC-DC Controller 2. Connect to VDDfor normal operation.

SHDN2

15

Low-Battery Output. An open-drain N-channel MOSFET output. Sinks current when the voltage on LBI
drops below 1.25V. If unused, connect to GND.

LBO13

Gate-Drive Output of DC-DC Controller 2. Drives an external N-channel power MOSFET.EXT29

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

6 _______________________________________________________________________________________

The flip-flop resets and turns off the MOSFET when
either a) the voltage across the current-sense resistor
exceeds 100mV, or b) the 17.5µs maximum on-time
one-shot trips. When the MOSFET turns off, the mag­netic field begins to collapse, and forces current into
the output capacitor and load. As the stored energy is
transferred to the output, the inductor current ramps
down. The output capacitor smoothes out the energy
transfer by storing charge when the diode current is

high, then supplying current to the load during the first
half of each cycle, maintaining a steady output voltage.
Resetting the flip-flop sets the off-time one-shot, dis­abling the error-comparator output and forcing the
MOSFET off for at least 2µs to enforce a minimum time
for energy transfer to the output. The MAX863 waits
until the output voltage drops again before beginning
another cycle. The MAX863’s switching frequency
depends on the load current and input voltage.

Figure 1. Functional Diagram

FB2

CS2

CS1

SENSE1

FB1

V

DD

100mV

- 100mV

REF

100mV

ERROR

COMPARATOR

CURRENT-SENSE

COMPARATOR

TRIG

MIN ON-TIME

ONE-SHOT

100mV

REF

Q

Q

MAX ON-TIME

ONE-SHOT

CURRENT-

SENSE

COMPARATOR

ERROR

COMPARATOR

TRIG

S

Q

R

OSCILLATOR

TIMING
BLOCK

LOW-

VOLTAGE

TIMING
BLOCK

MAX863

UVLO

EXT2

PGND

BOOT

EXT1

V

DD

LBO

LBI

REF

N

N

V

DD

REF

1.25V

GND SHDN2 SHDN1REF

BIAS

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

_______________________________________________________________________________________ 7

Continuous/Discontinuous-Conduction

Modes

Each converter in the MAX863 determines from moment
to moment whether to switch or not, waiting until the out­put voltage drops before initiating another cycle. Under
light loads, the inductor current ramps to zero before the
next cycle; this is discontinuous-conduction mode.
Continuous-conduction mode occurs when the next
switching cycle begins while current is still flowing
through the inductor. The transition point between dis­continuous- and continuous-conduction mode is deter­mined by input and output voltages, and by the size of
the inductor relative to the peak switching current. In
general, reducing inductance toward the minimum rec­ommended value pushes the transition point closer to
the maximum load current. If the inductor value is low
enough or the output/input voltage ratio high enough,
the DC-DC converter may remain in discontinuous-con­duction mode throughout its entire load range.

The MAX863 transitions into continuous-conduction
mode in two ways, depending on whether preset or
adjustable mode is used and how the external feed­back network is compensated. Under light loads, the IC
switches in single pulses (Figure 3a). The threshold of
transition into continuous-conduction mode is reached
when the inductor current waveforms are adjacent to
one another, as shown in Figure 3b. As the load
increases, the transition into continuous-conduction
mode progresses by raising the minimum inductor cur­rent (Figures 3c, 3d). Depending on feedback compen­sation, transition into continuous-conduction mode may
also progress with grouped pulses (Figures 3e, 3f).
Pulse groups should be separated by less than two or
three switching cycles. Output ripple should not be
significantly more than the single-cycle no-load case.

V

Figure 2. Bootstrapped Typical Operating Circuit

= 1.5V TO THE LOWER OF V

IN

V

= 5V

OUT1

C1

220µF

10V

≤0.1Ω

R5

OR V

OUT1

OUT2

C4

100µF

10V

≤0.1Ω

BOOT

DD

EXT2

CS2

50mΩ

R2

L2
10µH
2A

MBRS340T3

N1B
IRF7301

D2

V

= 3.3V

OUT2

C5
330µF
10V
≤0.1Ω

R3

165k

1%

C6
10pF

C2

0.1µF

100k

R7

D1

MBRS340T3

L1
10µH
2A

R1
50mΩ

N1A

C3

100µF

10V

≤0.1Ω

SENSE1 V

EXT1

CS1

MAX863

LOW-BATTERY
DETECTOR OUTPUT

R6

LBO

LBI

FB1

PGND

FB2

SHDN1

SHDN2

REF

GND

ON/OFF

C7

0.1µF

R4
100k
1%

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

8 _______________________________________________________________________________________

Figure 4a. Non-Bootstrapped Typical Operating Circuit

Figures 3a–3f. MAX863 Switching Waveforms During Transition into Continuous Conduction

A

B

C

20µs/div

= 287mA

a) I

OUT1

V

= 3.3V

OUT1

PLOTS a-d: INTERNAL FEEDBACK
PLOTS e-f: UNCOMPENSATED,
EXTERNAL FEEDBACK
A: MOSFET DRAIN, 2V/div

1, 100mV/div, 3.3V DC OFFSET

B: V

OUT

C: INDUCTOR CURRENT, 1A/div

b) I

20µs/div

= 608mA

OUT1

A

B

C

c) I

e) I

20µs/div

= 767mA

OUT1

20µs/div

= 757mA

OUT1

= 2.7V TO THE LOWER OF V

V

IN

= 5V

V

OUT1

C1

220µF

10V

≤0.1Ω

R5

R6

OR V

OUT1

OUT2

D1

MBRS340T3

R7

100k

LOW-BATTERY
DETECTOR OUTPUT

N1.A

L1
10µH
2A

R1
50mΩ

100µF

10V

≤0.1Ω

C3

SENSE1

EXT1

CS1

LBO

LBI

FB1

MAX863

BOOT

PGND

C2

0.1µF

V

DD

EXT2

CS2

FB2

SHDN1

SHDN2

REF

GND

100µF

10V

≤0.1Ω

C4

50mΩ

R2

N1.B
IRF7301

ON/OFF

C7

0.1µF

L2
10µH
2A

MBRS340T3

1M

OV

3.3V

0A

20µs/div

= 1.01A

d) I

OUT1

OV

3.3V

0A

20µs/div

= 881mA

f) I

OUT1

D2

R3
1M
1%

R4
115k
1%

V

C6
10pF

OUT2

= 12V

C5
100µF
20V
≤0.1Ω

C8
82pF

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

_______________________________________________________________________________________ 9

Low-Voltage Start-Up Oscillator

(BOOT Pin)

The MAX863 features a low-voltage start-up oscillator
that guarantees start-up in bootstrapped configuration
down to 1.5V. At these low supply voltages, the error
comparator and internal biasing of the IC are locked
out. The low-voltage oscillator switches the external
MOSFET with around 30% duty cycle until the voltage
at VDDrises above 2.7V. At this point, the error com­parator and one-shot timing circuitry turn on. The low­voltage oscillator is enabled by connecting the BOOT
pin to VDD. When BOOT is high, VDDmust be connect­ed to V

OUT1

.

Use the start-up oscillator in the bootstrapped configu­ration only, since the MAX863 operates in an open-loop
state while the start-up oscillator is active. When using

a non-bootstrapped circuit configuration, connect
BOOT to GND to disable the start-up oscillator. This

prevents the output from rising too high when V

DD

is
between 1.5V and 2.7V, such as during power-up and
low-battery conditions.

Bootstrapped/Non-Bootstrapped Modes

Figures 2 and 4 show standard applications in boot­strapped and non-bootstrapped modes. In boot-

strapped mode, the IC is powered from the output (V

DD

is connected to OUT1, BOOT is connected to VDD).
Bootstrapped-mode operation is useful for increasing
the gate drive to the MOSFETs in low-input-voltage
applications, since EXT1 and EXT2 swing from VDDto
GND. Increasing the gate-drive voltage reduces MOS­FET on-resistance, which improves efficiency and
increases the load range. For supply voltages below
5V, bootstrapped mode is recommended. In boot­strapped mode, the output connected to VDDmust not
exceed 11V. If BOOT is high, VDDmust be connect­ed to OUT1.

In non-bootstrapped mode, the IC is powered by a
direct connection from the input voltage to VDD. Since
the voltage swing applied to the gate of the external
MOSFET is derived from VDD, the external MOSFET on­resistance increases at low input voltages. The mini­mum input voltage is 2.7V. For operation down to 4V,
use logic-level MOSFETs. For lower input voltages, use
low-threshold logic-level MOSFETs. When both output
voltages are set above 11V, non-bootstrapped mode is
mandatory.

Figure 4b. Adjustable Non-Bootstrapped Typical Operating Circuit

V

= 2.7V TO 11V

IN

C3

R1
50mΩ

100µF
20V
≤0.1Ω

EXT1

CS1

LBO

FB1

SENSE1

BOOT

V

DD

MAX863

V

100µF

16V

≤0.1Ω

10pF

82pF

OUT1

C1

C8

C9

= 12V

R8
1M
1%

R9
115k
1%

L1

10µH

2A

D1

MBRS340T3

N1.A

R7
100k

LOW-BATTERY
DETECTOR OUTPUT

PGND

C2

0.1µF

LBI

EXT2

CS2

FB2

SHDN1

SHDN2

REF

GND

R6R5

C4
100µF
20V
≤0.1Ω

100mΩ

R2

L2
10µH
1A

N1.B
IRF7301

ON/OFF

C7

0.1µF

D2

MBRS140

R3

1M

1%

R4
56k
1%

V

OUT2

C6
15pF

C10
270pF

= 24V

C5
22µF
35V

0.1Ω

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

10 ______________________________________________________________________________________

Shutdown Mode

The MAX863 has two shutdown inputs useful for con­serving power and extending battery life. Driving
SHDN1 or SHDN2 low turns off the corresponding DC­DC controller and reduces quiescent current. Driving
both shutdown pins low turns off the reference, control,
and biasing circuitry, putting the MAX863 in a 1µA
shutdown mode. Connect SHDN1 and SHDN2 to V

DD

for normal operation.

__________________Design Procedure

Boost DC-DC converters using the MAX863 can be
designed in a few simple steps to yield a working first­iteration design. All designs should be prototyped and

tested prior to production. Table 1 provides a list of
component suppliers.

Two design methods are included. The first uses
graphs for selecting the peak current required for 3.3V,
5V, 12V, and 24V outputs. The second uses equations
for selecting the peak current and inductor value in cir­cuits with other outputs. When designing high-voltage,
flyback, SEPIC, and autotransformer boost circuits,
contact Maxim’s Applications Department for the
appropriate design equations.

Specify Design Objectives

For each of the two outputs, specify the output voltage
and maximum load current, as well as maximum and

Figure 5d. Maximum Output Current vs. Input Voltage and
I

PEAK

(V

OUT

= 24V)

Figure 5b. Maximum Output Current vs. Input Voltage and
I

PEAK

(V

OUT

= 5V)

Figure 5c. Maximum Output Current vs. Input Voltage and
I

PEAK

(V

OUT

= 12V)

Figure 5a. Maximum Output Current vs. Input Voltage and
I

PEAK

(V

OUT

= 3.3V)

2.5
V

= 3.3V, L = 1.5 L

OUT

A: I

= 3A

PEAK

2.0

1.5

1.0

0.5

MAXIMUM OUTPUT CURRENT (A)

0

= 2A

B: I

PEAK

= 1.47A

C: I

PEAK

= 1A

D: I

PEAK

= 0.67A

E: I

PEAK

= 0.5A

F: I

PEAK

1.4 2.4

1.0 3.0

Cond: Single +5V
Code = FFFhex

MIN

2.0

INPUT VOLTAGE (V)

A

MAX863 FIG05A

B

C

D

E

F

2.81.81.2 1.6 2.2

2.6

2.0
V

= 5V, L = 1.5 L

OUT

1.8
A: I

PEAK

B: I

PEAK

1.6
C: I

PEAK

1.4
D: I

PEAK

E: I

PEAK

1.2
F: I

PEAK

1.0

0.8

0.6

0.4

MAXIMUM OUTPUT CURRENT (A)

0.2

0

1.0 4.5

Cond: Single +5V
Code = FFFhex

= 3A
= 2A

= 1.47A

= 1A
= 0.67A
= 0.5A

2.51.5 2.0 3.0

INPUT VOLTAGE (V)

MIN

A

MAX863 FIG05B

B

C

D

E

F

3.5

4.0

2.5
V
A: I
B: I

2.0
C: I

D: I
E: I

1.5
F: I

1.0

0.5

MAXIMUM OUTPUT CURRENT (A)

0

012

Cond: Single +5V
Code = FFFhex

= 12V, L = 1.5 L

OUT

= 3A

PEAK

= 2A

PEAK

= 1.47A

PEAK

= 1A

PEAK

= 0.67A

PEAK

= 0.5A

PEAK

24 6

INPUT VOLTAGE (V)

MIN

A

MAX863 FIG05C

B

C

D

E

F

8

10

1.0
V

0.9
A: I

B: I

0.8
C: I

0.7
D: I

E: I

0.6
F: I

0.5

0.4

0.3

0.2

MAXIMUM OUTPUT CURRENT (A)

0.1

0

012

Cond: Single +5V
Code = FFFhex

= 24V, L = 1.5 L

OUT

= 3A

PEAK

= 2A

PEAK

= 1.47A

PEAK

= 1A

PEAK

= 0.67A

PEAK

= 0.5A

PEAK

24 6

INPUT VOLTAGE (V)

MIN

8

A

MAX863 FIG05D

B

C

D

E

F

10

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

______________________________________________________________________________________ 11

minimum input voltages. Estimate the maximum input
currents for each output based on the minimum input
voltage and desired output power:

where 0.8 is chosen as a working value for the nominal
efficiency. The power source must be capable of deliv­ering the sum of the maximum input currents of both
DC-DC converters.

Determine the Peak Switching Current

(Graphical Method)

The peak switching current set by R

SENSE

determines
the amount of energy transferred from the input on
each cycle. For 3.3V, 5V, 12V, and 24V output circuits,
the peak current can be selected using the output cur­rent curves shown in Figures 5a–5d.

Determine the Peak Switching Current and

Inductance (Analytical Method)

The following boost-circuit equations are useful when
the desired output voltage differs from those listed in
Figure 5. They allow trading off peak current and induc­tor value in consideration of component availability,
size, and cost.

Begin by calculating the minimum allowable ratio of
inductor AC ripple current to peak current, ξ

MIN

(Figure 6):

where t

OFF(MIN)

= 2µs and t

ON(MAX)

= 17.5µs.

Select a value for ξ greater than ξ

MIN

. If ξ

MIN

is less

than 1, an acceptable choice is (ξ

MIN

+ 1) / 2. If ξ

MIN

is

greater than 1, values between ξ

MIN

and 2 x ξ

MIN

are

acceptable (1.5 x ξ

MIN

, for example). Values greater
than 1 represent designs with full-load operation in dis­continuous-conduction mode.

Now calculate the peak switching current and induc­tance. If ξ

MIN

≤ξ≤1, use:

For ξ≥1%, use:

The suggested inductor value is:

Round L up to the next standard inductor value.

Choose R

SENSE

The peak switching current is set by R

SENSE

(R1 and

R2 in Figure 2):

Verify that you’ve selected the correct R

SENSE

by test­ing the prototype using the minimum input voltage
while supplying the maximum output current. If the out­put voltage droops, then decrease the value of the cur­rent-sense resistor and adjust the other components as
necessary.

The current-sense resistor must be a small, low-induc­tance type such as a surface-mount metal-strip resistor.
Do not use wire-wound resistors, since their high induc­tance will corrupt the current feedback signal. In order
to allow use of standard resistor values, round R

SENSE

to the next lowest value.

The current-sense resistor’s power rating should be
higher than:

Figure 6. Ratio of Inductor AC Ripple Current to Peak Current

I

IN,DC MAX

()

V x I

OUT OUT

≅

0.8 x V

IN MIN

()

ξ

MIN

=

t

OFF MIN

()

t

ON MAX

x

()

VV

−

OUT IN MIN

()

V

IN MIN

()

∆I

L

ξ

=

MIN

I

L

INDUCTOR CURRENT, I

L

≅

PEAK

∆I

L

I

PEAK

t



V - V x t

OUT IN MIN OFF MIN





() ()



Ix

PEAK

ξ

R

SENSE

V

CS MIN

()

≤=

I

PEAK

85mV

I

PEAK

I = 2 x I x

PEAK IN,DC MAX

I = I x

PEAK IN,DC MAX

()

()

V + V x

OUT IN

2-

V

ξ

OUT

2

−

ξ 1

()

R

POWER RATING

2

V

CS MAX

=

()

R

SENSE

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

12 ______________________________________________________________________________________

Select the Inductor Component

Two essential parameters are required for selecting the
inductor: inductance and current rating.

Inductance should be low enough to allow the MAX863
to reach the peak current limit during each cycle before
the 17.5µs maximum on-time. Conversely, if the induc­tance is too low, the current will ramp up to a high level
before the current-sense comparator can turn the
switch off. A practical minimum on-time (t

ON(MIN)

) is

1.5µs.

and:

When selecting I

PEAK

using the graphs in Figure 5,
choose inductance values between 1.3 and 1.7 times
the minimum inductance value to provide a good trade­off between switching frequency and efficiency.

The lower of the inductor saturation current rating or
heating current rating should be greater than I

PEAK

:

I

SATURATION

and I

HEATING

> I

PEAK

The saturation current limit is the current level where
the magnetic field in the inductor has reached the max­imum the core can sustain, and inductance starts to
fall. The heating current rating is the maximum DC cur­rent the inductor can sustain without overheating.
Disregarding the inductor’s saturation current rating is
a common mistake that results in poor efficiency, bad
regulation, component overheating, or other problems.
The resistance of the inductor windings should be com­parable to or less than that of the current-sense
resistor. To minimize radiated noise in sensitive
applications, use a toroid, pot core, or shielded bobbin
core inductor.

Choose the MOSFET Power Transistor

Use N-channel MOSFETs with the MAX863. When
selecting an N-channel MOSFET, five important para­meters are gate-drive voltage, drain-to-source break­down voltage, current rating, on-resistance (R

DS(ON)

),

and total gate charge (Qg).

The MAX863’s EXT1 and EXT2 outputs swing from
GND to VDD. To ensure the external N-channel MOS­FET is turned on sufficiently, use logic-level MOSFETs
when VDDis less than 8V and low-threshold logic-level

MOSFETs when starting from input voltages below 4V.
This also applies in bootstrapped mode to ensure
start-up.

The MOSFET in a simple boost converter must with­stand the output voltage plus the diode forward volt­age. Voltage ratings in SEPIC, flyback, and
autotransformer-boost circuits are more stringent.
Choose a MOSFET with a maximum continuous drain­current rating higher than the current limit set by CS.

The two most significant losses contributing to the
MOSFET’s power dissipation are I2R losses and switch­ing losses. Reduce I

2

R losses by choosing a MOSFET

with low R

DS(ON)

, preferably near the current-sense

resistor value or lower.

A MOSFET with a gate charge (Qg) of 50nC or smaller
is recommended for rise and fall times less than 100ns
on the EXT pins. Exceeding this limit results in slower
MOSFET switching speeds and higher switching loss­es, due to a longer transition time through the linear
region as the MOSFET turns on and off.

Select the Output Diode

Schottky diodes, such as the 1N5817–1N5822 family or
surface-mount equivalents, are recommended. Ultra­fast silicon rectifiers with reverse recovery times around
60ns or faster, such as the MUR series, are acceptable
but have greater forward voltage drop. Make sure that
the diode’s peak current rating exceeds the current
limit set by R

SENSE

, and that its breakdown voltage

exceeds V

OUT

. Schottky diodes are preferred for heavy
loads, especially in low-voltage applications, due to
their low forward voltage. For high-temperature applica­tions, some Schottky diodes may be inadequate due to
high leakage currents. In such cases, ultra-fast silicon
rectifiers are recommended, although acceptable per­formance can often be achieved by using a Schottky
diode with a higher reverse voltage rating.

Determine Input and Output Filter

Capacitors

Low-ESR capacitors are recommended for both input
bypassing and output filtering. Capacitor equivalent
series resistance (ESR) is a major contributor to output
ripple—typically 60% to 90%. Low-ESR tantalum
capacitors offer a good tradeoff between price and
performance. Ceramic and Sanyo OS-CON capacitors
have the lowest ESR. Ceramic capacitors are often a
good choice in high-output-voltage applications where
large capacitor values may not be needed. Low-ESR
aluminum-electrolytic capacitors are tolerable and can
be used when cost is the primary consideration; how­ever, standard aluminum-electrolytic capacitors should
be avoided.

L

MIN

L

MAX

≥

≥

V

IN MAX

V

IN MIN

xt

() ()

() ( )

ON MIN

I

PEAK

xt

ON MAX

I

PEAK

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

______________________________________________________________________________________ 13

Voltage ripple is the sum of contributions associated
with ESR and the capacitor value, as shown below:

V

RIPPLE

≅ V

RIPPLE,ESR

+ V

RIPPLE,C

To simplify selection, assume that 75% of the ripple
results from ESR and that 25% results from the capaci­tor value. Voltage ripple as a consequence of ESR is
approximated by:

V

RIPPLE,ESR

≅ R

ESR

x I

PEAK

so:

Estimate input and output capacitor values for a given
voltage ripple as follows:

where V is the input or output voltage, depending on
which capacitor is being calculated.

Choose input capacitors with working voltage ratings
over the maximum input voltage, and output capacitors
with working voltage ratings higher than their respec­tive outputs.

Add VDDand REF Bypass Capacitors

Bypass the MAX863 with 0.1µF or higher value ceramic
capacitors placed as close to the VDD, REF, and GND
pins as possible.

Set the Output Voltage

DC-DC converter 1 operates with a 3.3V, 5V, or
adjustable output. For a preset output, connect
SENSE1 to OUT1 (Figures 2 and 4a), then set FB1 to
VDDfor 3.3V operation or to GND for 5V operation. For
an adjustable output, connect a resistor voltage divider
to the FB1 pin (Figure 7). In adjustable output circuits,
connect SENSE1 to GND.

DC-DC converter 2 can be adjusted from very high
voltages down to VINusing external resistors connect­ed to the FB2 pin, as shown in Figure 7. Select feed­back resistor R2 in the 10kΩ to 500kΩ range. R1 is
given by:

where 1.25V is the voltage of the internal reference.

Figure 7. Adjustable Output Circuit

Table 1. Component Suppliers

PHONE

Inductors

SUPPLIER

Marcon/United
Chemi-Con

[phone]

TDK [phone]

Vishay/Vitramon [phone]

[phone]

[phone]

Large-Value Ceramic Capacitors

[phone]

Motorola [phone]

AVX [phone]

Sanyo USA [phone]

Sprague [phone]

[phone]

[phone]

Electrolytic Capacitors

[phone]

Dale/Vishay [phone]

IRC [phone]

[phone]

[phone]

[phone]

Current-Sense Resistors

Sumida USA [phone]

Central Semiconductor [phone]

International Rectifier [phone]

[phone]

[phone]

[phone]

MOSFETs and Diodes

Coiltronics [phone]

Dale Inductors [phone]

[phone]

[phone]

FAX

[phone]Coilcraft [phone]

OR V

V

OUT1

OUT2

C

R1

MAX863

FB1 OR FB2

R2

C1
(OPTIONAL)

C2
(OPTIONAL FOR HIGH­VOLTAGE CIRCUITS)

OUT

R

ESR

V

≤

RIPPLE,ESR

I

PEAK

C

≥

Vx V

0.5L x I

RIPPLE,C

2

PEAK

R1 = R2



V

OUT



1.25V



−



1




MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

14 ______________________________________________________________________________________

Set Feedback Compensation

External voltage feedback to the MAX863 should be
compensated for stray capacitance and EMI in the
feedback network. Proper compensation is achieved
when the MAX863 switches evenly, rather than in wide­ly spaced bursts of pulses with large output ripple.
Typically, lead compensation consisting of a 10pF to
220pF ceramic capacitor (C1 in Figure 7) across the
upper feedback resistor is adequate. Circuits with
V

OUT

or VDDgreater than 7.5V may require a second
capacitor across the lower feedback resistor. Initially,
choose this capacitor so that R2C2 = R1C1. Set the
final values of the compensation capacitors based on
empirical analysis of a prototype.

PC Board Layout and Routing

High switching speeds and large peak currents make
PC board layout an important part of design. Poor lay­out can cause excessive EMI and ground-bounce, both
of which can cause instability or regulation errors by

corrupting the voltage and current-feedback signals.
Place power components as close together as possi­ble, and keep their traces short, direct, and wide. Keep
the extra copper on the board and integrate it into
ground as an additional plane. On multi-layer boards,
avoid interconnecting the ground pins of the power
components using vias through an internal ground
plane. Instead, place the ground pins of the power
components close together and route them in a “star”
ground configuration using component-side copper,
then connect the star ground to the internal ground
plane using multiple vias.

The current-sense resistor and voltage-feedback net­works should be very close to the MAX863. Noisy
traces, such as from the EXT pins, should be kept away
from the voltage-feedback networks and isolated from
them using grounded copper. Consult the MAX863
evaluation kit manual for a full PC board example.

Figure 8. Bootstrapped 3.3V Logic and 24V LCD Bias Supply

V

= 1.8V TO V

V

OUT1

220µF

10V

≤0.1Ω

IN

= 5V

C1

R5

OUT1

C2

0.1µF

100k

R7

D1

MBRS340T3

L1
10µH
2A

N1A

R1
50mΩ

C3

100µF

10V

≤0.1Ω

SENSE1

EXT1

CS1

C4

V

DD

MAX863

100µF

10V

≤0.1Ω

BOOT

EXT2

CS2

100mΩ

L2
10µH
1A

D2

MBRS140

N1B
IRF7103

R2

R3

909k

1%

V

OUT2

= 24V, 35mA

C5
22µF
35V

0.1Ω

C6
15pF

GND

FB2

SHDN1

SHDN2

REF

ON/OFF

C7

0.1µF

49.9k
1%

R4

C8
270pF

LOW-BATTERY

LBO

DETECTOR OUTPUT

LBI

R6

PGNDFB1

MAX863

Dual, High-Efficiency, PFM, Step-Up

DC-DC Controller

______________________________________________________________________________________ 15

Figure 9. 3-Cell to 3.3V Step-Up/Step-Down Logic Supply with 12V for Flash Memory or Analog Functions

__________Applications Information

Low-Input-Voltage Operation

When the voltage at VDDfalls and EXT1 or EXT2
approaches the MOSFET gate-to-source threshold volt­age, the MOSFET may operate in its linear region and
dissipate excessive power. Prolonged operation in this
mode may damage the MOSFET if power dissipation
ratings are inadequate. This effect is more significant in
non-bootstrapped mode, but can occur in boot­strapped mode if the input voltage drops so low that it
cannot support the load and causes the output voltage
to collapse. To avoid this condition, use logic-level or
low-threshold MOSFETs.

Starting Up Under Load

The Typical Operating Characteristics show the
Bootstrapped-Mode Minimum Start-Up Input Voltage
vs. Output Current graph. The MAX863 is not intended
to start up under full load in bootstrapped mode with
low input voltages.

________________Application Circuits

Bootstrapped 5V Logic and

24V LCD Bias Supply

The circuit in Figure 8 operates from two AA or AAA
cells, and generates 5V (up to 750mA) for logic and
24V (up to 35mA) for an LCD bias supply. OUT1 is
used to bootstrap the MAX863 for better MOSFET gate
drive. V

OUT1

can be set to 3.3V if low threshold

MOSFETs are used.

= 2.0V TO 11V OR V

V

IN

FLYBACK OR SEPIC

OUTPUT
= 3.3V, 600mA

V

OUT1

C1

330µF

10V

≤0.1Ω

R7

100k

OUT2

10µH, 2.5A

CTX10-4

D1

MBRS340T3

T1

10µF

50mΩ

C9

N1A

R1

C3

100µF

10V

≤0.1Ω

CMPSH-3C

SENSE1

EXT1

CS1

R5

LBI

R6

C4

100µF

10V

V

DD

SHDN1

SHDN2

≤0.1Ω

C2

1µF

BOOT

EXT2

CS2

FB2

D3

FB1

MAX863

L2
10µH
2A

D2

50mΩ

R2

ON/OFF

MBRS340T3

N1B
IRF7301

R3
1M
1%

R4

115k

1%

V

OUT2

C6
10pF

C8
82pF

= 12V

C5
100µF
20V
≤0.1Ω

LOW-BATTERY
DETECTOR OUTPUT

LBO

PGND

REF

C7

0.1µF

GND

MAX863

Dual, High-Efficiency, PFM, Step-Up
DC-DC Controller

16 ______________________________________________________________________________________

___________________Chip Information

TRANSISTOR COUNT: 858

SUBSTRATE CONNECTED TO GND

Step-Up/Down SEPIC Converter

and 12V Supply

The circuit in Figure 9 provides a buck/boost function for
applications where the input voltage range can be
greater than or less than V

OUT1

. It provides 3.3V (up to
600mA) or 5V, as well as 12V (up to 200mA at VIN= 2.4V)
for powering flash memory or analog functions.

The main output employs a SEPIC topology using a
coupled inductor and a capacitor to transfer energy to
the output. C2 must be a low-ESR type capable of
withstanding high ripple current. Ceramic and Sanyo
OS-CONs work well, but low-ESR aluminum electrolyt­ics (which are less costly) are tolerable. Do not use a
tantalum capacitor for C2. C2’s voltage rating must be
higher than the maximum input voltage. The MOSFET
must withstand a voltage equal to the sum of the input
and output voltages; i.e., when converting 11V to 3.3V,
the MOSFET must withstand 14.3V. The dual Schottky
diode D3 bootstraps power to the MAX863, allowing
use of the low-voltage start-up oscillator, as well as
improved gate-drive voltages during normal operation.

________________________________________________________Package Information

QSOP.EPS