Rainbow Electronics MAX767 User Manual

________________General Description

The MAX767 is a high-efficiency, synchronous buck
controller IC dedicated to converting a fixed 5V supply
into a tightly regulated 3.3V output. Two key features set
this device apart from similar, low-voltage step-down
switching regulators: high operating frequency and all
N-channel construction in the application circuit. The
300kHz operating frequency results in very small, low­cost external surface-mount components.

The inductor, at 3.3µH for 5A, is physically at least five
times smaller than inductors found in competing solu­tions. All N-channel construction and synchronous rectifi­cation result in reduced cost and highest efficiency.
Efficiency exceeds 90% over a wide range of loading,
eliminating the need for heatsinking. Output capacitance
requirements are low, reducing board space and cost.

The MAX767 is a monolithic BiCMOS IC available in
20-pin SSOP packages. For other fixed output voltages
and package options, please consult the factory.

________________________Applications

Local 5V-to-3.3V DC-DC Conversion

Microprocessor Daughterboards

Power Supplies up to 10A or More

____________________________Features

♦ >90% Efficiency
♦ 700µA Quiescent Supply Current
♦ 120µA Standby Supply Current
♦ 4.5V-to-5.5V Input Range
♦ Low-Cost Application Circuit
♦ All N-Channel Switches
♦ Small External Components
♦ Tiny Shrink-Small-Outline Package (SSOP)
♦ Predesigned Applications:

Standard 5V to 3.3V DC-DC Converters up to 10A
High-Accuracy Pentium P54C VR-Spec Supply

♦ Fixed Output Voltages Available:

3.3V (Standard)

3.45V (High-Speed Pentium™)

3.6V (PowerPC™)

_______________Ordering Information

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

________________________________________________________________ Maxim Integrated Products 1

__________________Pin Configuration

OUTPUT

3.3V

AT 5A

INPUT

4.5V TO 5.5V

LX

DL

GND

MAX767

V

CC

REF

3.3µH

BST

DH

PGND

CS

FB

ON

_________Typical Application Circuit

19-0224; Rev 3; 7/00

PART TEMP. RANGE

PIN­PACKAGE

MAX767CAP 0°C to +70°C 20 SSOP

MAX767RCAP 0°C to +70°C 20 SSOP

MAX767SCAP 0°C to +70°C 20 SSOP

Pentium is a trademark of Intel. PowerPC is a trademark of IBM.

MAX767C/D 0°C to +70°C Dice*

REF
TOL

±1.8%

±1.8%

±1.8%

—

Ordering Information continued at end of data sheet.

* Contact factory for dice specifications.

V

OUT

(V)

3.3

3.45

3.6

—

MAX767TCAP 0°C to +70°C 20 SSOP ±1.2% 3.3

EVALUATION KIT MANUAL

FOLLOWS DATA SHEET

For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.

TOP VIEW

CS

1

SS

2

ON

GND

GND

GND

GND

REF

SYNC

V

3

MAX767

4

5

6

7

8

9

CC

10

SSOP

FB

20

DH

19

LX

18

BST

17

DL

16

V

15

CC

V

14

CC

PGND

13

N.C.

12

GND

11

MAX767

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

VCCto GND.................................................................-0.3V, +7V

PGND to GND ........................................................................±2V

BST to GND ...............................................................-0.3V, +15V

LX to BST.....................................................................-7V, +0.3V

Inputs/Outputs to GND

(ON, REF, SYNC, CS, FB, SS) .....................-0.3V, V

CC

+ 0.3V

DL to PGND .....................................................-0.3V, V

CC

+ 0.3V

DH to LX...........................................................-0.3V, BST + 0.3V

REF Short to GND.......................................................Momentary

REF Current.........................................................................20mA

Continuous Power Dissipation (T

A

= +70°C)

20-Pin SSOP (derate 8.00mW/°C above +70°C) ..........640mW

Operating Temperature Ranges:

MAX767CAP/MAX767_CAP.................................0°C to +70°C

MAX767EAP/MAX767_EAP ..............................-40°C to +85°C

Lead Temperature (soldering, 10s) .................................+300°C

PARAMETER

Oscillator Frequency

DH Sink/Source Current

MIN TYP MAX

1

260 300 340

UNITS

SS Source Current

A

2.50 4 6.5

(BST - LX) = 4.5V, DH = 2V

DL On Resistance

µA

Current-Limit Voltage

200

7

80 100 120 mV

VCCFault Lockout Voltage

Ω

3.80 4.20

kHz

V

Line Regulation

High or low

0.1 %

Oscillator SYNC Range

SS Fault Sink Current

DH On Resistance

2 mA

7

3.24 3.30 3.36

Ω

240 350 kHz

SYNC High Pulse Width

VCCStandby Current

High or low, (BST - LX) = 4.5V

120 200

200

µA

VCCQuiescent Current

ns

0.7 1.0 mA

SYNC Low Pulse Width 200 ns

SYNC Rise/Fall Time

Output Voltage (FB)

200 ns

Oscillator Maximum Duty Cycle

VCCInput Supply Range

89 92

4.5 5.5 V

95

%

Input Low Voltage

3.17 3.35 3.46

V

0.8 V

Input High Voltage

2.40

3.32 3.50 3.60

V

CC

- 0.5

V

Input Current ±1 µA

DL Sink/Source Current 1 A

CONDITIONS

SYNC = 3.3V

SYNC = 0V or 5V

CS - FB

Falling edge, hysteresis = 1%

VCC= 4.5V to 5.5V

MAX767, MAX767R, MAX767S

ON = 0V, VCC= 5.5V

FB = CS = 3.5V

Not tested

SYNC = 3.3V

SYNC = 0V

SYNC, ON

ON

0mV < (CS - FB) < 80mV,

4.5V < V

CC

< 5.5V
(includes load and
line regulation)

SYNC

SYNC, ON = 0V or 5V

DL = 2V

Load Regulation 2.5 %(CS - FB) = 0mV to 80mV

3.46 3.65 3.75

MAX767R

MAX767S

MAX767, MAX767T

ELECTRICAL CHARACTERISTICS

(VCC= ON = 5V, GND = PGND = SYNC = 0V, I

REF

= 0mA, TA= T

MIN

to T

MAX

, unless otherwise noted. Typical values are at TA= +25°C.)

Reference Voltage (REF)

3.26 3.30 3.34

V

MAX767T

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 3

__________________________________________Typical Operating Characteristics

(Circuit of Figure 1 (5A configuration), VIN= 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

EFFICIENCY vs. OUTPUT CURRENT

(1.5A CIRCUIT)

100

90

80

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

EFFICIENCY vs. OUTPUT CURRENT

(7A CIRCUIT)

100

90

80

MAX767-01

EFFICIENCY (%)

MAX767-04

EFFICIENCY vs. OUTPUT CURRENT

(3A CIRCUIT)

100

90

80

70

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

EFFICIENCY vs. OUTPUT CURRENT

(10A CIRCUIT)

100

90

80

100

MAX767-02

EFFICIENCY (%)

1000

MAX767-05

100

EFFICIENCY vs. OUTPUT CURRENT

(5A CIRCUIT)

90

80

70

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

SWITCHING FREQUENCY vs.

PERCENT OF FULL LOAD

SYNC = REF (300kHz)

10

MAX767-03

MAX767-06

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

IDLE-MODE WAVEFORMS

I

= 300mA

LOAD

5µs/div

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

3.3V OUTPUT
50mV/div, AC COUPLED

LX
5V/div

I

LOAD

1

0.1

SWITCHING FREQUENCY (kHz)

0.01

0.001 1 100

PWM-MODE WAVEFORMS

= 5A

1µs/div

0.01 0.1 10

LOAD CURRENT (% FULL LOAD)

3.3V OUTPUT
50mV/div, AC COUPLED

LX
5V/div

MAX767

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

4 _______________________________________________________________________________________

_____________________________Typical Operating Characteristics (continued)

(Circuit of Figure 1 (5A configuration), VIN= 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

1.5A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

1.5A

3.3V OUTPUT
50mV/div
AC-COUPLED

0A

LOAD CURRENT

3A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

3A

3.3V OUTPUT
50mV/div
AC-COUPLED

0A

LOAD CURRENT

5A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

5A

3.3V OUTPUT
50mV/div
AC-COUPLED

0A

LOAD CURRENT

10A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

10A

3.3V OUTPUT
50mV/div
AC-COUPLED

0A

LOAD CURRENT

7A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

7A

3.3V OUTPUT
50mV/div
AC-COUPLED

0A

LOAD CURRENT

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 5

______________________________________________________________Pin Description

16

NAME FUNCTION

1

CS Current-sense input: +100mV = nominal current-limit level referred to FB.

2 SS Soft-start input. Ramp time to full current limit is 1ms/nF of capacitance to GND.

3 ON

ON/O—F—F–control input to disable the PWM. Tie directly to VCCfor automatic start-up.

PIN

DL Gate-drive output for the low-side synchronous rectifier MOSFET

4–7, 11 GND Low-current analog ground. Feedback reference point for the output.

8 REF 3.3V internal reference output. Bypass to GND with 0.22µF minimum capacitor.

9 SYNC

Oscillator control/synchronization input. Connect to VCCor GND for 200kHz; connect to REF for
300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition
causes a new cycle to start.

10, 14, 15 V

CC

Supply voltage input: 4.5V to 5.5V

17 BST Boost capacitor connection (0.1µF)

12 N.C. No internal connection

13 PGND Power ground

18 LX Inductor connection. Can swing 2V below GND without latchup.

19 DH Gate-drive output for the high-side MOSFET

20 FB Feedback and current-sense input for the PWM

Figure 1. Standard Application Circuit

R2

INPUT

4.5V TO 5.5V

SHUTDOWN

ON/OFF

C5

(OPTIONAL)

C6

0.22µF

0.01µF

4.7µF

C4

ON

SS

SYNC

REF

V

CC

MAX767

GND

10

BST

PGND

Ω

DH

LX

DL

CS

FB

D1
SMALL­SIGNAL
SCHOTTKY

C3

N1

0.1µF

D2

N2

C1

OUTPUT

L1

R1

3.3V

C2

MAX767

_____Standard Application Circuits

This data sheet shows five predesigned circuits with
output current capabilities from 1.5A to 10A. Many
users will find one of these standard circuits appropri­ate for their needs. If a standard circuit is used, the
remainder of this data sheet (Detailed Description and
Applications Information and Design Procedure) can
be bypassed.

Figure 1 shows the Standard Application Circuit. Table 1
gives component values and part numbers for five dif­ferent implementations of this circuit: 1.5A, 3A, 5A, 7A,
and 10A output currents.

Each of these circuits is designed to deliver the full
rated output load current over the temperature range
listed. In addition, each will withstand a short circuit of
several seconds duration from the output to ground. If
the circuit must withstand a continuous short circuit,
refer to the Short-Circuit Duration section for the
required changes.

Layout and Grounding

Good layout is necessary to achieve the designed out­put power, high efficiency, and low noise. Good layout
includes the use of a ground plane, appropriate com­ponent placement, and correct routing of traces using
appropriate trace widths. The following points are in
order of decreasing importance.

1. A ground plane is essential for optimum perfor­mance. In most applications, the circuit will be
located on a multilayer board and full use of the four
or more copper layers is recommended. Use the
top and bottom layers for interconnections and the
inner layers for an uninterrupted ground plane.

2. Because the sense resistance values are similar to
a few centimeters of narrow traces on a printed cir­cuit board, trace resistance can contribute signifi­cant errors. To prevent this, Kelvin connect CS and
FB to the sense resistor; i.e., use separate traces
not carrying any of the inductor or load current, as
shown in Figure 2. These signals must be carefully
shielded from DH, DL, BST, and the LX node.

Important: place the sense resistor as close as pos­sible to and no further than 10mm from the MAX767.

3. Place the LX node components N1, N2, L1, and D2
as close together as possible. This reduces resis­tive and switching losses and confines noise due to
ground inductance.

4. The input filter capacitor C1 should be less than
10mm away from N1’s drain. The connecting cop­per trace carries large currents and must be at least
2mm wide, preferably 5mm.

5. Keep the gate connections to the MOSFETs short
for low inductance (less than 20mm long and more
than 0.5mm wide) to ensure clean switching.

6. To achieve good shielding, it is best to keep all
switching signals (MOSFET gate drives DH and DL,
BST, and the LX node) on one side of the board
and all sensitive nodes (CS, FB, and REF) on the
other side.

7. Connect the GND and PGND pins directly to the
ground plane, which should ideally be an inner
layer of a multilayer board.

_______________Detailed Description

Note: The remainder of this document contains the
detailed information necessary to design a circuit that
differs substantially from the five standard application
circuits. If you are using one of the predesigned stan­dard circuits, the following sections are provided only
for your reading pleasure.

The MAX767 converts a 4.5V to 5.5V input to a 3.3V
output. Its load capability depends on external compo­nents and can exceed 10A. The 3.3V output is generat­ed by a current-mode, pulse-width-modulation (PWM)
step-down regulator. The PWM regulator operates at
either 200kHz or 300kHz, with a corresponding trade­off between somewhat higher efficiency (200kHz) and
smaller external component size (300kHz). The
MAX767 also has a 3.3V, 5mA reference voltage. Fault­protection circuitry shuts off the output should the refer­ence lose regulation or the input voltage go below 4V
(nominally).

External components for the MAX767 include two N­channel MOSFETs, a rectifier, and an LC output filter.
The gate-drive signal for the high-side MOSFET, which
must exceed the input voltage, is provided by a boost
circuit that uses a 0.1µF capacitor. The synchronous
rectifier keeps efficiency high by clamping the voltage
across the rectifier diode. An external low-value cur­rent-sense resistor sets the maximum current limit, pre­venting excessive inductor current during start-up or
under short-circuit conditions. An optional external
capacitor sets the programmable soft-start, reducing
in-rush surge currents upon start-up and providing
adjustable power-up time.

The PWM regulator is a direct-summing type, lacking a
traditional integrator-type error amplifier and the phase
shift associated with it. It therefore does not require
external feedback-compensation components, as long
as you follow the ESR guidelines in the Applications
Information and Design Procedure sections.

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

6 _______________________________________________________________________________________

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 7

Part 1.5A Circuit 3A Circuit 5A Circuit 7A Circuit 10A Circuit

L1

10µH
Sumida CDR74B-100

5µH
Sumida CDR125
DRG# 4722-JPS-001

3.3µH
Coilcraft
DO3316-332

2.1µH, 5mΩ
Coiltronics
CTX03-12338-1

1.5µH, 3.5mΩ
Coiltronics
CTX03-12357-1

R1

0.04Ω
IRC LR2010-01-R040
or DD WSL-2512-R040

0.02Ω
IRC LR2010-01-R020
or DD WSL-2512-R020

0.012Ω
DD WSL-2512-R012
or 2 x 0.025Ω
IRC LR2010-01-R025
(in parallel)

3 x 0.025Ω
IRC LR2010-01-R025
or DD WSL-2512-R025
(in parallel)

3 x 0.020Ω
IRC LR2010-01-R020
or 2 x 0.012Ω
DD WSL-2512-R012
(in parallel)

N1,

N2

International
Rectifier IRF7101,
Siliconix Si9936DY
or Motorola
MMDF3N03HD
(dual N-channel)

Siliconix Si9410DY,
International
Rectifier IRF7101
or Motorola
MMDF3N03HD
(both FETs in parallel)

Motorola
MTD20N03HDL

Motorola
MTD75N03HDL
(N1)
MTD20N03HDL
(N2)

Motorola
MTD75N03HDL

C1

47µF, 20V
AVX TPSD476K020R

2 x 47µF, 20V
AVX TPSD476K020R

220µF, 10V
Sanyo
OS-CON 10SA220M

2 x 100µF, 10V
Sanyo
OS-CON 10SA100M

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

C2

220µF, 6.3V
Sprague
595D227X06R3D2B

2 x 150µF, 10V
Sprague
595D157X0010D7T

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

4 x 220µF, 10V
Sanyo
OS-CON 10SA220M

D2

1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3

1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

Temp.
Range

to +85°C to +85°C to +85°C to +85°C to +85°C

Table 1. Component Values

Table 2. Component Suppliers

Company Factory Fax [Country Code] USA Telephone

AVX [1] [phone]

[phone]
[phone]

Coilcraft [1] [phone] [phone]

Coiltronics [1] [phone] [phone]

DD [1] [phone] [phone]

IRC [1] [phone] [phone]

International
Rectifier

[1] [phone] [phone]

Nihon [81] 3-3494-7414 [phone]

Sanyo [81] 7-2070-1174 [phone]

Siliconix [1] [phone] [phone]

Sprague [1] [phone] [phone]

Sumida [81] 3-3607-5144 [phone]

Motorola [1] [phone] [phone]

MAX767

The main gain block is an open-loop comparator that
sums four signals: output voltage error signal, current­sense signal, slope-compensation ramp, and the 3.3V
reference. This direct-summing method approaches
the ideal of cycle-by-cycle control of the output voltage.
Under heavy loads, the controller operates in full PWM
mode. Every pulse from the oscillator sets the output
latch and turns on the high-side switch for a period
determined by the duty factor (approximately
V

OUT

/ VIN).

As the high-side switch turns off, the synchronous recti­fier latch is set; 60ns later, the low-side switch turns on.
The low-side switch stays on until the beginning of the
next clock cycle (in continuous-conduction mode) or
until the inductor current reaches zero (in discontinu­ous-conduction mode). Under fault conditions where
the inductor current exceeds the 100mV current-limit
threshold, the high-side latch resets and the high-side
switch turns off.

At light loads, the inductor current fails to exceed the
25mV threshold set by the minimum-current compara­tor. When this occurs, the PWM goes into Idle-Mode™,
skipping most of the oscillator pulses to reduce the
switching frequency and cut back switching losses.
The oscillator is effectively gated off at light loads

because the minimum-current comparator immediately
resets the high-side latch at the beginning of each
cycle, unless the FB signal falls below the reference
voltage level.

Soft-Start

Connecting a capacitor from the soft-start pin (SS) to
ground allows a gradual build-up of the 3.3V output
after power is applied or ON is driven high. When ON is
low, the soft-start capacitor is discharged to GND.
When ON is driven high, a 4µA constant current source
charges the capacitor up to 4V. The resulting ramp volt­age on SS linearly increases the current-limit compara­tor set-point, increasing the duty cycle to the external
power MOSFETs. With no soft-start capacitor, the full
output current is available within 10µs (see Applications
Information and Design Procedure section).

Synchronous Rectifier

Synchronous rectification allows for high efficiency by
reducing the losses associated with the Schottky rectifi­er. Also, the synchronous-rectifier MOSFET is neces­sary for correct operation of the MAX767’s boost gate­drive supply.

When the external power MOSFET (N1) turns off, ener­gy stored in the inductor causes its terminal voltage to
reverse instantly. Current flows in the loop formed by
the inductor (L1), Schottky diode (D2), and the load—
an action that charges up the output filter capacitor
(C2). The Schottky diode has a forward voltage of
about 0.5V which, although small, represents a signifi­cant power loss and degrades efficiency. The synchro­nous-rectifier MOSFET parallels the diode and is turned
on by DL shortly after the diode conducts. Since the
synchronous rectifier’s on resistance (r

DS(ON)

) is very
low, the losses are reduced. The synchronous-rectifier
MOSFET is turned off when the inductor current falls to
zero.

The MAX767’s internal break-before-make timing
ensures that shoot-through (both external switches
turned on at the same time) does not occur. The
Schottky rectifier conducts during the time that neither
MOSFET is on, which improves efficiency by preventing
the synchronous-rectifier MOSFET’s lossy body diode
from conducting.

The synchronous rectifier works under all operating
conditions, including discontinuous-conduction mode
and idle-mode.

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

8 _______________________________________________________________________________________

Figure 2. Kelvin Connections for the Current-Sense Resistor

Idle-Mode is a trademark of Maxim Integrated Products.

FAT, HIGH-CURRENT TRACES

MAIN CURRENT PATH

SENSE RESISTOR

MAX767

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

_______________________________________________________________________________________ 9

Figure 3. MAX767 Block Diagram

2.8V

CS

1X

MAIN PWM
COMPARATOR

MINIMUM
CURRENT

(IDLE-MODE)

3.3V

N

60kHz

LPF

30R

R

Q

S

V

CC

CURRENT

LIMIT

0mV TO 100mV

200kHz/300kHz

OSCILLATOR

1R

ON

LEVEL
SHIFT

SHOOT­THROUGH
CONTROL

V

REF

CC

+3.3V

REFERENCE

SLOPE COMP

4V

25mV

4µA

FAULT

SS

ON

FB

BST

DH

LX

MAX767

SYNCHRONOUS

RECTIFIER CONTROL

V

R

Q

S

SYNC

LEVEL
SHIFT

CC

DL

PGND

MAX767

Gate-Driver Boost Supply

Gate-drive voltage for the high-side N-channel switch is
generated with the flying-capacitor boost circuit shown
in Figure 4. The capacitor (C3) is alternately charged
from the 5V input via the diode (D1) and placed in par­allel with the high-side MOSFET’s gate-source termi­nals. On start-up, the synchronous rectifier (low-side)
MOSFET (N2) forces LX to 0V and charges the BST
capacitor to 5V. On the second half-cycle, the PWM
turns on the high-side MOSFET (N1); it does this by
closing an internal switch between BST and DH, which
connects the capacitor to the MOSFET gate. This pro­vides the necessary enhancement voltage to turn on
the high-side switch, an action that “boosts” the 5V
gate-drive signal above the input voltage.

Ringing seen at the high-side MOSFET gates (DH) in
discontinuous-conduction mode (light loads) is a natur­al operating condition. It is caused by the residual
energy in the tank circuit, formed by the inductor and
stray capacitance at the LX node. The gate-driver neg­ative rail is referred to LX, so any ringing there is direct­ly coupled to the gate-drive supply.

Modes of Operation

PWM Mode

Under heavy loads—over approximately 25% of full
load—the supply operates as a continuous-current
PWM supply (see Typical Operating Characteristics).
The duty cycle, %ON, is approximately:

V

OUT

%ON = ________

V

IN

Current flows continuously in the inductor: first, it ramps
up when the power MOSFET conducts; second, it
ramps down during the flyback portion of each cycle as
energy is put into the inductor and then discharged into
the load. Note that the current flowing into the inductor
when it is being charged is also flowing into the load,
so the load is continuously receiving current from the
inductor. This minimizes output ripple and maximizes
inductor use, allowing very small physical and electrical
sizes. Output ripple is primarily a function of the filter
capacitor’s effective series resistance (ESR), and is
typically under 50mV (see Design Procedure section).

Idle-Mode

Under light loads (<25% of full load), the MAX767
enhances efficiency by turning the drive voltage on and
off for only a single clock period, skipping most of the
clock pulses entirely. Asynchronous switching, seen as
“ghosting” on an oscilloscope, is thus a normal operat­ing condition whenever the load current is less than
approximately 25% of full load.

At certain input voltage and load conditions, a transition
region exists where the controller can pass back and
forth from idle-mode to PWM mode. In this situation,
short pulse bursts occur, which make the current wave­form look erratic but do not materially affect the output
ripple. Efficiency remains high.

Current Limiting

The voltage between CS and FB is continuously moni­tored. An external, low-value shunt resistor is connect­ed between these pins, in series with the inductor,
allowing the inductor current to be continuously mea­sured throughout the switching cycle. Whenever this
voltage exceeds 100mV, the drive voltage to the exter­nal high-side MOSFET is cut off. This protects the MOS­FET, the load, and the input supply in case of short cir­cuits or temporary load surges. The current-limiting
resistance is typically 20mΩ for 3A.

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

10 ______________________________________________________________________________________

Figure 4. Boost Supply for High-Side Gate Driver

V

MAX767

V

CC

LEVEL

TRANSLATOR

PWM

BST

DH

LX

V

CC

DL

D1

C3

N1

L1

N2

IN

C1

Oscillator Frequency

The SYNC input controls the oscillator frequency.
Connecting SYNC to GND or to VCCselects 200kHz
operation; connecting it to REF selects 300kHz opera­tion. SYNC can also be driven with an external 240kHz
to 350kHz CMOS/TTL source to synchronize the inter­nal oscillator. Normally, 300kHz operation is chosen to
minimize the inductor and output filter capacitor sizes,
but 200kHz operation may be chosen for a small (about
1%) increase in efficiency at heavy loads.

Internal Reference

The internal 3.3V bandgap reference (REF) remains
active, even when the switching regulator is turned off.
It can furnish up to 5mA, and can be used to supply
memory keep-alive power or for other purposes.
Bypass REF to GND with 0.22µF, plus 1µF/mA of load
current.

Applications Information and

__________________Design Procedure

Most users will be able to work with one of the standard
application circuits; others may want to implement a
circuit with an output current rating that lies between or
beyond the standard values.

If you want an output current level that lies between two
of the standard application circuits, you can interpolate
many of the component values from the values given
for the two circuits. These components include the
input and output filter capacitors, the inductor, and the
sense resistor. The capacitors must meet ESR and rip­ple current requirements (see Input Filter Capacitor and
Output Filter Capacitor sections). The inductor must
meet the required current rating (see Inductor section).

You may use the rectifier and MOSFETs specified for
the circuit with the greater output current capability, or
choose a new rectifier and MOSFETs according to the
requirements detailed in the Rectifier and MOSFET
Switches sections. For more complete information, or
for output currents in excess of 10A, refer to the design
information in the following sections.

Inductor, L1

Three inductor parameters are required: the inductance
value (L), the peak inductor current (I

LPEAK

), and the

coil resistance (RL). The inductance is:

1.32

L1 = ______________

f x I

OUT

x LIR

where:

f = switching frequency, normally 300kHz

I

OUT

= maximum 3.3V DC load current (A)

LIR = ratio of inductor peak-to-peak AC

current to average DC load current,
typically 0.3.

A higher LIR value allows smaller inductance, but
results in higher losses and ripple.

The highest peak inductor current (I

LPEAK

) equals the

DC load current (I

OUT

) plus half the peak-to-peak AC

inductor current (I

LPP

). The peak-to-peak AC inductor
current is typically chosen as 30% of the maximum DC
load current, so the peak inductor current is 1.15 x I

OUT

.

The peak inductor current at any load is given by:

1.32

I

LPEAK

= I

OUT

+ __________

2 x f x L1

The coil resistance should be as low as possible,
preferably in the low milliohms. The coil is effectively in
series with the load at all times, so the wire losses alone
are approximately:

Power Loss = I

OUT

2

x R

L

In general, select a standard inductor that meets the L,
I

LPEAK

, and RLrequirements. If a standard inductor is
unavailable, choose a core with an LI2parameter
greater than L x I

LPEAK

2

, and use the largest wire that

will fit the core.

Current-Sense Resistor, R1

The current-sense resistor must carry the peak current
in the inductor, which exceeds the full DC load current.
The internal current limiting starts when the voltage
across the sense resistors exceeds 100mV nominally,
80mV minimum. Use the minimum value to ensure ade­quate output current capability: R1 = 80mV / I

LPEAK

.

The low VIN/V

OUT

ratio creates a potential problem with
start-up under full load or with load transients from no­load to full load. If the supply is subjected to these con­ditions, reduce the sense resistor:

70mV

R1 = ———

I

LPEAK

Since the sense-resistance values are similar to a few
centimeters of narrow traces on a printed circuit board,
trace resistance can contribute significant errors. To
prevent this, Kelvin connect the CS and FB pins to the
sense resistors; i.e., use separate traces not carrying
any of the inductor or load current, as shown in Figure 2.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 11

MAX767

Place R1 as close as possible to the MAX767, prefer­ably less than 10mm. Run the traces at minimum spac­ing from one another. If they are longer than 20mm,
bypass CS to FB with a 1nF capacitor placed as close
as possible to these pins. The wiring layout for these
traces is critical for stable, low-ripple outputs (see
Layout and Grounding section).

Input Filter Capacitor, C1

Use at least 6µF per watt of output power for C1. If the
5V input is some distance away or comes through a PC
bus, greater capacitance may be desirable to improve
the load-transient response. Use a low-ESR capacitor
located no further than 10mm from the MOSFET switch
(N1) to prevent ringing. The ripple current rating must
be at least I

RMS

= 0.5 x I

OUT

. For high-current applica­tions, two or more capacitors in parallel may be needed
to meet these requirements.

The ESR of C1 is effectively in series with the input. The
resistive dissipation of C1, I

RMS

2

x ESRC1, can signifi-

cantly impact the circuit’s efficiency.

Output Filter Capacitor, C2

The output filter capacitor determines the loop stability,
output voltage ripple, and output load-transient
response.

Stability

To ensure stability, stay above the minimum capaci­tance value and below the maximum ESR value. These
values are:

3Ω

C2 > —— µF

R1

and

ESR

C2

< R1

Be sure to satisfy both these requirements. To achieve
the low ESR required, it may be appropriate to parallel
two or more capacitors and/or use a total capacitance
2 or 3 times larger than the calculated minimum.

Output Ripple

The output ripple in continuous-conduction mode is:

V

OUT(RPL)

= I

OUT

(max) x LIR x

1

(ESR

C2

+ ———————)

2 x π x f x C2

where f is the switching frequency (200kHz or 300kHz).

In idle-mode, the ripple has a capacitive and a resistive
component:

.

0.0004 x L

V

OUT(RPL)

(C) = _____________ x 0.89 Volts

R12x C2

0.02 x ESR

C2

V

OUT(RPL)

(R) = _____________

R1

The total ripple, V

OUT(RPL)

, can be approximated as

follows:

if

V

OUT(RPL)

(R) < 0.5 V

OUT(RPL)

(C)

then

V

OUT(RPL)

= V

OUT(RPL)

(C)

otherwise

V

OUT(RPL)

= 0.5 V

OUT(RPL)

(C) +

V

OUT(RPL)

(R)

Load-Transient Performance

In response to a large step increase in load current, the
output voltage will sag for several microseconds unless
C2 is increased beyond the values that satisfy the
above requirements. Note that an increase in capaci­tance is all that’s required to improve the transient
response, and that the ESR requirements don’t change.
Therefore, the added capacitance can be supplied by
an additional low-cost bulk capacitor in parallel with the
normal low-ESR switching-regulator capacitor. The
equation for voltage sag under a step load change is:

I

STEP

2

x L

V

SAG

= ________________________________

2 x C2 x (VIN(min) x DMAX - 3.3V)

where DMAX is the maximum duty cycle. Higher duty
cycles are possible when the oscillator frequency is
reduced to 200kHz, since fixed propagation delays
through the PWM comparator become a lesser part of
the whole period. The tested worst-case limit for DMAX
is 92% at 200kHz or 89% at 300kHz. Lower inductance
values can reduce the filter capacitance requirement,
but only at the expense of increased output ripple (due
to higher peak currents).

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

12 ______________________________________________________________________________________

RC Filter for V

CC

R2 and C4 form a lowpass filter to remove switching
noise from the V

CC

input to the MAX767. C4 must have
fairly low ESR (<5Ω). Switching noise can interfere with
proper output voltage regulation, resulting in an exces­sive output voltage decrease (>100mV) at full load.

Overheating during soldering can damage the surface­mount capacitors specified for C4, causing the regula­tion problems described above. Take care to heat the
capacitor for as short a time as possible, especially if it
is soldered by hand.

Rectifier, D2

Use a 1N5817 or similar Schottky diode for applications
up to 3A, or a 1N5820 for up to 10A. Surface-mount
equivalents are available from N.I.E.C. with part num­bers EC10QS02 and NSQ03A02, or from Motorola with
part numbers MBRS120T3 and MBRS320T3. D2 must
be a Schottky diode to prevent the lossy MOSFET body
diode from turning on.

Soft-Start

A capacitor connected from GND to SS causes the
supply’s current-limit level to ramp up slowly. The ramp
time to full current limit is approximately 1ms for every
nF of capacitance on SS, with a minimum value of
10µs. Typical values for the soft-start capacitor are in
the 10nF to 100nF range; a 5V rating is sufficient.

The time required for the output voltage to ramp up to
its rated value depends upon the output load, and is
not necessarily the same as the time it takes for the cur­rent limit to reach full capacity.

Duty Cycle

The duty cycle for the high-side MOSFET (N1) in con­tinuous-conduction mode is:

100% x ( V

OUT

+ VN2)

___________________

V

IN

- V

N1

where:

V

OUT

= 3.3V

VIN= 5V

VN1and VN2= I

LOAD

x r

DS(ON)

for each MOSFET.

It is apparent that, in continuous-conduction mode, N1
will conduct for about twice the time as N2. Under short­circuit conditions, however, N2 can conduct as much
90% of the time. If there is a significant chance of short
circuiting the output, select N2 to handle the resulting
duty cycle (see Short-Circuit Duration section).

MOSFET Switches, N1 and N2

The two N-channel MOSFETs must be “logic-level”
FETs; that is, they must be fully on (have low
r

DS(ON)

) with only 4V gate-source drive voltage. For
high-current applications, FETs with low gate­threshold voltage specifications (i.e., maximum
V

GS(TH)

= 2V rather than 3V) are preferred. In addi­tion, they should have low total gate charge (<70nC)
to minimize switching losses.

For output currents in excess of the five standard appli­cation circuits, placing MOSFETs with very low gate
charge in parallel increases output current and lowers
resistive losses. N2 does not normally require the same
current capacity as N1 because it conducts only about
33% of the time, while N1 conducts about 66% of the
time.

Short-Circuit Duration

At their highest rated temperatures (+70°C or +85°C),
each of the five standard application circuits will with­stand a short circuit of several seconds duration. In
most cases, the MAX767 will be used in applications
where long-term short circuiting of the output is unlikely.

If it is desirable for the circuit to withstand a continuous
short circuit, the MOSFETs must be able to dissipate
the required power. This depends on physical factors
such as the mounting of the transistor, any heat­sinking used, and ventilation provided, as well as the
actual current the transistor must deliver. The short­circuit current is approximately 100mV / R1, but may
vary by ±20%.

Cautious design requires that the transistors
withstand the maximum possible current, which is
ISC= 120mV / R1. N1 and N2 must withstand this
current scaled by their maximum duty factors. The
maximum duty factor for N1 occurs under high­load (but not short-circuit) conditions, and is approxi­mately V

OUT

/ VIN(min) or about 0.7. The max­imum duty factor for N2 occurs during short-circuit
conditions and is:

ISCx r

DS(ON)N2

1 - —————————————

VIN(max) - ISCx r

DS(ON)N1

which can exceed 0.9. The total power dissipated in
both MOSFETs together is I

SC

2

x r

DS(ON)

.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 13

MAX767

Proper circuit operation requires that the short-circuit
current be at least I

LOAD

x (1 + LIR / 2). However, the
standard application circuits are designed for a short­circuit current slightly in excess of this amount. This
excess design current guarantees proper start-up
under constant full-load conditions and proper full-load
transient response, and is particularly necessary with
low input voltages. If the circuit will not be subjected to
full-load transients or to loads approaching the full-load
at start-up, you can decrease the short-circuit current
by increasing R1, as described in the Current-Sense
Resistor section. This may allow use of MOSFETs with a
lower current-handling capability.

Heavy-Load Efficiency

Losses due to parasitic resistances in the switches,
coil, and sense resistor dominate at high load-current
levels. Under heavy loads, the MAX767 operates deep
in the continuous-conduction mode, where there is a
large DC offset to the inductor current (plus a small
sawtooth AC component) (see Inductor section). This
DC current is exactly equal to the load current, a fact
which makes it easy to estimate resistive losses via the
simplifying assumption that the total inductor current is
equal to this DC offset current. The major loss mecha­nisms under heavy loads, in usual order of importance,
are:

•I2R losses

• gate-charge losses

• diode-conduction losses

• transition losses

• capacitor-ESR losses

• losses due to the operating supply current of the IC.

Inductor-core losses, which are fairly low at heavy
loads because the AC component of the inductor cur­rent is small, are not accounted for in this analysis.

P

OUT

Efficiency = ______ x 100% =

P

IN

POUT

_______________ x 100%
P

OUT

+ PD

TOTAL

PD

TOTAL

= PD

(I2R)

+ PD

GATE

+ PD

DIODE

+

PD

TRAN

+ PD

CAP

+ PD

IC

I2R Losses

PD

(I2R)

= resistive loss = (I

LOAD

2

) x

(R

COIL

+ r

DS(ON)

+ R1)

where R

COIL

is the DC resistance of the coil and

r

DS(ON)

is the drain-source on resistance of the MOS-

FET. Note that the r

DS(ON)

term assumes that identical
MOSFETs are employed for both the synchronous recti­fier and high-side switch, because they time-share the
inductor current. If the MOSFETs are not identical, esti­mate losses by averaging the two individual r

DS(ON)

terms according to their duty factors: 0.66 for N1 and

0.34 for N2.

Gate-Charge Losses

PD

GATE

= gate driver loss = qGx f x 5V

where qGis the sum of the gate charge for low- and
high-side switches. Note that gate-charge losses are
dissipated in the IC, not the MOSFETs, and therefore
contribute to package temperature rise. For a pair of
matched MOSFETs, qGis simply twice the gate capaci­tance of a single MOSFET (a data sheet specification).

Diode Conduction Losses

PD

DIODE

= diode conduction losses =

I

LOAD

x VDx tDx f

where VDis the forward voltage of the Schottky diode
at the output current, tDis the diode’s conduction time
(typically 110ns), and f is the switching frequency.

Transition Losses

PD

TRAN

= transition loss =

V

IN

2

x C

RSS

x I

LOAD

x f

______________________

I

DRIVE

where C

RSS

is the reverse transfer capacitance of the
high-side MOSFET (a data sheet parameter), f is the
switching frequency, and I

DRIVE

is the peak current
available from the high-side gate driver output (approx­imately 1A).

Additional switching losses are introduced by other
sources of stray capacitance at the switching node,
including the catch-diode capacitance, coil interwind­ing capacitance, and low-side switch drain capaci­tance, and are given as PDSW= V

IN

2

x C

STRAY

x f, but

these are usually negligible compared to C

RSS

losses.
The low-side switch introduces only tiny switching loss­es, since its drain-source voltage is already low when it
turns on.

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

14 ______________________________________________________________________________________

Capacitor ESR Losses

PD

CAP

= capacitor ESR loss = I

RMS

2

x ESR

where I

RMS

= RMS AC input current, approximately

I

LOAD

/ 2.

Note that losses in the output filter capacitors are small
when the circuit is heavily loaded, because the current
into the capacitor is not chopped. The output capacitor
sees only the small AC sawtooth ripple current. Ensure
that the input bypass capacitor has a ripple current rat­ing that exceeds the value of I

RMS

.

IC Supply-Current Losses

PDICis the quiescent power dissipation of the IC and is
5V times the quiescent supply current (a data sheet
parameter), or about 5mW.

Light-Load Efficiency

Under light loads, the PWM will operate in discontinu­ous-conduction mode, where the inductor current dis­charges to zero at some point during each switching
cycle. New loss mechanisms, insignificant at heavy
loads, begin to become important. The basic difference
is that in discontinuous mode, the AC component of the
inductor current is large compared to the load current.
This increases losses in the core and in the output filter
capacitors. Ferrite cores are recommended over pow­dered-material types for best light-load efficiency.

At light loads, the inductor delivers triangular current
pulses rather than the nearly square waves found in
continuous-conduction mode. These pulses ramp up to
a point set by the idle-mode current comparator, which
is internally fixed at approximately 25% of the full-scale
current-limit level. This 25% threshold provides an opti­mum balance between low-current efficiency and out­put voltage noise (the efficiency curve would actually
look better with this threshold set at about 45%, but the
output noise would be too high).

____Additional Application Circuits

High-Accuracy Power Supplies

The standard application circuit’s accuracy is dominat­ed by reference voltage error (±1.8%) and load regula­tion error (-2.5%). Both of these parameters can be
improved as shown in Figures 5 and 6. Both circuits
rely on an external integrator amplifier to increase the
DC loop gain in order to reduce the load regulation
error to 0.1%. Reference error is improved in the first
circuit by employing a version of the MAX767 (“T”
grade) which has a ±1.2% reference voltage tolerance.

Reference error of the second circuit is further
improved by substituting a highly accurate external ref­erence chip (MAX872), which contributes ±0.38% total
error over temperature.

These two circuits were designed with the latest gener­ation of dynamic-clock µPs in mind, which place great
demands on the transient-response performance of the
power supply. As the µP clock starts and stops, the
load current can change by several amps in less than
100ns. This tremendous ∆i/∆t can cause output voltage
overshoot or sag that results in the CPU VCC going out
of tolerance unless the power supply is carefully
designed and located close to the CPU. These circuits
have excellent dynamic response and low ripple, with
transient excursions of less than 40mV under zero to
full-load step change. In particular, these two circuits
support the “VR” (voltage regulator) version of the Intel
P54C Pentium™ CPU, which requires that its supply
voltage, including noise and transient errors, be within
the 3.30V to 3.45V range.

To configure these circuits for a given load current
requirement, substitute standard components from
Table 1 for the power switching elements (N1, N2, L1,
C1, C2) or use the Design Procedure. R1 can also be
taken from Table 1, but must be adjusted approximate­ly 10% higher in order to maintain the correct current­limit threshold. This increased value is due to the 0.9
gain factor introduced by the H-bridge resistor divider
(R3–R6).

If the remote sense line must sense the output voltage
on the far side of a connector or jumper that has the
possibility of becoming disconnected while the power
supply is operating, an additional 10kΩ resistor should
connect the sense line to the output voltage in the con­nector’s power-supply side in order to prevent acciden­tal overvoltage at the CPU.

For applications that are powered from a fixed +12V or
battery input rather than from +5V, use a MAX797 IC
instead of the MAX767. The MAX797 is capable of
accepting inputs up to 30V. See the MAX796–MAX799
data sheet for a high-accuracy circuit schematic.

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 15

MAX767

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

16 ______________________________________________________________________________________

Figure 5. High-Accuracy CPU Power Supply with Internal Reference

INPUT

4.75V TO 5.5V

R2

Ω

10

C4

4.7µF

SHUTDOWN

ON/OFF

V

CC

ON

C10

0.01µF

(OPTIONAL)

MAX767T

SS

GND

DH

BST

PGND

CS

REF

SYNC

V

0.01µF

V

CC

OUT

C5

C8
620pF

MAX495

D1

N1

C3

0.1µF

LX

DL

FB

N2

R8

10k

C9

0.22µF

R9

332k, 1%

C1

L1

D2

TO MAX767

= V

REF

(

R10

+ 1

R9

R3
1k,
1%

R7

330k

R5
10k,
1%

R10

8.06k, 1%

)

3.38V OUTPUT

R11

5.1k
MIN

LOAD

3.427V MAX

3.330V MIN

C7

10µF CERAMIC

(LOCATE AT

µ

P PINS)

C2R1

R4
1k,
1%

C6

0.01µF

R6
10k,
1%

REMOTE SENSE LINE

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 17

Figure 6. High-Accuracy CPU Power Supply with External Reference

INPUT

4.75V TO 5.5V

R2

Ω

20

C4

22µF

SHUTDOWN

ON/OFF

0.22µF

V

ON

REF

SYNC

C10

0.01µF

(OPTIONAL)

R10

= V

REF

(

R9

R3
1k,
1%

330k

R5
10k,
1%

118k, 0.1%

+ 1

)

3.38V OUTPUT

R11

5.1k
MIN

LOAD

3.408V MAX

3.369V MIN

C7

10µF CERAMIC

(LOCATE AT

µ

P PINS)

C2R1

R4
1k,
1%

R7

R10

C6

0.01µF

R6
10k,
1%

REMOTE SENSE LINE

V

D1

C3

0.1µF

TO MAX767

V

CCC9

R8

10k

R9

332k, 0.1%

N1

N2

DH

GND

VIN

MAX872

GND

BST

LX

DL

PGND

CS

FB

VOUT

CC

MAX767

SS

C1

L1

D2

OUT

C5

0.01µF

C8
1000pF

MAX495

MAX767

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

18 ______________________________________________________________________________________

___________________Chip TopographyOrdering Information (continued)

TRANSISTOR COUNT: 1294

SUBSTRATE CONNECTED TO GND

BST

GND

GND

REF

ON

GND

V

CC

0.181"

(4.597mm)

0.109"

(2.769mm)

GND

GND

GND

SYNC

PGND

V

CC

V

CC

V

CC

DL

LX

SS CS FB DH

3.3±1.2%20 SSOP-40°C to +85°CMAX767TEAP

3.6

3.45

3.3

V

OUT

(V)

±1.8%

±1.8%

±1.8%

REF
TOL

20 SSOP-40°C to +85°CMAX767SEAP

20 SSOP-40°C to +85°CMAX767REAP

20 SSOP-40°C to +85°CMAX767EAP

PIN­PACKAGE

TEMP. RANGEPART

MAX767

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________________________________ 19

________________________________________________________Package Information

DIM

e

HE

A

A1

B
C
D
E
e
H
L

α

D

α

A

0.127mm

0.004in.

A1

B

C

L

INCHES MILLIMETERS

MIN

0.068

0.002

0.010

0.005

0.278

0.205

0.301

0.022

MAX

0.078

0.008

0.015

0.009

0.289

0.212

0.311

0.037

0˚

MIN

1.73

0.05

0.25

0.13

7.07

5.20

0.65 BSC0.0256 BSC

7.65

0.55

8˚

0˚

20-PIN PLASTIC

SHRINK

SMALL-OUTLINE

MAX

1.99

0.21

0.38

0.22

7.33

5.38

7.90

0.95
8˚

21-0003A

PACKAGE