Rainbow Electronics MAX746 User Manual

19-0192; Rev 1; 11/93

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

_______________General Description

The MAX746 is a high-efficiency, high-current, step-down
DC-DC power-supply controller that drives external N-chan­nel FETs. It provides 93% to 96% efficiency from a 6V supply
voltage with load currents ranging from 50mA up to 3A. It
uses a pulse-width-modulating (PWM) current-mode control
scheme to provide precise output regulation and low output
noise. The MAX746's 4V to 15V input voltage range, fixed
5V/adjustable (Dual-ModeTM) output, and adjustable current
limit make this device ideal for a wide range of applications.

High efficiency is maintained with light loads due to a propri­etary automatic pulse-skipping control (Idle-ModeTM) scheme
that minimizes switching losses by reducing the switching fre­quency at light loads. The low 950µA quiescent current and
ultra-low 1.4µA shutdown current further extend battery life.

External components are protected by the MAX746's cycle­by-cycle current limit. The MAX746 also features a 2V ±1.5%
reference, a comparator for low-battery detection or level
translating, and soft-start and shutdown capability.

The MAX747—discussed in aseparatedata sheet—
functions similarly to the MAX746, but drives P-channel logic
level FETs.

________________________Applications

5V-to-3.3V Green PC Applications
Notebook/Laptop Computers
Personal Digital Assistants
Battery-Operated Equipment
Cellular Phones

__________Typical Operating Circuit

INPUT 6V TO 15V

V+

AV+

40mΩ

OUTPUT

39µH

440µF

LOW-BATTERY
DETECTOR OUTPUT

5V

ON/OFF

LOW-BATTERY

DETECTOR INPUT

CP
HIGH
SHDN

LBI
REF

SS

MAX746

CC

FB

AGND

GND

CS

EXT

OUT
LBO

____________________________Features

♦ 93% to 96% Efficiency for 50mA to 3A

Output Currents

♦ 4V to 15V Input Voltage Range
♦ Low 950µA Supply Current
♦ 1.4µA Shutdown Current
♦ Drives External N-Channel FETs
♦ Fixed-Frequency Current-Mode PWM (Heavy Loads)
♦ Idle-Mode PFM (Light Loads)
♦ Cycle-by-Cycle Current Limiting
♦ 2V ±1.5% Accurate Reference Output
♦ Adjustable Soft-Start
♦ Undervoltage Lockout
♦ Precision Comparator for Power-Fail or

Low-Battery Warning

______________Ordering Information

PART TEMP. RANGE PIN-PACKAGE

MAX746CPE 0°C to +70°C 16 Plastic DIP
MAX746CSE 0°C to +70°C 16 Narrow SO
MAX746C/D 0°C to +70°C Dice*
MAX746EPE -40°C to +85°C 16 Plastic DIP
MAX746ESE -40°C to +85°C 16 Narrow SO
MAX746MJE -55°C to +125°C 16 CERDIP

* Contact factory for dice specifications.

__________________Pin Configuration

TOP VIEW

LBO

REF

SHDN

AV+

1

LBI

2

SS

3

MAX746

4
5

FB

6

CC

7
8

GND

16

V+

15

CP

14

HIGH

13

EXT

12

AGND

11

CS

10

OUT

9

MAX746

DIP/SO

™Dual-Mode and Idle-Mode are trademarks of Maxim Integrated Products.

________________________________________________________________

Maxim Integrated Products

Call toll free 1-800-998-8800 for free samples or literature.

1

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

ABSOLUTE MAXIMUM RATINGS

Supply Voltage V+, AV+ to GND..............................-0.3V to 17V

HIGH, EXT to GND....................................................-0.3V to 21V

AGND to GND..........................................................-0.3V to 0.3V

All Other Pins................................................-0.3V to (V+ + 0.3V)

Reference Current (I
Continuous Power Dissipation (T

Plastic DIP (derate 10.53mW/°C above +70°C) ..........842mW

MAX746

Narrow SO (derate 8.70mW/°C above +70°C) ............696mW

CERDIP (derate 10.00mW/°C above +70°C)...............800mW

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

)....................................................±2mA

REF

= +70°C)

A

ELECTRICAL CHARACTERISTICS

(V+ = 10V, I

Input Voltage 415V

Output Voltage

Feedback Voltage V

Line Regulation

Load Regulation 1.3 2.5 %

Efficiency 94 %

OUT Leakage Current 50 80
FB Input Logic Low 40 mV
FB Input Leakage Current 1 100 nAFB = 2V

Reference Voltage V

Reference Load Regulation 920mV
Soft-Start Source Current 0.5 1.0 1.5
Soft-Start Fault Current (Note 1) 100 500

Supply Current (Note 2) I

Oscillator Frequency f

= 0A, I

LOAD

PARAMETER SYMBOL MIN TYP MAX UNITS

= 0µA, TA= T

REF

to T

MIN

V+

V

OUT

FB

REF

SUPP

OSC

, unless otherwise noted.)

MAX

V+ = 6V to 15V, 0V < (V+ - CS) < 0.125V,
FB = 0V (includes line and load regulation)

(V+ - CS) = 0V,
external feedback mode

V+ = 6V to 15V, FB = 0V 0.05
V+ = 4V to 15V, external feedback mode
0V < (V+ - CS) < 0.125V

Circuit of Figure 1, I
V+ = 6V

V

= 5V

OUT

For dual-mode switchover

I

= 0µA

REF

I

= 0µA to 100µA

REF

SS = 0V
SS = 2V

Operating, V+ = 15V

Operating, V+ = 10V
Shutdown mode

Operating Temperature Ranges:

MAX746C_E........................................................0°C to +70°C

MAX746E_E.....................................................-40°C to +85°C

MAX746MJE ..................................................-55°C to +125°C

Junction Temperatures:

MAX746C_E/E_E..........................................................+150°C

MAX746MJE.................................................................+175°C

Storage Temperature Range.............................-65°C to +160°C

Lead Temperature (soldering, 10sec).............................+300°C

CONDITIONS

4.85 5.08 5.25 V

MAX746C 1.96 2.00 2.04
MAX746E/M 1.95 2.00 2.05

0.1

= 0.5A to 2.5A,

LOAD

MAX746C
MAX746E/M

MAX746C 1.1 1.4
MAX746E/M

MAX746E/M

1.97 2.00 2.03

1.96 2.00 2.04

1.7

0.95

1.4 20
85 100 115MAX746C
80 100 120

V

%/V

µA

V

µA
µA

mA

µA

kHz

2 _______________________________________________________________________________________

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

ELECTRICAL CHARACTERISTICS (continued)

(V+ = 10V, I

Maximum Duty Cycle 91 96 %
Charge-Pump Output Voltage
Current-Sense Amplifier

Current-Limit Threshold
EXT Output High V
EXT Output Low
EXT Sink Current 160 mA
EXT Source Current 270 mA
Compensation Pin Impedance 24

LBI Threshold Voltage

LBO Output Voltage Low V
LBI Input Leakage Current nA
LBO Output Leakage Current 1
SHDN Input Voltage Low V
SHDN Input Voltage High V
SHDN Input Leakage Current

Note 1: The soft-start fault current is the current sink capability of SS when V
Note 2: I

= 0A, I

LOAD

PARAMETER

is the supply current drawn by V+, which includes the current drawn by the charge pump. The charge pump

SUPP

doubles the current drawn by HIGH from the V+ input, so I

= 0µA, TA= T

REF

to T

MIN

SYMBOL MIN TYP MAX UNITS

V

HIGH

V

LIMIT

OL

IL
IH

, unless otherwise noted.)

MAX

CONDITIONS

V+ = 6V
I

= 0mA to 10mA

HIGH

V+ – CS

V

forced to 15V, I

HIGH

V

forced to 15V, I

HIGH

V

= 15V, V

HIGH

V

HIGH

LBI falling

I

= 0.5mA

SINK

LBI = 2.5V
V+ = 15V, LBO = 15V, LBI = 2.5V

SHDN = 10V

= 15V, V

EXT
EXT

EXT
EXT

= 12.5V
= 2.5V

SUPP

= -1mA
= 1mA

REF

= IV++ 2I

V+ + 4 V+ + 5 V+ + 6 V

125 150 175 mV

- 0.1 V

HIGH

MAX746C 1.97 2.00 2.03
MAX746E/M

< 1V or when the device is in shutdown.

HIGH

1.96 2.00 2.04

2.0 V

0.1 100

.

0.25 V

0.4 V

100

0.4 V

MAX746

kΩ

V

µA

nA

_______________________________________________________________________________________ 3

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

__________________________________________Typical Operating Characteristics

(Circuit of Figure 1a, TA = +25°C, unless otherwise noted.)

N0-LOAD SUPPLY CURRENT

1.2

MAX746

1.1

1.0

0.9

NO-LOAD SUPPLY CURRENT (mA)

0.8
579111315

EFFICIENCY vs. OUTPUT CURRENT

100

90

EFFICIENCY (%)

80

vs. SUPPLY VOLTAGE

SUPPLY VOLTAGE (V)



VIN = 6V

VIN = 9V

VIN = 12V

4

MAX746-01

3

2

1

NO-LOAD SUPPLY CURRENT (mA)

0

-75

100

MAX746-07

CIRCUIT OF FIGURE 1b
V
V+ = 5V

90

EFFICIENCY (%)

80

NO-LOAD SUPPLY CURRENT 

V+ = 9V

= 5V

V

OUT

-50 -25 0 25 50 75 100 125

EFFICIENCY vs. OUTPUT CURRENT

= 3.3V

OUT



vs. TEMPERATURE

ENTIRE
CIRCUIT

SCHOTTKY DIODE
LEAKAGE EXCLUDED

TEMPERATURE (°C)



MAX746-02

MAX746-08

CONTINUOUS-CONDUCTION MODE
BOUNDARY AND CORRESPONDING

PEAK INDUCTOR CURRENT

15

DISCONTINUOUS-

CONDUCTION REGION

13

PEAK

INDUCTOR

CURRENT

11

9

SUPPLY VOLTAGE (V)

7

5

0.9 1.1 1.3 1.5 1.7

0.7
OUTPUT CURRENT (A)

EFFICIENCY vs. OUTPUT CURRENT

100

CIRCUIT OF FIGURE 1c
V

= 5V

OUT

90

EFFICIENCY (%)

80

VIN = 6V

VIN = 12V

CONTINUOUS-

CONDUCTION

REGION



MAX746-09

MAX746-06

CIRCUIT OF FIGURE 1a

= 5V

V

70

0.01

4

CIRCUIT OF FIGURE 1a
V

3

2

1

PEAK INDUCTOR CURRENT (A)

VIN = 9V

0

0.01

OUT

0.1 1 10

OUTPUT CURRENT (A)

PEAK INDUCTOR CURRENT

vs. OUTPUT CURRENT

= 5V

OUT

VIN = 12V

VIN = 6V

0.1 1 10

OUTPUT CURRENT (A)

70

0.01

4

MAX746-03

CIRCUIT OF FIGURE 1b

V

OUT

3

V+ = 5V

2

1

PEAK INDUCTOR CURRENT (A)

0

0.01

0.1 1 10

OUTPUT CURRENT (A)

PEAK INDUCTOR CURRENT

vs. OUTPUT CURRENT

= 3.3V

0.1 1 10

OUTPUT CURRENT (A)

MAX746-05

PEAK INDUCTOR CURRENT (A)

70

0.01

1.5

1.0

0.5

0

0.01

0.1 1 10

OUTPUT CURRENT (A)

PEAK INDUCTOR CURRENT

vs. OUTPUT CURRENT

CIRCUIT OF FIGURE 1c
V

= 5V

OUT

VIN = 12V

VIN = 6V

0.1 1

OUTPUT CURRENT (A)

4 _______________________________________________________________________________________

MAX746-03

10

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

____________________________Typical Operating Characteristics (continued)

(Circuit of Figure 1a, TA = +25°C, unless otherwise noted.)

LOAD-TRANSIENT RESPONSE

200µs/div

A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
B: V

RIPPLE, 50mV/div, AC-COUPLED

OUT

V+ = 10V

CONTINUOUS-CONDUCTION MODE

WAVEFORMS

LOAD-TRANSIENT RESPONSE

A

B

A: LOAD CURRENT, 0.1A TO 1.5A, 1A/div
B: V

RIPPLE, 50mV/div, AC COUPLED

OUT

V+ = 10V

DISCONTINUOUS-CONDUCTION

IDLE-MODE WAVEFORMS

A

B

0V

C

A

B

1ms/div

A

B

C

LINE-TRANSIENT RESPONSE

8V

500ms/div

A: V+ = 8V TO 10V, 2V/div

RIPPLE, 100mV/div

B: V

OUT

I

= 3A

OUT

MODERATE-LOAD, IDLE-MODE

WAVEFORMS

10V
A

B

MAX746

A

B

C

5µs/div

A : EXT VOLTAGE, 20V/div
B : INDUCTOR CURRENT 1A/div
C : V

RIPPLE, 50mV/div

OUT

V+ = 10V, I

= 3A

OUT

_______________________________________________________________________________________

20µs/div

A: EXT VOLTAGE, 10V/div
B: INDUCTOR CURRENT, 500mA/div
C: V

RIPPLE, 50mV/div, AC-COUPLED

OUT

V+ = 10V, I

OUT

= 75mA

20µs/div

A: EXT VOLTAGE, 10V/div
B: INDUCTOR CURRENT, 500mA/div
C: V

RIPPLE, 50mV/div, AC-COUPLED

OUT

V+ = 6V, I

= 480mA

OUT

5

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

______________________________________________________________Pin Description

PIN NAME FUNCTION

1 LBO

2 LBI Input to the low-battery comparator. Tie to V+ or GND if not used.

MAX746

3 SS

4 REF

5 SHDN

6 FB

7 CC

8 AV+

9 OUT

10 CS
11 AGND Quiet analog ground.

12 EXT

13 HIGH Regulated high-side voltage, 5V above the V+ supply voltage.
14 CP Charge-pump output that generates a 0V to V+, 50kHz square wave (see
15 V+ High-current supply voltage for the charge pump.
16 GND High-current ground return for the output driver and charge pump.

Low-battery output is an open-drain output that goes low when LBI is less than 2V. Connect to V+ through a
pull-up resistor. Leave floating if not used. LBO is disabled in shutdown mode.

Soft-start limits start-up surge currents. On power-up, it charges the soft-start capacitor, slowly raising the peak
current limit to the level set by the sense resistor.

2V reference output can source 100µA for external loads. Bypass with 1µF. The reference is disabled in shutdown mode.
Active-high logic input. In shutdown mode, V

Connect to GND for normal operation.
Feedback input for adjustable-output operation. Connect to GND for fixed 5V output. Use a resistor-divider net-

work to adjust the output voltage (see
AC compensation input for the error amplifier. Connect a capacitor between CC and GND for fixed 5V-output

operation (see
Quiet supply voltage for sensitive analog circuitry. Also the noninverting input to the current-sense amplifier. A

separate bypass capacitor is not recommended for AV+.
Output voltage sense that connects to the internal resistor divider. Bypass with 0.1µF to AGND, close to the IC

for fixed output operation. Leave unconnected for adjustable-output operation.
Inverting input to the current-sense amplifier. Connect the current-sense resistor (R

Power MOSFET gate-drive output that swings between HIGH and GND. EXT is not protected against short cir­cuits to V+ or AGND.

Compensation Capacitor

Setting the Output Voltage

= 0V and the supply current is reduced to less than 20µA.

OUT

section).

section).

SENSE

Charge Pump

) from AV+ to CS.

section).

____________________Getting Started

Figure 1a shows the 5V-output 3A standard application
circuit, Figure 1b shows the 3.3V-output 3A standard
application circuit, and Figure 1c shows the 5V-output

1.5A standard application circuit. Most applications will
be served by these circuits. To learn more about compo­nent selection for particular applications, refer to the

Design Procedure

ation of the MAX746, refer to the

section. To learn more about the oper-

Detailed Description

.

_______________Detailed Description

The MAX746 monolithic, CMOS, step-down, switch­mode power-supply controller provides high-side drive
for external logic-level N-channel FETs. A charge pump
generates a voltage 5V above the supply voltage for
high-side drive capability. The MAX746 uses a unique

6 _______________________________________________________________________________________

current-mode pulse-width-modulating (PWM) control
scheme that results in tight output-voltage regulation,
excellent load- and line-transient response, low noise,
and high efficiency over a wide range of load currents.
Efficiency at light loads is further enhanced by a propri­etary idle-mode switching control scheme that skips
oscillator cycles in order to reduce switching losses.
Other features include undervoltage lockout, shutdown,
and a low-battery detection comparator.

Operating Principle

Figure 2 is the MAX746 block diagram. The MAX746
regulates using an inner current-feedback loop and an
outer voltage-feedback loop. A slope-compensation
scheme stabilizes the current loop; the dominant pole,
formed by the output filter capacitor and the load,
stabilizes the voltage loop.

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

VIN

6V TO 15V

C3

0.1µF

R2

R1

0.1µF

1.0µF

SEE TABLE 2 FOR DIODE SELECTION.

*

C2
100µF

15

LBI

SS

REF

FB

SHDN

AGND

V+

MAX746

GND

16

2

C5

3

C6

4

6

5

11

Figure 1a. 5V Standard Application Circuit (15W)

Discontinuous-/Continuous-

Conduction Modes

The MAX746 is designed to operate in continuous-con­duction mode (CCM) but can also operate in discontinu­ous-conduction mode (DCM), making it ideal for variable­load applications. In DCM, the current starts at zero and
returns to zero on each cycle. In CCM, the inductor current
never returns to zero; it consists of a small AC component
superimposed on a DC offset. This results in higher current
capability because the AC component in the inductor cur­rent waveform is small. It also results in lower output noise,
since the inductor does not exhibit the ringing that would
occur if the current reached zero (see inductor waveforms
in the

Typical Operating Characteristics

). To transfer equal
amounts of energy to the load in one cycle, the peak cur­rent level for the discontinuous waveform must be much
larger than the peak current for the continuous waveform.

Slope Compensation

Slope compensation stabilizes the inner current-feedback
loop by adding a ramp signal to the current-sense amplifier
output. Ideal slope compensation can be achieved by
adding a linear ramp, with the same slope as the declining
inductor current, to the rising inductor current-sense voltage.

D3

1N914

Si9410DY

C9

4.7µF

*

Q1

NSQ03A03

R



SENSE

40mΩ

L1

N

39µH

D1

R3
100k

5V

AT 3A

C1
430µF

HIGH

AV+

EXT

OUT

LB0

D4

1N5817

D2

1N914

*

C8

0.1µF

14

CP

13
8

10

CS

12

7

CC

C7

2.7nF

9

C4

0.1µF

1

Under these conditions, the inductor must be scaled to the
current-sense resistor value.

Overcompensation adds a pole to the outer voltage feed­back-loop response, degrading loop stability. This may cause
voltage-mode pulse-frequency-modulation instead of PWM
operation. Undercompensation results in inner current feed­back-loop instability, and may cause the inductor current to
staircase. Ideal matching between the sense resistor and
inductor is not required; it can differ by ±30% or more.

Oscillator and EXT Control

The oscillator frequency is nominally 100kHz, and the duty
cycle varies from 5% to 96%, depending on the input/out­put voltage ratio. EXT, which provides the gate drive for the
external logic-level N-FET, is switched between HIGH and
GND at the switching frequency. EXT is controlled by a
unique two-comparator control scheme consisting of a PWM
comparator and an idle-mode comparator (Figure 2). The
PWM comparator determines the cycle-by-cycle peak cur­rent with heavy loads, and the idle-mode comparator sets
the light-load peak current. As V

begins to drop, EXT

OUT

goes high and remains high until both comparators trip.
With heavy loads, the idle-mode comparator trips first and
the PWM control comparator determines the EXT on-time;

MAX746

_______________________________________________________________________________________ 7

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

V



IN

4.5V TO 6V

C3

0.1µF

C2
100µF

D2

1N914

D3

1N914

D5

1N914

D6

1N914

D4
1N5817

C11
1µF

R2

MAX746

R1

SUMIDA CDR125 22µH SURFACE-MOUNT INDUCTOR

*

C5

0.1µF
C6

1µF

2

LBI

3

SS

4

REF





5

SHDN

11

AGND

15

V+

MAX746

GND

16

HIGH

AV+

EXT

OUT

LB0

14

CP

13
8

10

CS

12
7

CC

9
6

FB

1

Figure 1b. 3.3V Standard Application Circuit (9.9W)

with light loads, the PWM comparator trips quickly and the
idle-mode comparator sets the EXT on-time.

Traditional PWM converters continue to switch on every
cycle, even when the inductor current is discontinuous
due to smaller loads, decreasing light-load efficiency.
In contrast, the MAX746’s idle-mode comparator increas­es the switch on-time, allowing more energy to be trans­ferred per cycle. Since fewer cycles are required, the
switching frequency is reduced, resulting in minimal
switching losses and increased efficiency.

The light-load output noise spectrum widens due to the
variable switching frequency in idle-mode, but output
ripple remains low. Using the Typical Operating Circuit,
with a 9V input and a 125mA load current, output ripple
is less than 40mV.

Charge Pump

The MAX746 contains all the control circuitry required
to provide a regulated charge-pump voltage 5V
above V+ for high-side driving N-channel logic FETs.
The charge pump operates with a nominal 50kHz fre-

C8

0.1µF

C7
2nF

C9

C10

1µF

0.1µF

R



SENSE

40mΩ

Q1

R5

13k (1%)

R4
20k (1%)

Si9410DY

NSQ03A03

L1*

N

22µH

D1

R3
100k

C4

0.1µF

quency. When the voltage at HIGH exceeds AV+ by
5V, the charge-pump oscillator is inhibited (Figure 2).
When the voltage at HIGH is less than 4.3V below V+,
undervoltage lockout occurs. Use the voltage tripler
(Figure 3b) when V+ ≤ 6V; otherwise, use the voltage
doubler (Figure 3a).

Soft-Start and Current Limiting

The MAX746 draws its highest current at power-up. If
the power source to the MAX746 cannot provide this
initial elevated current, the circuit may not function cor­rectly. For example, after prolonged use the increased
series resistance of a battery may prevent it from pro­viding adequate initial surge currents when the
MAX746 is brought out of shutdown. Using soft-start
(SS) minimizes the possibility of overloading the incom­ing supply at power-up by gradually increasing the
peak current limit. Connect an external capacitor from
SS to AGND to reduce the initial peak currents drawn
from the supply.

3.3V

AT 3A

C3
660µF

8 _______________________________________________________________________________________

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

V



IN

6V TO 15V

C3

0.1µF

R2

R1

0.1µF

1µF

SEE TABLE 2 FOR DIODE SELECTION.

*

SUMIDA CDR125 SURFACE-MOUNT INDUCTOR.

**

C2
47µF

2

LBI

C5

3

SS

C6

4

REF

6

FB

5

SHDN

11

AGND

Figure 1c. 5V Standard Application Circuit (7.5W)

V+

MAX746

GND

MAX746

D3

1N914

*

Si9410DY

NSQ03A03

C9

4.7µF

Q1

R



SENSE

75mΩ

L1**

N

82µH

D1

R3
100k

5V

AT 1.5A

C1
220µF

D4

1N5817

HIGH

AV+

EXT

OUT

LB0

D2

1N914

*

C8

0.1µF

14

CP

13
8

10

CS

12

7

CC

C7
1nF

9

C4

0.1µF

1

15

16

The steady-state SS pin voltage is typically 3.8V. On
power-up, SS sources 1µA until its voltage reaches

3.8V. The current-limit comparator inhibits EXT switch­ing until the SS voltage reaches 1.8V. The peak current
limit is set by:

V

where V

LIMIT

_________

IPK=

R

SENSE

is the differential voltage across the current-

LIMIT

150mV (typ)

___________

=

R

SENSE

sense amplifier inputs. Figure 4 shows how the SS peak
current limit increases as the voltage on SS rises for two
R

values.

SENSE

Undervoltage Lockout

Undervoltage lockout inhibits operation of EXT until the
charge pump is capable of generating a voltage greater
than 4.3V above the supply voltage (Figure 2). When
the undervoltage-lockout comparator detects an under­voltage condition, the switching action at EXT is halted.

_______________________________________________________________________________________ 9

When SHDN is high, the MAX746 is shut down. In this

Shutdown Mode

mode, the internal biasing circuitry (including EXT) is
turned off, V

drops to 0V, and the supply current

OUT

drops to 1.4µA (20µA max). This excludes external
component leakage, which may add several
microamps to the shutdown supply current for the
entire circuit. SHDN is a logic input. Connect SHDN to
GND for normal operation.

Low-Battery Detector

The MAX746 provides a low-battery comparator that
compares the voltage on LBI to the reference voltage.
LBO, an open-drain output, goes low when the LBI volt­age is below V

. Use a resistor-divider network, as

REF

shown in the Input Voltage Monitor Circuit (Figure 5),
to set the trip voltage (V
this circuit, LBO goes low when V+ ≤ V

) at the desired level. In

TRIP

. LBO is high

TRIP

impedance in shutdown mode.

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

LBI

MAX746

REF

OUT

CC

FB

100mV

AV+

CS

LOW-BATTERY
COMPARATOR

DUAL-MODE
COMPARATOR

CURRENT-SENSE
AMPLIFIER

COMPENSATION

SLOPE-

RAMP

LBO

N

+2V

REFERENCE

V

RAMP

EXT

ERROR
AMPLIFIER

Σ

50mV

HIGH

4.3V

5V

PWM
COMPARATOR

LIGHT-LOAD
COMPARATOR

FROM AV+

CHARGE-PUMP CONTROL
COMPARATOR

UNDERVOLTAGE-
LOCKOUT
COMPARATOR

EXT

CONTROL

T

T FLIP-
FLOP

100kHz

OSCILLATOR

V+ PUMP

Q

SHDN

SOFT-START

SS

CIRCUITRY

CURRENT-LIMIT
COMPARATOR

AGND

GND

Figure 2. Block Diagram

10 ______________________________________________________________________________________

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

V

IN

15

V+

D2

1N914

0.1µF

C8

MAX746

T FLIP-

T

FLOP

CLK

100kHz

OSCILLATOR

GND

MAX746

GND

CP

AV+

HIGH

HIGH

14

13

5V

D3

D2

1N914

1N914

C8

0.1µF

14

CP

13

Q

16

V

IN

15

V+

16

Figure 3. Charge-Pump Configurations

3a. CHARGE-PUMP
VOLTAGE DOUBLER

D4

D3

1N5817

1N914

3b. CHARGE-PUMP
VOLTAGE TRIPLER

D4

1N5817

D5

D6

1N914

1N914

C9

C10

1µF

0.1µF

C9
1µF

C11
1µF

PEAK CURRENT LIMIT

vs. SOFT-START VOLTAGE

3

R

= 50mΩ

SENSE

2

V+ - VCS = 150mV

1

PEAK CURRENT LIMIT (A)

R

= 100mΩ

0

01234

SOFT-START VOLTAGE (V)

SENSE

Figure 4. Peak Current Limit vs. Soft-Start Voltage

V

IN

15

R2

2

R1

V+

MAX746

LBI

GND

16

LBO

…TO V

OUT

R3
100k

1

LOW-BATTERY

R2 = R1 -1

= 2.0V

V

REF

OR VIN

( )

Figure 5. Input Voltage Monitor Circuit

MAX746-FG03

OUTPUT

V

TRIP

V

REF

MAX746



__________________Design Procedure

The MAX746’s dual-mode output voltage can be set
to 5V by grounding FB, or it can be adjusted from
2V to 14V using external resistors R4 and R5 config­ured as shown in Figure 6. Select feedback resistor
R4 in the 10kΩ to 60kΩ range. R5 is given by:

The MAX746 is designed to use either internal or exter­nal feedback mode, but should not be toggled between

Setting the Output Voltage

V

OUT

R5 = (R4)

______________________________________________________________________________________ 11

_______

(

2V

– 1

)

the two modes while operating. If two different output
voltages are required, use external feedback mode
with a resistor network similar to the 3.3V/5V adjustable
output circuit shown in Figure 7.

To select the sense-resistor value (R

Selecting R

SENSE

SENSE

), first
approximate the peak current assuming IPKis
(1.1) (I

LOAD

), where I

is the maximum load cur-

LOAD

rent. Once all component values have been deter­mined, the actual peak current is given by:

IPK= I

LOAD

___________

+

(

(2L) (f

V

OUT

OSC

V

OUT

_______

1–

)(

)V

)

IN

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

VIN

15

V+

MAX746

* SEE

Figure 6. Adjustable Output Circuit

MAX746

GND

16

COMPENSATION CAPACITOR

FB

OUT

SECTION.

6

C7*

9

R5

R4

R4 = 10kΩ TO 60kΩ

V



OUT

R5 = R4 -1

( )

V

REF

= 2.0V NOMINAL

V

REF

V

OUT

SELECT WITH FET OFF:

SELECT WITH FET OFF:

Figure 7. 3.3V/5V Ajustable Output Circuit

MAX746

V

= V

OUT

V

= V

OUT

= 2.0V NOMINAL

V

REF

12

EXT





6

FB

9

OUT

R5

+1

REF

( )

R4a

R5

REF +1

( )

R4a + R4b

5V/3.3V

N

D1

26.1k (1%)

17.4k (1%)

L

R4a

R5

N

V

OUT

C1

C7

R4b

22.6k (1%)

Next, determine the value of R

V

LIMIT(min)

R

SENSE

_____________

=

I

PK

SENSE

such that:

125mV

________

=

I

PK

For example, to obtain 5V at 3A, IPK= 3.3A and
R

= 125mV/3.3A = 38mΩ.

SENSE

The sense resistor should have a power rating greater

PK

2)

(R

) with an adequate safety margin.

SENSE

SENSE

= 38mΩ.

than (I
With a 3A load current, IPK= 3.3A and R
The power dissipated by the resistor (assuming an 80%
duty cycle) is 331mW. Metal-film resistors are recom­mended. Do not use wire-wound resistors because
their inductance will adversely affect circuit operation.

The duty cycle (for continuous conduction) is determined
from the following equation:

V

V

OUT

DIODE

Duty Cycle (%) =

_____________________

+

V+ - VSW+ V

DIODE

x 100%

where VSWis the voltage drop across the external
N-FET and sense resistor. VSWcan be approximated
as [I

LOAD

x (r

DS(ON)

+ R

SENSE

)].

Inductor Selection

Once the sense-resistor value is determined, calculate
the inductor value (L) using the following equation. The
correct inductor value ensures proper slope compen­sation. Continuing from the equations above:

(

______________________

L =

(V

_____________________

=

(50mV)(100kHz)

R

SENSE

RAMP(max)

(

38mΩ)(5V

)(

V

)(f

OUT

OSC

)

)

)

= 38µH

where V

RAMP(max)

is the 50mV peak value of the slope-

compensation linear ramp signal.
Although 38µH is the calculated value, the component

used may have a tolerance of ±30% or more.
Inductors with molypermalloy powder (MPP), Kool Mµ,

or ferrite are recommended. Inexpensive iron-powder
core inductors are not suitable, due to their increased
core losses, especially at switching frequencies in the
100kHz range. MPP and Kool Mµ cores have low per­meability, allowing larger currents.

For highest efficiency, use a coil with low DC resis­tance. To minimize radiated noise, use a toroid, a pot
core, or a shielded coil.

It is customary to select an inductor with a saturation
rating that exceeds the peak current set by R

SENSE

but inductors are often specified very conservatively.
If the inductor’s core losses do not cause excessive
temperature rise (inductor wire insulation is usually
rated for +125°C) and the associated efficiency loss­es are minimal, inductors with lower current ratings
are acceptable.

In the 3.3V Standard Application Circuit (Figure 1b), the
inductor selected has a 2.2A current rating even
though the peak current is 3.3A. This inductor was
selected for two reasons: it is the highest-rated readily
available surface-mount inductor of its size, and lab
tests have verified that the core-loss increase is mini­mal. With a 3A load current, the inductor current does
not begin showing significant losses due to saturation
until the supply voltage increases to 10V (the maximum
supply for this circuit is 6V).

,

12 ______________________________________________________________________________________

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

External Logic-Level N-FET Selection

To ensure the external N-FET is turned on hard, use
logic-level or low-threshold N-FETs. Three important
parameters to note when selecting the N-FET are the
total gate charge (Qg), on resistance (r
reverse transfer capacitance (C

Qgincludes all capacitances associated with charging
the gate. Use the typical Qgvalue for best results; the
maximum value is usually grossly overspecified, since
it is a guaranteed limit and not the measured value.
The typical total gate charge should be 50nC or less;
with larger numbers, EXT may not be able to ade­quately drive the gate. EXT sink/source capability
(I

) is typically 210mA.

EXT

The two most significant losses contributing to the
N-FET’s power dissipation are I2R losses and switching
losses. CCM power dissipation (PD), is approximated by:

PD= (Duty Cycle)(I

(V+2)(C

__________________________

where the duty cycle is approximately V
f

= 100kHz, and r

OSC

data sheet of the chosen N-FET. In the equation,
r
of temperature. The equation given does not account
for losses incurred by charging and discharging the
gate capacitance, because that energy is dissipated
by the gate-drive circuitry, not the N-FET.

The Standard Application Circuits (Figure 1) use an
8-pin, Si9410DY, surface-mount N-FET that has 0.05Ω
on resistance with a 4.5V VGS. Optimum efficiency is
obtained when the voltage at the source swings between
the supply rails (within a few hundred millivolts).

is assumed constant, but is actually a function

DS(ON)

2

PK

RSS

(I

EXT

DS(ON)

)(r

)(IPK)(f

RSS

DS(ON)

)

and C

).

) +

)

OSC

are given in the

RSS

DS(ON)

OUT

), and

/V+,

Diode Selection

The MAX746’s high switching frequency demands a
high-speed rectifier. Schottky diodes are recommend­ed. Ensure that the Schottky diode average current
rating exceeds the maximum load current.

Capacitor Selection

The output filter capacitor C1 should have a low effec­tive series resistance (ESR), and its capacitance should
remain fairly constant over temperature. This is espe­cially true when in CCM, since the output filter capaci­tor and the load form the dominant pole that
stabilizes the voltage loop.

Output Filter Capacitor

To ensure stability, the minimum capacitance and max­imum ESR values are:

C1

and,

where GBW = the loop gain-bandwidth product, 15kHz.
Sprague 595D surface-mount solid tantalum capacitors

and Sanyo OS-CON through-hole capacitors are rec­ommended due to their extremely low ESR. OS-CON
capacitors are particularly useful at low temperatures.
For best results when using other capacitors, increase
the output filter capacitor’s size or use capacitors in
parallel to reduce the ESR.

Bypass OUT with a 0.1µF (C4) capacitor to GND when using
a fixed 5V output (Figures 1a and 1c). With adjustable-output
operation, place C4 between the output voltage and AGND
as close to the IC as possible (Figure 1b).

The circuit load-step response is improved by using a
larger output filter capacitor or by placing a low-cost
bulk capacitor in parallel with the required low-ESR
output filter capacitor. The output voltage sag under a
load step (I

V

SAG

where DMAX is the maximum duty cycle (91% worst
case). The equation assumes an input/output voltage
differential of 2V or more. Table 1 gives measured val­ues of output voltage sag with a 30mA to 3A load step
for various input voltages and output filter capacitors.
Refer also to the

Differentials

The input bypass capacitor C2 reduces peak currents
drawn from the voltage source, and also reduces the
amount of noise at the voltage source caused by the
MAX746’s fast switching action (this is especially
important when other circuitry is operated from the
same source). The input capacitor ripple current rating
must exceed the RMS input ripple current.

= I

______________________________

>

(min)

(2π)(GBW)(V

ESRC1<

) is approximated by:

STEP

_____________________________________

=

(2)(C1)(V

AC Stability with Low Input/Output

section.

I

= RMS AC input current

RMS

(

LOAD

(5)(V

(V

OUT

___________________

(I

STEP

IN(MIN

(

V

OUT

√

_______________________

)

REF

)(R

OUT

)(R

SENSE

(V

)

REF

2

)(L)

)(D

MAX

Input Bypass Capacitor

)(

V

IN - VOUT

V

IN

SENSE

)

- V

OUT

)

)

)

)

MAX746

______________________________________________________________________________________ 13

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

Table 1. Measured Output Voltage Sag
with 30mA to 3A Load Step*

OUTPUT

FILTER

CAPACITOR

C1 (mF)

MAX746

440 400 250 210 140 90
660 260 190 160 70 50
880 200 100 90 40 25

*Circuit of Figure 1a.

For load currents up to 3A, 100µF (C2) in parallel with

0.1µF (C3) is adequate. Smaller bypass capacitors may
also be acceptable for lighter loads. The input voltage
source impedance determines the size of the capacitor
required at the V+ input. As with the output filter capaci­tor, a low-ESR capacitor (Sanyo OS-CON, Sprague 595D
or equivalent) is recommended for input bypassing.

Figure 3a shows the charge-pump doubler circuit con­figured with a 0.1µF charge-pump capacitor C8 and a

1.0µF reservoir capacitor C9. The ratio of the capaci­tors, along with the input voltage, determines the
amount of ripple on HIGH. If the input supply range
exceeds 12V, increase C9 to 4.7µF to reduce the
charge-pump ripple. C9 should be 10µF for less.
Figure 3b shows the charge-pump tripler circuit.

Refer to Table 2 to determine the proper charge-pump
configuration (which is based on the minimum expect­ed supply voltage at V+).

Some interaction occurs between the switch oscillator
and the charge-pump oscillator. This interaction modu­lates the inductor-current waveform, but has negligible
impact on the output.

Soft-start provides a ramp to the full current limit. A typi­cal value for the soft-start capacitor (C5) is 0.1µF,
which provides a 380ms soft-start time. Use values in
the 0.001µF to 1µF range. The nominal time for C5 to
reach its steady-state value is given by:

Note that tSSdoes NOT equal the time it takes for the
MAX746 to power-up, although it does affect the start­up time. The start-up time is also a function of the input

OUTPUT VOLTAGE SAG (mV)

FOR VARIOUS INPUT VOLTAGES

VIN=6V VIN=6.5V VIN=7V VIN=9V VIN=10V

Charge-Pump Capacitors

Soft-Start and Reference Capacitors

tSS(sec) = (C5) (3.8 x 106)

Table 2. Charge-Pump Configuration

V+ CHARGE-PUMP CONFIGURATION

V+ ≤ 6V

6V < V+ < 6.5V*

V+ ≥ 6.5V*

Voltage tripler with 1N914 diodes for D2,
D3, D5, and D6

Voltage doubler with 1N5817 Schottky
diodes for D2 and D3

Voltage doubler with 1N914 diodes for
D2 and D3

* When using the voltage-doubler circuit over the military

temperature range, increase the 6.5V limit to 7V.

voltage and load current. With a 3A load current, a 10V
input voltage, and a 0.1µF soft-start capacitor, it typi­cally takes 240ms for the MAX746 to power up. A

0.47µF soft-start capacitor increases the start-up time
to approximately 2.3sec.

Bypass REF with a 1µF capacitor (C6).

With a fixed 5V output, connect a compensation capac­itor (C7) between CC and AGND to optimize transient
response. Appropriate compensation is determined by
the size and ESR of the output filter capacitor (C1), and
by the load current.

In the standard 5V application circuit, 2.7nF is appro­priate for load currents up to 3A; for lighter loads,
C7’s value can be reduced. If 2.7nF does not com­pensate adequately, use the following equations to
determine C7.

For fixed 5V-output operation:

C7 =

For adjustable-output operation, FB becomes the
compensation input pin, and CC and OUT are left
unconnected. Connect C7 between FB and GND in
parallel with R4 (Figure 6). C7 is determined by:

___________________

C7 =

For example, with a fixed 5V output with C1 = 470µF
and an ESRC1of 0.04Ω (at a frequency of 100kHz):

(C1)(

_____________

C7 =

Compensation Capacitor

)

(

C1

_____________

12kΩ

(

ESR

C1

)

(2) (C1)(ESRC1)

R4  R5

)

ESR

C1

12kΩ

= 1560pF

14 ______________________________________________________________________________________

High-Efficiency, PWM, Step-Down,

N-Channel DC-DC Controller

Increasing C7 by up to 50% enhances outer-loop
stability by adding stability to the inductor current
waveform. But increasing C7 too much causes
FB’s response time to decrease (due to the larger
RC time constant caused by the feedback resistors
and the compensation capacitor), which reduces
load-transient stability.

Setting the Low-Battery

Detector Voltage

Select R1 between 10kΩ and 1MΩ. Determine R2 using
the following equation:

- V

REF

OUT

)

REF

)

(Figure 5).

(V

TRIP

R2 = R1

where V
(e.g., 100kΩ) between LBO and V

is typically 2.0V. Connect a pull-up resistor

REF

________________

(

V

Using a Second Supply in

Place of the Charge Pump

If a secondary power supply (a minimum of 5V above
the main supply) is available, it can be substituted for
the charge-pump high-side supply. In this case,
bypass HIGH with a 1µF capacitor and leave CP
unconnected. Since this secondary supply voltage
is applied to the gate, VGSmust not exceed the
gate-source breakdown voltage of the external N-FET.
Also, the voltage at HIGH must not exceed 20V. If

a secondary supply is used, the shutdown function
cannot be used because HIGH is internally tied to
V+ in shutdown mode. In this case, SHDN must be

tied low. With the main supply off and HIGH at 12V,
HIGH will typically sink 130µA.

Layout Considerations

Because high current levels and fast switching wave­forms radiate noise, proper PC board layout is essen­tial. Use a ground plane, and minimize ground noise by
connecting GND, the anode of the steering Schottky
diode, the input bypass-capacitor ground lead, and the
output filter capacitor ground lead to a single point (star
ground configuration). Also minimize lead lengths to
reduce stray capacitance, trace resistance, and radiat­ed noise. Place bypass capacitor C3 as close to V+
and GND as possible.

AV+ and CS are the inputs to the differential-input
current-sense amplifier. Use a Kelvin connection
across the sense resistor, as shown in Figure 8.
Although AV+ also functions as the supply voltage
for sensitive analog circuitry, a separate AV+ bypass
capacitor should not be used. By not using a capaci-

V

IN

V+

AV+

N

KELVIN SENSE

CONNECTION

L1

V

OUT

R

SENSE

MAX746

CS

EXT

Figure 8. Kelvin Connection for Current-Sense Amplifier

tor, any noise at the CS input will also appear at the
AV+ input, and will be interpreted by the current­sense amplifier as a common-mode signal . A sepa­rate AV+ capacitor causes the noise to appear on
only one input, and this differential noise will be
amplified, adversely affecting circuit operation.

Additional Notes

When probing the MAX746 circuit, avoid shorting
V+ to GND (the two pins are adjacent) as this may
cause the IC to malfunction because of large ground
currents. Because of its fast switching and high drive­capability requirements, EXT is a low-impedance point
that is not short-circuit protected. Therefore, do not
short EXT to any node (including AGND and V+, which
are adjacent to EXT).

Similarly, CC (or FB in adjustable-output operation) is a
sensitive input that should not be shorted to any node.
Avoid shorting CC when probing the circuit, as this may
damage the device.

The MAX746 may continue to operate with AV+ discon­nected, but erratic switching waveforms will appear at EXT.

Switching Waveforms

There is a region between CCM and DCM where the
inductor current operates in both modes, as shown
in the Idle-Mode Moderate Current EXT waveform in
the

Typical Operating Characteristics

put voltage varies, it is fed back into CC and the
duty cycle adjusts to compensate for this change.
The switch is considered off when V

. As the out-

is less than

EXT

MAX746

______________________________________________________________________________________ 15

High-Efficiency, PWM, Step-Down,
N-Channel DC-DC Controller

or equal to the N-FET’s VGSthreshold voltage. Once
the switch is off, the voltage at EXT is pulled to GND
and the N-FET source voltage is a Schottky diode
drop below GND. However, this is not always the case
in the “in-between” mode, due to the changing duty
cycle inherent with DCM. When the device is at maxi­mum duty cycle, EXT turns off at VGS, but the switch
sometimes turns on again after the minimum off-time

MAX746

before EXT can be pulled to GND. This results in short
spikes, which can be seen on the EXT waveform in the

Typical Operating Characteristics

.

Table 3. Component Suppliers

SUPPLIER PHONE FAX

INDUCTORS

Coiltronics [phone] [phone]
Gowanda [phone] [phone]
Sumida USA [phone] [phone]
Sumida Japan 81-3-3607-511 81-3-3607-5428

CAPACITORS

Kemet [phone] [phone]
Matsuo [phone] [phone]
Nichicon [phone] [phone]
Sprague [phone] [phone]
Sanyo USA [phone]
Sanyo Japan 81-3-3837-6242
United Chemi-Con [phone] [phone]

DIODES

Motorola [phone]
Nihon USA [phone] [phone]
Nihon Japan 81-3-3494-7411 81-3-3494-7414

POWER TRANSISTORS

Harris [phone] [phone]
International Rectifier [phone] [phone]
Siliconix [phone] [phone]

RESISTORS

IRC [phone] [phone]

AC Stability with Low Input/Output Differentials

At low input/output differentials, the inductor current
cannot slew quickly enough to respond to load
changes, so the output filter capacitor must hold up the
voltage as the load transient is applied. In Figure 1a’s
circuit, for V+ = 6V, increase the output filter capacitor
to 900µF (Sprague 595D low-ESR capacitors) to obtain
a transient response less than 250mV with a load step
from 0.1A to 3A. As V+ increases, the inductor current
slews faster, so the size of the output filter capacitor can
be reduced (see Table 1).

___________________Chip Topography

LBI

LBO GND

SS

REF

SHDN

FB

TRANSISTOR COUNT: 508;
SUBSTRATE CONNECTED TO HIGH.

AV+

0.080"

(2.03mm)

V+

CP

HIGH

EXT

0.130"

(3.30mm)

AGND

CS

OUT CC

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

16

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