Rainbow Electronics MAX5099 User Manual

General Description

The MAX5099 offers a dual-output, high-switching-fre­quency DC-DC buck converter with an integrated high­side switch. The MAX5099 integrates two low-side
MOSFET drivers to allow each converter to drive an
external synchronous-rectifier MOSFET. Converter 1
delivers up to 2A output current, and converter 2 can
deliver up to 1A of output current. The MAX5099 inte­grates load-dump protection circuitry that is capable of
handling load-dump transients up to 80V for automotive
applications. The load-dump protection circuit utilizes
an internal charge pump to drive the gate of an external
n-channel MOSFET. When an overvoltage or load­dump condition occurs, the series protection MOSFET
absorbs the high voltage transient to prevent damage
to lower voltage components.

The DC-DC converter operates over a wide 4.5V to 19V
operating voltage range. The MAX5099 operates 180°
out-of-phase with an adjustable switching frequency to
minimize external components while allowing the ability
to make trade-offs between the size, efficiency, and
cost. The high switching frequency also allows these
devices to operate outside the AM band for automotive
applications. These regulators can be protected
against high voltage transients such as a load-dump
condition by using the integrated overvoltage controller.

This device utilizes voltage-mode control for stable
operation and external compensation, so that the loop
gain is tailored to optimize component selection and
transient response. The MAX5099 has a maximum duty
cycle of 92.5% and is synchronized to an external clock
fed at the SYNC input.

Additional features include internal digital soft-start,
individual enable for each DC-DC regulator (EN1 and
EN2), open-drain power-good outputs (PGOOD1 and
PGOOD2), and shutdown input (ON/OFF).

Other features of the MAX5099 include overvoltage pro­tection and short-circuit (hiccup current limit) and ther­mal protection. The MAX5099 is available in a thermally
enhanced, exposed pad 5mm x 5mm, 32-pin TQFN
package and operates over the automotive -40°C to
+125°C temperature range.

Applications

Automotive AM/FM Radio Power Supply

Automotive Instrument Cluster Display

Features

♦ Wide 4.5V to 5.5V or 5.2V to 19V Input Voltage

Range with 80V Load-Dump Protection

♦ Dual-Output DC-DC Converter with Integrated

Power MOSFETs

♦ Adjustable Outputs from 0.8V to 0.9V

IN

♦ Output Current Capability Up to 2A and 1A

♦ Switching Frequency Programmable from 200kHz

to 2.2MHz

♦ Synchronization Input (SYNC)

♦ Individual Converter Enable Input and Power-

Good Output

♦ Low-I

Q

(7µA) Standby Current (ON/OFF)

♦ Internal Digital Soft-Start and Soft-Stop

♦ Short-Circuit Protection on Outputs and

Maximum Duty-Cycle Limit

♦ Overvoltage Protection on Outputs with Auto

Restart

♦ Thermal Shutdown

♦ Thermally Enhanced 32-Pin TQFN Package

Dissipates Up to 2.7W at +70°C

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

________________________________________________________________

Maxim Integrated Products

1

Ordering Information

19-4112; Rev 0; 5/08

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.

Pin Configuration appears at end of data sheet.

EVALUATION KIT

AVAILABLE

+

Denotes a lead-free package.

*

EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX5099ATJ+

-40°C to +125°C

32 TQFN-EP*

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND = SGND,
C

BYPASS

= 0.22μF (low ESR), CVL= 4.7μF (ceramic), CV+= 1μF (low ESR), C

IN_HIGH

= 1μF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

Note 1: Package thermal resistances were obtained using the method described in JEDEC specifications. For detailed information

on package thermal considerations refer to www.maxim-ic.com/thermal-tutorial

.

V+ to SGND............................................................-0.3V to +25V

V+ to IN_HIGH...........................................................-19V to +6V

IN_HIGH to SGND ..................................................-0.3V to +19V

IN_HIGH Maximum Input Current .......................................60mA

BYPASS to SGND..................................................-0.3V to +2.5V

GATE to V+.............................................................-0.3V to +12V

GATE to SGND .......................................................-0.3V to +36V

SGND to PGND .....................................................-0.3V to +0.3V

V

L

to SGND..................-0.3V to the Lower of +6V or (V+ + 0.3V)

VDRV to SGND .........................................................-0.3V to +6V

BST1/VDD1, BST2/VDD2, DRAIN_,

PGOOD_ to SGND ..............................................-0.3V to +30V

ON/OFF to SGND ...............................-0.3V to (IN_HIGH + 0.3V)

BST1/VDD1 to SOURCE1,

BST2/VDD2 to SOURCE2......................................-0.3V to +6V

SOURCE_ to SGND................................................-0.6V to +25V

EN_ to SGND............................................................-0.3V to +6V

OSC, FSEL_1, COMP_, SYNC,

FB_ to SGND..............................................-0.3V to (V

L

+ 0.3V)

DL_ to PGND ...........................................-0.3V to (VDRV + 0.3V)

SOURCE1, DRAIN1 Peak Current ..............................5A for 1ms

SOURCE2, DRAIN2 Peak Current ..............................3A for 1ms

V

L

, BYPASS to

SGND Short Circuit ................... Continuous, Internally Limited

Continuous Power Dissipation (T

A

= +70°C)

32-Pin TQFN-EP (derate 34.5mW/°C above +70°C)..2759mW

Package Junction-to-Ambient

Thermal Resistance (θ

JA

) (Note 1).............................29.0°C/W

Package Junction-to-Case

Thermal Resistance (θ

JC

) (Note 1) ..............................1.7°C/W

Operating Temperature Range .........................-40°C to +125°C

Storage Temperature Range ............................-65°C to +150°C

Junction Temperature......................................................+150°C

Lead Temperature (soldering, 10s) ................................+300°C

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

SYSTEM SPECIFICATIONS

Input Voltage Range V+

V+ Operating Supply Current I

V+ Standby Supply Current I

V+STBY

Efficiency η

Q

V+ = IN_HIGH 5.2 19

V

= V+ = IN_HIGH, Figure 6 (Note 3) 4.5 5.5

L

VL unloaded, no switching, V

V

= 0V, PGOOD_ unconnected,

EN_

V+ = V

V

OUT1

V

OUT2

= 300kHz

f

SW

OVERVOLTAGE PROTECTOR

IN_HIGH Clamp Voltage IN_HIGH I

IN_HIGH Clamp Load
Regulation

IN_HIGH Supply Current I

IN_HIGH Standby Supply
Current

V+ to IN_HIGH Overvoltage
Clamp

IN_HIGH

I

IN_HIGHSTBY

V

OV

= 10mA 19 20 21 V

SINK

1mA < I

V

= V

EN_

V

IN_HIGH

V

ON/OFF

unconnected, V

VOV = V+ - V

= 14V

IN_HIGH

= 5V at 1.5A,
= 3.3V at 0.75A,

= 1V 4.2 6.0 mA

FB_

V+ = VL = 5.2V 86

V+ = 12V 85

0.75 1.1 mA

V+ = 16V 85

< 50mA 160 mV

SINK

PGOOD_

= V

ON/OFF

= 0V , V

IN_HIGH

= V

PGOOD_

IN_HIGH

= 0V,

GATE

= 14V

= V + =

= 14V

, I

= -1mA 1.20 1.85 2.50 V

GATE

270 600 μA

79μA

V

%

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND = SGND,
C

BYPASS

= 0.22μF (low ESR), CVL= 4.7μF (ceramic), CV+= 1μF (low ESR), C

IN_HIGH

= 1μF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

IN_HIGH Startup Voltage

GATE Charge Current I

GATE Output Voltage

GATE Turn-Off Pulldown
Current

IN_HIGH

UVLO

GATE_CH

V

GATE

V

IN_HIGH

I

GATE_PD

Rising, ON/OFF = IN_HIGH, GATE rising 3.6 4.1

Falling, ON/OFF = IN_HIGH, GATE falling 3.45

V
V

V+ = V
I

GATE

-

V+ = V
I

GATE

V
V

IN_HIGH

= V+ = 0V

GATE

IN_HIGH

= 1μA

IN_HIGH

= 1μA

IN_HIGH

= 5V

GATE

= V

ON/OFF

= V

= V

= 14V, V

= 14V,

ON/OFF

ON/OFF

ON/OFF

= 4.5V,

= 14V,

= 0V, V+ = 0V,

20 45 80 μA

4.0 5.3 7.5

V

V

9

3.6 mA

STARTUP/VL REGULATOR

VL Undervoltage-Lockout Trip
Level

VL Undervoltage-Lockout
Hysteresis

VL Output Voltage V

VL LDO Short-Circuit Current I

VL LDO Dropout Voltage V

UVLO V

L

VL_SHORT

LDO

falling 3.9 4.1 4.3 V

L

180 mV

I

SOURCE_

V+ = V

I

SOURCE_

= 0 to 40mA, 5.5V ≤ V+ ≤ 19V 5.0 5.2 5.5 V

= 5.2V 130 mA

IN_HIGH

= 40mA, V+ = V

= 4.5V 300 550 mV

IN_HIGH

BYPASS OUTPUT

BYPASS Voltage V

BYPASS Load Regulation ΔV

BYPASS

BYPASS

I

= 0μA 1.98 2.00 2.02 V

BYPASS

0 < I

< 100μA (sourcing) 2 5 mV

BYPASS

SOFT-START/SOFT-STOP

f

Digital Ramp Period Soft­Start/Soft-Stop

Internal 6-bit DAC 2048

SW

Clock

Cycles

Soft-Start/Soft-Stop Steps 64 Steps

VOLTAGE-ERROR AMPLIFIER

FB_ Input Bias Current I

FB_ Input-Voltage Set Point V

FB_ to COMP_
Transconductance

FB_

FB_

g

-40°C ≤ TA ≤ +85°C 0.783 0.8 0.809

-40°C ≤ TA ≤ +125°C 0.785 0.814

M

1.4 2.4 3.4 mS

250 nA

V

INTERNAL MOSFETS

I

= 100mA, BST1/VDD1 to

On-Resistance High-Side
MOSFET Converter 1

R

ON1

SWITCH

V

SOURCE1

I

SWITCH

V

SOURCE1

= 5.2V

= 100mA, BST1/VDD1 to

= 4.5V

195

mΩ

208 355

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND = SGND,
C

BYPASS

= 0.22μF (low ESR), CVL= 4.7μF (ceramic), CV+= 1μF (low ESR), C

IN_HIGH

= 1μF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

I

SWITCH

V

On-Resistance High-Side
MOSFET Converter 2

Minimum Converter 1 Output
Current

Minimum Converter 2 Output
Current

Converter 1/Converter 2
MOSFET DRAIN_ Leakage
Current

R

ON2

I

OUT1

I

OUT2

I

LK12

SOURCE2

I

SWITCH

V

SOURCE2

V

OUT1

V

OUT2

V

= V

EN1

19V, V

= 100mA, BST2/VDD2 to

= 5.2V

= 100mA, BST2/VDD2 to

= 4.5V

280

300 520

= 5V, V+ = 12V (Note 4) 2 A

= 3.3V, V+ = 12V (Note 4) 1 A

= 0V, VDS = 19V, V

EN2

SOURCE_

= 0V

DRAIN_

=

20 μA

mΩ

Internal Weak Low-Side Switch
On-Resistance

R

ONLSSW_ILSSW

INTERNAL SWITCH CURRENT LIMIT

Internal Switch Current-Limit
Converter 1

Internal Switch Current-Limit
Converter 2

SWITCHING FREQUENCY

PWM Maximum Duty Cycle D

Switching Frequency Range f

Switching Frequency f

Switching Frequency Accuracy

SYNC Frequency Range f

SYNC High Threshold V

SYNC Low Threshold V

SYNC Input Leakage I

SYNC Input Minimum Pulse
Width

SYNCH

SYNCL

SYNC_LEAK

t

SYNCIN

Sync to Source 1 Phase Delay SYNC

I

CL1

I

CL2

MAX

SW

SW

SYNC

PHASEROSC

= 30mA 22 Ω

V+ = V
V

BST_/VDD_

V+ = V
V

BST_/VDD_

= 5.2V, VL = VDRV =

IN_HIGH

= 5.2V

= 5.2V, VL = VDRV =

IN_HIGH

= 5.2V

2.8 3.45 4.3 A

1.75 2.10 2.60 A

SYNC = SGND, fSW = 1.25MHz 90 92 100 %

200 2200 kHz

R

= 6.81kΩ, each converter 1.7 1.9 2.1 MHz

OSC

5.6kΩ < R

10kΩ < R

< 10kΩ, 1% 5

OSC

< 62.5kΩ, 1% 7

OSC

Each converter switching frequency is half
of the SYNC input frequency,
FSEL_1 = V

(see the Setting the

L

400 4400 kHz

Switching Frequency section)

2V

0.8 V

2μA

100 ns

= 62.5kΩ 90 Degrees

%

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________ 5

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND = SGND,
C

BYPASS

= 0.22μF (low ESR), CVL= 4.7μF (ceramic), CV+= 1μF (low ESR), C

IN_HIGH

= 1μF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

Note 2: 100% tested at TA= +25°C and TA= +125°C. Specifications at TA= -40°C are guaranteed by design and not production

tested.

Note 3: Operating supply range (V+) is guaranteed by V

L

line regulation test. Connect V+ to IN_HIGH and VLfor 5V operation.

Note 4: Output current is limited by the power dissipation of the package; see the

Power Dissipation

section in the

Applications

Information

section.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INTERNAL DL_ DRIVERS

R

R

DL_ Sink R

DS(ON)

DL_ Source R

DS(ON)

ONDLN

ONDLP

I

= 200mA

SINK

I

SOURCE

= 200mA

1.8

Break-Before-Make Time 50

FSEL_1

FSEL_1 Input High Threshold V

FSEL_1 Input Low Threshold V

FSEL_1 Input Leakage I

FSEL_1_LEAK

IH

IL

2V

ON/OFF

ON/OFF Input High Threshold V

ON/OFF Input Low Threshold V

ON/OFF Input Leakage Current I

ON/OFF_LEAKVON/OFF

IH

IL

= 5V 0.35 2 μA

2V

EN_ INPUTS

EN_ Input High Threshold V

EN_ Input Hysteresis V

EN_ Input Leakage Current I

EN_HYS

EN_LEAK

IH

EN_ rising 1.9 2.0 2.1 V

0.5 V

-1 +1 μA

POWER-GOOD OUTPUT (PGOOD1, PGOOD2)

1

Ω

Ω

ns

0.8 V

2μA

0.8 V

PGOOD_ Threshold V

PGOOD_ Output Voltage V

PGOOD_ Output Leakage
Current

TPGOOD_

PGOOD_ISINK

I

LKPGOOD_

Falling 90 92.5 95 % V

= 3mA 0.4 V

V+ = VL = V
V

PGOOD_

= 23V, V

IN_HIGH

= V

FB_

EN_

= 1V

= 5.2V,

2μA

OUTPUT OVERVOLTAGE PROTECTION

FB_ OVP Threshold Rising V

FB_ OVP Threshold Falling V

OVP_R

OVP_F

107 114 121 % V

112.5 % V

THERMAL PROTECTION

Thermal Shutdown T

Thermal Hysteresis T

SHDN

HYST

Rising +165 °C

20 °C

FB_

FB

FB

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

6 _______________________________________________________________________________________

Typical Operating Characteristics

(V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means that N1 is shorted externally.)

OUTPUT1 EFFICIENCY

vs. LOAD CURRENT

MAX5099 toc01

LOAD CURRENT (A)

OUTPUT1 EFFICIENCY (%)

1.81.61.2 1.40.6 0.8 1.00.4

10

20

30

40

50

60

70

80

90

100

0

0.2 2.0

VIN = 8V

VIN = 14V

VIN = 16V

V

OUT

= 5V

f

SW

= 1.85MHz

OUTPUT2 EFFICIENCY

vs. LOAD CURRENT

MAX5099 toc02

LOAD CURRENT (A)

OUTPUT2 EFFICIENCY (%)

0.90.7 0.80.4 0.5 0.60.3

10

20

30

40

50

60

70

80

90

100

0

0.2 1.0

VIN = 4.5V

VIN = 5.5V

V

OUT

= 3.3V

f

SW

= 1.85MHz

VIN = 14V

VIN = 8V

VIN = 16V

OUTPUT1 EFFICIENCY

vs. LOAD CURRENT

MAX5099 toc03

LOAD CURRENT (A)

OUTPUT1 EFFICIENCY (%)

1.81.61.2 1.40.6 0.8 1.00.4

10

20

30

40

50

60

70

80

90

100

0

0.2 2.0

VIN = 8V

VIN = 14V VIN = 16V

V

OUT

= 5V

f

SW

= 300kHz

L1 = 18μH

OUTPUT2 EFFICIENCY

vs. LOAD CURRENT

MAX5099 toc04

LOAD CURRENT (A)

OUTPUT2 EFFICIENCY (%)

0.90.80.6 0.70.4 0.50.30.2 1.0

10

20

30

40

50

60

70

80

90

100

0

VIN = 16V

VIN = 8V

VIN = 14V

VIN = 5.5V

VIN = 4.5V

V

OUT

= 3.3V

f

SW

= 300kHz

L2 = 27μH

OUTPUT1 VOLTAGE
vs. LOAD CURRENT

MAX5099 toc05

LOAD CURRENT (A)

OUTPUT1 VOLTAGE (V)

1.81.61.41.21.00.80.60.4

4.92

4.94

4.96

4.98

5.00

4.90

0.2 2.0

VIN = 8V

VIN = 14V

VIN = 16V

V

OUT

= 5V

f

SW

= 1.85MHz

OUTPUT2 VOLTAGE

vs. LOAD CURRENT

MAX5099 toc06

LOAD CURRENT (A)

OUTPUT2 VOLTAGE (V)

0.90.80.70.60.50.40.3

3.25

3.24

3.26

3.27

3.28

3.29

3.30

3.23

0.2 1.0

VIN = 4.5V

VIN = 16V

VIN = 14V

VIN = 8V

VIN = 5.5V

V

OUT

= 3.3V

f

SW

= 1.85MHz

VL OUTPUT VOLTAGE

vs. CONVERTER SWITCHING FREQUENCY

MAX5099 toc07

CONVERTER SWITCHING FREQUENCY (kHz)

V

L

OUTPUT VOLTAGE (V)

1600 19001300700 1000

4.2

4.4

4.6

4.8

5.0

5.2

5.4

4.0
400 2200

VIN = 4.5V

VIN = 5.5V

VIN = 8V

VIN = 19V

VIN = 5V

BOTH CONVERTERS SWITCHING
FSEL_1 = V

L

EACH CONVERTER SWITCHING

FREQUENCY vs. R

OSC

MAX5099 toc08

R

OSC

(kΩ)

SWITCHING FREQUENCY (MHz)

604020

1

0

80

10

0.1

CONVERTER 1, CONVERTER 2

CONVERTER 1

FSEL_1 = VL,
FSEL_1 = GND,

EACH CONVERTER SWITCHING

FREQUENCY vs. TEMPERATURE

MAX5099 toc09

TEMPERATURE (°C)

SWITCHING FREQUENCY (MHz)

-5 30 65 100

1

10

0.1

-40 135

0.3MHz

0.6MHz

1.25MHz

1.85MHz

2.2MHz

FSEL_1 = V

L

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________

7

Typical Operating Characteristics (continued)

(V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means that N1 is shorted externally.)

LINE-TRANSIENT RESPONSE

(BUCK CONVERTER)

CONVERTER 2

LOAD-TRANSIENT RESPONSE

0V

1ms/div

MAX5099 toc10

MAX5099 toc12

CONVERTER 1

LOAD-TRANSIENT RESPONSE

100μs/div

SOFT-START/SOFT-STOP FROM EN1

MAX5099 toc11

MAX5099 toc13

fSW = 1.85MHz

= 5.0V

V

OUT1

AC-COUPLED
200mV/div

I

OUT1

1A/div

0A

EN1
5V/div
0V

= 5V/2A

V

OUT1

5V/div
0V

P

GOOD1

5V/div
0V

V

IN

5V/div

0V

= 5.0V/1.5A

V

OUT1

AC-COUPLED
200mV/div

= 3.3V/0.75A

V

OUT2

AC-COUPLED
200mV/div

= 3.3V

V

OUT2

AC-COUPLED
200mV/div

I

OUT2

500mA/div

0A

100μs/div

1ms/div

OUT-OF-PHASE OPERATION

SOFT-START FROM ON/OFF

0V

0V

0V

0V

2ms/div

MAX5099 toc14

ON/OFF
5V/div

= EN1 = EN2

V

L

5V/div
GATE
10V/div
V+
10V/div

V

= 5V/2A

OUT1

5V/div

0V

0V

0V

0V

(FSEL_1 = V

200ns/div

)

L

MAX5099 toc15

SOURCE1
10V/div

SOURCE2
10V/div

DL1
10V/div
DL2
10V/div

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

8 _______________________________________________________________________________________

Typical Operating Characteristics (continued)

(V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means that N1 is shorted externally.)

FB_ VOLTAGE

vs. TEMPERATURE

MAX5099 toc20

TEMPERATURE (°C)

FB_ VOLTAGE (V)

10065-5 30

0.790

0.795

0.800

0.805

0.815

0.810

0.820

0.825

0.785

-40 135

VL = V+ = V

IN_HIGH

= 5.5V

BYPASS VOLTAGE

vs. TEMPERATURE

MAX5099 toc21

TEMPERATURE (°C)

BYPASS VOLTAGE (V)

10065-5 30

1.994

1.996

1.998

2.000

2.002

2.006

2.004

2.008

2.010

1.990

1.992

-40 135

VL = V+ = V

IN_HIGH

= 5.5V

EXTERNAL SYNCHRONIZATION

(FSEL_1 = SGND )

MAX5099 toc18

200ns/div

SOURCE2
10V/div

SOURCE1
10V/div

0V

SYNC
5V/div
0V

0V

OVP BEHAVIOR

MAX5099 toc19

1ms/div

PGOOD2
10V/div

VOUT1
10V/div

VOUT2
10V/div

GATE
10V/div

0V

0V

0V

V+
10V/div

0V

0V

EXTERNAL OVERVOLTAGE REMOVED

OUT-OF-PHASE OPERATION

(FSEL_1 = SGND)

MAX5099 toc16

200ns/div

SOURCE2
10V/div

DL1
10V/div
DL2
10V/div

SOURCE1
10V/div

0V

0V

0V

0V

EXTERNAL SYNCHRONIZATION

(FSEL_1 = V

L

)

MAX5099 toc17

200ns/div

SOURCE2
10V/div

SOURCE1
10V/div

0V

SYNC
5V/div
0V

0V

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________

9

Typical Operating Characteristics (continued)

(V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means that N1 is shorted externally.)

2.000

1.998

1.996

1.994

BYPASS VOLTAGE (V)

1.992

1.990

BYPASS VOLTAGE

vs. BYPASS CURRENT

TA = +85°C

TA = -40°C

0100705010 30 90

V+ SWITCHING SUPPLY CURRENT

vs. SWITCHING FREQUENCY

100

V+ = IN_HIGH = ON/OFF

80

60

40

TA = +125°C

TA = +25°C

BYPASS CURRENT (μA)

TA = +135°C

TA = +25°C

TA = +135°C

806020 40

MAX5099 toc22

MAX5099 toc24

SOURCE1, I

SOURCE1

200ns/div

V+ STANDBY SUPPLY CURRENT

vs. TEMPERATURE

4

V+ = IN_HIGH = ON/OFF
EN1 = EN2 = SGND

3

fSW = 1.85MHz

2

, DL1, I

IDL1

MAX5099 toc23

SOURCE1
10V/div
0V

DL1
10V/div
OV

I

SOURCE1

1A/div

0A

MAX5099 toc25

20

V+ SWITCHING SUPPLY CURRENT (mA)

0

300 2200

TA = -40°C

SWITCHING FREQUENCY (kHz)

IN_HIGH SHUTDOWN CURRENT

vs. TEMPERATURE

20

ON/OFF = SGND

16

IN_HIGH = 14V

12

IN_HIGH = 8V

8

4

IN_HIGH SHUTDOWN CURRENT (μA)

0

-50 150

TEMPERATURE (°C)

IN_HIGH = 16V

182014401060680

MAX5099 toc26

100500

1

V+ STANDBY SUPPLY CURRENT (mA)

0

-50 150

fSW = 300kHz

TEMPERATURE (°C)

IN_HIGH STANDBY CURRENT

vs. TEMPERATURE

ON/OFF = IN_HIGH

145

EN1 = EN2 = SGND

135

125

115

105

IN_HIGH = 8V

95

IN_HIGH STANDBY CURRENT (μA)

85

75

-50 150

IN_HIGH = 16V

IN_HIGH = 14V

TEMPERATURE (°C)

100500

100500

MAX5099 toc27

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

10 ______________________________________________________________________________________

Typical Operating Characteristics (continued)

(V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means that N1 is shorted externally.)

V+ TO IN_HIGH CLAMP VOLTAGE

vs. GATE SINK CURRENT

MAX5099 toc29

GATE SINK CURRENT (mA)

V+ TO IN_HIGH CLAMP VOLTAGE (V)

8642

1

2

3

4

5

0

010

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

(V

GATE

- V) vs. V

IN_HIGH

MAX5099 toc30

V

IN_HIGH

(V)

(V

GATE

- V) (V)

15.512.08.5

2

4

6

8

10

0

5.0 19.0

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

ON/OFF = IN_HIGH

SYSTEM TURN-ON FROM BATTERY

MAX5099 toc31

10ms/div

V

L

10V/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

V

IN

10V/div

0V

0V

0V

0V

0V

SYSTEM TURN-OFF FROM BATTERY

MAX5099 toc32

10ms/div

V

L

10V/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

V

IN

10V/div

0V

0V

0V

0V

0V

SYSTEM LOAD DUMP

MAX5099 toc33

100ms/div

V

OUT1

AC-COUPLED
100mV/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

0V

0V

0V

0V
0V

V

IN

50V/div

IN_HIGH CLAMP VOLTAGE

vs. CLAMP CURRENT

MAX5099 toc28

CLAMP CURRENT (mA)

IN_HIGH CLAMP VOLTAGE (V)

40302010

20.0

20.1

20.2

20.3

19.9
050

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 11

Pin Description

PIN NAME FUNCTION

1, 32 SOURCE2

2, 3 DRAIN2

4 PGOOD2

5 EN2 Converter 2 Active-High Enable Input. Connect to VL for always-on operation.

6 FB2

7 COMP2 Converter 2 Internal Transconductance Amplifier Output. See the Compensation section.

8 OSC

9 SYNC

Converter 2 Internal MOSFET Source Connection. For buck converter operation, connect SOURCE2 to the
switched side of the inductor. For boost operation, connect SOURCE2 to PGND (Figure 5).

Converter 2 Internal MOSFET Drain Connection. For buck converter operation, use the MOSFET as a high­side switch and connect DRAIN2 to the DC-DC converters supply input rail. For boost converter operation,
use the MOSFET as a low-side switch and connect DRAIN2 to the inductor and diode junction (Figure 5).

Converter Open-Drain Power-Good Output. PGOOD2 goes low when converter 2’s output falls below 92.5%
of its set regulation voltage. Use PGOOD2 and EN1 to sequence the converters.

Converter 2 Feedback Input. Connect FB2 to a resistive divider between converter 2’s output and SGND to
adjust the output voltage. To set the output voltage below 0.8V, connect FB2 to a resistive voltage-divider
from BYPASS to regulator 2’s output (Figure 2). See the Setting the Output Voltage section.

Oscillator Frequency Set Input. Connect a resistor from OSC to SGND (R
(see the Setting the Switching Frequency section). Set R
input frequency when using external synchronization. R
connected to the SYNC input. See the Synchronization (SYNC) section.

External Clock Synchronization Input. Connect SYNC to a 400kHz to 4400kHz clock to synchronize the
switching frequency with the system clock. Each converter frequency is 1/2
SYNC (FSEL_1 = V
SYNC frequency. Connect SYNC to SGND when not used.

). For FSEL_1 = SGND, the switching frequency of converter 1 becomes 1/4 of the

L

for an oscillator frequency equal to the SYNC

OSC

is still required when an external clock is

OSC

) to set the switching frequency

OSC

of the frequency applied to

10 GATE

11 ON/OFF

12 IN_HIGH

13 V+

14 V

L

Gate Drive Output. Connect to the gate of the external n-channel load-dump protection MOSFET. GATE =
IN_HIGH + 9V (typ) with IN_HIGH = 12V. GATE pulls to IN_HIGH by an internal n-channel MOSFET when V+
raises 2V above IN_HIGH. Leave GATE unconnected if the load-dump protection is not used (MOSFET not
installed).

n-Channel Switch Enable Input. Drive ON/OFF high for normal operation. Drive ON/OFF low to turn off the
external n-channel load-dump protection MOSFET and reduce the supply current to 7μA (typ). When
ON/OFF is driven low, both DC-DC converters are disabled and the PGOOD_ outputs are driven low.
Connect to V+ if the external load-dump protection is not used (MOSFET not installed).

Startup Input. IN_HIGH is protected by internally clamping to 21V (max). Connect a resistor (4kΩ max) from
IN_HIGH to the drain of the protection switch. Bypass IN_HIGH with a 4.7μF electrolytic or 1μF minimum
ceramic capacitor. Connect to V+ if the external load-dump protection is not used (MOSFET not installed).

Input Supply Voltage. V+ can range from 5.2V to 19V. Connect V+, IN_HIGH, and V

5.5V input operation. Bypass V+ to SGND with a 1μF minimum ceramic capacitor.

Internal Regulator Output. The VL regulator is used to supply the drive current at input VDRV. When driving
VDRV, use an RC lowpass filter to decouple switching noise from VDRV to the V
Application Circuit). Bypass V

to SGND with a 4.7μF minimum ceramic capacitor.

L

together for 4.5V to

L

regulator (see the Typical

L

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

12 ______________________________________________________________________________________

Pin Description (continued)

PIN NAME FUNCTION

15 SGND

16 BYPASS Reference Output Bypass Connection. Bypass to SGND with a 0.22μF or greater ceramic capacitor.

17 FSEL_1

18 COMP1 Converter 1 Internal Transconductance Amplifier Output. See the Compensation section.

19 FB1

20 EN1 Converter 1 Active-High Enable Input. Connect to VL for an always-on operation.

21 PGOOD1

22, 23 DRAIN1

Signal Ground. Connect SGND to exposed pad and to the board signal ground plane. Connect the board
signal ground and power ground planes together at a single point.

Converter 1 Frequency Select Input. Connect FSEL_1 to V
to reduce converter 1’s switching frequency to 1/2 of converter 2’s switching frequency (converter 1
switching frequency is 1/4 the SYNC frequency). Do not leave FSEL_1 unconnected.

Converter 1 Feedback Input. Connect FB1 to a resistive divider between converter 1’s output and SGND to
adjust the output voltage. To set the output voltage below 0.8V, connect FB1 to a resistive voltage-divider
from BYPASS to regulator 1’s output (Figure 2). See the Setting the Output Voltage section.

Converter 1 Power-Good Output. Open-drain output goes low when converter 1’s output falls below 92.5%
of its set regulation voltage. Use PGOOD1 and EN2 to sequence the converters (converter 1 starts first).

Converter 1 Internal MOSFET Drain Connection. For buck converter operation, use the MOSFET as a high­side switch and connect DRAIN1 to the DC-DC converters supply input rail. For boost converter operation,
use the MOSFET as a low-side switch and connect DRAIN1 to the inductor and diode junction (Figure 5).

for normal operation. Connect FSEL_1 to SGND

L

24, 25 SOURCE1

26 BST1/VDD1

27 VDRV

28 DL1 Converter 1 Low-Side Synchronous-Rectifier Gate Driver Output

29 PGND Power Ground. Connect to the board power ground plane.

30 DL2 Converter 2 Low-Side Synchronous-Rectifier Gate Driver Output

31 BST2/VDD2

—EP

Converter 1 Internal MOSFET Source Connection. For buck operation, connect SOURCE1 to the switched
side of the inductor. For boost operation, connect SOURCE1 to PGND (Figure 5).

Converter 1 Bootstrap Flying-Capacitor Connection. For buck converter operation, connect BST1/VDD1 to a

0.1μF ceramic capacitor and diode according to the Typical Application Circuit. For boost converter
operation, driver bypass capacitor connection. Connect to VDRV and bypass with a 0.1μF ceramic
capacitor to PGND (Figure 5).

Low-Side Driver Supply Input. Connect VDRV to VL through an RC filter to bypass switching noise to the
internal VL regulator. For buck converter operation, connect anode terminals of external bootstrap diodes to
VDRV. For boost converter operation, connect VDRV to BST1/VDD1 and BST2/VDD2.
Bypass with a minimum 2.2μF ceramic capacitor to PGND (see the Typical Application Circuit). Do not
connect to an external supply.

Converter 2 Bootstrap Flying-Capacitor Connection. For buck converter operation, connect BST2/VDD2 to a

0.1μF ceramic capacitor and diode according to the Typical Application Circuit. For boost converter
operation, driver bypass capacitor connection. Connect to VDRV and bypass with a 0.1μF ceramic
capacitor from BST2/VDD2 to PGND (Figure 5).

Exposed Pad. Connect EP to SGND. For enhanced thermal dissipation, connect EP to a copper area as
large as possible. Do not use EP as the sole ground connection.

MAX5099

Functional Diagram

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 13

IN_HIGH

ON/OFF

BYPASS

FSEL_1

V+

MAX5099

1.8V

GATE

V

L

BST1/VDD1

DRAIN1

SOURCE1

VDRV

DL1

PGOOD1

CLK1

FREQUENCY

DIVIDER

CONVERTER 1

OSCILLATOR

MAX DUTY-CYCLE

CONTROL

FREQUENCY
CONTROL

PWM

COMPARATOR

CHARGE

PUMP

20V SHUNT
REGULATOR

S

R

Q

Q

STARTUP CIRCUIT/

PROTECTION CIRCUIT/

CHARGE PUMP

FSW/4

OVERVOLTAGE

CURRENT

LIMIT

LDO

VL

EN1

SYNC

OSC

EN2

TRANSCONDUCTANCE

DIGITAL

SOFT-START

MAIN

OSCILLATOR

CLK2

OVERVOLTAGE

ERROR AMPLIFIER

CONVERTER 2

0.8V

0.2V

0.9V

0.74V

VLVDRV

PGND

FB1

COMP1

SGND

PGOOD2
DRAIN2

BST2/VDD2
SOURCE2
FB2

COMP2
PGND

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

14 ______________________________________________________________________________________

Detailed Description

PWM Controller

The MAX5099 dual DC-DC converters use a pulse-width­modulation (PWM) voltage-mode control scheme. On
each converter the device includes one integrated n­channel MOSFET switch and requires an external low-for­ward-drop Schottky diode for output rectification. The
controller generates the clock signal by dividing down
the internal oscillator (f

OSC

) or the SYNC input when dri­ven by an external clock; therefore, each controller’s
switching frequency equals half the oscillator frequency
(fSW= f

OSC

/2) or half of the SYNC input frequency (fSW=

f

SYNC

/2). An internal transconductance error amplifier
produces an integrated error voltage at COMP_, provid­ing high DC accuracy. The voltage at COMP_ sets
the duty cycle using a PWM comparator and a ramp
generator. At each rising edge of the clock, converter
1’s MOSFET switch turns on and remains on until either
the appropriate or maximum duty cycle is reached, or the
maximum current limit for the switch is reached.
Converter 2 operates 180° out-of-phase, so its MOSFET
switch turns on at each falling edge of the clock.

In the case of buck operation (see the

Typical Application

Circuit

), the internal MOSFET is used in high-side config­uration. During each MOSFET’s on-time, the associated
inductor current ramps up. During the second half of the
switching cycle, the high-side MOSFET turns off and for­ward biases the Schottky rectifier. During this time, the
SOURCE_ voltage is clamped to a diode drop (VD) below
ground. A low-forward-voltage-drop (0.4V) Schottky
diode must be used to ensure the SOURCE_ voltage
does not go below -0.6V absolute max. The inductor
releases the stored energy as its current ramps down,
and provides current to the output. The bootstrap capaci­tor is also recharged when the SOURCE_ voltage goes
low during the high-side MOSFET off-time. The maximum
duty-cycle limits ensure proper bootstrap charging at
startup or low input voltages. The circuit goes in discon­tinuous conduction mode operation at light load, when
the inductor current completely discharges before the
next cycle commences. Under overload conditions, when
the inductor current exceeds the peak current limit of the
respective switch, the high-side MOSFET turns off quickly
and waits until the next clock cycle.

Synchronous-Rectifier Output

The MAX5099 is intended mostly for synchronous buck
operation with an external synchronous-rectifier MOSFET.
During the internal high-side MOSFET on-time, the induc­tor current ramps up. When the high-side MOSFET turns
off, the inductor reverses polarity and forward biases
the Schottky rectifier in parallel with the low-side external

synchronous MOSFET. The SOURCE_ voltage is
clamped to 0.5V below ground until the adaptive break­before-make time (t

BBM

) of 25ns is over. After t

BBM

, the
synchronous-rectifier MOSFET turns on, thus bypassing
the Schottky rectifier and reducing the conduction loss
during the inductor freewheeling time. The synchronous­rectifier MOSFET keeps the circuit in continuous conduc­tion mode operation even at light load because the
inductor current is allowed to go negative.

The MAX5099, with the synchronous-rectifier driver out­put (DL_), has an adaptive break-before-make circuit
to avoid cross-conduction between the internal power
MOSFET and the external synchronous-rectifier MOSFET.
When the synchronous-rectifier MOSFET is turning off,
the internal high-side power MOSFET is kept off until V

DL

falls below 0.97V. Similarly, DL_ does not go high until the
internal power MOSFET gate voltage falls below 1.24V.

Load-Dump Protection

Most automotive applications are powered by a multi­cell, 12V lead-acid battery with a voltage from 9V to
16V (depending on load current, charging status, tem­perature, battery age, etc.). The battery voltage is dis­tributed throughout the automobile and is locally
regulated down to voltages required by the different
system modules. Load dump occurs when the alterna­tor is charging the battery and the battery becomes
disconnected. Power in the alternator inductance flows
into the distributed power system and elevates the volt­age seen at each module. The voltage spikes have rise
times typically greater than 5ms and decays within sev­eral hundred milliseconds but can extend out to 1s or
more depending on the characteristics of the charging
system. These transients are capable of destroying
sensitive electronic equipment on the first fault event.

During load dump, the MAX5099 provides the ability to
clamp the input-voltage rail of the internal DC-DC con­verters to a safe level, while preventing power disconti­nuity at the DC-DC converters’ outputs.

The load-dump protection circuit utilizes an internal
charge pump to drive the gate of an external n-channel
MOSFET. This series-protection MOSFET absorbs the
load-dump overvoltage transient and operates in satu­ration over the normal battery range to minimize power
dissipation. During load dump, the gate voltage of the
protection MOSFET is regulated to prevent the source
terminal from exceeding 19V.

The DC-DC converters are powered from the source
terminal of the load-dump protection MOSFET, so that
their input voltage is limited during load dump and can
operate normally.

MAX5099

ON/OFF

The MAX5099 provide an input (ON/OFF) to turn on and
off the external load-dump protection MOSFET. Drive
ON/OFF high for normal operation. Drive ON/OFF low to
turn off the external n-channel load-dump protection
MOSFET and reduce the supply current to 7μA (typ).
When ON/OFF is driven low, both converters are also
turned off, and the PGOOD_ outputs are driven, low. V+
will be self-discharged through the converters’ output
currents and the IC supply current.

Internal Oscillator/

Out-of-Phase Operation

The internal oscillator generates the 180° out-of-phase
clock signal required by each regulator. The switching
frequency of each converter (fSW) is programmable
from 200kHz to 2.2MHz using a single 1% resistor at
R

OSC

. See the

Setting the Switching Frequency

section.

With dual-synchronized out-of-phase operation, the
MAX5099’s internal MOSFETs turn on 180° out-of­phase. The instantaneous input current peaks of both
regulators do not overlap, resulting in reduced RMS rip­ple current and input-voltage ripple. This reduces the
required input capacitor ripple current rating, allows for
fewer or less expensive capacitors, and reduces
shielding requirements for EMI.

Synchronization (SYNC)

The main oscillator can be synchronized to the system
clock by applying an external clock (f

SYNC

) at SYNC.

The f

SYNC

frequency must be twice the required operat­ing frequency of an individual converter. Use a TTL logic
signal for the external clock with at least a 100ns pulse
width. R

OSC

is still required when using external syn­chronization. Program the internal oscillator frequency to
have fSW= 1/2 f

SYNC

. The device is properly synchro-

nized if the SYNC frequency f

SYNC

varies within the

range ±20%.

Short SYNC to SGND if unused.

Input Voltage (V+)/

Internal Linear Regulator (V

L

)

All internal control circuitry operates from an internally
regulated nominal voltage of 5.2V (VL). At higher input
voltages (V+) of 5.2V to 19V, VLis regulated to 5.2V. At

5.2V or below, the internal linear regulator operates in
dropout mode, where VLfollows V+. Depending on the
load on VL, the dropout voltage can be high enough to
reduce VLbelow the undervoltage-lockout (UVLO)
threshold. Do not use VLto power external circuitry.

For input voltages less than 5.5V, connect V+ and V

L

together. The load on VLis proportional to the switching
frequency of converter 1 and converter 2. See the V

L

Output Voltage vs. Converter Switching Frequency
graph in the

Typical Operating Characteristics

. For

input voltage ranges higher than 5.5V, disconnect V

L

from V+.

Bypass V+ to SGND with a 1μF or greater ceramic
capacitor placed close to the MAX5099. Bypass VLwith
a low-ESR 4.7μF ceramic capacitor to SGND.

Undervoltage Lockout/

Soft-Start/Soft-Stop

The MAX5099 includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter
turn-on and monotonic rise of the output voltage. The
falling UVLO threshold is internally set to 4.1V (typ) with
180mV hysteresis. Hysteresis at UVLO eliminates “chat­tering” during startup. When VLdrops below UVLO, the
internal MOSFET switches are turned off.

The MAX5099 digital soft-start reduces input inrush
currents and glitches at the input during turn-on. When
UVLO is cleared and EN_ is high, digital soft-start slow­ly ramps up the internal reference voltage in 64 steps.
The total soft-start period is 4096 internal oscillator
switching cycles.

Driving EN_ low initiates digital soft-stop that slowly
ramps down the internal reference voltage in 64 steps.
The total soft-stop period is equal to the soft-start period.

To calculate the soft-start/soft-stop period, use the fol­lowing equation:

where f

OSC

is the internal oscillator and f

OSC

is twice

each converter’s switching frequency (FSEL_1 = VL).

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 15

tms

()

SS

4096

=

f kHz

OSC

()

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

16 ______________________________________________________________________________________

Enable (EN1, EN2)

The MAX5099 dual converter provides separate enable
inputs, EN1 and EN2, to individually control or sequence
the output voltages. These active-high enable inputs are
TTL compatible. Driving EN_ high initiates soft-start of the
converter, and PGOOD_ goes logic-high when the con­verter output voltage reaches the V

TPGOOD_

threshold.
Driving EN_ low initiates a soft-stop of the converter. Use
EN1, EN2, and PGOOD1 for sequencing (see Figure 1).
Connect PGOOD1 to EN2 to make sure converter 1’s out­put is within regulation before converter 2 starts. Add an
RC network from VLto EN1 and EN2 to delay the individ­ual converter. Sequencing reduces input inrush current
and possible chattering. Connect EN_ to VLfor always-on
operation.

PGOOD_

Converter 1 and converter 2 include power-good flags,
PGOOD1 and PGOOD2, respectively. Since PGOOD_
is an open-drain output and can sink 3mA while provid­ing the TTL logic-low signal, pull PGOOD_ to a logic
voltage to provide a logic-level output. PGOOD1 goes
low when converter 1’s feedback (FB_) drops to 92.5%
(V

TPGOOD_

) of its nominal set point. The same is true
for converter 2. Connect PGOOD_ to SGND or leave
unconnected, if not used.

Current Limit

The internal high-side MOSFET switch current of each
converter is monitored during its on-time. When the
peak switch current crosses the current-limit threshold

of 3.45A (typ) and 2.1A (typ) for converter 1 and con­verter 2, respectively, the on-cycle is terminated imme­diately and the inductor is allowed to discharge. The
MOSFET switch is turned on at the next clock pulse ini­tiating a new clock cycle.

In deep overload or short-circuit conditions when V

FB

drops below 0.2V, the switching frequency is reduced to
1/4 x fSWto provide sufficient time for the inductor to dis­charge. During overload conditions, if the voltage across
the inductor is not high enough to allow for the inductor
current to properly discharge, current runaway may
occur. Current runaway can destroy the device in spite of
internal thermal-overload protection. Reducing the
switching frequency during overload conditions prevents
current runaway.

Output Overvoltage Protection

The MAX5099 outputs are protected from output volt­age overshoots due to input transients and shorting the
output to a high voltage. When the output voltage rises
over the overvoltage threshold, 114% (typ) nominal FB,
the overvoltage condition is triggered. When the over­voltage condition is triggered on either channel, both
converters are immediately turned off, 20Ω pulldown
switches from SOURCE_ to PGND are turned on to help
the output-voltage discharge, and the gate of the load­dump protection external MOSFET is pulled low. The
device restarts as soon as both converter outputs dis­charge, bringing both FB_ input voltages below 12.5%
of their nominal set points.

Figure 1. Power-Supply Sequencing Configurations

V

L

VLV+

OUTPUT2 OUTPUT1

V

SEQUENCING—OUTPUT 2 DELAYED WITH RESPECT TO OUTPUT 1. R1/C1 AND R2/C2 ARE SIZED FOR REQUIRED SEQUENCING.

N

L

DRAIN2

SOURCE2

DL2

MAX5099

DRAIN1

SOURCE1

DL1

FB1FB2

EN1EN2

PGOOD1

N

V

L

V

IN

V

L

VLV+

OUTPUT2

DRAIN2

SOURCE2

N

DL2

V

L

MAX5099

DRAIN1

SOURCE1

DL1

FB1FB2

EN1EN2

N

R1R2

C1C2

V

L

V

IN

OUTPUT1

MAX5099

Thermal-Overload Protection

During continuous short circuit or overload at the output,
the power dissipation in the IC can exceed its limit. The
MAX5099 provides thermal shutdown protection with
temperature hysteresis. Internal thermal shutdown is
provided to avoid irreversible damage to the device.
When the die temperature exceeds +165°C (typ), an on­chip thermal sensor shuts down the device, forcing the
internal switches to turn off, allowing the IC to cool. The
thermal sensor turns the part on again with soft-start
after the junction temperature cools by +20°C. During
thermal shutdown, both regulators shut down, PGOOD_
goes low, and soft-start resets. The internal 20V zener
clamp from IN_HIGH to SGND is not turned off during
thermal shutdown because this clamping action must
always be active.

Applications Information

Setting the Switching Frequency

The controller generates the clock signal by dividing
down the internal oscillator f

OSC

or the SYNC input sig­nal when driven by an external oscillator. The switching
frequency equals half the internal oscillator frequency
(fSW= f

OSC

/2). The internal oscillator frequency is set

by a resistor (R

OSC

) connected from OSC to SGND. To

find R

OSC

for each converter switching frequency fSW,

use the formulas:

A rising clock edge on SYNC is interpreted as a syn­chronization input. If the SYNC signal is lost, the inter­nal oscillator takes control of the switching rate,
returning the switching frequency to that set by R

OSC

.

When an external synchronization signal is used, R

OSC

must be selected such that fSW= 1/2 f

SYNC

.

Buck Converter

Effective Input Voltage Range

Although the MAX5099 converter operates from input
supplies ranging from 5.2V to 19V, the input voltage
range can be effectively limited by the MAX5099 duty­cycle limitations for a given output voltage. The maximum

input voltage is limited by the minimum on-time
(t

ON(MIN)

):

where t

ON(MIN)

is 100ns. The minimum input voltage is

limited by the maximum duty cycle (D

MAX

= 0.92):

where V

DROP1

is the total parasitic voltage drops in the
inductor discharge path, which includes the forward
voltage drop (VDS) of the low-side n-channel MOSFET,
the series resistance of the inductor, and the PCB resis­tance. V

DROP2

is the total resistance in the charging
path that includes the on-resistance of the high-side
switch, the series resistance of the inductor, and the
PCB resistance.

Setting the Output Voltage

For 0.8V or greater output voltages, connect a voltage­divider from OUT_ to FB_ to SGND (Figure 2). Select
RB (FB_ to SGND resistor) to between 1kΩ and 20kΩ.
Calculate RA(OUT_ to FB_ resistor) with the following
equation:

where V

FB_

= 0.8V (see the

Electrical Characteristics

table).

For output voltages below 0.8V, set the MAX5099 out­put voltage by connecting a voltage-divider from OUT_
to FB_ to BYPASS (Figure 2). Select RC (FB_ to
BYPASS resistor) higher than a 50kΩ range. Calculate
RAwith the following equation:

where V

FB_

= 0.8V, V

BYPASS

= 2V (see the

Electrical

Characteristics

table), and V

OUT_

can range from 0V to

V

FB_

.

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 17

10 721

Rk

Rk

()

OSC

()

OSC

Ω

=

Ω

=

.

f MHz

()

SW

12 184

.

f MHz

()

SW

125

.

f MHz

≥

()

SW

0 920

.

125

f MHz

()

0 973

.

.

<

SW

V

IN MIN

V

V

IN MAX

()

≤

OUT

tf

ON MIN SW

×

()

⎡

VV

+

OUT DROP

=

⎢

D

⎣

MAX

⎤

1

VV

+ −

⎥
⎦

DROP DROP()

211

RR

AB

⎡

⎛

V

OUT

⎢

=

⎜

V

⎢

⎝

FB

⎣

⎤

⎞

_

⎥

−

1

⎟

⎥

⎠

_

⎦

RR

AC

⎡

VV

−

FB OUT

__

⎢

=

VV

⎢

⎣

BYPA SS FB

⎤
⎥

−

⎥

⎦

_

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

18 ______________________________________________________________________________________

Inductor Selection

Three key inductor parameters must be specified for
operation with the MAX5099: inductance value (L),
peak inductor current (IL), and inductor saturation cur­rent (I

SAT

). The minimum required inductance is a func­tion of operating frequency, input-to-output voltage
differential, and the peak-to-peak inductor current (ΔIL).
A good compromise is to choose ΔILequal to 30% of
the full load current. To calculate the inductance, use
the following equation:

where VINand V

OUT

are typical values (so that efficien­cy is optimum for typical conditions). The switching fre­quency is set by R

OSC

(see the

Setting the Switching

Frequency

section). The peak-to-peak inductor current,
which reflects the peak-to-peak output ripple, is worse
at the maximum input voltage. See the

Output

Capacitor

section to verify that the worst-case output
ripple is acceptable. The inductor saturation current is
also important to avoid runaway current during output
overload and continuous short circuit. Select the I

SAT

to
be higher than the maximum peak current limits of 4.3A
and 2.6A for converter 1 and converter 2.

Input Capacitor

The discontinuous input current waveform of the buck
converter causes large ripple currents at the input. The
switching frequency, peak inductor current, and allow­able peak-to-peak voltage ripple dictate the input

capacitance requirement. Note that the two converters
of the MAX5099 run 180° out-of-phase, thereby effec­tively doubling the switching frequency at the input.

The input ripple waveform would be unsymmetrical due
to the difference in load current and duty cycle between
converter 1 and converter 2. The worst-case mismatch
is when one converter is at full load while the other is at
no load or in shutdown. The input ripple is comprised of
ΔV

Q

(caused by the capacitor discharge) and ΔV

ESR

(caused by the ESR of the capacitor). Use ceramic
capacitors with high ripple-current capability at the input
connected between DRAIN_ and PGND. Assume the
contribution from the ESR and capacitor discharge
equal to 50%. Calculate the input capacitance and ESR
required for a specified ripple using the following equa­tions:

where

and

where

where I

OUT

is the maximum output current from either
converter 1 or converter 2, and D is the duty cycle for
that converter. fSWis the frequency of each individual
converter. For example, at VIN= 12V, V

OUT

= 3.3V at

I

OUT

= 2A, and with L = 3.3μH, the ESR and input
capacitance are calculated for a peak-to-peak input rip­ple of 100mV or less, yielding an ESR and capacitance
value of 20mΩ and 6.8μF for 1.25MHz frequency. At low
input voltages, also add one electrolytic bulk capacitor
of at least 100μF on the converters’ input voltage rail.
This capacitor acts as an energy reservoir to avoid pos­sible undershoot below the undervoltage-lockout thresh­old during power-on and transient loading.

Figure 2. Adjustable Output Voltage

V

SOURCE_

FB_

MAX5099

V

≥ 0.8V

OUT_

OUT_

R

A

R

B

BYPASS

MAX5099

SOURCE_

FB_

R

C

R

A

V

< 0.8V

OUT_

VVV

()

OUT IN OUT

L

=

××−Δ

Vf I

IN SW L

V

OUT_

ESR

IN

V

Δ

ESR

=

I

OUT

I

Δ

L

+

2

−

VV V

()

ΔI

IN OUT OUT

=

L

Vf L

IN SW

×

××

IDD

OUT

C

=

IN

Δ

−1

×

()

×

Vf

QSW

V

OUT

D

=

V

IN

MAX5099

Output Capacitor

The allowable output ripple voltage and the maximum
deviation of the output voltage during step load cur­rents determine the output capacitance and its ESR.
The output ripple is comprised of ΔV

Q

(caused by the

capacitor discharge) and ΔV

ESR

(caused by the ESR of
the capacitor). Use low-ESR ceramic or aluminum elec­trolytic capacitors at the output. For aluminum elec­trolytic capacitors, the entire output ripple is
contributed by ΔV

ESR

. Use the ESR

OUT

equation to cal­culate the ESR requirements and choose the capacitor
accordingly. If using ceramic capacitors, assume the
contribution to the output ripple voltage from the ESR
and the capacitor discharge are equal. Calculate the
output capacitance and ESR required for a specified
ripple using the following equations:

where

ΔI

L

is the peak-to-peak inductor current as calculated
above and fSWis the individual converter’s switching
frequency.

The allowable deviation of the output voltage during
fast transient loads also determines the output capaci­tance and its ESR. The output capacitor supplies the
step load current until the controller responds with a
greater duty cycle. The response time (t

RESPONSE

)
depends on the closed-loop bandwidth of the converter.
The high switching frequency of the MAX5099 allows
for higher closed-loop bandwidth, reducing t

RESPONSE

and the output capacitance requirement. The resistive
drop across the output capacitor ESR and the capaci­tor discharge causes a voltage droop during a step
load. Use a combination of low-ESR tantalum or poly­mer and ceramic capacitors for better transient load
and ripple/noise performance. Keep the maximum out­put-voltage deviation within the tolerable limits of the
electronics being powered. When using a ceramic
capacitor, assume 80% and 20% contribution from the
output capacitance discharge and the ESR drop,
respectively. Use the following equations to calculate
the required ESR and capacitance value:

where I

STEP

is the load step and t

RESPONSE

is the
response time of the controller. Controller response
time depends on the control-loop bandwidth.

Boost Converter

The MAX5099 can be configured for step-up conver­sion since the internal MOSFET can be used as a low­side switch. Use the following equations to calculate
the values for the inductor (L

MIN

), input capacitor (CIN),

and output capacitor (C

OUT

) when using the converter

in boost operation.

Inductor

Choose the minimum inductor value so the converter
remains in continuous mode operation at minimum out­put current (I

OMIN

).

where

VDis the forward voltage drop of the external Schottky
diode, D is the duty cycle, and V

DS

is the voltage drop
across the internal MOSFET switch. Select the inductor
with low DC resistance and with a saturation current
(I

SAT

) rating higher than the peak switch current limit of

4.3A (I

CL1

) and 2.6A (I

CL2

) of converter 1 and converter 2,

respectively.

Input Capacitor

The input current for the boost converter is continuous,
and the RMS ripple current at the input is low. Calculate
the capacitor value and ESR of the input capacitor
using the following equations:

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 19

V

Δ

ESR

=

I

Δ

L

I

Δ

L

Vf

××

Δ8

QSW

C

ESR

OUT

OUT

=

ΔΔΔVVV

O RIPPLE ESR Q_

≅+

V

Δ

ESR

=

I

STEP

×

It

STEP RESPONSE

=

Δ

V

Q

ESR

OUT

C

OUT

L

MIN

VD

=

IN

2

fVI

×××

SW O OMIN

2

×

VVV

ODIN

D

=

VVV

ODDS

−

+

−

+

I

Δ

C

=

IN

ESR

L

fV

××

8

=

Δ

SW Q

V

Δ

ESR

I

Δ

L

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

20 ______________________________________________________________________________________

where

where VDSis the voltage drop across the internal MOSFET
switch. ΔI

L

is the peak-to-peak inductor ripple current
as calculated above. ΔVQis the portion of input ripple
due to the capacitor discharge, and ΔV

ESR

is the con-

tribution due to ESR of the capacitor.

Output Capacitor

For the boost converter, the output capacitor supplies
the load current when the main switch is on. The
required output capacitance is high, especially at high­er duty cycles. Also, the output capacitor ESR needs to
be low enough to minimize the voltage drop due to the
ESR while supporting the load current. Use the follow­ing equation to calculate the output capacitor for a
specified output ripple tolerance:

where I

PK

is the peak inductor current as defined in the

following

Power Dissipation

section, IOis the load cur­rent, ΔVQis the portion of the ripple due to the capaci­tor discharge, and ΔV

ESR

is the contribution due to the

ESR of the capacitor. D

MAX

is the maximum duty cycle
at minimum input voltage.

Power Dissipation

The MAX5099 includes two internal power MOSFET
switches. The DC loss is a function of the RMS current in
the switch while the switching loss is a function of switch­ing frequency and instantaneous switch voltage and cur­rent. Use the following equations to calculate the RMS
current, DC loss, and switching loss of each converter.
The MAX5099 is available in a thermally enhanced pack­age and can dissipate up to 2.7W at +70°C ambient
temperature. The total power dissipation in the package
must be limited so that the operating junction tempera­ture does not exceed its absolute maximum rating of
+150°C at maximum ambient temperature.

For the buck converter:

where

See the

Electrical Characteristics

table for the

R

ON(MAX)

maximum value.

For the boost converter:

where VDSis the drop across the internal MOSFET and
η is the efficiency. See the

Electrical Characteristics

table for the R

ON(MAX)

value.

where tRand tFare rise and fall times of the internal
MOSFET. The tRand tFcan be measured in the actual
application.

The supply current in the MAX5099 is dependent on
the switching frequency. See the

Typical Operating

Characteristics

to find the supply current of the
MAX5099 at a given operating frequency. The power
dissipation (PS) in the device due to supply current
(I

SUPPLY

) is calculated using following equation:

PS = V

INMAX

x I

SUPPLY

The total power dissipation PTin the device is:

PT = P

DC1

+ P

DC2

+ P

SW1

+ P

SW2

+ P

S

where P

DC1

and P

DC2

are DC losses in converter 1 and

converter 2, respectively. P

SW1

and P

SW2

are switching

losses in converter 1 and converter 2, respectively.

L

=

IN DS

×

Lf

SW

ΔI

×

−

VV D

()

V

Δ

ESR

=

I

PK

ID

×

OMAX

=

Vf

×

Δ

QSW

ESR

C

OUT

IIIII

RMS DC PK DC PK

22

=++×

()

PI R

=×

DC RMS

()

2

OON MAX()

D

MAX

×

3

I

Δ

L

−

2

I

Δ

L

2

()

×

WW

4

II

=

DC O

II

=+

PK O

VI tt f

×× +

P

IN O R F S

=

SW

IIIII

RMS DC PK DC PK

22

=++×

()

I

I

=

Δ

L

II

DC IN

II

PK IN

=×

PI R

DC RMS ON MAX

()

VI

×

OO

=

IN

V

×

η

NN

I

VV D

()

IN DS

Lf

=

=

2

−

×

SW

Δ

−

ΔI

++

×

I

L

2

L

2

()

D

MAX

×

3

P

=

SW

VI tt f

OIN R F SW

×× +

()

×

4

MAX5099

Calculate the temperature rise of the die using the fol­lowing equation:

TJ = TCx (PT x θJC)

where θJCis the junction-to-case thermal impedance of
the package equal to +1.7°C/W. Solder the exposed
pad of the package to a large copper area to minimize
the case-to-ambient thermal impedance. Measure the
temperature of the copper area near the device at a
worst-case condition of power dissipation, and use
+1.7°C/W as θJCthermal impedance.

Compensation

The MAX5099 provides an internal transconductance
amplifier with its inverting input and its output available
for external frequency compensation. The flexibility of
external compensation for each converter offers wide
selection of output filtering components, especially the
output capacitor. For cost-sensitive applications, use
high-ESR aluminum electrolytic capacitors; for compo­nent size-sensitive applications, use low-ESR tantalum,
polymer, or ceramic capacitors at the output. The high
switching frequency of the MAX5099 allows the use of
ceramic capacitors at the output.

Choose all the passive power components that meet
the output ripple, component size, and component cost
requirements. Choose the small-signal components for
the error amplifier to achieve the desired closed-loop
bandwidth and phase margin. Use a simple pole-zero
pair (Type II) compensation if the output capacitor ESR
zero frequency is below the unity-gain crossover
frequency (fC). Type III compensation is necessary
when the ESR zero frequency is higher than fCor when
compensating for a continuous-mode boost converter
that has a right-half-plane zero.

Use procedure 1 to calculate the compensation
network components when f

ZERO,ESR

< fC.

Buck Converter Compensation

Procedure 1 (See Figure 3)

1) Calculate the f

ZERO,ESR

and LC double-pole

frequencies:

2) Select the unity-gain crossover frequency:

If the f

ZERO,ESR

is lower than fCand close to fLC, use a
Type II compensation network where RFCFprovides a
midband zero f

MID,ZERO

, and RFCCFprovides a high-

frequency pole.

3) Calculate modulator gain GMat the crossover
frequency.

where V

OSC

is a peak-to-peak ramp amplitude equal

to 1V.

The transconductance error-amplifier gain is:

G

E/A

= gMx R

F

The total loop gain at fCshould be equal to 1:

GM x G

E/A

= 1

or

4) Place a zero at or below the LC double-pole:

5) Place a high-frequency pole at fP= 0.5 x fSW.

Figure 3. Type II Compensation Network

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 21

f

,

ZERO ESR

f

LC

=

2

=

2ππ

1

ESR C

××

OUT

1

LC

×

OUT OUT

f

SW

f

≤

C

20

G

V

IN

=×

M

V

ESR f L V

OSC C OUT OUT

ESR

+××

208π

()

.

×

V ESR f L V

OSC C OUT OUT

R

=

F

+××

208π

()

V g ESR

×××

.

IN M

×

C

=

F

××

2π

1

Rf

FLC

C

C

=

CF

R

R

1

2

×××

()

V

OUT

FB_

V

REF

F

fRC

SW F F

-

g

M

+

R

F

C

F

−205 1π .

COMP_

C

CF

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

22 ______________________________________________________________________________________

Procedure 2 (See Figure 4)

If the output capacitor used is a low-ESR ceramic type,
the ESR frequency is usually far away from the targeted
unity crossover frequency (f

C

). In this case, Type III
compensation is recommended. Type III compensation
provides two-pole zero pairs. The locations of the zero
and poles should be such that the phase margin peaks
around f

C

. It is also important to place the two zeros at
or below the double pole to avoid the conditional stabil­ity issue.

1) Select a crossover frequency:

2) Calculate the LC double-pole frequency, fLC:

3) Place a zero

where

and RF≥ 10kΩ.

4) Calculate CIfor a target unity crossover frequency, fC.

5) Place a pole

6) Place a second zero, f

Z2

, at 0.2 x fCor at fLC,

whichever is lower.

7) Place a second pole at 1/2 the switching frequency.

Boost Converter Compensation

The boost converter compensation gets complicated
due to the presence of a right-half-plane zero
f

ZERO,RHP

. The right-half-plane zero causes a drop in
phase while adding positive (+1) slope to the gain
curve. It is important to drop the gain significantly below
unity before the RHP frequency. Use the following pro­cedure to calculate the compensation components:

1) Calculate the LC double-pole frequency, fLC, and
the right-half-plane-zero frequency.

where

Target the unity-gain crossover frequency for:

Figure 4. Type III Compensation Network

f

SW

f

≤

SW

20

1

LC

××

OUT OUT

RC

2

××

π

1

at f

FF

..

1

×××

fR

LC F

075=

×

f

LC

C

F

=

2π

f

Z

=

2075π .

fL C V

×× × ×

2π

C

I

=

C OUT OUT OSC

VR

×

IN F

2=××π

1

RC

II

at f

.

f

P

ZERO ESR1

,

V

OUT

C

CF

R

I

R1

C

I

R2

FB_

V

REF

C

F

R

F

-

g

M

+

COMP_

C

C

=

CF

LC1

×× ××

()

F

fRC

SW F F

−205 1π .

−1

f

LC

=

2π

××

D

LC

OUT OUT

DR

−

1

R

=

I

fC

××

2π

ZERO ESR I

R

1

=

π

2

××

1

,

1

fC

ZI

2

R

−

I

f

ZERO RHP

,

R

MIN

()

()

=

D

=

−

1

=

I

OUT MAX

×

22π

V

IN

V

OUT

V

OUT

()

L

f

ZERO RHP

f

≤

C

,

5

MIN

()

OUT

MAX5099

2) Place a zero

where RF≥ 10kΩ.

3) Calculate C

I

for a target crossover frequency, fC:

where ω

C

= 2π x f

C

.

4) Place a pole

or 5 x fC, whichever is lower.

5) Place the second zero

6) Place the second pole at 1/2 the switching frequency.

Load-Dump Protection MOSFET

Select the external MOSFET with an adequate voltage
rating, V

DSS

, to withstand the maximum expected load­dump input voltage. The on-resistance of the MOSFET,
R

DS(ON)

, should be low enough to maintain a minimal
voltage drop at full load, limiting the power dissipation
of the MOSFET.

During regular operation, the power dissipated by the
MOSFET is:

P

NORMAL

= I

LOAD

2

x R

DS(ON)

where I

LOAD

is equal to the sum of both converters’

input currents.

The MOSFET operates in a saturation region during
load dump, with both high voltage and current applied.

Choose a suitable power MOSFET that can safely oper­ate in the saturation region. Verify its capability to sup­port the downstream DC-DC converters

’

input current
during the load-dump event by checking its safe oper­ating area (SOA) characteristics.

Since the transient peak power dissipation on the
MOSFET can be very high during the load-dump event,
also refer to the thermal impedance graph given in the
data sheet of the power MOSFET to make sure its tran­sient power dissipation is kept within the recommended
limits.

Improving Noise Immunity

In applications where the MAX5099 is subject to noisy
environments, adjust the controller’s compensation to
improve the system’s noise immunity. In particular, high­frequency noise coupled into the feedback loop causes
jittery duty cycles. One solution is to lower the crossover
frequency (see the

Compensation

section).

Figure 5. Boost Application

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 23

f

Z

2

××

C

=

F

×××

2075π .

⎡

−1

()

OSC C O O

⎢

⎣

2=××π

C

=

I

VDLC

f

P

R

=

I

2π

1

RC

FF

1

fR

LC F

2

+

ω

ω

RV

CFIN

1

at f

RC

II

1

fC

××

f

=

Z

21

at f

075=

2

ZERO RHP1

,

I

1

RC

××π

×π..

LC1

⎤
⎥

⎦

at f

.

LC2

I

MAX5099

V

L

R

1

1

π

fC

2=××

LC I

R

−

I

C

C

=

CF

×× ××

()

F

fRC

SW F F

−205 1π .

VDRV

BST_/VDD_

PGND

DRAIN_

DRAIN_

*DL_

SOURCE_

SOURCE_

SGND

FB_

*LEAVE DL_ UNCONNECTED.

V+

VOUT_

C

OUT

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

24 ______________________________________________________________________________________

PCB Layout Guidelines

Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. This is especially
true for dual converters where one channel can affect
the other. Refer to the MAX5098A/MAX5099 Evaluation
Kit data sheet for a specific layout example. Use a mul­tilayer board whenever possible for better noise immu­nity. Follow these guidelines for good PCB layout:

1) For SGND, use a large copper plane under the IC
and solder it to the exposed paddle. To effectively
use this copper area as a heat exchanger between
the PCB and ambient, expose this copper area on
the top and bottom side of the PCB. Do not make a
direct connection from the exposed pad copper
plane to SGND underneath the IC.

2) Isolate the power components and high-current
path from the sensitive analog circuitry.

3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta­ble, jitter-free operation.

4) Connect SGND and PGND together at a single
point. Do not connect them together anywhere else
(refer to the MAX5099 Evaluation Kit data sheet for
more information).

5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full­load efficiency.

6) Ensure that the feedback connection to C

OUT

is

short and direct.

7) Route high-speed switching nodes (BST_/VDD_,
SOURCE_) away from the sensitive analog areas
(BYPASS, COMP_, and FB_). Use the internal PCB
layer for SGND as an EMI shield to keep radiated
noise away from the IC, feedback dividers, and
analog bypass capacitors.

Layout Procedure

1) Place the power components first, with ground ter­minals adjacent (inductor, C

IN_

, and C

OUT_

). Make
all these connections on the top layer with wide,
copper-filled areas (2oz copper recommended).

2) Group the gate-drive components (bootstrap
diodes and capacitors, and VLbypass capacitor)
together near the controller IC.

3) Make the DC-DC controller ground connections as
follows:

a) Create a signal ground plane underneath the IC.

b) Connect this plane to SGND and use this plane

for the ground connection for the reference
(BYPASS), enable, compensation components,
feedback dividers, and OSC resistor.

c) Connect SGND and PGND together (this is the

only connection between SGND and PGND).
Refer to the MAX5098A/MAX5099 Evaluation Kit
data sheet for more information.

MAX5099

Figure 6. 4.5V to 5.5V Operation

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 25

VIN

= 4.5V

IN

V

C1

TO 5.5V

PGND

PGND

VOUT2

R23

C5

R15

L2

D5

6
5

C14

C6

1

32

SOURCE2

SOURCE1

25

L1

SOURCE2

SOURCE1

24

2

4

1

N3

3

30

29

DL2

PGND

MAX5099

DL1

28

3

1

N2

2

4

5
6

D2

C7

D4

VDRV

22 23 2 3

13

10

11

12

VDRV

31

DRAIN2
DRAIN2

BST2/VDD2

DRAIN1
DRAIN1

V+

GATE

ON/OFF

IN_HIGH

BST1/VDD1

26

D1

C15

C4

C19

C16

R16

C17

R18

6

7

FB2

COMP2

FB1

COMP1

19

18

R9

C9

R7

C8

R6

SGND

R17

C21

4

5

9

EN2

SYNC

PGOOD2

PGOOD1

21

C20

R8

L

1427

V

IN

V

VDRV

SGND

BYPASS

OSC

FSEL_1

EN1

20

17

R11

L

V

C12 C13

15

C11

16

R12

8

JU4

1

2

VOUT1

PGND

R22

SGND

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck
Converter with 80V Load-Dump Protection

26 ______________________________________________________________________________________

Typical Application Circuit

VOUT2 = 3.3V

AT 1A

VOUT2

PGND

R23

10kΩ

1%

SGND

C5

22μF

1%

R15

37.4kΩ

C16

L2

4.7μH

D5

6
5

C14

0.1μF

1

BST1/VDD1

C6

0.1μF

32

SOURCE2

SOURCE1

SOURCE1

2
1

SOURCE2

1
2
5
6

N3

3

32

29

DL2

DL1

28

3

N2

D4

VDRV

C15

10μF

25V

C4

10μF

25V

C19

1μF

25V

C3

150μF

25V

N1

C2

4.7μF

R1

3.9kΩ

C1

22μF

100V

VIN

PGND

= 5.2V

TO 19V

IN

V

31
DRAIN2
DRAIN2

BST2/VDD2

DRAIN1
DRAIN1

22 23 2 3

V+

13

GATE

10

35V

ON/OFF

11

12

IN_HIGH

262524

VDRV

D1

4

PGND

MAX5099

4

R16

C17

2700pF

R18

7.15Ω

6

FB2

FB1

19

R9

12.7Ω

C9

2700pF

270pF

7

18

12.1kΩ

COMP2

COMP1

1%

R17

976Ω

1%

C21

56pF

4

9

5

EN2

SYNC

PGOOD2

PGOOD1

EN1

20

21

C20

33pF

L

V

C13

L

V

VDRV

SGND

BYPASS

OSC

FSEL_1

17

R11

L

V

4.7μF

1427

1Ω

R21

C12

2.2μF

VDRV

15

C11

0.22μF

16

R12

6.49Ω

8

JU4

1

2

L1

4.7μH

VOUT1

VOUT1 = 5V

D2

C7

22μF

PGND

AT 2A

R7

1%

10kΩ

C8

R6

1%

52.3kΩ

R22

R8

270pF

1%

976Ω

1%

10kΩ

SGND

MAX5099

Dual, 2.2MHz, Automotive Synchronous Buck

Converter with 80V Load-Dump Protection

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________

27

© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

Pin Configuration

Chip Information

PROCESS: BiCMOS

Package Information

For the latest package outline information, go to

www.maxim-ic.com/packages

.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

32 TQFN T3255+4

21-0140

TOP VIEW

SOURCE1

BST1/VDD1

BST2/VDD2

SOURCE2

VDRV

DL1

PGND

DL2

DRAIN1

SOURCE1

2324 22 20 19 18

25

26

27

28

29

30

+

12

DRAIN2

SOURCE2

*EP

31

32

(5mm x 5mm)

PGOOD1

EN1

FB1

EN2

FB2

COMP1

17

8

COMP2

FSEL_1

16

15

14

13

12

11

10

9

OSC

BYPASS

SGND

V

V+

IN_HIGH

ON/OFF

GATE

SYNC

DRAIN1

21

MAX5099

4567

3

DRAIN2

PGOOD2

TQFN

L

*CONNECT EXPOSED PAD TO GROUND PLANE AND TO SGND.