Rainbow Electronics MAX5098A User Manual

General Description

The MAX5098A is a dual-output, high-switching-frequen­cy DC-DC converter with integrated n-channel switches
that can be used either in high-side or low-side configura­tion. Each output can be configured either as a buck con­verter or a boost converter. In the buck configuration, this
device delivers up to 2A from converter 1 and 1A from
converter 2. The MAX5098A also integrates a load-dump
protection circuitry that is capable of handling load-dump
transients up to 80V for automotive applications. The
load-dump protection circuit utilizes an internal charge­pump to drive the gate of an external n-channel MOSFET.
When an overvoltage or load-dump condition occurs, the
series protection MOSFET absorbs the high voltage tran­sient to prevent damage to lower voltage components.

The DC-DC converters operate over a wide operating
voltage range from 4.5V to 19V. The MAX5098A oper­ates 180° out-of-phase with an adjustable switching fre­quency to minimize external components while allowing
the ability to make trade-offs between the size, efficiency,
and cost. The high switching frequency (up to 2.2MHz)
also allows this device to operate outside the AM band
for automotive applications.

This device utilizes voltage-mode control for stable oper­ation and external compensation, thus the loop gain is
tailored to optimize component selection and transient
response. This device can be synchronized to an exter­nal clock fed at the SYNC input. Also, a clock output
(CKO) allows a master-slave connection of two devices
with a four-phase synchronized switching sequence.
Additional features include internal digital soft-start, indi­vidual enable for each DC-DC regulator (EN1 and EN2),
open-drain power-good outputs (PGOOD1 and
PGOOD2), and a shutdown input (ON/OFF).

Other features of the MAX5098A include overvoltage pro­tection, short-circuit (hiccup current limit) and thermal
protection. The MAX5098A is available in a thermally
enhanced, exposed pad, 5mm x 5mm, 32-pin TQFN
package and is fully specified over the automotive

-40°C to +125°C temperature range.

Applications

Automotive AM/FM Radio Power Supply

Automotive Instrument Cluster Display

Features

o Wide 4.5V to 5.5V or 5.2V to 19V Input Voltage

Range (with Up to 80V Load-Dump Protection)

o Dual-Output DC-DC Converter with Integrated

Power MOSFETs

o Each Output Configurable in Buck or Boost Mode

o Adjustable Outputs from 0.8V to 0.85V

IN

Buck
Configuration) and from VINto 28V (Boost
Configuration)

o I

OUT1

and I

OUT2

of 2A and 1A (Respectively) in

Buck Configuration

o Switching Frequency Programmable from 200kHz

to 2.2MHz

o Synchronization Input (SYNC)

o Clock Output (CKO) for Four-Phase Master-Slave

Operation

o Individual Converter Enable Input and Power-

Good Output

o Low-IQ(7µA) Standby Current (ON/OFF)

o Internal Digital Soft-Start and Soft-Stop

o Short-Circuit Protection on Outputs and

Maximum Duty-Cycle Limit

o Overvoltage Protection on Outputs with Auto

Restart

o Thermal Shutdown

o Thermally Enhanced 32-Pin TQFN Package

Dissipates up to 2.7W at +70°C

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

________________________________________________________________

Maxim Integrated Products

1

Ordering Information

19-4111; Rev 0; 5/08

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.

Pin Configuration appears at end of data sheet

EVALUATION KIT

AVAILABLE

+

Denotes a lead-free package.

*

EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX5098AATJ+

-40°C to +125°C 32 TQFN-EP*

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND_ = SGND,
C

BYPASS

= 0.22µF (low ESR), CVL= 4.7µF (ceramic), CV+= 1µF (low ESR), C

IN_HIGH

= 1µF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

Note 1: Package thermal resistances were obtained using the method described in JEDEC specifications. For detailed information

on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial

.

V+ to SGND............................................................-0.3V to +25V

V+ to IN_HIGH...........................................................-19V to +6V

IN_HIGH to SGND ..................................................-0.3V to +19V

IN_HIGH Maximum Input Current .......................................60mA

BYPASS to SGND..................................................-0.3V to +2.5V

GATE to V+.............................................................-0.3V to +12V

GATE to SGND .......................................................-0.3V to +36V

SGND to PGND_ ...................................................-0.3V to +0.3V

V

L

to SGND..................-0.3V to the Lower of +6V or (V+ + 0.3V)

VDRV to SGND .........................................................-0.3V to +6V

BST1/VDD1, BST2/VDD2, DRAIN_,

PGOOD_ to SGND ..............................................-0.3V to +30V

ON/OFF to SGND ...............................-0.3V to (IN_HIGH + 0.3V)

BST1/VDD1 to SOURCE1,

BST2/VDD2 to SOURCE2......................................-0.3V to +6V

SOURCE_ to SGND................................................-0.6V to +25V

SOURCE_ to PGND_.................................................-1V for 50ns

EN_ to SGND............................................................-0.3V to +6V

OSC, FSEL_1, COMP_, SYNC,

FB_ to SGND..............................................-0.3V to (V

L

+ 0.3V)

CKO to SGND..........................................-0.3V to (VDRV + 0.3V)

SOURCE1, DRAIN1 Peak Current ..............................5A for 1ms

SOURCE2, DRAIN2 Peak Current ..............................3A for 1ms

V

L

, BYPASS to

SGND Short Circuit ................... Continuous, Internally Limited

Continuous Power Dissipation (T

A

= +70°C)

32-Pin TQFN-EP (derate 34.5mW/°C above +70°C)..2759mW

Package Junction-to-Ambient

Thermal Resistance (θ

JA

) (Note 1).............................29.0°C/W

Package Junction-to-Case

Thermal Resistance (θ

JC

) (Note 1) ..............................1.7°C/W

Operating Temperature Range .........................-40°C to +125°C

Storage Temperature Range ............................-65°C to +150°C

Junction Temperature......................................................+150°C

Lead Temperature (soldering, 10s) ................................+300°C

SYSTEM SPECIFICATIONS

Input Voltage Range V+

V+ Operating Supply Current I

V+ Standby Supply Current I

Efficiency η

OVERVOLTAGE PROTECTOR

IN_HIGH Clamp Voltage IN_HIGH I

IN_HIGH Clamp Load
Regulation

IN_HIGH Supply Current I

IN_HIGH Standby Supply
Current

V+ to IN_HIGH Overvoltage
Clamp

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

V+ = IN_HIGH 5.2 19

V

= V+ = IN_HIGH (Note 3) 4.5 5.5

L

Q

V+STBY

IN_HIGH

I

IN_HIGHSTBY

V

OV

VL unloaded, no switching 4.2 mA

V

= 0V, PGOOD_ unconnected, V+ =

EN_

V

IN_HIGH

(V

OUT1

V

OUT2

= 1.85MHz

f

SW

= 10mA 19 20 21 V

SINK

1mA < I

V

= V

EN_

V

IN_HIGH

V

ON/OFF

unconnected, V
to +85°C

VOV = V+ - V
(sinking)

= 14V

= 5V at 1.5A,

= 3.3V at 0.75A,

< 50mA 160 mV

SINK

= V

PGOOD_

= V

= 0V , P GOOD _ = V + =

ON/OFF

IN_HIGH

IN_HIGH

GATE

= 14V

= 14V , TA = -40°C

, I

GATE

V+ = VL = 5.2V 78

V+ = 12V 76

V+ = 16V 70

= 0V,

= 0mA

1.2 1.85 2.5 V

0.75 1.1 mA

270 600 µA

79µA

V

%

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND_ = SGND,
C

BYPASS

= 0.22µF (low ESR), CVL= 4.7µF (ceramic), CV+= 1µF (low ESR), C

IN_HIGH

= 1µF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

IN_HIGH Startup Voltage

GATE Charge Current I

IN_HIGH

UVLO

GATE_CH

Rising, ON/OFF = IN_HIGH, GATE rising 3.6 4.1

Falling, ON/OFF = IN_HIGH, GATE falling 3.45

V

IN_HIGH

V

GATE

V+ = V
I

GATE Output Voltage

GATE Turn-Off Pulldown
Current

V

GATE

V

IN_HIGH

I

GATE_PD

GATE

-

V+ = V
I

GATE

V

IN_HIGH

V

GATE

STARTUP/VL REGULATOR

VL Undervoltage Lockout Trip
Level

UVLO V

falling 3.9 4.1 4.3 V

L

VL Undervoltage Lockout
Hysteresis

VL Output Voltage V

VL LDO Short-Circuit Current I

VL_SHORT

VL LDO Dropout Voltage V

L

LDO

I

SOURCE_

V+ = V

I

SOURCE_

BYPASS OUTPUT

BYPASS Voltage V

BYPASS Load Regulation ΔV

BYPASS

BYPASS

I

BYPASS

0 < I

BYPASS

SOFT-START/SOFT-STOP

Digital Ramp Period Soft­Start/Soft-Stop

Internal 6-bit DAC 2048

Soft-Start/Soft-Stop

VOLTAGE-ERROR AMPLIFIER

FB_ Input Bias Current I

FB_ Input Voltage Set Point V

FB_ to COMP_
Transconductance

FB_

FB_

g

-40°C ≤ TA ≤ +85°C 0.783 0.8 0.809

-40°C ≤ TA ≤ +125°C 0.785 0.814

M

INTERNAL MOSFETS

I

SWITCH

V

On-Resistance High-Side
MOSFET Converter 1

R

ON1

SOURCE1

I

SWITCH

V

SOURCE1

= V

ON/OFF

= 14V,

= V+ = 0V

= V

IN_HIGH

ON/OFF

= 1µA, sourcing

= V

IN_HIGH

ON/OFF

= 1µA, sourcing

= 14V, V

ON/OFF

= 5V, sinking

= 4.5V,

= 14V,

= 0V, V+ = 0V,

20 45 80 µA

4.0 5.3 7.5

9

3.6 mA

180 mV

= 0 to 40mA, 5.5V ≤ V+ ≤ 19V 5.0 5.2 5.5 V

= 5.2V 130 mA

IN_HIGH

= 40mA, V+ = V

= 4.5V 300 550 mV

IN_HIGH

= 0µA 1.98 2.00 2.02 V

< 100µA (sourcing) 2 5 mV

64 Steps

250 nA

1.4 2.4 3.4 mS

= 100mA, BST1/VDD1 to

= 5.2V

= 100mA, BST1/VDD1 to

= 4.5V

195

208 355

V

V

f

SW

Clock

Cycles

V

mΩ

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND_ = SGND,
C

BYPASS

= 0.22µF (low ESR), CVL= 4.7µF (ceramic), CV+= 1µF (low ESR), C

IN_HIGH

= 1µF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

On-Resistance High-Side
MOSFET Converter 2

Minimum Converter 1 Output
Current

Minimum Converter 2 Output
Current

Converter 1/Converter 2
MOSFET DRAIN_ Leakage
Current

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

I

SWITCH

V

R

ON2

I

OUT1

I

OUT2

I

LK12

SOURCE2

I

SWITCH

V

SOURCE2

V

OUT1

V

OUT2

V

= V

EN1

V

SOURCE_

= 100mA, BST2/VDD2 to

= 5.2V

= 100mA, BST2/VDD2 to

= 4.5V

280

300 520

= 5V, V+ = 12V (Note 4) 2 A

= 3.3V, V+ = 12V (Note 4) 1 A

EN2

= 0V

= 0V, V

DRAIN_

= 19V,

20 µA

mΩ

Internal Weak Low-Side Switch
On-Resistance

R

ONLSSW_ILSSW

INTERNAL SWITCH CURRENT LIMIT

Internal Switch Current-Limit
Converter 1

Internal Switch Current-Limit
Converter 2

SWITCHING FREQUENCY

PWM Maximum Duty Cycle D

Switching Frequency Range f

Switching Frequency f

Switching Frequency Accuracy

SYNC Frequency Range f

SYNC High Threshold V

SYNC Low Threshold V

SYNC Input Leakage I

SYNC Input Minimum Pulse
Width

SYNC

SYNCH

SYNCL

SYNC_LEAK

t

SYNCIN

Clock Output Phase Delay CKO

SYNC to Source 1 Phase Delay SYNC

Clock Output High Level V

Clock Output Low Level V

I

CL1

I

CL2

MAX

SW

SW

PHASE

PHASEROSC

CKOH

CKOL

= 30mA 22 Ω

V+ = V
V

BST_/VDD_

V+ = V
V

BST_/VDD_

= 5.2V, VL = VDRV =

IN_HIGH

= 5.2V

= 5.2V, VL = VDRV =

IN_HIGH

= 5.2V

2.8 3.45 4.3 A

1.75 2.1 2.6 A

SYNC = SGND, fSW = 1.25MHz 82 90 95 %

200 2200 kHz

R

= 6.81kΩ, each converter

OSC

(FSEL_1 = V

5.6kΩ < R

10kΩ < R

OSC

OSC

)

L

< 10kΩ, 1% 5

< 62.5kΩ, 1% 7

1.7 1.9 2.1 MHz

SYNC input frequency is twice the
individual converter frequency,
FSEL_1 = V

(see the Setting the

L

400 4400 kHz

Switching Frequency section)

2V

0.8 V

2µA

100 ns

R

= 62.5kΩ, with respect to converter

OSC

2/SOURCE2 waveform

40 D eg r ees

= 62.5kΩ 90 D eg r ees

VL = 5.2V, sourcing 5mA 3.6 V

VL = 5.2V, sinking 5mA 0.6 V

%

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________ 5

ELECTRICAL CHARACTERISTICS (continued)

(VDRV = VL, V+ = VL= IN_HIGH = 5.2V or V+ = IN_HIGH = 5.2V to 19V, EN_ = VL, SYNC = GND, IVL= 0mA, PGND_ = SGND,
C

BYPASS

= 0.22µF (low ESR), CVL= 4.7µF (ceramic), CV+= 1µF (low ESR), C

IN_HIGH

= 1µF (ceramic), R

IN_HIGH

= 3.9kΩ, R

OSC

= 10kΩ,

T

J

= -40°C to +125°C, unless otherwise noted.) (Note 2)

Note 2: 100% tested at TA= +25°C and TA= +125°C. Specifications at TA= -40°C are guaranteed by design and not production

tested.

Note 3: Operating supply range (V+) is guaranteed by V

L

line regulation test. Connect V+ to IN_HIGH and VLfor 5V operation.

Note 4: Output current is limited by the power dissipation of the package; see the

Power Dissipation

section in the

Applications

Information

section.

FSEL_1

FSEL_1 Input High Threshold V

FSEL_1 Input Low Threshold V

FSEL_1 Input Leakage I

ON/OFF

ON/OFF Input High Threshold V

ON/OFF Input Low Threshold V

ON/OFF Input Leakage Current I

EN_ INPUTS

EN_ Input High Threshold V

EN_ Input Hysteresis V

EN_ Input Leakage Current I

POWER-GOOD OUTPUT (PGOOD1, PGOOD2)

PGOOD_ Threshold V

PGOOD_ Output Voltage V

PGOOD_ Output Leakage
Current

OUTPUT OVERVOLTAGE PROTECTION

FB_ OVP Threshold Rising V

FB_ OVP Threshold Falling V

THERMAL PROTECTION

Thermal Shutdown T

Thermal Hysteresis T

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

IH

IL

FSEL_1_LEAK

IH

IL

ON/OFF_LEAKVON/OFF

IH

EN_HYS

EN_LEAK

TPGOOD_

PGOOD_ISINK

I

LKPGOOD_

OVP_R

OVP_F

SHDN

HYST

EN_ rising 1.9 2.0 2.1 V

Falling 90 92.5 95 % V

= 3mA 0.4 V

V+ = VL = V
V

PGOOD_

Rising +165 °C

= 5V 0.26 2.00 µA

= V

IN_HIGH

= 23V, V

FB_

= 1V

= 5.2V,

EN_

2V

2V

-1 +1 µA

107 114 121 % V

0.5 V

12.5 V

20 °C

0.8 V

2µA

0.8 V

2µA

FB_

FB_

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

6 _______________________________________________________________________________________

Typical Operating Characteristics

(See the

Typical Application Circuit

, unless otherwise noted. V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means

that N1 is shorted externally.)

OUTPUT1 EFFICIENCY

vs. LOAD CURRENT

MAX5098A toc01

LOAD (A)

OUTPUT1 EFFICIENCY (%)

1.81.61.2 1.40.6 0.8 1.00.4

10

20

30

40

50

60

70

80

90

100

0

0.2 2.0

VIN = 8V

VIN = 14V

VIN = 16V

V

OUT

= 5V

f

SW

= 1.85MHz

OUTPUT2 EFFICIENCY

vs. LOAD CURRENT

MAX5098A toc02

LOAD (A)

OUTPUT2 EFFICIENCY (%)

0.90.80.6 0.70.4 0.50.30.2 1.0

10

20

30

40

50

60

70

80

90

100

0

VIN = 16V

VIN = 8V

VIN = 14V

VIN = 5.5V

VIN = 4.5V

V

OUT

= 3.3V

f

SW

= 1.85MHz

OUTPUT1 EFFICIENCY

vs. LOAD CURRENT

MAX5098A toc03

LOAD (A)

OUTPUT1 EFFICIENCY (%)

1.81.61.2 1.40.6 0.8 1.00.4

10

20

30

40

50

60

70

80

90

100

0

0.2 2.0

VIN = 8V

VIN = 14V VIN = 16V

V

OUT

= 5V

f

SW

= 300kHz

L1 = 18μH

OUTPUT2 EFFICIENCY

vs. LOAD CURRENT

MAX5098A toc04

LOAD (A)

OUTPUT2 EFFICIENCY (%)

0.90.80.6 0.70.4 0.50.30.2 1.0

10

20

30

40

50

60

70

80

90

100

0

VIN = 16V

VIN = 8V

VIN = 14V

VIN = 5.5V

VIN = 4.5V

V

OUT

= 3.3V

f

SW

= 300kHz

L2 = 27μH

OUTPUT1 VOLTAGE
vs. LOAD CURRENT

MAX5098A toc05

LOAD (A)

OUTPUT1 VOLTAGE (V)

1.81.61.41.21.00.80.60.4

4.92

4.94

4.96

4.98

5.00

4.90

0.2 2.0

VIN = 8V

VIN = 14V

VIN = 16V

V

OUT

= 5V

f

SW

= 1.85MHz

OUTPUT2 VOLTAGE
vs. LOAD CURRENT

MAX5098A toc06

LOAD (A)

OUTPUT2 VOLTAGE (V)

0.90.80.70.60.50.40.3

3.22

3.24

3.26

3.28

3.30

3.20

0.2 1.0

VIN = 5.5V

VIN = 16V

VIN = 14V

V

OUT

= 3.3V

f

SW

= 1.85MHz

VL OUTPUT VOLTAGE

vs. CONVERTER SWITCHING FREQUENCY

MAX5098A toc07

CONVERTER SWITCHING FREQUENCY (kHz)

V

L

OUTPUT VOLTAGE (V)

17001200700

4.2

4.4

4.6

4.8

5.0

5.2

5.4

4.0
200 2200

VIN = 4.5V

VIN = 5.5V

VIN = 8V

VIN = 19V

VIN = 5V

BOTH CONVERTERS SWITCHING
FSEL_1 = V

L

EACH CONVERTER SWITCHING

FREQUENCY vs. R

OSC

MAX5098A toc08

R

OSC

(kΩ)

SWITCHING FREQUENCY (MHz)

604020

1

0

80

10

0.1

CONVERTER 1, CONVERTER 2

CONVERTER 1

FSEL_1 = VL,
FSEL_1 = GND,

EACH CONVERTER SWITCHING

FREQUENCY vs. TEMPERATURE

MAX5098A toc09

TEMPERATURE (°C)

SWITCHING FREQUENCY (MHz)

-5 30 65 100

1

10

0.1

-40 135

0.3MHz

0.6MHz

1.25MHz

1.85MHz

2.2MHz

FSEL_1 = V

L

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________

7

Typical Operating Characteristics (continued)

(See the

Typical Application Circuit

, unless otherwise noted. V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means

that N1 is shorted externally.)

LINE-TRANSIENT RESPONSE

(BUCK CONVERTER)

1ms/div

CONVERTER 2

LOAD-TRANSIENT RESPONSE

MAX5098A toc10

V

IN

5V/div

0V

= 5.0V/1.5A

V

OUT1

AC-COUPLED
200mV/div

= 3.3V/0.75A

V

OUT2

AC-COUPLED
200mV/div

MAX5098A toc12

V

= 3.3V

OUT2

AC-COUPLED
200mV/div

I

OUT2

500mA/div

0A

CONVERTER 1

LOAD-TRANSIENT RESPONSE

100μs/div

SOFT-START/SOFT-STOP FROM EN1

MAX5098A toc11

MAX5098A toc13

fSW = 1.85MHz

= 5.0V

V

OUT1

AC-COUPLED
200mV/div

I

OUT1

1A/div

0A

EN1
5V/div

0V

= 5V/2A

V

OUT1

5V/div
0V

P

GOOD1

5V/div
0V

100μs/div

1ms/div

OUT-OF-PHASE OPERATION

SOFT-START FROM ON/OFF

0V

0V

0V

0V

2ms/div

MAX5098A toc14

ON/OFF
5V/div

= EN1 = EN2

V

L

5V/div
GATE
10V/div
V+
10V/div

= 5V/2A

V

OUT1

5V/div

(FSEL_1 = V

200ns/div

)

L

MAX5098A toc15

CKO
5V/div
0V

SOURCE1
10V/div
0V

SOURCE2
10V/div
0V

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

8 _______________________________________________________________________________________

Typical Operating Characteristics (continued)

(See the

Typical Application Circuit

, unless otherwise noted. V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means

that N1 is shorted externally.)

OUT-OF-PHASE OPERATION

(FSEL_1 = SGND)

EXTERNAL SYNCHRONIZATION

(FSEL_1 = SGND )

200ns/div

MAX5098A toc16

MAX5098A toc18

CKO
5V/div
0V

SOURCE1
10V/div
0V

SOURCE2
10V/div
0V

SYNC
5V/div
0V

CKO
5V/div
0V

SOURCE1
10V/div
0V

SOURCE2
10V/div
0V

EXTERNAL SYNCHRONIZATION

(FSEL_1 = V

FOUR-PHASE OPERATION

(FSEL_1 = V

0V

0V

0V

0V

0V

200ns/div

)

L

MAX5098A toc17

)

L

MAX5098A toc19

SYNC
5V/div
0V

CKO
5V/div
0V

SOURCE1
10V/div
0V

SOURCE2
10V/div
0V

MASTER
CKO
5V/div

MASTER SOURCE1
20V/div

MASTER SOURCE2
20V/div

SLAVE SOURCE1
20V/div

SLAVE SOURCE2
20V/div

200ns/div

200ns/div

FB_ VOLTAGE

VL = V+ = V

vs. TEMPERATURE

= 5.5V

IN_HIGH

TEMPERATURE (°C)

10065-5 30

OVP BEHAVIOR

0V

EXTERNAL OVERVOLTAGE REMOVED

0V

0V

0V

0V

1ms/div

MAX5098A toc20

V+
10V/div

GATE
10V/div

V

OUT2

10V/div
V

OUT1

10V/div
PGOOD2
10V/div

0.825

0.820

0.815

0.810

0.805

0.800

FB_ VOLTAGE (V)

0.795

0.790

0.785

-40 135

MAX5098A toc21

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

_______________________________________________________________________________________

9

Typical Operating Characteristics (continued)

(See the

Typical Application Circuit

, unless otherwise noted. V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means

that N1 is shorted externally.)

V+ SWITCHING SUPPLY CURRENT

vs. SWITCHING FREQUENCY

MAX5098A toc25

SWITCHING FREQUENCY (kHz)

V+ SWITCHING SUPPLY CURRENT (mA)

182014401060680

10

20

30

40

50

0

300 2200

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

V+ = IN_HIGH = ON/OFF

V+ STANDBY SUPPLY CURRENT

vs. TEMPERATURE

MAX5098A toc26

TEMPERATURE (°C)

V+ STANDBY SUPPLY CURRENT (mA)

100500

1

2

3

4

0

-50 150

fSW = 1.85MHz

fSW = 300kHz

V+ = IN_HIGH = ON/OFF
EN1 = EN2 = SGND

IN_HIGH SHUTDOWN CURRENT

vs. TEMPERATURE

MAX5098A toc27

TEMPERATURE (°C)

IN_HIGH SHUTDOWN CURRENT (μA)

100500

4

8

12

16

20

0

-50 150

IN_HIGH = 8V

IN_HIGH = 14V

IN_HIGH = 16V

ON/OFF = SGND

IN_HIGH STANDBY CURRENT

vs. TEMPERATURE

MAX5098A toc28

TEMPERATURE (°C)

IN_HIGH STANDBY CURRENT (μA)

100500

85

95

105

115

125

135

145

75

-50 150

IN_HIGH = 8V

IN_HIGH = 14V

IN_HIGH = 16V

ON/OFF = IN_HIGH
EN1 = EN2 = SGND

BYPASS VOLTAGE

vs. TEMPERATURE

2.010
VL = V+ = V

2.008

2.006

2.004

2.002

2.000

1.998

BYPASS VOLTAGE (V)

1.996

1.994

1.992

1.990

-40 135

= 5.5V

IN_HIGH

TEMPERATURE (°C)

BYPASS VOLTAGE

2.000

1.998

MAX5098A toc22

1.996

1.994

BYPASS VOLTAGE (V)

1.992

10065-5 30

1.990

vs. BYPASS CURRENT

TA = +85°C

TA = -40°C

0 100705010 30 90

BYPASS CURRENT (μA)

TA = +125°C

SOURCE1, SOURCE1 INDICATOR CURRENT,

SOURCE2, SOURCE2 INDICATOR CURRENT

TA = +135°C

TA = +25°C

806020 40

MAX5098A toc23

0V

0A

0V

0A

1μs/div

MAX5098A toc24

SOURCE1
20V/div
NO LOAD

I
500mA/div
NO LOAD

SOURCE2
20V/div

I
1A/div

SOURCE1

SOURCE2

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

10 ______________________________________________________________________________________

Typical Operating Characteristics (continued)

(See the

Typical Application Circuit

, unless otherwise noted. V+ = V

IN_HIGH

= 14V, unless otherwise noted. V+ = V

IN_HIGH

means

that N1 is shorted externally.)

IN_HIGH CLAMP VOLTAGE

vs. CLAMP CURRENT

MAX5098A toc29

CLAMP CURRENT (mA)

IN_HIGH CLAMP VOLTAGE (V)

40302010

20.0

20.1

20.2

20.3

19.9
050

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

V+ TO IN_HIGH CLAMP VOLTAGE

vs. GATE SINK CURRENT

MAX5098A toc30

GATE SINK CURRENT (mA)

V+ TO IN_HIGH CLAMP VOLTAGE (V)

8642

1

2

3

4

5

0

010

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

(V

GATE

- V) vs. V

IN_HIGH

MAX5098A toc31

V

IN_HIGH

(V)

(V

GATE

- V) (V)

15.512.08.5

2

4

6

8

10

0

5.0 19.0

TA = +25°C

TA = +135°C

TA = +125°C

TA = +85°C

TA = -40°C

ON/OFF = IN_HIGH

SYSTEM TURN-ON FROM BATTERY

MAX5098A toc32

10ms/div

V

L

10V/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

V

IN

10V/div

0V

0V

0V

0V

0V

SYSTEM TURN-OFF FROM BATTERY

MAX5098A toc33

10ms/div

V

L

10V/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

V

IN

10V/div

0V

0V

0V

0V

0V

SYSTEM LOAD-DUMP

MAX5098A toc34

100ms/div

V

OUT1

AC-COUPLED
100mV/div

V+
10V/div

GATE
10V/div

IN_HIGH
10V/div

0V

0V

0V

0V
0V

V

IN

50V/div

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 11

Pin Description

PIN NAME FUNCTION

1, 32 SOURCE2

2, 3 DRAIN2

4 PGOOD2

5 EN2 Converter 2 Active-High Enable Input. Connect to VL for always-on operation.

6 FB2

7 COMP2 Converter 2 Internal Transconductance Amplifier Output. See the Compensation section.

8 OSC

9 SYNC

10 GATE

11 ON/OFF

Converter 2 Internal MOSFET Source Connection. For buck converter operation, connect SOURCE2 to the
switched side of the inductor. For boost operation, connect SOURCE2 to PGND_ (Figure 6).

Converter 2 Internal MOSFET Drain Connection. For buck converter operation, use the MOSFET as a high­side switch and connect DRAIN2 to the DC-DC converters supply input rail. For boost converter operation,
use the MOSFET as a low-side switch and connect DRAIN2 to the inductor and diode junction (Figure 6).

Converter 2 Open-Drain Power-Good Output. PGOOD2 goes low when converter 2’s output falls below

92.5% of its set regulation voltage. Use PGOOD2 and EN1 to sequence the converters. Converter 2 starts
first.

Converter 2 Feedback Input. Connect FB2 to a resistive divider between converter 2’s output and SGND to
adjust the output voltage. To set the output voltage below 0.8V, connect FB2 to a resistive voltage-divider
from BYPASS to regulator 2’s output (Figure 3). See the Setting the Output Voltage section.

Oscillator Frequency Set Input. Connect a resistor from OSC to SGND (R
(see the Setting the Switching Frequency section). Set R
input frequency when using external synchronization. R
connected to the SYNC input. See the Synchronization (SYNC)/Clock Output (CKO) section.

External Clock Synchronization Input. Connect SYNC to a 400kHz to 4400kHz clock to synchronize the
switching frequency with the system clock. Each converter frequency is 1/2 of the frequency applied to
SYNC (FSEL_1 = V
SYNC frequency. Connect SYNC to SGND when not used.

Gate Drive Output. Connect to the gate of the external n-channel load-dump protection MOSFET. GATE =
IN_HIGH + 9V (typ) with IN_HIGH = 12V. GATE pulls to IN_HIGH by an internal n-channel MOSFET when V+
raises 2V above IN_HIGH. Leave gate unconnected if the load-dump protection is not used (MOSFET not
installed).

n-Channel Switch Enable Input. Drive ON/OFF high for normal operation. Drive ON/OFF low to turn off the
external n-channel load-dump protection MOSFET and reduce the supply current to 7µA (typ). When
ON/OFF is driven low, both DC-DC converters are disabled and the PGOOD_ outputs are driven low.
Connect to V+ if the external load-dump protection is not used (MOSFET not installed).

). For FSEL_1 = SGND, the switching frequency of converter 1 becomes 1/4 of the

L

for an oscillator frequency equal to the SYNC

OSC

is still required when an external clock is

OSC

) to set the switching frequency

OSC

12 IN_HIGH

13 V+

14 V

15 SGND

L

Startup Input. IN_HIGH is protected by internally clamping to 21V (max). Connect a resistor (4kΩ max) from
IN_HIGH to the drain of the protection switch. Bypass IN_HIGH with a 4.7µF electrolytic or 1µF minimum
ceramic capacitor. Connect to V+ if the external load-dump protection is not used (MOSFET not installed).

Input Supply Voltage. V+ can range from 5.2V to 19V. Connect V+, IN_HIGH, and V

5.5V input operation. Bypass V+ to SGND with a 1µF minimum ceramic capacitor.

Internal Regulator Output. The VL regulator is used to supply the drive current at input VDRV. When driving
VDRV, use an RC lowpass filter to decouple switching noise from VDRV to the V
Application Circuit). Bypass V

Signal Ground. Connect SGND to exposed pad and to the board signal ground plane. Connect the board
signal ground and power ground planes together at a single point.

to SGND with a 4.7µF minimum ceramic capacitor.

L

together for 4.5V to

L

regulator (see the Typical

L

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

12 ______________________________________________________________________________________

Pin Description (continued)

PIN NAME FUNCTION

16 BYPASS Reference Output Bypass Connection. Bypass to SGND with a 0.22µF or greater ceramic capacitor.

for normal operation. Connect FSEL_1 to SGND

L

17 FSEL_1

18 COMP1 Converter 1 Internal Transconductance Amplifier Output. See the Compensation section.

19 FB1

20 EN1 Converter 1 Active-High Enable Input. Connect to VL for an always-on operation.

21 PGOOD1

22, 23 DRAIN1

Converter 1 Frequency Select Input. Connect FSEL_1 to V
to reduce converter 1’s switching frequency to 1/2 of converter 2’s switching frequency (converter 1
switching frequency is 1/4 the CKO frequency). Do not leave FSEL_1 unconnected.

Converter 1 Feedback Input. Connect FB1 to a resistive divider between converter 1’s output and SGND to
adjust the output voltage. To set the output voltage below 0.8V, connect FB1 to a resistive voltage-divider
from BYPASS to regulator 1’s output (Figure 3). See the Setting the Output Voltage section.

Converter 1 Open-Drain Power-Good Output. PGOOD1 output goes low when converter 1’s output falls
below 92.5% of its set regulation voltage. Use PGOOD1 and EN2 to sequence the converters. Converter 1
starts first.

Converter 1 Internal MOSFET Drain Connection. For buck converter operation, use the MOSFET as a high­side switch and connect DRAIN1 to the DC-DC converters supply input rail. For boost converter operation,
use the MOSFET as a low-side switch and connect DRAIN1 to the inductor and diode junction (Figure 6).

24, 25 SOURCE1

26 BST1/VDD1

27 VDRV

28 CKO

29, 30

31 BST2/VDD2

—EP

PGND1,

PGND2

Converter 1 Internal MOSFET Source Connection. For buck operation, connect SOURCE1 to the switched
side of the inductor. For boost operation, connect SOURCE1 to PGND_ (Figure 6).

Converter 1 Bootstrap Flying-Capacitor Connection. For buck converter operation, connect BST1/VDD1 to a

0.1µF ceramic capacitor and diode according to the Typical Application Circuit. For boost converter
operation, driver bypass capacitor connection. Connect to VDRV and bypass with a 0.1µF ceramic
capacitor to PGND_ (Figure 6).

Low-Side Driver Supply Input. Connect VDRV to V
internal V
VDRV. For boost converter operation, connect VDRV to BST1/VDD1 and BST2/VDD2. Bypass with a
minimum 2.2µF ceramic capacitor to PGND_ (see the Typical Application Circuit). Do not connect to an
external supply.

Clock Output. CKO is an output with twice the frequency of each converter (FSEL_1 = V
phase with respect to converter 1. Connect CKO to the SYNC input of another MAX5098A for a four-phase
converter.

Power Ground. Connect both PGND1 and PGND2 together and to the board power ground plane.

Converter 2 Bootstrap Flying-Capacitor Connection. For buck converter operation, connect BST2/VDD2 to a

0.1µF ceramic capacitor and diode according to the Typical Application Circuit. For boost converter
operation, driver bypass capacitor connection. Connect to VDRV and bypass with a 0.1µF ceramic
capacitor from BST2/VDD2 to PGND_ (Figure 6).

Exposed Pad. Connect EP to SGND. For enhanced thermal dissipation, connect EP to a copper area as
large as possible. Do not use EP as the sole ground connection.

regulator. For buck converter operation, connect anode terminals of external bootstrap diodes to

L

through an RC filter to bypass switching noise to the

L

) and 90° out-of-

L

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 13

Functional Diagram

IN_HIGH

ON/OFF

BYPASS

FSEL_1

V+

MAX5098A

1.8V

GATE

V

L

BST1/VDD1

DRAIN1

SOURCE1

PGOOD1

CKO1

FREQUENCY

DIVIDER

CONVERTER 1

OSCILLATOR

20V SHUNT

REGULATOR

MAXIMUM DUTY-CYCLE

CONTROL

FREQUENCY
CONTROL

PWM

COMPARATOR

CHARGE

PUMP

S

R

Q

Q

STARTUP CIRCUIT/

PROTECTION CIRCUIT/

CHARGE PUMP

f

/4

SW

OVERVOLTAGE

CURRENT

LIMIT

LDO

VL

EN1

SYNC

OSC

VDRV

CKO

EN2

TRANSCONDUCTANCE

DIGITAL

SOFT-START

MAIN

OSCILLATOR

CKO2

OVERVOLTAGE

ERROR AMPLIFIER

CONVERTER 2

0.8V

0.2V

0.9V

0.74V

VL

PGND_

FB1

COMP1

SGND

PGOOD2
DRAIN2

BST2/VDD2
SOURCE2
FB2
COMP2
PGND_

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

14 ______________________________________________________________________________________

Detailed Description

PWM Controller

The MAX5098A dual DC-DC converter uses a pulse­width-modulation (PWM) voltage-mode control scheme.
On each converter the device includes one integrated
n-channel MOSFET switch and requires an external
low-forward-drop Schottky diode for output rectifica­tion. The controller generates the clock signal by divid­ing down the internal oscillator (f

CKO

) or the SYNC
input when driven by an external clock, therefore each
controller’s switching frequency equals half the oscilla­tor frequency (fSW= f

CKO

/2) or half of the SYNC input

frequency (fSW= f

SYNC

/2). An internal transconduc­tance error amplifier produces an integrated error volt­age at COMP_, providing high DC accuracy. The
voltage at COMP_ sets the duty cycle using a PWM
comparator and a ramp generator. At each rising edge
of the clock, converter 1’s MOSFET switch turns on and
remains on until either the appropriate or maximum
duty cycle is reached, or the maximum current limit for
the switch is reached. Converter 2 operates 180° out­of-phase, so its MOSFET switch turns on at each falling
edge of the clock.

In the case of buck operation (see the

Typical

Application Circuit

), the internal MOSFET is used in
high-side configuration. During each MOSFET’s on­time, the associated inductor current ramps up. During
the second half of the switching cycle, the high-side
MOSFET turns off and forward biases the Schottky rec­tifier. During this time, the SOURCE_ voltage is
clamped to a diode drop (VD) below ground. A low for­ward voltage drop (0.4V) Schottky diode must be used
to ensure the SOURCE_ voltage does not go below

-0.6V abs max. The inductor releases the stored energy
as its current ramps down, and provides current to the
output. The bootstrap capacitor is also recharged when
the SOURCE_ voltage goes low during the high-side
MOSFET off-time. The maximum duty-cycle limit
ensures proper bootstrap charging at startup or low
input voltages. The circuit goes in discontinuous con­duction mode operation at light load, when the inductor
current completely discharges before the next cycle
commences. Under overload conditions, when the
inductor current exceeds the peak current limit of the
respective switch, the high-side MOSFET turns off
quickly and waits until the next clock cycle.

In the case of boost operation, the MOSFET is a low­side switch (Figure 6). During each on-time, the induc­tor current ramps up. During the second half of the
switching cycle, the low-side switch turns off and for-

ward biases the Schottky diode. During this time, the
DRAIN_ voltage is clamped to a diode drop (VD) above
V

OUT_

and the inductor provides energy to the output

as well as replenishes the output capacitor charge.

Load-Dump Protection

Most automotive applications are powered by a multi­cell, 12V lead-acid battery with a voltage from 9V to
16V (depending on load current, charging status, tem­perature, battery age, etc.). The battery voltage is dis­tributed throughout the automobile and is locally
regulated down to voltages required by the different
system modules. Load dump occurs when the alterna­tor is charging the battery and the battery becomes
disconnected. Power in the alternator inductance flows
into the distributed power system and elevates the volt­age seen at each module. The voltage spikes have rise
times typically greater than 5ms and decays within sev­eral hundred milliseconds but can extend out to 1s or
more depending on the characteristics of the charging
system. These transients are capable of destroying
sensitive electronic equipment on the first fault event.

During load dump, the MAX5098A provides the ability
to clamp the input-voltage rail of the internal DC-DC
converters to a safe level, while preventing power dis­continuity at the DC-DC converters’ outputs.

The load-dump protection circuit utilizes an internal
charge pump to drive the gate of an external n-channel
MOSFET. This series protection MOSFET absorbs the
load-dump overvoltage transient and operates in satu­ration over the normal battery range to minimize power
dissipation. During load dump, the gate voltage of the
protection MOSFET is regulated to prevent the source
terminal from exceeding 19V.

The DC-DC converters are powered from the source
terminal of the load-dump protection MOSFET, so that
their input voltage is limited during load-dump and can
operate normally.

ON/OFF

The MAX5098A provides an input (ON/OFF) to turn on
and off the external load-dump protection MOSFET.
Drive ON/OFF high for normal operation. Drive ON/OFF
low to turn off the external n-channel load-dump protec­tion MOSFET and reduce the supply current to 7µA (typ).
When ON/OFF is driven low, the converter also turns off,
and the PGOOD_ outputs are driven low. V+ will be self
discharged through the converters output currents and
the IC supply current.

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 15

Internal Oscillator/Out-of-Phase Operation

The internal oscillator generates the 180° out-of-phase
clock signal required by each regulator. The switching
frequency of each converter (f

SW

) is programmable
from 200kHz to 2.2MHz using a single 1% resistor at
R

OSC

. See the

Setting the Switching Frequency

section.

With dual synchronized out-of-phase operation, the
MAX5098A’s internal MOSFETs turn on 180° out-of­phase. The instantaneous input current peaks of both
regulators do not overlap, resulting in reduced RMS rip­ple current and input-voltage ripple. This reduces the
required input capacitor ripple current rating, allows for
fewer or less expensive capacitors, and reduces
shielding requirements for EMI.

Synchronization (SYNC)/

Clock Output (CKO)

The main oscillator can be synchronized to the system
clock by applying an external clock (f

SYNC

) at SYNC.

The f

SYNC

frequency must be twice the required oper­ating frequency of an individual converter. Use a TTL
logic signal for the external clock with at least 100ns
pulse width. R

OSC

is still required when using external
synchronization. Program the internal oscillator fre­quency to have f

SW

= 1/2 f

SYNC.

The device is properly

synchronized if the SYNC frequency, f

SYNC

, varies

within ±20%.

Two MAX5098As can be connected in the master-slave
configuration for four ripple-phase operation (Figure 1).
The MAX5098A provides a clock output (CKO) that is
45° phase-shifted with respect to the internal switch
turn-on edge. Feed the CKO of the master to the SYNC
input of the slave. The effective input ripple switching
frequency is four times the individual converter’s switch­ing frequency. When driving the master converter using
an external clock at SYNC, set the f

SYNC

clock duty
cycle to 50% for effective 90° phase-shifted interleaved
operation. When a SYNC is applied (and FSEL_1 = 0),
converter 1 duty cycle is limited to 75% (max).

Input Voltage (V+)/

Internal Linear Regulator (V

L

)

All internal control circuitry operates from an internally
regulated nominal voltage of 5.2V (VL). At higher input
voltages (V+) of 5.2V to 19V, VLis regulated to 5.2V. At

5.2V or below, the internal linear regulator operates in
dropout mode, where VLfollows V+. Depending on the
load on VL, the dropout voltage can be high enough to
reduce VLbelow the undervoltage lockout (UVLO)
threshold. Do not use VLto power external circuitry.

For input voltages less than 5.5V, connect V+ and V

L

together. The load on VLis proportional to the switching
frequency of converter 1 and converter 2. See the V

L

Output Voltage vs. Converter Switching Frequency
graph in the

Typical Operating Characteristics

. For

input voltage ranges higher than 5.5V, disconnect V

L

from V+.

Bypass V+ to SGND with a 1µF or greater ceramic
capacitor placed close to the MAX5098A. Bypass V

L

with a 4.7µF ceramic capacitor to SGND.

Undervoltage Lockout/

Soft-Start/Soft-Stop

The MAX5098A includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter
turn-on and monotonic rise of the output voltage. The
falling UVLO threshold is internally set to 4.1V (typ) with
180mV hysteresis. Hysteresis at UVLO eliminates “chat­tering” during startup. When VLdrops below UVLO, the
internal MOSFET switches are turned off.

The MAX5098A digital soft-start reduces input inrush
currents and glitches at the input during turn-on. When
UVLO is cleared and EN_ is high, digital soft-start slow­ly ramps up the internal reference voltage in 64 steps.
The total soft-start period is 4096 internal oscillator
switching cycles.

Driving EN_ low initiates digital soft-stop that slowly
ramps down the internal reference voltage in 64 steps.
The total soft-stop period is equal to the soft-start period.

To calculate the soft-start/soft-stop period, use the fol­lowing equation:

where f

CKO

is the internal oscillator and f

CKO

is twice

each converters’ switching frequency (FSEL_1 = VL)

Enable (EN1, EN2)

The MAX5098A dual converter provides separate
enable inputs, EN1 and EN2, to individually control or
sequence the output voltages. These active-high enable
inputs are TTL compatible. Driving EN_ high initiates
soft-start of the converter, and PGOOD_ goes logic-high
when the converter output voltage reaches the
V

TPGOOD_

threshold. Driving EN_ low initiates a soft­stop of the converter, and immediately forces PGOOD_
low. Use EN1, EN2, and PGOOD1 for sequencing (see
Figure 2). Connect PGOOD1 to EN2 to make sure con­verter 1’s output is within regulation before converter 2
starts. Add an RC network from VLto EN1 and EN2 to
delay the individual converter. Sequencing reduces
input inrush current and possible chattering. Connect
EN_ to VLfor always-on operation.

tms

()

SS

4096

=

f kHz

CKO

()

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

16 ______________________________________________________________________________________

Figure 1. Synchronized Controllers

V

IN

C

IN

OUTPUT2

DUTY CYCLE = 50%

SYNC

CKO

(MASTER)

CKO

(SLAVE)

SOURCE1

(MASTER)

SYNC

PHASE

CLKIN

DRAIN2

SOURCE2

V+

DRAIN1

SOURCE1

CKO

CKO

PHASE

OUTPUT1

OUTPUT4

DRAIN2

SOURCE2

SYNCSYNC

SLAVEMASTER

V+

DRAIN1

SOURCE1

OUTPUT3

SOURCE2

(MASTER)

SOURCE1

(SLAVE)

SOURCE2

(SLAVE)

CIN (RIPPLE)

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 17

PGOOD_

Converter 1 and converter 2 include a power-good flag,
PGOOD1 and PGOOD2, respectively. Since PGOOD_
is an open-drain output and can sink 3mA while provid­ing the TTL logic-low signal, pull PGOOD_ to a logic
voltage to provide a logic-level output. PGOOD1 goes
low when converter 1’s feedback FB1 drops to 92.5%
(V

TPGOOD_

) of its nominal set point. The same is true for
converter 2. Connect PGOOD_ to SGND or leave
unconnected if not used.

Current Limit

The internal MOSFET switch current of each converter is
monitored during its on-time. When the peak switch cur­rent crosses the current-limit threshold of 3.45A (typ) and

2.1A (typ) for converter 1 and converter 2, respectively,
the on-cycle is terminated immediately and the inductor
is allowed to discharge. The MOSFET is turned on at the
next clock pulse, initiating a new switching cycle.

In deep overload or short-circuit conditions when the
V

FB_

voltage drops below 0.2V, the switching frequen­cy is reduced to 1/4 x fSWto provide sufficient time for
the inductor to discharge. During overload conditions, if
the voltage across the inductor is not high enough to
allow for the inductor current to properly discharge,
current runaway may occur. Current runaway can
destroy the device in spite of internal thermal-overload
protection. Reducing the switching frequency during
overload conditions allows more time for inductor dis­charge and prevents current runaway.

Output Overvoltage Protection

The MAX5098A outputs are protected from output volt­age overshoots due to input transients and shorting the
output to a high voltage. When the output voltage rises
above the overvoltage threshold, 110% (typ) nominal
FB_, the overvoltage condition is triggered. When the
overvoltage condition is triggered on either channel,
both converters are immediately turned off, 20Ω pull-
down switches from SOURCE_ to PGND_ are turned on
to help the output-voltage discharge, and the gate of
the load-dump protection external MOSFET is pulled
low. The device restarts as soon as both converter out­puts discharge, bringing both FB_ input voltages below

12.5V of their nominal set points.

Thermal-Overload Protection

During continuous short circuit or overload at the out­put, the power dissipation in the IC can exceed its limit.
The MAX5098A provides thermal shutdown protection
with temperature hysteresis. Internal thermal shutdown
is provided to avoid irreversible damage to the device.
When the die temperature exceeds +165°C (typ), an
on-chip thermal sensor shuts down the device, forcing
the internal switches to turn off, allowing the IC to cool.
The thermal sensor turns the part on again with soft­start after the junction temperature cools by +20°C.
During thermal shutdown, both regulators shut down,
PGOOD_ goes low, and soft-start resets. The internal
20V zener clamp from IN_HIGH to SGND is not turned
off during thermal shutdown because clamping action
must be always active.

Figure 2. Power-Supply Sequencing Configurations

V

IN

V

L

V

L

V

IN

VLV+

OUTPUT2 OUTPUT1

V

L

SEQUENCING—OUTPUT 2 DELAYED WITH RESPECT TO OUTPUT 1. R1/C1 AND R2/C2 ARE SIZED FOR REQUIRED SEQUENCING.

DRAIN2

SOURCE2

MAX5098A

DRAIN1

SOURCE1

FB1FB2

EN1EN2

PGOOD1

V

L

OUTPUT2 OUTPUT1

V

L

VLV+

DRAIN2

SOURCE2

MAX5098A

DRAIN1

SOURCE1

EN1EN2

FB1FB2

R1R2

V

L

C1C2

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

18 ______________________________________________________________________________________

Applications Information

Setting the Switching Frequency

The controller generates the clock signal by dividing
down the internal oscillator f

OSC

or the SYNC input sig­nal when driven by an external oscillator. The switching
frequency equals half the internal oscillator frequency
(fSW= f

OSC

/2). The internal oscillator frequency is set

by a resistor (R

OSC

) connected from OSC to SGND. To

find R

OSC

for each converter switching frequency fSW,

use the formulas:

A rising clock edge on SYNC is interpreted as a syn­chronization input. If the SYNC signal is lost, the inter­nal oscillator takes control of the switching rate,
returning the switching frequency to that set by R

OSC

.

When an external synchronization signal is used, R

OSC

must be selected such that fSW= 1/2 f

SYNC

. When

f

SYNC

clock signal is applied, f

CKO

equals f

SYNC

wave­form, phase shifted by 180°. If the MAX5098A is run­ning without external synchronization, f

CKO

equals the

internal oscillator frequency f

OSC

.

Buck Converter

Effective Input Voltage Range

Although the MAX5098A converter can operate from
input supplies ranging from 5.2V to 19V, the input volt­age range can be effectively limited by the MAX5098A
duty-cycle limitations for a given output voltage. The
maximum input voltage is limited by the minimum on­time (t

ON(MIN)

):

where t

ON(MIN)

is 100ns. The minimum input voltage is

limited by the maximum duty cycle (D

MAX

= 0.82):

where V

DROP1

is the total parasitic voltage drops in the
inductor discharge path, which includes the forward
voltage drop (VD) of the rectifier, the series resistance

of the inductor, and the PCB resistance. V

DROP2

is the
total resistance in the charging path that includes the
on-resistance of the high-side switch, the series resis­tance of the inductor, and the PCB resistance.

Setting the Output Voltage

For 0.8V or greater output voltages, connect a voltage­divider from OUT_ to FB_ to SGND (Figure 3). Select R

B

(FB_ to SGND resistor) to between 1kΩ and 20kΩ.
Calculate RA(OUT_ to FB_ resistor) with the following
equation:

where V

FB_

= 0.8V (see the

Electrical Characteristics

table) and V

OUT_

can range from V

FB_

to 28V (boost

operation).

For output voltages below 0.8V, set the MAX5098A out­put voltage by connecting a voltage-divider from OUT_
to FB_ to BYPASS (Figure 3). Select RC(FB_ to BYPASS
resistor) in the 50kΩ range. Calculate RAwith the fol­lowing equation:

where V

FB_

= 0.8V, V

BYPASS

= 2V (see the

Electrical

Characteristics

table), and V

OUT_

can range from 0V to

V

FB_

.

⎣

⎦

Figure 3. Adjustable Output Voltage

Rk

()

OSC

Rk

()

OSC

10 721

Ω

=

Ω

=

.

f MHz

()

SW

12 184

.

f MHz

()

SW

V

IN MAX

()

≤

125

.

f MHz

≥

()

SW

0 920

.

125

.

f MHz

<

()

SW

0 973

.

V

OUT

tf

ON MIN SW

×

()

⎡

⎛

V

RR

⎢

=

AB

⎜

⎢

⎝

⎣

OUT

V

FB

⎤

⎞

_

⎥

−

1

⎟

⎥

⎠

_

⎦

SOURCE_

FB_

⎡

RR

AC

VV

=

⎢

VV

⎢

⎣

V

OUT_

R

A

−

FB OUT

__

BYPASS FB

BYPASS

⎤
⎥

−

⎥

_

⎦

FB_

R

C

V

IN MIN

⎡

VV

+

OUT DROP

=

⎢

D

MAX

⎤

1

+

VV

⎥

DROP DROP()

−

21

MAX5098A

V

OUT_

≥ 0.8V

R

B

MAX5098A

SOURCE_

R

A

V

OUT_

V

< 0.8V

OUT_

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 19

Inductor Selection

Three key inductor parameters must be specified for
operation with the MAX5098A: inductance value (L),
peak inductor current (IL), and inductor saturation cur­rent (I

SAT

). The minimum required inductance is a func­tion of operating frequency, input-to-output voltage
differential and the peak-to-peak inductor current (ΔIL).
A good compromise is to choose ΔILequal to 30% of
the full load current. To calculate the inductance, use
the following equation:

where VINand V

OUT

are typical values (so that efficien­cy is optimum for typical conditions). The switching fre­quency is set by R

OSC

(see the

Setting the Switching

Frequency

section). The peak-to-peak inductor current,
which reflects the peak-to-peak output ripple, is worse
at the maximum input voltage. See the

Output

Capacitor

section to verify that the worst-case output
ripple is acceptable. The inductor saturation current is
also important to avoid runaway current during output
overload and continuous short circuit. Select the I

SAT

to
be higher than the maximum peak current limits of 4.3A
and 2.6A for converter 1 and converter 2.

Input Capacitor

The discontinuous input current waveform of the buck
converter causes large ripple currents at the input. The
switching frequency, peak inductor current, and the
allowable peak-to-peak voltage ripple dictate the input
capacitance requirement. Note that the two converters
of the MAX5098A run 180° out-of-phase, thereby effec­tively doubling the switching frequency at the input.

The input ripple waveform would be unsymmetrical due
to the difference in load current and duty cycle
between converter 1 and converter 2. The worst-case
mismatch is when one converter is at full load while the
other converter is at no load or in shutdown. The input
ripple is comprised of ΔVQ(caused by the capacitor
discharge) and ΔV

ESR

(caused by the ESR of the
capacitor). Use ceramic capacitors with high ripple­current capability at the input, connected between
DRAIN_ and PGND_. Assume the contribution from the
ESR and capacitor discharge equal to 50%. Calculate
the input capacitance and ESR required for a specified
ripple using the following equations:

where

and

where

where I

OUT

is the maximum output current from either
converter 1 or converter 2, and D is the duty cycle for
that converter. The frequency of each individual con­verter is fSW. For example, at VIN= 12V, V

OUT

= 3.3V at

I

OUT

= 2A, and with L = 3.3µH, the ESR and input
capacitance are calculated for a peak-to-peak input rip­ple of 100mV or less, yielding an ESR and capacitance
value of 20mΩ and 6.8µF for 1.25MHz frequency. At low
input voltages, also add one electrolytic bulk capacitor
of at least 100µF on the converters’ input voltage rail.
This capacitor acts as an energy reservoir to avoid pos­sible undershoot below the undervoltage lockout thresh­old during power-on and transient loading.

Output Capacitor

The allowable output ripple voltage and the maximum
deviation of the output voltage during step load cur­rents determines the output capacitance and its ESR.
The output ripple is comprised of ΔVQ(caused by the
capacitor discharge) and ΔV

ESR

(caused by the ESR of
the capacitor). Use low-ESR ceramic or aluminum elec­trolytic capacitors at the output. For aluminum elec­trolytic capacitors, the entire output ripple is
contributed by ΔV

ESR

. Use the ESR

OUT

equation to cal­culate the ESR requirement and choose the capacitor
accordingly. If using ceramic capacitors, assume the
contribution to the output ripple voltage from the ESR
and the capacitor discharge are equal. Calculate the
output capacitance and ESR required for a specified
ripple using the following equations:

VVV

()

OUT IN OUT

L

=

××−Δ

Vf I

IN SW L

ESR

V

Δ

ESR

=

IN

I

OUT

I

Δ

L

+

2

−

VV V

()

ΔI

IN OUT OUT

=

L

Vf L

IN SW

IDD

OUT

C

=

IN

Δ

D

=

×

××

−1

×

()

×

Vf

QSW

V

OUT

V

IN

ESR

OUT

=

C

OUT

Δ

V

ESR

=

Δ

I

L

Δ

I

L

××

Δ8

Vf

QSW

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

20 ______________________________________________________________________________________

where

ΔILis the peak-to-peak inductor current as calculated
above and fSWis the individual converter’s switching
frequency.

The allowable deviation of the output voltage during
fast transient loads also determines the output capaci­tance and its ESR. The output capacitor supplies the
step load current until the controller responds with a
greater duty cycle. The response time (t

RESPONSE

)
depends on the closed-loop bandwidth of the convert­er. The high switching frequency of the MAX5098A
allows for higher closed-loop bandwidth, reducing
t

RESPONSE

and the output capacitance requirement.
The resistive drop across the output capacitor ESR and
the capacitor discharge causes a voltage droop during
a step load. Use a combination of low-ESR tantalum or
polymer and ceramic capacitors for better transient
load and ripple/noise performance. Keep the maximum
output voltage deviation within the tolerable limits of the
electronics being powered. When using a ceramic
capacitor, assume 80% and 20% contribution from the
output capacitance discharge and the ESR drop,
respectively. Use the following equations to calculate
the required ESR and capacitance value:

where I

STEP

is the load step and t

RESPONSE

is the
response time of the controller. Controller response
time depends on the control-loop bandwidth.

Boost Converter

The MAX5098A can be configured for step-up conver­sion since the internal MOSFET can be used as a low­side switch. Use the following equations to calculate
the values for the inductor (L

MIN

), input capacitor (CIN),

and output capacitor (C

OUT

) when using the converter

in boost operation.

Inductor

Choose the minimum inductor value so the converter
remains in continuous mode operation at minimum out­put current (I

OMIN

).

where

The V

D

is the forward voltage drop of the external
Schottky diode, D is the duty cycle, and VDSis the volt­age drop across the internal MOSFET switch. Select
the inductor with low DC resistance and with a satura­tion current (I

SAT

) rating higher than the peak switch

current limit of 4.3A (I

CL1

) and 2.6A (I

CL2

) of converter

1 and converter 2, respectively.

Input Capacitor

The input current for the boost converter is continuous
and the RMS ripple current at the input is low. Calculate
the capacitor value and ESR of the input capacitor
using the following equations.

where

where VDSis the voltage drop across the internal
MOSFET switch. ΔILis the peak-to-peak inductor ripple
current as calculated above. ΔVQis the portion of input
ripple due to the capacitor discharge and ΔV

ESR

is the

contribution due to ESR of the capacitor.

Output Capacitor

For the boost converter, the output capacitor supplies
the load current when the main switch is ON. The
required output capacitance is high, especially at high­er duty cycles. Also, the output capacitor ESR needs to
be low enough to minimize the voltage drop due to the
ESR while supporting the load current. Use the follow­ing equation to calculate the output capacitor for a
specified output ripple tolerance.

where I

PK

is the peak inductor current as defined in the

Power Dissipation

section for the boost converter, IOis

the load current, ΔVQis the portion of the ripple due to

ΔΔΔVVV

O RIPPLE ESR Q_

≅+

V

Δ

ESR

=

I

STEP

×

It

STEP RESPONSE

Δ

V

Q

ESR

OUT

=

C

OUT

L

MIN

VD

=

IN

2

fVI

×××

SW O OMIN

2

×

−

+

−

+

VVV

OD IN

D

=

VVV

ODDS

Δ

I

C

IN

ESR

=

8

=

L

××

Δ

V

Δ

fV

SW Q

ESR

Δ

I

L

ΔI

IN DS

=

L

×

Lf

SW

×

−

VV D

()

V

Δ

=

I

=

ESR

PK

×

ID

O MAX

×

Δ

Vf

QSW

ESR

C

OUT

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 21

the capacitor discharge, and ΔV

ESR

is the contribution

due to the ESR of the capacitor. D

MAX

is the maximum

duty cycle at minimum input voltage.

Power Dissipation

The MAX5098A includes two internal power MOSFET
switches. The DC loss is a function of the RMS current in
the switch while the switching loss is a function of switch­ing frequency and instantaneous switch voltage and cur­rent. Use the following equations to calculate the RMS
current, DC loss, and switching loss of each converter.
The MAX5098A is available in a thermally enhanced
package and can dissipate up to 2.7W at +70°C ambient
temperature. The total power dissipation in the package
must be limited so that the operating junction tempera­ture does not exceed its absolute maximum rating of
+150°C at maximum ambient temperature.

For the buck converter:

where

See the

Electrical Characteristics

table for the

R

ON(MAX)

maximum value.

For the boost converter:

where V

DS

is the drop across the internal MOSFET and

η is the efficiency. See the

Electrical Characteristics

table for the R

ON(MAX)

value.

where t

R

and tFare rise and fall times of the internal

MOSFET. tFcan be measured in the actual application.

The supply current in the MAX5098A is dependent on
the switching frequency. See the

Typical Operating

Characteristics

to find the supply current of the
MAX5098A at a given operating frequency. The power
dissipation (P

S

) in the device due to supply current

(I

SUPPLY

) is calculated using following equation.

PS = V

INMAX

x I

SUPPLY

The total power dissipation PTin the device is:

PT = P

DC1

+ P

DC2

+ P

SW1

+ P

SW2

+ P

S

where P

DC1

and P

DC2

are DC losses in converter 1 and

converter 2, respectively. P

SW1

and P

SW2

are switching

losses in converter 1 and converter 2, respectively.

Calculate the temperature rise of the die using the fol­lowing equation:

TJ = TCx (PT x θJC)

where θJCis the junction-to-case thermal impedance of
the package equal to +1.7°C/W. Solder the exposed
pad of the package to a large copper area to minimize
the case-to-ambient thermal impedance. Measure the
temperature of the copper area near the device at a
worst-case condition of power dissipation and use
+1.7°C/W as θJCthermal impedance.

Compensation

The MAX5098A provides an internal transconductance
amplifier with its inverting input and its output available
for external frequency compensation. The flexibility of
external compensation for each converter offers wide
selection of output filtering components, especially the
output capacitor. For cost-sensitive applications, use
aluminum electrolytic capacitors; for component size­sensitive applications, use low-ESR tantalum, polymer,
or ceramic capacitors at the output. The high switching
frequency of MAX5098A allows use of ceramic capaci­tors at the output.

Choose all the passive power components that meet
the output ripple, component size, and component cost
requirements. Choose the small-signal components for
the error amplifier to achieve the desired closed-loop

22

IIIII

RMS DC PK DC PK

⎛

=++×

⎝

PI R

=×

DC RMS DS ON MAX

II

II

VI tt f

IN O R F SW

=

SW

P

IIIII

RMS

22

=++×

DC PK

()

Δ

I

L

=×

PI R

DC RMS DS ON MAX

()

2

=

DC O

=+

PK O

×× +

I

=

IN

VV D

()

=

=

II

DC IN

=+

II

PK IN

2

()

Δ

I

L

−

2

Δ

I

L

2

()

4

()

DC PK

VI

×

OO

η

V

×

IN

×

−

IN DS

×

Lf

SW

Δ

I

L

−

2

Δ

I

L

2

()( )

D

MAX

⎞

×

⎠

3

×

D

MAX

×

3

P

SW

VI tt f

×× +

OIN R F SW

=

()

4

×

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

22 ______________________________________________________________________________________

bandwidth and phase margin. Use a simple pole-zero
pair (Type II) compensation if the output capacitor ESR
zero frequency is below the unity-gain crossover fre­quency (fC). Type III compensation is necessary when
the ESR zero frequency is higher than f

C

or when com­pensating for a continuous mode boost converter that
has a right-half-plane zero.

Use procedure 1 to calculate the compensation net­work components when f

ZERO,ESR

< fC.

Buck Converter Compensation

Procedure 1 (See Figure 4)

1) Calculate the f

ZERO,ESR

and LC double-pole fre-

quencies:

2) Select the unity-gain crossover frequency:

If the f

ZERO,ESR

is lower than fCand close to fLC, use a
Type II compensation network where RFCFprovides a
midband zero f

MID,ZERO

, and RFCCFprovides a high-

frequency pole.

3) Calculate modulator gain GMat the crossover fre­quency.

where V

OSC

is a peak-to-peak ramp amplitude equal to

1V.

The transconductance error amplifier gain is:

G

E/A

= gMx R

F

The total loop gain at fCshould be equal to 1:

GM x G

E/A

= 1

or

4) Place a zero at or below the LC double pole:

5) Place a high-frequency pole at f

P

= 0.5 x fSW.

Procedure 2 (See Figure 5)

If the output capacitor used is a low-ESR ceramic type,
the ESR frequency is usually far away from the targeted
unity crossover frequency (fC). In this case, Type III
compensation is recommended. Type III compensation
provides two-pole zero pairs. The locations of the zero
and poles should be such that the phase margin peaks
around f

C

. It is also important to place the two zeros at
or below the double pole to avoid the conditional stabil­ity issue.

1) Select a crossover frequency:

2) Calculate the LC double-pole frequency, f

LC

:

Figure 4. Type II Compensation Network

f

ZERO ESR

,

f

=

LC

=

2

1

ESR C

××

OUT

1

LC

2ππ

×

OUT OUT

f

SW

f

≤

C

20

V

G

IN

=×

M

V

ESR f L V

OSC C OUT OUT

ESR

+××

208π

()

×

V ESR f L V

OSC C OUT OUT

R

=

F

208π

+××

()

×××

.

V g ESR

IN M

×

V

OUT

R

1

FB_

-

g

M

+

V

R

REF

2

R

F

C

F

COMP_

C

CF

C

=

F

C

=

CF

×××

()

1

Rf

××

2π

FLC

C

F

fRC

SW F F

−205 1π .

.

f

SW

f

≤

C

20

f

LC

=

2π

1

LC

××

OUT OUT

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 23

3) Place a zero

where

and R

F

≥ 10kΩ.

4) Calculate CIfor a target unity crossover frequency, fC.

5) Place a pole
or 5 x f

C

, whichever

is lower,

6) Place a second zero, fZ2, at 0.2 x fCor at fLC,
whichever is lower.

7) Place a second pole at 1/2 the switching frequency.

Boost Converter Compensation

The boost converter compensation gets complicated
due to the presence of a right-half-plane zero
f

ZERO,RHP

. The right-half-plane zero causes a drop in
phase while adding positive (+1) slope to the gain
curve. It is important to drop the gain significantly below
unity before the RHP frequency. Use the following pro­cedure to calculate the compensation components:

1) Calculate the LC double-pole frequency, f

LC

, and

the right-half-plane-zero frequency.

where

Figure 5. Type III Compensation Network

Figure 6. Boost Application

f

Z

2

××

π

1

RC

FF

at f

..

075=

×

LC1

1

×××

fR

LC F

C

=

F

2075π .

fL C V

×× × ×

2π

C

=

I

C OUT OUT OSC

VR

×

IN F

2=××π

1

fC

××

π

PI

1

1

RC

II

at f

f

P

R

=

I

2

ZERO ESR1

,

R

1

=

fC

××

π

2

ZI

1

2

R

−

I

C

C

=

CF

×× ××

()

V

OUT

F

fRC

SW F F

C

CF

−205 1π .

V

L

MAX5098A

VDRV

BST_/VDD_

PGND_
PGND_

DRAIN_

DRAIN_

SOURCE_

SOURCE_

SGND

FB_

V+

V

OUT_

C

OUT

−1

f

LC

=

2π

××

D

LC

OUT OUT

R

I

R1

C

I

R2

FB_

V

REF

C

R

F

-

g

M

+

F

COMP_

f

ZERO RHP

,

D

R

MIN

()

=

DR

−

1

()

=

1

=

×

22π

V

IN

−

V

OUT

V

OUT

I

OUT MAX

()

L

OUT

MIN

()

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

24 ______________________________________________________________________________________

Target the unity-gain crossover frequency for:

2) Place a zero

where R

F

≥ 10kΩ.

3) Calculate CIfor a target crossover frequency, fC:

where ω

C

= 2π x fC:

4) Place a pole

5) Place the second zero

where

6) Place the second pole
the switching frequency.

Load-Dump Protection MOSFET

Select the external MOSFET with an adequate voltage
rating, V

DSS

, to withstand the maximum expected load­dump input voltage. The on-resistance of the MOSFET,
R

DS(ON)

, should be low enough to maintain a minimal
voltage drop at full load, limiting the power dissipation
of the MOSFET.

During regular operation, the power dissipated by the
MOSFET is:

P

NORMAL

= I

LOAD

2

x R

DS(ON)

where I

LOAD

is equal to the sum of both converters’

input currents.

The MOSFET operates in a saturation region during
load dump, with both high voltage and current applied.
Choose a suitable power MOSFET that can safely oper­ate in the saturation region. Verify its capability to sup­port the downstream DC-DC converters input current
during the load-dump event by checking its safe oper­ating area (SOA) characteristics. Since the transient
peak power dissipation on the MOSFET can be very
high during the load-dump event, also refer to the ther­mal impedance graph given in the data sheet of the
power MOSFET to make sure its transient power dissi­pation is kept within the recommended limits.

Improving Noise Immunity

In applications where the MAX5098A is subject to noisy
environments, adjust the controller’s compensation to
improve the system’s noise immunity. In particular, high­frequency noise coupled into the feedback loop causes
jittery duty cycles. One solution is to lower the crossover
frequency (see the

Compensation

section).

f

ZERO RHP

f

f

Z

C

=

F

≤

C

2

π

2075π .

,

5

1

RC

××

FF

at f

075=

. .

1

×××

fR

LC F

2

⎡

−1

()

⎢

⎣

f

P

2=××π

R

=

I

2π

C

=

I

VDLC

OSC C OUT OUT

ω

RC

fC

××

2

+

ω

RV

CFIN

1

at f

.

II

ZERO RHP1

1

ZERO RHP I

,

f

Z

1

=

RC

21

××π

,

R

1

1

fC

π

2=××

LC I

R

−

I

×

at f

I

LC1

⎤
⎥

⎦

.

LC2

C

=

CF

×× ××

()

2

××π

1

RC

FCF

−205 1π .

f

P

2

C

F

fRC

SW F F

12=

/

at

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 25

PCB Layout Guidelines

Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. This is especially
true for dual converters where one channel can affect
the other. Refer to the MAX5099 Evaluation Kit data
sheet for a specific layout example. Use a multilayer
board whenever possible for better noise immunity.
Follow these guidelines for good PCB layout:

1) For SGND, use a large copper plane under the IC
and solder it to the exposed paddle. To effectively
use this copper area as a heat exchanger between
the PCB and ambient, expose this copper area on
the top and bottom side of the PCB. Do not make a
direct connection from the exposed pad copper
plane to SGND underneath the IC.

2) Isolate the power components and high-current
path from the sensitive analog circuitry.

3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta­ble, jitter-free operation.

4) Connect SGND and PGND_ together at a single
point. Do not connect them together anywhere else
(refer to the MAX5099 Evaluation Kit data sheet for
more information).

5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full­load efficiency.

6) Ensure that the feedback connection to C

OUT

is

short and direct.

7) Route high-speed switching nodes (BST_/VDD_,
SOURCE_) away from the sensitive analog areas
(BYPASS, COMP_, and FB_). Use the internal PCB
layer for SGND as an EMI shield to keep radiated
noise away from the IC, feedback dividers, and
analog bypass capacitors.

Layout Procedure

1) Place the power components first, with ground ter­minals adjacent (inductor, C

IN_

, and C

OUT_

). Make
all these connections on the top layer with wide,
copper-filled areas (2oz copper recommended).

2) Group the gate-drive components (bootstrap
diodes and capacitors, and VLbypass capacitor)
together near the controller IC.

3) Make the DC-DC controller ground connections as
follows:

a) Create a small, signal ground plane underneath

the IC.

b) Connect this plane to SGND and use this plane

for the ground connection for the reference
(BYPASS), enable, compensation components,
feedback dividers, and OSC resistor.

c) Connect SGND and PGND_ together (this is the

only connection between SGND and PGND_).
Refer to the MAX5099 Evaluation Kit data sheet
for more information.

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

26 ______________________________________________________________________________________

Figure 7. 4.5V to 5.5V Operation

VIN

= 4.5V

IN

V

C1

TO 5.5V

PGND

PGND

VOUT2

R23

C5

L2

D5

C14

D4

VDRV

31

1

BST2/VDD2

BST1/VDD1

26

C6

32

SOURCE2

SOURCE1

25

24

L1

SOURCE2

SOURCE1

D2

C15

DRAIN2
DRAIN2

C4

DRAIN1
DRAIN1

22 23 2 3

C19

V+

13

GATE

10

ON/OFF

11

12

IN_HIGH

VDRV

D1

29

CKO

28

30

CLOCK OUT

R15

PGND1

MAX5098A

PGND2

C16

R16

C17

R18

6

7

FB2

COMP2

FB1

COMP1

19

18

R9

C9

R7

SGND

R17

C21

4

5

9

EN2

PGOOD2

PGOOD1

21

C20

SYNC

L

1427

V

IN

V

VDRV

SGND

BYPASS

OSC

FSEL_1

EN1

20

17

L

V

C12 C13

15

C11

16

R12

8

C8

R8

R22

SGND

VOUT1

C7

R6

PGND

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost

Converter with 80V Load-Dump Protection

______________________________________________________________________________________ 27

Typical Application Circuit

VOUT2 = 3.3V

AT 1A

PGND

VOUT2

R23

10kΩ

1%

C5

22μF

1%

R15

37.4kΩ

C16

270pF

L2

4.7μH

C14

D4

VDRV

C15

10μF

25V

31
DRAIN2
DRAIN2

22 23 2 3

13

10

11

12

VDRV

V+

GATE

DRAIN1
DRAIN1

ON/OFF

D1

BST2/VDD2

IN_HIGH

BST1/VDD1

262524

C4

10μF

25V

C19

1μF

25V

25V

C3

150μF

N1

C2

4.7μF

35V

R1

3.9kΩ

C1

22μF

100V

VIN

PGND

= 5.2V

TO 19V

IN

V

D5

0.1μF

1

32

29

SOURCE2

SOURCE1

D2

CKO

28

30

CLOCK OUT

PGND1

PGND2

SOURCE2

SOURCE1

C6

0.1μF

L1

4.7μH

6

MAX5098A

19

R16

12.1kΩ

1%

C17

2700pF

R18

7.15Ω

7

FB2

COMP2

FB1

COMP1

18

R9

12.7Ω

C9

2700pF

R7

1%

10kΩ

SGND

1%

R17

976Ω

C21

56pF

4

5

9

EN2

SYNC

PGOOD2

L

V

VDRV

SGND

BYPASS

OSC

PGOOD1

FSEL_1

EN1

20

17

21

L

V

C20

33pF

C13

4.7μF

1427

1Ω

R21

C12

2.2μF

VDRV

15

C11

0.22μF

16

R12

6.49Ω

8

C8

VOUT1

VOUT1 =

R6

1%

52.3kΩ

C7

22μF

R22

PGND

5V AT 2A

R8

270pF

1%

976Ω

1%

10kΩ

SGND

MAX5098A

Dual, 2.2MHz, Automotive Buck or Boost
Converter with 80V Load-Dump Protection

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

28

____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

Pin Configuration

Chip Information

PROCESS: BiCMOS

Package Information

For the latest package outline information, go to

www.maxim-ic.com/packages

.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

32 TQFN T3255+4

21-0140

TOP VIEW

SOURCE1

SOURCE1

BST1/VDD1

BST2/VDD2

SOURCE2

*EP = EXPOSED PAD.

VDRV

CKO

PGND1

PGND2

25

26

27

28

29

30

31

32

12

SOURCE2

DRAIN1

PGOOD1

EN1

FB1

EN2

COMP1

FB2

COMP2

DRAIN1

2324 22 20 19 18

21

MAX5098A

*EP

+

4567

3

DRAIN2

DRAIN2

PGOOD2

TQFN

(5mm x 5mm)

FSEL_1

17

8

OSC

16

15

14

13

12

11

10

9

BYPASS

SGND

V

L

V+

IN_HIGH

ON/OFF

GATE

SYNC