Rainbow Electronics MAX1791 User Manual

General Description

The MAX1762/MAX1791 PWM step-down controllers
provide high efficiency, excellent transient response,
and high DC output accuracy needed for stepping
down high-voltage batteries to generate low-voltage
CPU core, I/O, and chipset RAM supplies in notebook
computers and PDAs.

Maxim’s proprietary Quick-PWM™ pulse-width modula­tor is a free-running constant on-time type with input
feed-forward. Its high operating frequency (300kHz)
allows small external components to be utilized in PC
board area-critical applications such as subnotebook
computers and smart phones. PWM operation occurs
at heavy loads, and automatic switchover to pulse-skip­ping operation occurs at lighter loads. The external
high-side P-channel and low-side N-channel MOSFETs
require no bootstrap components. The MAX1762/
MAX1791 are simple, easy to compensate, and do not
have the noise sensitivity of conventional fixed-frequen­cy current-mode PWMs.

These devices achieve high efficiency at a reduced
cost by eliminating the current-sense resistor found in
traditional current-mode PWMs. Efficiency is further
enhanced by their ability to drive synchronous-rectifier
MOSFETs. The MAX1762/MAX1791 come in a 10-pin
µMAX package and offer two fixed voltages (Dual
Mode™) for each device, 1.8V/2.5V/adj (MAX1762) and

3.3V/5.0V/adj (MAX1791).

________________________Applications

Notebooks Handy-Terminals

Subnotebooks PDAs

Digital Cameras Smart Phones

1.8V/2.5V Logic
and I/O Supplies

Features

♦ High Operating Frequency (300kHz)

♦ No Current-Sense Resistor

♦ Accurate Current Limit

♦ ±1% Total DC Error over Line and During

Continuous Conduction

♦ Dual Mode Fixed Output

1.8V/2.5V/adj (MAX1762)

3.3V/5.0V/adj (MAX1791)

♦ 0.5V to 5.5V Output Adjust Range

♦ 5V to 20V Input Range

♦ Automatic Light-Load Pulse Skipping Operation

♦ Free-Running On-Demand PWM

♦ Foldback Mode™ UVLO

♦ PFET/NFET Synchronous Buck

♦ 4.65V at 25mA Linear Regulator Output

♦ 5µA Shutdown Supply Current

♦ 230µA Quiescent Supply Current

♦ 10-Pin µMAX Package

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

________________________________________________________________ Maxim Integrated Products 1

19-1923; Rev 0; 1/01

Ordering Information

Typical Operating Circuit

1

2

3

4

5

10

9

8

7

6

DH

CS

DLOUT

FB

REF

VL

MAX1762
MAX1791

µ

MAX

TOP VIEW

GNDSHDN

VP

Pin Configuration

For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.

Quick-PWM, Dual Mode, and Foldback Mode are a trade­marks of Maxim Integrated Products.

V

BATT

(5V TO 20V)

PART TEMP. RANGE PIN-PACKAGE

MAX1762EUB -40°C to +85°C 10 µMAX
MAX1791EUB -40°C to +85°C 10 µMAX

VL

MAX1762
MAX1791

REF

FB

OUT

SHDN

VP

DH

CS

DL

GND

V

OUT

1.8V/3.3V

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

VP, SHDN to GND ..................................................-0.3V to +22V

VP to VL ..................................................................-0.3V to +22V

OUT, VL to GND .......................................................-0.3V to +6V

DL, FB, REF to GND ....................................-0.3V to (VL + 0.3V)

DH to GND....................................................-0.3V to (VP + 0.3V)

CS to GND ....................................................-2.0V to (VP + 0.3V)

REF Short Circuit to GND ...........................................Continuous

Continuous Power Dissipation (T

A

= +70°C)

10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW

Operating Temperature .......................................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature.........................................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

ELECTRICAL CHARACTERISTICS

(VVP= 15V, VL enabled, CVL= 1µF, C

REF

= 0.1µF, TA= 0 to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 1)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

VP Input Voltage Range V

VL Input Voltage Range V

OUT Output Voltage
(MAX1762, 1.8V Fixed)

OUT Output Voltage
(MAX1762, 2.5V Fixed)

OUT Output Voltage
(MAX1791, 3.3V Fixed)

OUT Output Voltage
(MAX1791, 5V Fixed)

OUT Output Voltage (Adj Mode)

Output Voltage Adjust Range 0.5 5.5 V
OUT Input Resistance Adjustable-output mode 300 800 1700 kΩ

FB Input Bias Current VFB = 1.3V -0.1 0.1 µA

Soft-Start Ramp Time Zero to full I

On-Time (Note 2) t

Minimum Off-Time (Note 2) t

VL Quiescent Supply Current

VP Quiescent Supply Current

VL Shutdown Supply Current VVL = 5V, SHDN = GND 2 15 µA
VP Shutdown Supply Current S HD N = GN D , m easur ed at V P , V

VL Output Voltage I

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

VP

VL (overdriven) 4.75 5.25 V

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode

VVP = 7V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode

V

= 5V to 20V, VVL = 4.75V to 5.25V,

VP

FB = OUT, continuous conduction mode

LIM

V

= 1.25V, VVP = 6V 666 740 814

OUT

V

= 5V, VVP = 6V 2550 2830 3110

OUT

FB = GND, V
regulation point

FB = GND, OUT forced
above the regulation point,
V

= 20V

VP

= 0 to 25mA, VVP = 5V to 20V 4.5 4.65 4.75 V

LOAD

= 5V, OUT forced above the

VL

VVL = float 227 410

V

= 5V 93 200

VL

= 0 or 5V 4 12 µA

V L

V

V

V

V

VL

OUT

OUT

OUT

OUT

ON

OFF

520V

1.773 1.8 1.827 V

2.463 2.5 2.538 V

3.250 3.3 3.350 V

4.925 5 5.075 V

1.231 1.250 1.269 V

1700 µs

300 400 500 ns

153 260 µA

ns

µA

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VVP= 15V, VL enabled, CVL= 1µF, C

REF

= 0.1µF, TA= 0 to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 1)

Reference Voltage V

Reference Load Regulation I
REF Sink Current REF in regulation 10 µA

REF Fault Lockout Voltage Falling edge 1.6 V

Output Undervoltage Threshold
(Foldback)

Output Undervoltage Lockout
Time (Foldback)

Current Limit Threshold V

Thermal Shutdown Threshold Hysteresis = 10oC 160

VL Undervoltage Lockout
Threshold

DH Gate Driver On-Resistance VVP = 6V to 20V, measure at 50mA 5 8 Ω

DL Gate Driver On-Resistance
(Pullup)

DL Gate Driver On-Resistance
(Pulldown)

DH Gate Driver Source/Sink
Current

DL Gate Driver Sink Current VDL = 2.5V 0.9 A

DL Gate Driver Source Current V

SHDN Logic Input High
Threshold Voltage

SHDN Logic Input Low
Threshold Voltage

Dual Mode Threshold Voltage

SHDN Logic Input Current SHDN = 0 or 5V -2 2 µA

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

= 4.75V to 5.25V, no load 1.98 2 2.02 V

VL

= 0 to 50µA 0.01 V

REF

With respect to regulation point, no load 60 70 80 %

ILIM

V

V

From SHDN signal going high V
regulation point

Rising edge, hysteresis = 20mV, PWM
disabled below this level

DL, high state, measure at 50mA 5 8 Ω

DL, low state, measure at 50mA 1 5 Ω

V

= 3V, VVP = 6V 0.6 A

DH

= 2.5V 0.5 A

DL

IH

IL

MAX1762 V

MAX1791 V

MAX1762 V

MAX1791 V

= 1.8V fixed

OUT

= 3.3V fixed

OUT

= 2.5V fixed

OUT

= 5V fixed

OUT

OUT

< 0.6 x

10 20 42 ms

-90 -100 -110 mV

4.1 4.4 V

1.6 V

0.6 V

50 100 150 mV

2.5 3.25 4 V

o

C

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS

(VVP= 15V, VL enabled, CVL= 1µF, C

REF

= 0.1µF, TA= -40 to +85°C, unless otherwise noted.) (Note 1)

VP Input Voltage Range V

VL Input Voltage Range V

OUT Output Voltage
(MAX1762, 1.8V Fixed)

OUT Output Voltage
(MAX1762, 2.5V Fixed)

OUT Output Voltage
(MAX1791, 3.3V Fixed)

OUT Output Voltage
(MAX1791, 5V Fixed)

OUT Output Voltage (adj Mode)

FB Input Bias Current VFB = 1.3V -0.2 0.2 µA

On-Time (Note 2) t

Minimum Off-Time (Note 2) t

VL Quiescent Supply Current

VP Quiescent Supply Current

VL Shutdown Supply Current VVL = 5V, SHDN = GND 15 µA
VP Shutdown Supply Current S HD N = GN D , m easur ed at V P , V

VL Output Voltage I

Reference Voltage VVL = 4.75V to 5.25V, no load 1.98 2.02 V

Reference Load Regulation I
REF Sink Current REF in regulation 10 µA

Output Undervoltage Threshold
(Foldback)

Output Undervoltage Lockout
Time (Foldback)

Current-Limit Threshold V

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

VP

VL (overdriven) 4.75 5.25 V

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode

VVP = 5V to 20V, VVL = 4.75V to 5.25V,
FB = GND, continuous conduction mode

VVP = 7V to 20V, VVL = 4.75V to 5.25V,
FB = VL, continuous conduction mode

V

= 5V to 20V, VVL = 4.75V to 5.25V,

VP

FB = OUT, continuous conduction mode

V

= 1.25V, VVP = 6V 666 814

OUT

= 5V, V

V

OUT

FB = GND, V
regulation point

FB = GND, OUT forced above
the regulation point V

= 0 to 25mA, VVP = 5V to 20V 4.5 4.75 V

LOAD

= 0 to 50µA 0.01 V

REF

With respect to regulation point, no load 60 80 %

From SHDN signal going high, V
regulation point

= 6V 2550 3110

VP

= 5V, OUT forced above the

VL

V

= float 410

V L

VP

= 20V

V

= 5V 200

V L

= 0 or 5V 12 µA

V L

< 0.6 x

OUT

V

V

V

V

VL

OUT

OUT

OUT

OUT

ON

OFF

ILIM

520V

1.773 1.827 V

2.463 2.538 V

3.250 3.350 V

4.925 5.075 V

1.231 1.269 V

250 550 ns

10 42 ms

-90 -110 mV

ns

260 µA

µA

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

_______________________________________________________________________________________ 5

Note 1: Specifications to -40°C are guaranteed by design, not production tested.
Note 2: One-shot times are measured at the DH pin (VP = 15V, C

DH

= 400pF, 90% point to 90% point; see drawing below for

measurement details).

ELECTRICAL CHARACTERISTICS (continued)

(VVP= 15V, VL enabled, CVL= 1µF, C

REF

= 0.1µF, TA= -40 to +85°C, unless otherwise noted.) (Note 1)

VL Undervoltage Lockout
Threshold

SHDN Logic Input High
Threshold Voltage

SHDN Logic Input Low
Threshold Voltage

Dual Mode Threshold Voltage

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

DH

90% 90%

Rising edge, hysteresis = 20mV, PWM
disabled below this level

V

IH

V

IL

MAX1762 V

MAX1791 V

MAX1762 V

MAX1791 V

t

ON

= 1.8V fixed

OUT

= 3.3V fixed

OUT

= 2.5V fixed

OUT

= 5V fixed

OUT

4.1 4.4 V

1.6 V

0.6 V

50 150 mV

2.5 4 V

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

6 _______________________________________________________________________________________

Typical Operating Characteristics

(TA = +25°C, unless otherwise noted.)

100

0

0.1 1 10 100 1000 10,000

MAX1762

EFFICIENCY vs. LOAD (2.5V)

20

MAX1762/91 toc01

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 5V

VVP = 12V

VVP = 7V

VVP = 18V

100

0

0.1 1 10 100 1000 10,000

MAX1791

EFFICIENCY vs. LOAD (3.3V)

20

MAX1762/91 toc05

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 18V

VVP = 12V

VVP = 7V

VVP = 5V

100

0

0.1 1 10 100 1000 10,000

MAX1791

EFFICIENCY vs. LOAD (3.0V)

20

MAX1762/91 toc06

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 7V

VVP = 18V

VVP = 12V

VVP = 5V

-0.6

-0.4

-0.5

-0.2

-0.3

0

-0.1

0.1

0105 152025

VL VOLTAGE ERROR vs. OUTPUT CURRENT

MAX1762/91 toc08

VL CURRENT (mA)

VL VOLTAGE ERROR (%)

VVP = 12V

V

OUT

= 2.5V

100

0

0.1 1 10 100 1000 10,000

MAX1791

EFFICIENCY vs. LOAD (5V)

20

MAX1762/91 toc04

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 7V

VVP = 18V

VVP = 12V

250

275

325

300

350

375

FREQUENCY vs. SUPPLY VOLTAGE

MAX1762/91 toc07

SUPPLY VOLTAGE (V)

FREQUENCY (kHz)

51181417

V

OUT

= 3.3V

V

OUT

= 5.0V

V

OUT

= 2.5V

V

OUT

= 1.8V

LOAD = 1A

100

0

0.1 1 10 100 1000 10,000

MAX1762

EFFICIENCY vs. LOAD (1.8V)

20

MAX1762/91 toc02

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 18V

VVP = 12V

VVP = 5V

VVP = 7V

100

0

0.1 1 10 100 1000 10,000

MAX1762

EFFICIENCY vs. LOAD (1V)

20

MAX1762/91 toc03

LOAD CURRENT (mA)

EFFICIENCY (%)

40

60

80

10

30

50

70

90

VVP = 7V

VVP = 12V

VVP = 5V

VVP = 18V

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

_______________________________________________________________________________________ 7

Typical Operating Characteristics (continued)

(TA = +25°C, unless otherwise noted.)

LOAD-TRANSIENT RESPONSE

MAX1762/91 toc12

V

OUT

AC-COUPLED
100mV/div

I

L

2A/div

100µs/div

V

VP

= 12V, IL = 0 TO 2A, V

OUT

= 2.5V

LOAD-TRANSIENT RESPONSE

MAX1762/91 toc13

V

OUT

AC-COUPLED
100mV/div

I

LOAD

2A/div

100µs/div

V

VP

= 12V, I

LOAD

= 0 TO 2A, V

OUT

= 1.8V

LINE-TRANSIENT RESPONSE

MAX1762/91 toc11

VP
1V/div

7V

V

OUT

AC-COUPLED
20mV/div

100µs/div

V

VP

= 7.5V TO 8V, I

LOAD

= 0, V

OUT

= 2.5V

SHUTDOWN AND STARTUP WAVEFORMS

(I

L

= 300mA)

MAX1762/91 toc15

V

OUT

2V/div

I

LX

1A/div

SHDN
5V/div

2ms/div

V

VP

= 8V, I

LOAD

= 300mA, V

OUT

= 2.5V

SHUTDOWN AND STARTUP WAVEFORMS

(I

L

= 2.5A)

MAX1762/91 toc16

I

LX

2A/div

SHDN
5V/div

V

OUT

2V/div

2ms/div

V

VP

= 8V, I

LOAD

= 2.5A, V

OUT

= 2.5V

OUTPUT OVERLOAD WAVEFORMS

MAX1762/91 toc14

100µs/div

V

VP

= 12V, I

LOAD

= 0 TO 3A, V

OUT

= 2.5V

V

OUT

AC-COUPLED

I

LOAD

2A/div

4.0

4.6

4.4

4.2

4.8

5.0

5.2

5.4

5.6

5.8

6.0

51181417

SUPPLY CURRENT vs. INPUT VOLTAGE

(SHUTDOWN)

MAX1762/91 toc10

VP (V)

SUPPLY CURRENT (µA)

NO LOAD

0

100

50

200

150

250

300

51181417

SUPPLY CURRENT vs. INPUT VOLTAGE

MAX1762/91 toc09

VP (V)

SUPPLY CURRENT (µA)

V

OUT

= 3.3V

NO LOAD

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

8 _______________________________________________________________________________________

Standard Application Circuit

The standard application circuit (Figure 1) generates a
low-voltage output for general-purpose use in notebook
computers (I/O supply, fixed CPU, core supply, and
DRAM supply). This DC-DC converter steps down bat­tery voltage from 5V to 20V with high efficiency and
accuracy to a fixed voltage of 1.8V/2.5V/adj (MAX1762)
or 3.3V/5.0V/adj (MAX1791). Both the MAX1762 and
MAX1791 can be configured for adjustable output volt­ages (V

OUT

> 1.25V), using a resistive voltage-divider

from V

OUT

to FB to adjust the output voltage (Figure 2).

Similarly, Figure 3 shows an application circuit for V

OUT

< 1.25V, where a resistive voltage-divider from REF to
FB is used to set the output voltage. Figure 4 shows
how to set the regulator’s current limit with an external
sense resistor from CS to GND. Table 1 lists the com­ponents for each application circuit, and Table 2 con­tains contact information for the component
manufacturers.

Detailed Description

The MAX1762/MAX1791 step-down controllers are tar­geted at low-voltage chipsets and RAM power supplies
for notebook and subnotebook computers, with addi­tional applications in digital cameras, PDAs, and
handy-terminals. Maxim’s proprietary Quick-PWM
pulse-width modulator (Figure 5) is specifically
designed for handling fast load steps while maintaining
a relatively constant operating frequency (300kHz) over
a wide range of input voltages (5V to 20V). The
MAX1762 has fixed 1.8V or 2.5V outputs, while the
MAX1791 has fixed 3.3V or 5.0V output voltages. Using
an external resistive divider, V

OUT

can be set between

0.5V and 5.5V on either device. Quick-PWM architec­ture circumvents the poor load-transient response of
fixed-frequency current-mode PWMs. This type of
design avoids the problems commonly encountered
with conventional constant-on-time and constant-off­time PWM schemes.

Pin Description

PIN NAME FUNCTION

+4.65V Linear Regulator Output. Serves as the supply input for the DL gate driver and supplies up to

1VL

2 REF

3FB

4 OUT

5 SHDN

6 GND Analog and Power Ground

7 DL Low-Side Gate Driver Output. DL swings between VL and GND.

8CS

9 DH High-Side Gate Driver Output. DH swings between VP and GND.

10 VP

25mA to external loads. VL can be overdriven using an external 5V supply. Bypass VL to GND with
at least a 1µF ceramic capacitor.

2V Reference Voltage Output. Bypass to GND with 0.1µF ceramic capacitor. REF can deliver up to
50µA for external loads.

Feedback Input. Connect to an external resistive divider from OUT to GND in adjustable version.
Regulates to 1.25V. FB also serves as Dual Mode select pin. Connect FB to GND for a fixed 1.8V
( M AX 1762) or 3.3V (M AX 1791) outp ut, or to VL for a fi xed 2.5V (M AX 1762) or 5.0V ( M AX 1791) outp ut.

Output Voltage Connection. OUT is used for sensing the output voltage to determine the on-time and
also serves as the feedback input in fixed-output modes.

Shutdown Input. Connect to a voltage less than V
voltage greater than V

Current-Sense Connection. For lossless current sensing, connect CS to the junction of the MOSFETs
and inductor. For more accurate current sensing, connect CS to a current-sense resistor from the
source of the low-side switch to GND.

Battery Voltage Supply Input. Used for PWM one-shot timing and as the input for the VL regulator
and DH gate drivers.

(>1.6V) for normal operation.

IH

(<0.6V) to shut down the device. Connect to a

IL

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

_______________________________________________________________________________________ 9

Figure 1. Typical Application Circuit for Fixed Voltage

Figure 2. Typical Application Circuit for Adjustable Output V

OUT

> 1.25V

Figure 3. Typical Application Circuit for V

OUT

< 1.25V

V

VP

C2

C3

0.1µF

1µF

VL

REF

FB

OUT

SHDN

MAX1762
MAX1791

DH

CS

GND

1µF

10Ω

Q1

L1

7µH

Q2

220µF

V

OUT

C4

VP

DL

10µF

C1

V

10µF

10µF

C1

V

VP

C1

VP

C2

1µF

C3

0.1µF

0.1µF

C2

1µF

VL

REF

C3

FB

OUT

SHDN

VL

REF

R1

FB

R2

OUT

MAX1762
MAX1791

MAX1762
MAX1791

VP

DH

CS

DL

GND

VP

DH

CS

DL

1µF

1µF

10Ω

10Ω

Q1

L1

7µH

C4

Q2

220µF

Q1

L1

7µH

C4

220µF

Q2

V

OUT

R1

R2

V

OUT

SHDN

GND

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

10 ______________________________________________________________________________________

Figure 4. Operation with External Current-Sense Resistor

Table 1. Component Selection for Standard Applications

V

VP

C2

1µF

VL

MAX1762
MAX1791

10Ω

VP

1µF

10µF

C1

DH

CS

DL

GND

C3

0.1µF

REF

FB

OUT

SHDN

COMPONENT 1.8V/2.5V/3.3V/5.0V AT 2A 1V AT 2A

Input Voltage Range 5V to 20V 5V to 20V

Inductor (µH) 7 5.2

L1 Inductor

Q1 MOSFETS

C1 Input Capacitor

C2 VL Cap

C3 REF Cap

C4 Output Cap

CDRH104-7R0NC

Sumida

NDS8958A

Fairchild

TMK432BJ106KM

Taiyo Yuden

EMK3160J105KL

Taiyo Yuden

UMK316BI104KH

Taiyo Yuden

10TPB220M

Sanyo

Q1

L1

10µH

150µF

Q2

R

S

V

OUT

C4

CDRH104-5R2NC

Sumida

SI4539ADY

Fairchild

TMK432BJ106

Taiyo Yuden

LMK316BJ475

Taiyo Yuden

UMK316BI104KH

Taiyo Yuden

6TPB150M

Sanyo

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

______________________________________________________________________________________ 11

VP Input and VL Logic Supply

An internal linear regulator supplied by VP produces
the +4.65V supply (VL) that powers the PWM controller,
logic, reference, and other blocks within the
MAX1762/MAX1791. This +4.65V low-dropout linear
regulator can supply up to 25mA for external loads.
Bypass VL to GND with at least a 1µF ceramic capaci­tor. VVPcan range between 5V and 20V. VL is turned

off when the device is in shutdown and drops by
approximately 500mV during a fault condition, such as
when the output is short circuited to ground, and recov­ers when SHDN is cycled or power is reset. If VL is not
driven externally, then V

VP

should be at least 5V to
ensure operation. If VVPis running from a 5V (±10%)
supply, VVPshould be externally connected to VL.

Table 2. Component Manufacturers

Figure 5. Functional Block Diagram

MANUFACTURER USA PHONE WEBSITE INFO

Coiltronics 561-241-7876 www.coiltronics.com

Fairchild Semiconductor 408-822-2181 www.fairchildsemi.com

Sanyo 619-661-6835 www.secc.co.jp

USA 847-956-0666

Sumida

www.sumida.com

Japan 81-3-3607-5111

Taiyo Yuden 408-573-4150 www.t-yuden.com

10Ω

1µF

SHDN

REF

DH

DRIVER

VOS

-100mV

DL

DRIVER

VP

DH

CS

DL

GND

OUT

VP

ON-TIME

COMPUTE

TON

TRIG

1-SHOT

VP

VL

C

VL

C

REF

LINEAR

ON/OFF

CONTROL

VP

REG

V

2V

REF

OUT

Q

REF

TIMER

MAX1762
MAX1791

Q

Q

REF

-30%

T

T

1-SHOT

UVP

LATCH

OFF

ON

TRIG

S

Q

FEEDBACK

MUX

(FIGURE 9)

S

Q

R

ILIM

OUT

FB

VL

R

V

IN

C

IN

Q1

OUT

C

OUT

Q2

FB

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

12 ______________________________________________________________________________________

Overdriving the VL regulator with an external 5V supply
also increases the MAX1762/MAX1791s’ efficiency.

The MAX1762/MAX1791 include an input undervoltage
lockout (UVLO) circuit that prevents the device from
switching until VL > 4.4V (max). UVLO ensures there is
a sufficient drive for the external MOSFETs, prevents
the high-side MOSFET from being turned on for near
100% duty cycle, and keeps the output in regulation.

Voltage Reference (REF)

The 2V reference (REF) is accurate to ±1% over tem­perature, making REF useful as a precision system ref­erence. Bypass REF to GND with a 0.1µF (min) ceramic
capacitor. REF can supply up to 50µA for external
loads. However, if tight-accuracy specs for either V

OUT

or REF are essential, avoid loading REF. Loading slight­ly reduces the main output voltage by an amount that
tracks the reference-voltage load regulation error.

Free-Running Constant On-Time PWM

Controller with Input Feed-Forward

The PWM control architecture is a quasi-fixed-frequen­cy constant on-time current-mode type with voltage
feed-forward. This architecture relies on the output rip­ple voltage to provide the PWM ramp signal; thus, the
output filter capacitor’s ESR acts as a feedback resis­tor. The control algorithm is very simple. The high-side
switch on-time is determined solely by a one-shot
whose period is inversely proportional to input voltage
and directly proportional to output voltage. There is
another one-shot that sets a minimum amount of off­time (500ns max). The on-time one-shot triggers when
all of the following conditions are met: the error com­parator is low, the low-side switch current is below the
current-limit threshold, and the minimum off-time one­shot has timed out.

On-Time One-Shot

The on-time of the one-shot is inversely proportional to
the battery voltage as measured by the VP input, and
directly proportional to the output voltage sensed at
OUT:

where K is internally fixed at 3.349µs, and 0.075V is a
factor that accounts for the expected drop across the
synchronous switch. This arrangement maintains a
switching frequency that is nearly constant as V

BATT

,

I

LOAD

, and V

OUT

are changed. Table 3 shows the oper-

ating frequency range for the MAX1762/MAX1791.

Note that the output voltage adjust range for continu­ous-conduction operation is restricted by the non-

adjustable 0.5µs (max) minimum off-time. Worst-case
dropout performance is determined by the minimum
on-time spec. The worst-case duty factor limit is:

with V

BATT

= 6V and V

OUT

= 5V. Therefore, with IR volt­age drops in the loop included, the minimum input volt­age to achieve V

OUT

= 5V is about 6.1V, using the
step-down transfer function equation for duty cycle (DC
= V

OUT/VIN

). Typical units exhibit better performance.
Note that transient response is somewhat degraded
near dropout, and the circuit may need additional bulk
output capacitance to support fast load changes.

Automatic Pulse-Skipping Switchover

This PWM control algorithm automatically switches over
to pulse-skipping operation at light loads. The
MAX1762/MAX1791 truncates the low-side switch’s on­time when the inductor current drops to zero. The load
current level at which pulse-skipping/PWM crossover
occurs is equal to 1/2 the peak-to-peak ripple current,
which is a function of the inductor value (Figure 6).

The inductor current is never allowed to go negative. If
the output voltage is above its regulation point and the
inductor current reaches zero, the low-side driver is
switched off. Once the output voltage falls below its
regulation point, the high-side driver is switched on.
This causes a dead time in between when the high­side and low-side drivers are on, skipping pulses and
resulting in the switching frequency slowing at light
loads, thereby improving efficiency.

MOSFET Gate Drivers

The DH and DL drivers are optimized for driving moder­ate-size power MOSFETs. This is consistent with the
low duty factor seen in the notebook CPU environment
where a large V

BATT

- V

OUT

differential exists. The high­side driver (DH) is rated for 0.6A source/sink capability
and swings from VP to GND. The low-side driver (DL) is

() ( )

Table 3. Operating Frequency

t

ON

V +0.075V

OUT

()

=×

K

V

BATT

t

ON MIN

()

tt

ON MIN OFF MAX

+

=

2.55 s+0.5 s

µ

2.55 s

µµ

I

LOAD(SKIP)

KV2LV-V

×

=



OUT VP OUT



V



VP






DEVICE

MAX1762/MAX1791 3.349 268.7 298.5 328

K

(µs)

MIN

(kHz)

TYP

(kHz)

=

84%

MAX

(kHz)

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

______________________________________________________________________________________ 13

rated for +0.5A, -0.9A source/sink capability and
swings from VL to GND.

The internal pulldown transistor that drives DL low is
robust, with a 1Ω typical on-resistance. This helps pre­vent DL from being pulled up during the fast rise time of
the inductor node, due to capacitive coupling from the
drain to the gate of the low-side synchronous-rectifier
MOSFET. However, for high-current applications, some
combinations of high-and low-side FETS may cause
excessive gate-drain coupling, which can lead to poor
efficiency, EMI, and shoot-through currents.

An adaptive dead-time circuit monitors the DL output
and prevents the high-side FET from turning on until DL
is fully turned off. The dead time at the other edge (DH
turning off) is determined by a fixed 35ns (typ) internal
delay.

Low-Side Current-Limit Sensing (ILIM)

The current-limit circuit employs a unique “valley” cur­rent-sensing algorithm that uses the on-state resistance
of the low-side MOSFET as a current-sensing element.
If the current-sense signal is below the current-limit
threshold (-100mV from CS to GND), the PWM is not
allowed to initiate a new cycle (Figure 7). The actual
peak current is greater than the current-limit threshold
by an amount equal to the inductor ripple current.
Therefore, the exact current-limit characteristic and
maximum load capability are a function of the MOSFET
on-resistance, inductor value, and battery voltage.

If greater current-limit accuracy is desired, CS must be
connected to the junction of the low-side switch source
and a current-sense resistor to GND. The current limit
will be 0.1V/R

SENSE

, and the accuracy will be ±10%.

A resistive voltage-divider from the inductor’s switching
mode to ground can be used to adjust the current-limit

sense voltage that appears at CS (Figure 8). Keep the
impedance at this mode low to avoid errors at CS.

POR and Soft-Start

Power-on reset (POR) occurs when V

BATT

rises above
approximately 2V, resetting the fault latch and soft-start
counter and preparing the PWM for operation. UVLO
circuitry inhibits switching until VVPrises above 4.1V,
whereupon an internal digital soft-start timer begins to
ramp up the maximum allowed current limit. The ramp
occurs in five steps: 20%, 40%, 60%, 80%, and 100%;
100% current is available after approximately 1.7ms.

Output Undervoltage Protection

The output UVLO function is similar to foldback current
limiting but employs a timer rather than a variable cur­rent limit. The output undervoltage protection is
enabled 20ms after POR or when coming out of shut­down. If the output is under 70% of the nominal value,

Figure 6. Pulse-Skipping/Discontinuous Crossover Point

Figure 7. “Valley” Current-Limit Threshold Point

Figure 8. Using a Resistive Voltage-Divider to Adjust Current­Limit Sense Voltage to 200mV

∆

i

- V

V

BATT

=

∆

t

INDUCTOR CURRENT

ON-TIME0 TIME

OUT

L

I

PEAK

I

LOAD

= I

/2

PEAK

I

PEAK

I

LOAD

I

LIMIT

INDUCTOR CURRENT

0 TIME

V

P

DH

MAX1762
MAX1791

CS

DL

1.0kΩ

1.0kΩ

V

OUT

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

14 ______________________________________________________________________________________

then the PWM is latched off and will not restart until VP
power is cycled, or SHDN is toggled low then high.

Design Procedure

Begin by establishing the input voltage range and max­imum load current before choosing an inductor and its
associated ripple-current ratio (LIR). The following four
factors dictate the rest of the design:

1) Input voltage range. The maximum value (V

VP

(MAX)

) must accommodate the maximum AC

adapter voltage. The minimum value (V

VP(MIN)

)
must account for the lowest input voltage after
drops due to connectors, fuses, and battery selec­tor switches. If there is a choice at all, lower input
voltages result in better efficiency.

2) Maximum load current. There are two values to
consider. The peak load current (I

LOAD(MAX)

) deter­mines the instantaneous component stress and fil­tering requirements and thus drives output
capacitor selection, inductor saturation rating, and
the design of the current-limit circuit. The continu­ous load current (I

LOAD

) determines the thermal
stress and thus drives the selection of input capaci­tors, MOSFETs, and other critical heat-contributing
components. Modern notebook CPUs generally
exhibit, I

LOAD

= I

LOAD(MAX)

x 0.8.

3) Switching frequency. The MAX1762/MAX1791
have a nominal switching frequency of 300kHz.

4) Inductor ripple-current ratio (LIR). LIR is the ratio
of the peak-to-peak ripple current to the average
inductor current. Size and efficiency trade-offs must
be considered when setting the inductor ripple-cur­rent ratio. Low inductor values cause large ripple
currents, resulting in the smallest size but poor effi­ciency and high output noise. The minimum practi­cal inductor value is one that causes the circuit to
operate at critical conduction (where the inductor
current just touches zero with every cycle). Inductor
values lower than this grant no further size-reduc­tion benefit.

The MAX1762/MAX1791s’ pulse-skipping algorithm ini­tiates skip mode at the critical conduction point. So, the
inductor operating point also determines the load-cur­rent value at which switchover occurs. The optimum
point is usually found between 20% and 50% ripple
current.

The inductor ripple current also impacts transient­response performance, especially at low V

VP

- V

OUT

difference. Low inductor values allow the inductor cur­rent to slew faster, replenishing charge removed from
the output filter capacitors by a sudden load step. The

peak amplitude of the output transient (V

SAG

) is also a
function of the maximum duty factor, which can be cal­culated from the on-time and minimum off-time:

where minimum off-time = 0.5µs (max).

Inductor Selection

The switching frequency (on-time) and operating point
(% ripple or LIR) determine the inductor value as fol­lows:

Example: I

LOAD(MAX)

= 2A, VVP= 7V, V

OUT

= 1.6V, f =

300kHz, 35% ripple current or LIR = 0.35:

Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice. The core must be large
enough not to saturate at the peak inductor current
(I

PEAK

):

I

PEAK

= I

LOAD(MAX)

+ [(LIR/2) ✕I

LOAD(MAX)

]

Determining Current Limit

The minimum current-limit threshold must be great
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The val­ley of the inductor current occurs at I

LOAD(MAX)

minus

half of the ripple current; therefore:

I

VALLEY

> I

LOAD(MAX)

- [(LIR/2) ✕I

LOAD(MAX)

]

where I

VALLEY

= minimum current-limit threshold volt-

age divided by the R

DS(ON)

of Q2. For the MAX1762/
MAX1791, the minimum current-limit threshold is 90mV.
Use the worst-case maximum value for R

DS(ON)

from
the MOSFET Q2 data sheet, and add some margin for
the rise in R

DS(ON)

with temperature. A good general
rule is to allow 0.5% additional resistance for each °C of
temperature rise.

(∆IK

LOAD(MAX)

V

=

SAG

××

CVK2

OUT OUT



2

)L

×








V-V

VP OUT











V

V

V

OUT

VP

VP

+t

OFF(MIN)



-t




V(V-V)

OUT VP OUT

×ƒ× ×

V LIR I

VP LOAD(MAX)

L =

L

1.6V(7V -1.6V)
kHz A

×××

7 300 0 35 2

=

59.. µ

H=






OFF(MIN)







MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

______________________________________________________________________________________ 15

Examining the 2A circuit example with a maximum
R

DS(ON)

= 52mΩ at +85°C temperature reveals the fol-

lowing:

I

VALLEY

= 90mV / 52mΩ = 1.73A

Checking the corresponding I

LOAD(MAX) reveals:

A current-sense resistor can be connected from CS to
GND to set the current limit for the device. The
MAX1762/MAX1791 will use the sense resistor instead
of the R

DS(ON)

of Q2 to limit the current. The maximum
value of the sense resistor can be calculated with the
equation:

I

LIMIT

= 90mV / R

SENSE

Output Capacitor Selection

The output filter capacitor must have low enough effec­tive series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements. In CPU V

CORE

convert­ers and other applications where the output is subject
to large load transients, the output capacitor’s size
depends on how much ESR is needed to prevent the
output from dipping too low under a load transient.
Ignoring the sag due to finite capacitance:

where V

DIP

is the maximum tolerable transient voltage
drop. In non-CPU applications, the output capacitor’s
size depends on how much ESR is needed to maintain
an acceptable level of output voltage ripple:

where Vp-p is the peak-to-peak output voltage ripple.
The actual microfarad capacitance value required
relates to the physical size needed to achieve low ESR,
as well as to the chemistry of the capacitor technology.
Thus, the capacitor is usually selected by ESR and volt­age rating rather than by capacitance value (this is true
of tantalum, SP, POS, and other electrolytic-type
capacitors).

When using low-capacity filter capacitors such as
ceramics, capacitor size is usually determined by the
capacity needed to prevent V

SAG

and V

SOAR

from

causing problems during load transients. Generally,

once enough capacitance is added to meet the over­shoot requirement, undershoot at the rising load edge
is no longer a problem (see the V

SAG

equation in the

Design Procedure section).

The amount of overshoot due to stored inductor energy
can be calculated as:

where I

PEAK

is the peak inductor current.

Stability Considerations

Stability is determined by the value of the ESR zero
(f

ESR

) relative to the switching frequency (f). The point

of instability is given by the following equation:

where:

For a typical 300kHz application, the ESR zero frequen­cy must be well below 95kHz, preferably below 50kHz.
Tantalum, Sanyo POSCAP, and Panasonic SP capaci­tors in widespread use at the time of publication have
typical ESR zero frequencies of 20kHz. In the design
example used for inductor selection, the ESR needed
to support a specified ripple voltage is found by the
equation:

where LIR is the inductor ripple current ratio, and I

LOAD

is the average DC load. Using a LIR = 0.35 and an
average load current of 2A, the ESR needed to support
50mVp-p ripple is 71mΩ.

Do not use high-value ceramic capacitors directly
across the fast feedback inputs (FB to GND) without
taking precautions to ensure stability. Large ceramic
capacitors can have a high-ESR zero frequency and
cause erratic, unstable operation. However, it’s easy to
add enough series resistance by placing the capaci­tors a couple of inches downstream from the junction of
the inductor and FB pin.

Unstable operation manifests itself in two related but dis­tinctly different ways: double-pulsing and fast-feedback
loop instability. Double pulsing occurs due to noise on
the output or because the ESR is so low that there isn’t
enough voltage ramp in the output voltage signal. This

I

LOAD(MAX)

I

VALLEY

==×=

1- 0.5 LIR

1.73A

1- 0.5 0.35

2.1A

R

ESR

V

DIP

≤

I

LOAD(MAX)

R

ESR

≤

×

LIR I

Vp - p

LOAD(MAX)

2

LI

PEAK

∆V

≤

2

CV

OUT

IJ

IJ

ESR

2 π RC

×× ×

ESR

ESR OUT

ƒ

π

1

R

ESR

V

RIPPLE(p-p)

=

LIR

× I

LOAD

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

16 ______________________________________________________________________________________

“fools” the error comparator into triggering a new cycle
immediately after the 500ns minimum off-time period has
expired. Double pulsing is more annoying than harmful,
resulting in nothing worse than increased output ripple.
However, it can indicate the possible presence of loop
instability, which is caused by insufficient ESR. Loop
instability can result in oscillations at the output after line
or load perturbations that can cause the output voltage
to fall below the tolerance limit.

The easiest method for checking stability is to apply a
very fast zero-to-max load transient (refer to the
MAX1762/MAX1791 EV kit manual) and carefully
observe the output voltage ripple envelope for over­shoot and ringing. It can help to simultaneously monitor
the inductor current with an AC current probe. Don’t
allow more than one cycle of ringing after the initial
step-response under- or overshoot.

Input Capacitor Selection

The input capacitor must meet the ripple-current require­ment (I

RMS

) imposed by the switching currents.
Nontantalum chemistries (ceramic or OS-CON™) are pre­ferred due to their resilience to power-up surge currents:

Power MOSFET Selection

DC bias and output power considerations dominate the
selection of the power MOSFETs used with the
MAX1762/MAX1791. Take care not to exceed the
device’s maximum voltage ratings. In general, both
switches are exposed to the supply voltage, so select
MOSFETs with VDS(max) greater than VP (max). Gate
drives to the N-channel and P-channel MOSFETs are
not symmetrical. The N-channel device is driven from
ground to the logic supply VL, while the P-channel
device is driven from VP to ground. The maximum rat­ing for VGSfor the N-channel device is usually not an
issue; however, V

GS

(max) for the P-channel must be at

least VP (max). Since V

GS

(max) is usually lower than

V

DS

(max), gate drive constraints often dictate the

required P-channel breakdown rating.

For moderate input-to-output differentials, the high-side
MOSFET (Q1) can be sized smaller than the low-side
MOSFET (Q2) without compromising efficiency. The
high-side switch operates at a very low duty cycle
under these conditions, so most conduction losses
occur in Q2. For maximum efficiency, choose a high­side MOSFET (Q1) that has conduction losses (I2R x

Duty Cycle) equal to the switching losses (CV

VP

2

f).
Make sure that the conduction losses at the minimum
input voltage do not exceed the package thermal limits
or violate the overall thermal budget. Conduction losses
plus switching losses at the maximum input voltage
should not exceed the package ratings or violate the
overall thermal budget (see MOSFET Power Dis-
sipation).

In addition to efficiency considerations, the selection of
the R

DS(ON)

of the low-side MOSFET must account for
the regulator’s required current limit. Choose a MOS­FET that has a low enough resistance over the operat­ing temperature range such that the device will not
enter current limit during normal operation (see
Determining Current Limit). Conversely, ultra-low
R

DS(ON)

devices may set the current limit too high and
may result in only incremental improvements in efficien­cy. Some large N-channel FETs also have substantial
interelectrode capacitance. Verify that the MAX1762/
MAX1791 DL driver can hold the gate off when the high
side switch turns on. Cross-conduction problems can
occur when the high-side switch turns on due to cou­pling through the N-channel’s parasitic drain-to-gate
capacitance.

The MAX1762/MAX1791 have adaptive dead-time cir­cuitry that prevents the high-side and low-side
MOSFETs from conducting at the same time (see MOS-
FET Gate Drivers). Even with this protection, it is still
possible for delays internal to the MOSFET to prevent
one MOSFET from turning off while the other is turned
on. The maximum mismatch time that can be tolerated
is 60ns. Select devices that have low turn-off times, and
make sure that NFET(tD(off,max)) - PFET(tD(on,min)) <
60ns, and PFET(tD(off,max)) - NFET(tD(on,min)) < 60ns.
Failure to do so may result in efficiency-killing shoot­through currents.

MOSFET Power Dissipation

Worst-case conduction losses occur at the duty factor
extremes. For the high-side MOSFET, the worst-case
power dissipation (PD) due to resistance occurs at min­imum battery voltage:

Generally, a small high-side MOSFET is desired to
reduce switching losses at high input voltage. However,
the R

DS(ON)

required to stay within package power-dis­sipation limits often limits how small the MOSFET can
be. Again, the optimum occurs when the switching (AC)
losses equal the conduction (R

DS(ON)

) losses. High-

OS-CON is a trademark of Sanyo.

I

≤=

I

RMS LOAD



V(V-V)

OUT VP OUT





I

VP







=



V




V

VP(MIN)



PD(Q1 resistance)

OUT



××





2

IR

LOAD DS(ON)

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

______________________________________________________________________________________ 17

side switching losses don’t usually become an issue
until the input is greater than approximately 15V.

Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum battery volt­age is applied, due to the squared term in the CV2f
switching loss equation. If the high-side MOSFET cho­sen for adequate R

DS(ON)

at low battery voltages
becomes extraordinarily hot when subjected to
V

VP(MAX)

, reconsider your choice of high-side MOS-

FET.

Calculating the power dissipation in Q1 due to switch­ing losses is difficult since it must allow for difficult
quantifying factors that influence the turn-on and turn­off times. These factors include the internal gate resis­tance, gate charge, threshold voltage, source induc­tance, and PC board layout characteristics. The follow­ing switching loss calculation provides only a very
rough estimate and is no substitute for breadboard
evaluation, preferably including a verification using a
thermocouple mounted on Q1:

where C

RSS

is the reverse transfer capacitance of Q1,

and I

GATE

is the peak gate-drive source/sink current.

For the low-side MOSFET, the worst-case power dissi­pation always occurs at maximum battery voltage:

The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
I

LOAD(MAX)

but are not quite high enough to exceed
the current limit and cause the fault latch to trip. To pro­tect against this possibility, the circuit must be overde­signed to tolerate:

I

LOAD

= I

LIMIT(HIGH)

+ (LIR / 2 ) ✕I

LOAD(MAX)

where I

LIMIT(HIGH)

is the maximum valley current
allowed by the current-limit circuit, including threshold
tolerance and on-resistance variation. This means that
the MOSFET must be very well heatsinked. If short-cir­cuit protection without overload protection is enough, a
normal I

LOAD

value can be used for calculating compo-

nent stresses.

During the period when the high-side switch is off, cur­rent circulates from ground to the junction of both FETs
and the inductor. As a consequence, the polarity of the
switching node is negative with respect to ground. If

unchanged, this voltage will be approximately 0.7V (a
diode drop) at both transition edges while both switch­es are off. In between the edges, the low-side switch
conducts; the drop is I

L

✕

R

DS(ON)

. If a Schottky clamp
is connected across the low-side switch, the initial and
final voltage drops will be reduced, improving efficien­cy slightly.

Choose a Schottky diode (D1) having a forward voltage
low enough to prevent the Q2 MOSFET body diode
from turning on during the dead time. As a general rule,
a diode having a DC current rating equal to 1/3 of the
load current is sufficient. This diode is optional and can
be removed if efficiency isn’t critical.

Applications Issues

Dropout Performance

The output voltage adjust range for continuous-conduc­tion operation is restricted by the nonadjustable 500ns
(max) minimum off-time one-shot. When working with
low input voltages, the duty-factor limit must be calcu­lated using worst-case values for on- and off-times.
Manufacturing tolerances and internal propagation
delays introduce an error to the tONK-factor. Also,
keep in mind that transient response performance of
buck regulators operating close to dropout is poor, and
bulk output capacitance must often be added.

Dropout design example: VIN= 7V (min), V

OUT

= 5V, f

= 300kHz. The required duty cycle is :

The worst-case on-time is:

The maximum IC duty factor based on timing con­straints of the MAX1762/MAX1792 is:

which meets the required duty cycle. Remember to
include inductor resistance and MOSFET on-state volt­age drops (V

SW

) when doing worst-case dropout duty-

factor calculations.

Fixed Output Voltages

The MAX1762/MAX1791 Dual Mode operation allows
the selection of common voltages without requiring
external components (Figure 9). Connect FB to GND for

PD (Q1 switching)



CV I

×׃×

RSS VP(MAX) LOAD

=





I

GATE

2







PD(Q2) -







V

V

VP(MAX)

=

OUT



2

IR

××1



LOAD DS




DC

V+V

OUT SW

===074.

REQ

V-V

VP SW

5V+ 0.1V

7V - 0.1V

t

ON(MIN)

V + 0.075

OUT

=×=×

V

VP

×=

ss335 90 218.%.µµ

5V+ 0.075

K

7V

Duty

==

t

ON(MIN)

t+t

ON(MIN) OFF(MAX)

218

.

µ

s

+

218 05

..

µµ

ss

=

082

.

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

18 ______________________________________________________________________________________

a fixed +1.8V (MAX1762) or 3.3V (MAX1791) output.
Connect FB to VL for a fixed 2.5V (MAX1762) or 5.0V
(MAX1791) output. Otherwise, connect FB to a resistive
voltage-divider for an adjustable output.

Setting the Output Voltage

Select V

OUT

> 1.25V for the MAX1762/MAX1791 by
connecting FB to a resistive voltage-divider between
V

OUT

and GND (Figure 2). Choose R2 to be about

10kΩ, and solve for R1 using the equation:

where VFB= 1.25V. For a V

OUT

= 3.0V, R2 = 10kΩ and

R1 = 14kΩ.

For a desired V

OUT

< 1.25V, connect FB to a resistive

voltage-divider between REF and OUT (Figure 3).
Choose R1 to be about 50kΩ, and solve for R2 using
the equation:

where V

FB

= 1.25V and V

REF

= 2.0V. For a V

OUT

=

1.0V, R1 = 50kΩ and R2 = 16.5kΩ. Under these condi­tions, a minimum load of V

REF

- VFB/ R1 >15µA is

required.

PC Board Layout Guidelines

Careful PC board layout is critical to achieve low
switching losses and clean, stable operation. This is
especially true when multiple converters are on the
same PC board where one circuit can affect the other.
The switching power stages require particular attention

(Figure 10). Refer to the MAX1791 EV kit manual for a
specific layout example.

If possible, mount all of the power components on the
top side of the board, with their ground terminals flush
against one another. Follow these guidelines for good
PC board layout:

• Isolate the power components on the top side from
the sensitive analog components on the bottom
side with a ground shield. Use a separate GND
plane under OUT. Avoid the introduction of AC cur­rents into the GND ground planes. Run the power
plane ground currents on the top side only, if possi­ble.

• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta­ble, jitter-free operation.

• Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PC boards (2oz vs. 1oz) can enhance
full-load efficiency by 1% or more. Correctly routing
PC board traces is a difficult task that must be
approached in terms of fractions of centimeters,
where a single milliohm of excess trace resistance
causes a measurable efficiency penalty.

• Inductor and GND connections to the synchronous
rectifiers for current limiting must be made using
Kelvin sensed connections to guarantee the cur­rent-limit accuracy. With SO-8 MOSFETs, this is
best done by routing power to the MOSFETs from
outside using the top copper layer, while connect­ing GND and CS inside (underneath) the µMAX
package.

• When trade-offs in trace lengths must be made, it’s
preferable to allow the inductor charging path to be
made longer than the discharge path. For example,
it’s better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
allow distance between the inductor and the low­side MOSFET or between the inductor and the out­put filter capacitor.

• Ensure that the OUT connection to C

OUT

is short
and direct. However, in some cases it may be desir­able to deliberately introduce some trace length
between the OUT connector node and the output
filter capacitor (see Stability Considerations).

• Route high-speed switching nodes (CS, DH, and
DL) away from sensitive analog areas (FB). Use
GND as an EMI shield to keep radiated switching
noise away from the IC’s feedback divider and ana­log bypass capacitors.

Figure 9. Feedback MUX

OUT

TO ERROR

FIXED

1.8V

FB

0.150V

2.5V

FIXED

3.3V

AMP

MAX1762

R1



VV

=×+

OUT FB



1









R2

R2



=




V-V

OUT FB

V-V

FB REF



×

R1




MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down

Controllers for Notebooks

______________________________________________________________________________________ 19

Layout Procedure

1) Place the power components first, with ground ter­minals adjacent (Q1 source, CIN, C

OUT

). If possi­ble, make all these connections on the top layer
with wide, copper-filled areas.

2) Mount the controller IC adjacent to the synchro­nous-rectifier MOSFETs, preferably on the back
side in order to keep CS, GND, and the DL gate
drive lines short and wide. The DL gate trace must
be short and wide (measuring 50mils to 100mils
wide if the MOSFET is 1in from the controller IC).

3) Place the V

L

bypass capacitor near the controller

IC.

4) Make the DC-DC controller ground connections as
follows: Near the IC, create a small analog ground
plane. Connect this plane to GND, and use this
plane for the ground connection for the REF and
V

VP

bypass capacitors and FB dividers.

5) On the board’s top side (power planes), make a
star ground to minimize crosstalk between the two

sides. The top-side star ground is a star connection
of the input capacitors, side 1 low-side MOSFET.
Keep the resistance low between the star ground
and the source of the low-side MOSFETs for accu­rate current limit. Connect the top-side star ground
(used for MOSFET, input, and output capacitors) to
the small island with a single short, wide connection
(preferably just a via).

6) Connect the output power planes directly to the out­put filter capacitor positive and negative terminals
with multiple vias.

Chip Information

TRANSISTOR COUNT: 3520

PROCESS: S8E1FP

Figure 10. PC Board Layout Example

USE AGND PLANE TO:

- BYPASS V

- TERMINATE EXTERNAL FB
DIVIDER (IF USED)

- PIN-STRAP CONTROL
INPUTS

AND REF

CC

CONNECT PGND TO AGND
BENEATH THE MAX1762/MAX1791 AT
ONE POINT ONLY AS SHOWN.

USE PGND PLANE TO:

- BYPASS V

- CONNECT PGND TO THE TOPSIDE STAR GROUND

AGND

VP

PGND

VIA TO GROUND

NOTE: EXAMPLE SHOWN IS FOR DUAL N-CHANNEL MOSFET.

VOUT

C2

GND

VBATT

L1

D1

VL

P1

N1

C1

MAX1762/MAX1791

High-Efficiency, 10-Pin µMAX, Step-Down
Controllers for Notebooks

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2001 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

Package Information

10LUMAX.EPS