Rainbow Electronics MAX17031 User Manual

General Description

The MAX17031 is a dual Quick-PWM™ step-down power-supply (SMPS) controller with synchronous rectification, intended for main 5V/3.3V power generation in battery-powered systems. Low-side MOSFET sensing provides a simple low-cost, highly efficient current sense for valley current-limit protection. Combined with the output overvoltage and undervoltage protection features, this current limit ensures robust output supplies.

The 5V/3.3V SMPS outputs can save power by operating in pulse-skipping mode or in ultrasonic mode to avoid audible noise. Ultrasonic mode forces the controller to maintain switching frequencies greater than 20kHz at light loads. The SKIP input also has an accurate logic threshold, allowing it to be used as a secondary feedback input to refresh an external charge pump or secondary winding without overcharging the output voltages.

An internal 100mA linear regulator generates the 5V bias needed for power-up or other low-power “alwayson” suspend supplies. An internal bypass circuitry allows automatic bypassing of the linear regulator when the 5V SMPS is active.

The device includes independent shutdown controls with well-defined logic thresholds to simplify power-up and power-down sequencing. To prevent current surges at startup, the internal voltage target is slowly ramped up from zero to the final target over a 1ms period. To prevent the output from ringing below ground in shutdown, the internal voltage target is ramped down from its previous value to zero over a 1ms period. A combined power-good (PGOOD) output simplifies the interface with external controllers. The MAX17031 is available in a 24-pin thin QFN (4mm x 4mm) package.

Applications

Notebook Computers

Ultra-Mobile PC

Main System Supply (5V and 3.3V Supplies)

2 to 4 Li+ Cells Battery-Powered Devices

Telecommunication

Features

o Dual Quick-PWM

o Preset 5V and 3.3V Outputs

o Internal 100mA, 5V Linear Regulator

o Internal OUT1 LDO5 Bypass Switch

o Secondary Feedback (SKIP Input) Maintains

Charge Pump

o 3.3V, 5mA Real-Time Clock (RTC) Power (Always

On)

o 2V ±1% 50µA Reference

o 6V to 24V Input Range

o Pulse-Skipping/Forced-PWM/Ultrasonic Mode

Control

o Independent SMPS and LDO5 Enable Controls

o Combined SMPS PGOOD Outputs

o Minimal Component Count

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

________________________________________________________________

Maxim Integrated Products

Pin Configuration

Ordering Information

19-4305; Rev 0; 10/08

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

Denotes a lead-free/RoHS-compliant package.

EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX17031ETG+ -40°C to +85°C 24 TQFN-EP*

Quick-PWM is a trademark of Maxim Integrated Products, Inc.

TOP VIEW

LX2

DH2

ON2

SKIP

OUT2

ILIM2

*EXPOSED PAD.

BST2

DL2

1718 16 14 13

MAX17031

REF

ONLDO

THIN QFN

4mm × 4mm

GND

456

RTC

*EP

DL1

BST1

LDO5

LX1

DH1

ON1

PGOOD

ILIM1

OUT1

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

IN to GND ...............................................................-0.3V to +28V

, VCCto GND .....................................................-0.3V to +6V

RTC, LDO5, ONLDO to GND ...................................-0.3V to +6V

OUT2 to GND ...........................................................-0.3V to +6V

ON1, ON2, PGOOD to GND.....................................-0.3V to +6V

OUT1 to GND..........................................-0.3V to (V

LDO5

+ 0.3V)

SKIP to GND...............................................-0.3V to (V

+ 0.3V)

REF, ILIM1, ILIM2 to GND..........................-0.3V to (V

+ 0.3V)

DL_ to GND ................................................-0.3V to (V

+ 0.3V)

BST_ to GND ..........................................................-0.3V to +36V

BST_ to V

............................................................-0.3V to +30V

DH1 to LX1 ..............................................-0.3V to (V

BST1

+ 0.3V)

BST1 to LX1..............................................................-0.3V to +6V

DH2 to LX2 ..............................................-0.3V to (V

BST2

+ 0.3V)

BST2 to LX2..............................................................-0.3V to +6V

LDO5, RTC, REF Short Circuit to GND.......................Momentary

RTC Current Continuous.....................................................+5mA

LDO5 Current (Internal Regulator) Continuous ..............+100mA

LDO5 Current (Switched Over) Continuous ...................+200mA

Continuous Power Dissipation (T

= +70°C) 24-Pin, 4mm x 4mm Thin QFN (T2444-3)

(derate 27.8mW/°C above +70°C).................................2.22W

Operating Temperature Range ...........................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Note: Measurements are valid using a 20MHz bandwidth limit.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INPUT SUPPLIE S

IN Input Voltage Range LDO5 in regulation 6 24 V

= 6V to 24V, ON1 = ON2 = GND,

IN Standby Supply Current

IN Shutdown Supply Current

IN Supply Current I

VCC Bias Supply Current I

VCC

PWM CO NTROLLERS

OUT1 Output-Voltage Accuracy V

OUT2 Output-Voltage Accuracy V

OUT1

OUT2

Load Regulation Error

Line Regulation Error Either SMPS, IN = 6V to 28V 0.005 %/V

DH1 On-Time t

DH2 On-Time t

Minimum Off-Time t

Soft-Start Slew Rate t

Ultrasonic Operating Frequenc y f

ON1

ON2

OFF(MIN)

SW(USONIC) VSKIP

ONLDO = RTC

= 4.5V to 24V,

ON1 = ON2 = ONLDO = GND

ON1 = ON2 = VCC, V V

= 5.3V, V

OUT1

ON1 = ON2 = VCC, V

= 5.3V, V

OUT1

= 1.8V 4.95 5.00 5.05 V

SKIP

= 1.8V 3.267 3.30 3.333 V

SKIP

Either SMPS, V

= 5.0V (Note 1) 895 1052 1209 ns

OUT1

= 3.3V (Note 1) 833 925 1017 ns

OUT2

(Note 1) 300 400 ns

Rising/fal ling edge on ON1 or ON2 1 m s

= GND 20 34 kHz

85 175 µA

40 70 µA

= VCC;

OUT2

SKI P

SKIP

= 3.5V

SKIP

= 3.5V

= 1.8V, I

= GND, I

= VCC, I

= VCC;

= 0 to 5A -0.1

LOAD

= 0 to 5A -1.7

LOAD

= 0 to 5A -1.5

LOAD

0.1 0.2 mA

0.7 1.5 mA

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

LINEAR REGULATOR (LDO5)

LDO5 Output-Voltage Accuracy V

LDO5 Short-Circuit Current LDO5 = GND 100 260 mA

LDO5 Regulation Reduction/ Bootstrap Switchover Threshold

LDO5 Bootstrap Switch Resistance LDO5 to OUT1, V

VCC Undervoltage Lockout Threshold

Thermal-Shutdown Thresho ld T

3.3V ALWAYS-ON LINEAR REGULATOR (RTC)

RTC Output-Voltage Accuracy V

RTC Short-Circuit Current RTC = GND 5 22 mA

REFERENCE (REF)

Reference Voltage V

Reference Load Regulation Error V

REF Lockout Voltage V

OUT1 FAULT DETECTION

OUT1 Overvoltage and PGOOD Trip Threshold

OUT1 Overvoltage Fault Propagation De la y

OUT1 Undervoltage Protect ion Trip Threshold

OUT1 Output Undervoltage Fault Propagation Delay

OUT2 FAULT DETECTION

OUT2 Overvoltage and PGOOD Trip Threshold

OUT2 Overvoltage Fault Propagation De la y

OUT2 Undervoltage Protect ion Trip Threshold

OUT2 Output Undervoltage Fault Propagation De la y

LDO5

SHDN

RTC

REF

REF IREF

REF(UVLO)

OVP

UVP

OVP

UVP

VIN = 6V to 24V, ON1 = GND, 0 < I

Fal ling edge of OUT1 -11.0 -8.8 -6.0

Rising edge of OUT1 -7.0

Fal ling edge of VCC, PWM disabled below this threshold

Rising edge of V

Hysteresis = 10°C 160 °C

ON1 = ON2 = GND, VIN = 6V to 24V, 0 < I

ON1 = ON2 = ONLDO = GND, V

VCC = 4.5V to 5.5V, I

Rising edge, 350mV (typ) hysteresis 1.95 V

With respect to error comparator thresho ld 10 13 16 %

OUT1 forced 50mV above trip threshold 10 µs

With respect to error comparator thresho ld 65 70 75 %

10 µs

With respect to error comparator thresho ld 10 13 16 %

OUT2 forced 50mV above trip threshold 10 µs

With respect to error comparator thresho ld 65 70 75 %

10 µs

< 100mA

LDO5

= 5V (Note 3) 1.9 4.5 

OUT1

4.2

< 5mA

RTC

= 6V to 24V, 0 < I

= -20µA to +50µA -10 +10 mV

< 5mA

RTC

= 0 1.980 2.00 2.020 V

REF

4.90 5.0 5.10 V

3.8 4.0 4.3

3.23 3.33 3.43

3.19 3.47

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

PO WER-GOOD

PGOOD Lower Trip Thresho ld

PGOOD Prop agation De lay t

PGOOD Output Low Voltage

PGOOD Leakage Current I

CURRENT LIMIT

ILIM_ Adjustment Range 0.2 2 V

ILIM_ Current 5 µA

Valley Current-Limit Threshold (Adjustable)

Current-Limit Threshold (Negative)

Ultrasonic Current-Limit Threshold V

Current-Limit Threshold (Zero Crossing)

GATE DRIVERS

DH_ Gate-Driver On-Resistance R

DL_ Gate-Driver On-Resistance R

DH_ Gate-Driver Source/Sink Current

DL_ Gate-Driver Source Current

DL_ Gate-Driver Sink Current I

Dead Time t

Internal BST_ Switch On-Resistance

BST_Leakage Current

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

With respect to either error comparator threshold, falling edge, hysteresis = 1%

PGOOD

LIM_ (VAL) VAGND

NEG

NEG(US)

(SOURCE)

DL (SINK)

DEAD

BST IBST

OUT1 or OUT2 forced 50mV beyond PGOOD trip threshold, falling edge

ON1 or ON2 = GND (PGOOD low impedance), I

OUT1 and OUT2 in regulation (PGOOD high impedance), PGOOD forced to 5.5V, T

= +25°C

- VLX_

With respect to valley current-lim it threshold, V

= 3.5V, V

OUT2

- VLX_,

AGND

= VCC or GND

SKIP

BST1 - LX1 and BST2 - LX2 forced to 5V 1.5 3.5 

DL1, DL2; high state 1.4 4.5

DL1, DL2; low state 0.5 1.5

DH1, DH2 forced to 2.5V, BST1 - LX1 and BST2 - LX2 forced to 5V

DL1, DL2 forced to 2.5V 1.7 A

DL1, DL2 forced to 2.5V 3.3 A

DL1, DL2 rising (Note 4) 30

DH1, DH2 ri sing (Note 4) 35

_ = 10mA, VDD = 5V 5.5 

_ = 26V, TA = +25°C;

BST

OUT1 and OUT2 above regulation threshold

SINK

SKIP

= 4mA

_ = 100k

ILIM

_ = 500mV)

ILIM

_ = 200k

ILIM

_ = 1.00V)

ILIM

_ = 400k

ILIM

_ = 2.00V)

ILIM

= V

REF

= 5.3V 20 mV

OUT1

-16 -13 -10 %

10 µs

0.3 V

1 µA

44 50 56

90 100 110

180 200 220

-120 %

1.5 mV

2 A

0.1 5 µA



MAX17031

_______________________________________________________________________________________ 5

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= -40°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

INPUTS AND OUTPUTS

SKIP Input Thresholds

SKIP Leakage Current V

ON_ Input-Logic Le ve l s ONLDO, ON1, ON2

ON_ Leakage Current

OUT_ Leakage Current V

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

Upper SKIP/PWM threshold falling edge, 33mV hysteresis

Lower PWM/ultrasonic threshold 0.4 1.6

= 0 or 5V, TA = +25°C -1 +1 µA

SKIP

High (SMPS on) 2.4

Low (SMPS off) 0.8

= V

ON1

= +25°C

= V

ON1

ON2

= V

ONLDO

= 0 or 5V,

= 5.3V 15 65

OUT1

= 3.5V 5 30

OUT2

1.94 2.0 2.06

-2 +2 µA

µA

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INPUT SUPPLIE S

IN Input Voltage Range LDO5 in regulation 6 24 V

= 6V to 24V, ON1 = ON2 = GND,

IN Standby Supply Current

IN Shutdown Supply Current

IN Supply Current I

VCC Bias Supply Current I

PWM CO NTROLLERS

OUT1 Output-Voltage Accuracy V

OUT2 Output-Voltage Accuracy V

DH1 On-Time t

DH2 On-Time t

Minimum Off-Time t

Ultrasonic Operating Frequenc y f

OFF(MIN)

SW(USONIC) VSKIP

VCC

OUT1

OUT2

ON1

ON2

ONLDO = RTC

= 4.5V to 24V,

ON1 = ON2 = ONLDO = GND

ON1 = ON2 = VCC, V V

= 5.3V, V

OUT1

ON1 = ON2 = VCC, V V

= 5.3V, V

OUT1

= 1.8V 4.90 5.10 V

SKIP

= 1.8V 3.234 3.366 V

SKIP

= 5.0V (Note 1) 895 1209 ns

OUT1

= 3.3V (Note 1) 833 1017 ns

OUT2

(Note 1) 400 ns

= GND 18 kH z

OUT2

= VCC,

SKIP

= 3.5V

= VCC,

SKIP

= 3.5V

200 µA

70 µA

0.2 mA

1.5 mA

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

6 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= -40°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

LINEAR REGULATOR (LDO5)

LDO5 Output-Voltage Accuracy V

LDO5 Short-Circuit Current LDO5 = GND 260 mA

LDO5 Regulation Reduction/ Bootstrap Switchover Threshold

LDO5 Bootstrap Switch Resistance

VCC Undervoltage Lockout Threshold

3.3V ALWAYS-ON LINEAR REGULATOR (RTC)

RTC Output-Voltage Accuracy V

RTC Short-Circuit Current RTC = GND 5 22 mA

REFERENCE (REF)

Reference Voltage V

Reference Load Regulation Error V

OUT1 FAULT DETECTION

OUT1 Overvoltage and PGOOD Trip Threshold

OUT1 Undervoltage Protection Trip Threshold

OUT2 FAULT DETECTION

OUT2 Overvoltage and PGOOD Trip Threshold

OUT2 Undervoltage Protection Trip Threshold

PO WER-GOOD

PGOOD Lower Trip Thresho ld

PGOOD Output Low Voltage

LDO5

RTC

REF

REF IREF

VIN = 6V to 24V, ON1 = GND; 0mA < I

Fal ling edge of OUT1 -12.0 -5.0 %

LDO5 to OUT1, V

Fal ling edge of VCC, PWM disabled below this threshold

ON1 = ON2 = GND, VIN = 6V to 24V, 0 < I

RTC

ON1 = ON2 = ONLDO = GND, VIN = 6V to 24V, 0 < I

VCC = 4.5V to 5.5V, I

= -20µA to +50µA -10 +10 mV

With respect to error comparator threshold 10 16 %

With respect to error comparator thresho ld 63 77 %

With respect to error comparator threshold 10 16 %

With respect to error comparator threshold 63 77 %

With respect to either error comparator threshold, falling edge, hysteresis = 1%

ON1 or ON2 = GND (PGOOD low impedance), I

< 100mA

LDO5

< 5mA

RTC

SINK

= 5V (Note 3) 4.5 

OUT1

= 0 1.975 2.025 V

REF

= 4mA

4.85 5.15 V

3.8 4.3 V

3.18 3.45

3.16 3.50

-16 -10 %

0.3 V

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

_______________________________________________________________________________________ 7

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 2, no load on LDO5, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SKIP

= 5V, ONLDO = RTC, ON1 = ON2

= V

, TA= -40°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

Note 1: On-time and off-time specifications are measured from 50% point to 50% point at the DH pin with LX = GND, V

BST

= 5V, and a 500pF capacitor from DH to LX to simulate external MOSFET gate capacitance. Actual in-circuit times might be different due to MOSFET switching speeds.

Note 2: Specifications to T

= -40°C are guaranteed by design and not production tested.

Note 3: Specification increased by 1Ω to account for test measurement error. Note 4: Production tested for functionality only.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

CURRENT LIMIT

ILIM_ Adjustment Range 0.2 2 V

_ = 100k

ILIM

_ = 500mV)

ILIM

Valley Current-Limit Threshold (Adjustable)

GATE DRIVERS

DH_ Gate-Driver On-Resistance R

DL_ Gate-Driver On-Resistance R

INPUTS AND OUTPUTS

SKIP Input Thresholds

ON_ Input-Logic Le ve l s ONLDO, ON1, ON2

LIM_ (VAL) VAGND

- VLX_

BST1 - LX1 and BST2 - LX2 forced to 5V 3.5 

DL1, DL2; high state 4.5

DL1, DL2; low state 1.5

Upper SKIP/PWM threshold falling edge, 33mV hysteresis

Lower PWM/ultrasonic threshold 0.4 1.6

_ = 200k

ILIM

_ = 1.00V)

ILIM

_ = 400k

ILIM

_ = 2.00V)

ILIM

High (SMPS on) 2.4

Low (SMPS off) 0.8

40 60

85 115

164 236

1.94 2.06



MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

8 _______________________________________________________________________________________

Typical Operating Characteristics

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TA= +25°C, unless otherwise noted.)

5V OUTPUT EFFICIENCY

vs. LOAD CURRENT

100

EFFICIENCY (%)

0.01 10 LOAD CURRENT (A)

3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT

100

SKIP MODE

EFFICIENCY (%)

0.01 10

ULTRASONIC MODE

LOAD CURRENT (A)

12V

20V

10.1

PWM MODE

10.1

SKIP MODE

PWM MODE

12V INPUT

100

SKIP MODE

MAX17031 toc01

EFFICIENCY (%)

0.01 10

SMPS OUTPUT-VOLTAGE DEVIATION

MAX17031 toc04

ULTRASONIC MODE

-1

PWM MODE

OUTPUT-VOLTAGE DEVIATION (%)

-2

-3

0.01 10

5V OUTPUT EFFICIENCY

vs. LOAD CURRENT

PWM MODE

ULTRASONIC MODE

10.1

LOAD CURRENT (A)

vs. LOAD CURRENT

LOW-NOISE

SKIP MODE

10.1

LOAD CURRENT (A)

12V INPUT

100

MAX17031 toc02

EFFICIENCY (%)

0.01 10

1000

MAX17031 toc05

100

SWITCHING FREQUENCY (kHz)

0.01 10

3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT

LOAD CURRENT (A)

SWITCHING FREQUENCY

vs. LOAD CURRENT

PWM MODE

ULTRASONIC MODE

SKIP MODE

LOAD CURRENT (A)

MAX17031 toc03

12V

20V

SKIP MODE

PWM MODE

10.1

MAX17031 toc06

LOW-NOISE

12V INPUT

10.1

5V LDO OUTPUT VOLTAGE

3.3V RTC OUTPUT VOLTAGE

vs. LOAD CURRENT

5.2

5.1

5.0

4.9

OUTPUT VOLTAGE (V)

4.8

4.7 0160

LOAD CURRENT (mA)

100 120 14020 40 60 80

MAX17031 toc07

OUTPUT VOLTAGE (V)

3.5

3.4

3.3

3.2

3.1

3.0 012

vs. LOAD CURRENT

MAX17031 toc08

810246

LOAD CURRENT (mA)

NO-LOAD INPUT SUPPLY CURRENT

vs. INPUT VOLTAGE

100

ICC + I

SUPPLY CURRENT (mA)

0.1

SKIP MODE

0.01 025

INPUT VOLTAGE (V)

PWM MODE

LOW-NOISE

ULTRASONIC MODE

MAX17031 toc09

2051015

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

_______________________________________________________________________________________ 9

Typical Operating Characteristics (continued)

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TA= +25°C, unless otherwise noted.)

STANDBY AND SHUTDOWN INPUT

SUPPLY CURRENT vs. INPUT VOLTAGE

0.1 ICC + I

(ONLDO = RTC, ON1 = ON2 = GND)

0.01

SUPPLY CURRENT (mA)

0.001 025

LDO AND RTC POWER-UP

12V

STANDBY

SHUTDOWN

(ONLDO = ON1 = ON2 = GND)

2051015

INPUT VOLTAGE (V)

MAX17031 toc13

MAX17031 toc10

SAMPLE PERCENTAGE (%)

12V

B 5V

3.3V

3.3V D

2.0V 2V

-20 20

REFERENCE OFFSET

VOLTAGE DISTRIBUTION

TA = +85°C

= +25°C

2V OFFSET VOLTAGE (mV)

SAMPLE SIZE = 150

LDO AND RTC POWER REMOVAL

12-12 -4 4

MAX17031 toc14

MAX17031 toc11

SAMPLE PERCENTAGE (%)

A 12V

B 5V C

3.3V

0.1A

2.0V 0A

100mV ILIM THRESHOLD VOLTAGE DISTRIBUTION

TA = +85°C

= +25°C

90 110

ILIM THRESHOLD VOLTAGE (mV)

5V LDO LOAD TRANSIENT

SAMPLE SIZE = 150

10694 98 102

MAX17031 toc15

MAX17031 toc12

200µs/div

A. INPUT SUPPLY, 5V/div B. 5V LDO, 2V/div

5V SMPS STARTUP AND SHUTDOWN

A. 5V LDO OUTPUT, 0.2V/div B. 5V SMPS OUTPUT, 2V/div

200µs/div

C. 3.3V RTC, 2V/div D. 1.0 REF, 1V/div

MAX17031 toc16

C. ON1, 5V/div

200µs/div

A. INPUT SUPPLY, 5V/div B. 5V LDO, 2V/div

C. 3.3V RTC, 2V/div D. 2.0 REF, 1V/div

STARTUP WAVEFORMS

(SWITCHING REGULATORS)

A 5V

B 5V

0V 5V

SKIP MODE

A. ON1, 5V/div B. 5V SMPS OUTPUT, 2V/div

200µs/div

C. PGOOD, 5V/div D. INDUCTOR CURRENT, 5A/div

MAX17031 toc17

B 5V

A. LDO OUTPUT, 100mV/div B. LDO CURRENT, 100mA/div

SHUTDOWN WAVEFORMS

(SWITCHING REGULATORS)

0V 5V

A. ON1, 5V/div B. 5V SMPS OUTPUT, 2V/div

4µs/div

200µs/div

C. PGOOD, 2V/div D. INDUCTOR CURRENT, 5A/div

MAX17031 toc18

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

10 ______________________________________________________________________________________

Pin Description

Typical Operating Characteristics (continued)

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TA= +25°C, unless otherwise noted.)

5V SMPS LOAD TRANSIENT

(1A TO 4A)

MAX17031 toc19

3.3V SMPS LOAD TRANSIENT (1A TO 4A)

3.3V

MAX17031 toc20

12V

POWER REMOVAL

(SMPS UVLO RESPONSE)

MAX17031 toc21

A 0V

B 0V

C 0V

D 0V

40µs/div

A. LOAD CURRENT, 2A/div B. 5V SMPS OUTPUT, 100mV/div

C. INDUCTOR CURRENT, 2A/div

A. LOAD CURRENT, 2A/div B. 3.3V SMPS OUTPUT, 100mV/div

40µs/div

C. INDUCTOR CURRENT, 2A/div

PIN NAME FUNCTION

2V Reference Voltage Output. Bypas s REF to analog ground with a 0.22µF or greater ceramic

1 REF

capacitor. The reference can source up to 50µA for external loads. Loading REF degrades output voltage accuracy according to the REF load regulation error (see Typical Operating Characteristics). The reference shuts down when ON1, ON2, and ONLDO are all pulled low.

Enable Input for LDO5. Drive ONLDO high (pull up to RTC) to enable the linear regulator (LDO5)

2 ONLDO

3 V

4 RTC

output. Drive ONLDO low to shut down the linear regulator output. When ONLDO i s high, LDO5 must supply V

Analog Supply Voltage Input. Connect V

and VDD.

to the system supply voltage with a series 50

resistor, and bypass to analog ground using a 1µF or greater ceramic capacitor.

3.3V Alwa ys-On Linear Regulator Output for RTC Power. Bypass RTC with a 1µF or greater ceramic capacitor to analog ground. RTC can source up to 5mA for external loads.

Power Input Supply. Bypas s IN with a 0.1µF or greater ceramic capacitor to GND. IN powers the

5 IN

linear regulators (RTC and LDO5) and sense s the input vo ltage for the Quick-PWM on-time oneshot timer. The DH on-time is inversel y proportiona l to input vo ltage.

6 LDO5

5V Linear Regulator Output. Bypass LDO5 with a 4.7µF or greater ceramic capacitor to GND. LDO5 can source 100mA for external load support. LDO5 is powered from IN.

Output-Voltage Sense Input for SMPS1 and Linear Regulator Bypass Input. OUT1 is an input to the Quick-PWM on-time one-shot timer. OUT1 also serves as the feedback input for the SMPS1.

7 OUT1

When OUT1 exceeds 93.5% of the LDO5 voltage, the controller bypas ses the LDO5 output to OUT1. The bypass sw itch is disab led if the OUT1 voltage drops by 8.5% from LDO5 nomina l regulation threshold.

Valley Current-Limit Adjustment for SMPS1. The GND - LX1 current-limit threshold is 1/10 the

8 ILIM1

voltage present on ILIM1 over a 0.2V to 2V range. An internal 5µA current source allow s thi s voltage to be set w ith a single resi stor between ILIM1 and analog ground.

10ms/div

A. INPUT VOLTAGE, 5V/div B. 5V LDO OUTPUT, 2V/div

C. 5V SMPS, 2V/div D. PGOOD, 5V/div

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 11

Pin Description (continued)

PIN NAME FUNCTION

Open-Drain Power-Good Output for SMPS1 and SMPS2. PGOOD is low when either output voltage is

9 PGOOD

10 ON1 Enable Input for SMPS1. Drive ON1 high to enable SMPS1. Drive ON1 low to shut down SMPS1.

11 DH1 High-Side Gate-Driver Output for SMPS1. DH1 swings from LX1 to BST1.

12 LX1

13 BST1

14 DL1 Low-Side Gate-Driver Output for SMPS1. DL1 sw ings from power GND to V

15 V

16 GND Analog and Power Ground

17 DL2 Low-Side Gate-Driver Output for SMPS2. DL2 sw ings from power GND to V

18 BST2

19 LX2

20 DH2 High-Side Gate-Driver Output for SMPS2. DH2 swings from LX2 to BST2.

21 ON2 Enable Input for SMPS2. Drive ON2 high to enable SMPS2. Drive ON2 low to shut down SMPS2.

22 SKIP

23 OUT2

24 ILIM2

— EP Exposed Pad. Connect backside exposed pad to analog GND and power GND.

more than 15% (typ) below the nominal regulation threshold, during soft-start, in shutdown, when either SMPS is disabled, and after the fault latch ha s been tripped. After the soft-start circuit ha s terminated, PGOOD becomes high impedance if both output s are in regulation.

Inductor Connection for SMPS1. Connect LX1 to the switched side of the inductor. LX1 is the lower suppl y rail for the DH1 high-side gate driver.

Boost Flying Capacitor Connection for SMPS1. Connect to an external capacitor as shown in Figure 1. An optional resistor in series with BST1 allows the DH1 turn-on current to be adjusted.

DD.

Supply Voltage Input for the DL_ Gate Drivers. V BST diode sw itch. Connect to a 5V supply, and bypass V ceramic capacitor.

Boost Flying Capacitor Connection for SMPS2. Connect to an external capacitor as shown in Figure 1. An optional resistor in series with BST2 allows the DH2 turn-on current to be adjusted.

Inductor Connection for SMPS2. Connect LX2 to the switched side of the inductor. LX2 is the lower suppl y rail for the DH2 high-side gate driver.

Pulse-Skipping Control Input. This three-leve l input determines the operating mode for the switching regulators:

High (> 2V) = pulse-skipping mode Middle (1.8V) = forced-PWM mode GND = ultrasonic mode

Output-Voltage Sense Input for SMPS2. OUT2 is an input to the Quick-PWM on-time one-shot timer. OUT2 also serves as the feedback input for the preset 3.3V.

Valley Current-Limit Adjustment for SMPS2. The GND - LX2 current-limit threshold is 1/10 the voltage present on ILIM2 over a 0.2V to 2V range. An internal 5µA current source allow s thi s voltage to be set w ith a single resi stor between ILIM2 and analog ground.

is internally connected to the drain of the HVPV

to power GND with a 1µF or greater

DD.

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

12 ______________________________________________________________________________________

Figure 1. Standard Application Circuit—Main Supply

INPUT (V

7V TO 24V

IN_PIN

0.1µF

4x 10µF 25V

5V OUTPUT

OUT1

12V TO 15V

CHARGE PUMP

0.1µF

5V LDO OUTPUT

POWER GROUND

ANALOG GROUND

0.1µF

1MΩ

200kΩ

1.0µF

47Ω

BST1

0.1µF

DH1

BST1

LX1

DL1

DH2

BST2

LX2

DL2 D2

BST2

0.1µF

OUT2

3.3V OUTPUT

MAX17031

ILIM1

OUT1

SKIP

LDO5

ILIM1

10nF

4.7µF

PAD

OUT2

PGOOD

RTC

REF

GND

ON1 ON2

ONLDO

ILIM2

0.1µF

C3 1µF

ILIM2

R6 100kΩ

COMBINED POWER-GOOD

RTC SUPPLY

ON OFF

*NOTE: LOWER INPUT VOLTAGES REQUIRE ADDITIONAL INPUT CAPACITANCE. IF OPERATING NEAR DROPOUT, COMPONENT SELECTION MUST BE CAREFULLY DONE TO ENSURE PROPER OPERATION.

Detailed Description

The MAX17031 step-down controller is ideal for highvoltage, low-power supplies for notebook computers. Maxim’s Quick-PWM pulse-width modulator in the MAX17031 is specifically designed for handling fast load steps while maintaining a relatively constant operating frequency and inductor operating point over a wide range of input voltages. The Quick-PWM architecture circumvents the poor load-transient timing problems of fixed-frequency current-mode PWMs, while also avoiding the problems caused by widely varying switching frequencies in conventional constant-on-time and constant-off-time PWM schemes. Figure 2 is the functional diagram overview and Figure 3 is the QuickPWM core functional diagram

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 13

Table 1. Component Selection for Standard Applications

Table 2. Component Suppliers

400kHz/300kH z

COMPONENT

Input Voltage VIN = 7V to 24V

Input Capacitor

)

SMPS 1

Output Capac itor (C

)

OUT1

Inductor (L1)

High-Side MOSFET

)

Low-Side MOSFET

)

Current-Limit Res istor

)

ILIM1

SMPS 2

Output Capac itor (C

)

OUT2

Inductor (L2)

High-Side MOSFET (N

)

Low-Side MOSFET (N

)

Current-Limit Res istor (R

)

ILIM2

SMPS1: 5V AT 5A

SMPS2: 3.3V AT 8A

4X 10µF, 25V Taiyo Yuden TMK432BJ106KM

2x 100µF, 6V, 35m SANYO 6TPE100MAZB

4.3µH, 11.4m, 11A Sumida CEP125U

Siliconix Si4800BDY 23m/30m 30V

Siliconix Si4812BDY

16.5m/20m 30V

71k

2x 150µF, 4V, 35m SANYO 4TPE150MAZB

2.2µH, 5.4m, 14A Sumida CEP125U

Siliconix Si4684DY

9.2m/11.5m, 30V

Siliconix Si4430BDY

4.8m/6.0m, 30V

71k

SUPPLIER WEBSITE

AVX Corp. www.avx.com

Central Semiconductor Corp.

Fairch ild Semiconductor www.fairchildsem i.com International Rect ifier www.irf.com KEMET Corp. www.kemet.com NEC/TOKIN America, Inc. www.nec-tokinamerica.com Panason ic Corp. www.panasonic.coml

Philips/nxp Semiconductor www.semiconductors.philips.com

Pulse Engineering www.pul seeng.com

Renesas Technology Corp.

SANYO Electric Co., Ltd. www,sanyode vice.com

Sumida Corp. www.sumida.com

Taiyo Yuden www.t-yuden.com

TDK Corp. www.component.tdk.com

TOKO America, Inc. www.tokoam.com

Vishay (Dale, Siliconix) www.vishay.com

Würth Elektronik GmbH & Co. KG

www.centralsemi.com

www.renesas.com

www.we-online.com

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

14 ______________________________________________________________________________________

Figure 2. Functional Diagram Overview

SKIP

RTC

TON

ILIM1 OUT1

BST1

DH1

LX1

DL1

ON1

5V LINEAR

REGULATOR

3.3V LINEAR REGULATOR

LDO BYPASS

CIRCUITRY

BYP

SECFB

PWM2

CONTROLLER

PWM1

CONTROLLER

(FIGURE 3)

FB1 SELECT (PRESET 5V)

UVLO

FAULT1

(FIGURE 3)

FAULT2

FB2 SELECT

(PRESET 3.3V)

UVLO

ONLDO LDO5

ILIM2 OUT2 V

BST2

DH2

LX2

DL2

ON2

POWER-GOOD

PGOOD

POWER-GOOD

AND FAULT

PROTECTION

MAX17031

PAD

AND FAULT

PROTECTION

REF

GND

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 15

Figure 3. Functional Diagram—Quick-PWM Core

FB INT PRESET OR EXT ADJ

REFIN

DH DRIVER

TON IN

AGND

ILIM

INTEGRATOR

GND

REF

NEG CURRENT

LIMIT

VALLEY

CURRENT LIMIT

AGND

∆ FB

SLOPE COMP

ANALOG

SOFT-START/

SOFT-STOP

OFF(MIN)

Q TRIG

ONE-SHOT

*RESET DOMINATE

Q TRIG

ONE-SHOT

ON-TIME

COMPUTE

ZERO

CROSSING

GND

REFIN

GND

SKIP

ULTRASONIC

THRESHOLD

THREE-LEVEL

DECODE

ULTRASONIC

Q TRIG

ONE-SHOT

DL DRIVER

MAX17031

The MAX17031 includes several features for multipurpose notebook functionality, and is specifically designed for 5V/3.3V main power-supply rails. The MAX17031 includes a 100mA, 5V linear regulator (LDO5) ideal for initial power-up of the notebook and main supply. Additionally, the MAX17031 includes a

3.3V, 5mA RTC supply that remains always enabled, which can be used to power the RTC supply and system pullups when the notebook shuts down. The MAX17031 also includes a SKIP mode control input with an accurate threshold that allows an unregulated charge pump or secondary winding to be automatically refreshed—ideal for generating the low-power 12V to 15V load switch supply.

3.3V RTC Power

The MAX17031 includes a low-current (5mA) linear regulator that remains active as long as the input supply (IN) exceeds 2V (typ). The main purpose of this “always-enabled” linear regulator is to power the RTC when all other notebook regulators are disabled. The RTC regulator sources at least 5mA for external loads.

Preset 5V, 100mA Linear Regulator

The MAX17031 includes a high-current (100mA) 5V linear regulator. This LDO5 is required to generate the 5V bias supply necessary to power up the switching regulators. Once the 5V switching regulator (MAX17031 OUT1) is enabled, LDO5 is bypassed to OUT1. The MAX17031 LDO5 sources at least 100mA of supply current.

Bypass Switch

The MAX17031 includes an LDO5 bypass switch that allows the LDO5 to be bypassed to OUT1. When OUT1 exceeds 93.5% of the LDO5 output voltage for 500µs, then the MAX17031 reduces the LDO5 regulation threshold and turns on an internal p-channel MOSFET to short OUT1 to LDO5. Instead of disabling the LDO5 when the MAX17031 enables the bypass switch, the controller reduces the LDO5 regulation voltage, which effectively places the linear regulator in a standby state while switched over, allowing a fast recovery if the OUT1 drops by 8.5% from LDO5 nominal regulation threshold.

5V Bias Supply (VCC/VDD)

The MAX17031 requires an external 5V bias supply (VDDand VCC) in addition to the battery. Typically, this 5V bias supply is generated by the internal 100mA LDO5 or from the notebook’s 95%-efficient 5V main supply. Keeping these bias supply inputs independent improves the overall efficiency. When ONLDO is enabled, VDDand VCCmust be supplied from LDO5.

The VDDbias supply input powers the internal gate drivers and the VCCbias supply input powers the analog control blocks. The maximum current required is dominated by the switching losses of the drivers and can be estimated as follows:

BIAS(MAX)

= I

CC(MAX)

+ fSWQG≈ 30mA to 60mA (typ)

Free-Running Constant-On-Time PWM

Controller with Input Feed-Forward

The Quick-PWM control architecture is a pseudo-fixedfrequency, constant on-time, current-mode regulator with voltage feed-forward. This architecture relies on the output filter capacitor’s ESR to act as a currentsense resistor, so the feedback ripple voltage provides the PWM ramp signal. The control algorithm is simple: the high-side switch on-time is determined solely by a one-shot whose pulse width is inversely proportional to input voltage and directly proportional to output voltage. Another one-shot sets a minimum off-time (400ns typ). The on-time one-shot is triggered if the error comparator is low, the low-side switch current is below the valley current-limit threshold, and the minimum off-time one-shot has timed out.

On-Time One-Shot

The heart of the PWM core is the one-shot that sets the high-side switch on-time. This fast, low-jitter, adjustable one-shot includes circuitry that varies the on-time in response to battery and output voltage. The high-side switch on-time is inversely proportional to the battery voltage as sensed by IN, and proportional to the feedback voltage:

where K (switching period) is set 2.5µs for side 1 and

3.3µs for side 2. For continuous conduction operation, the actual switching frequency can be estimated by:

where V

DROP1

is the sum of the parasitic voltage drops in the inductor discharge path, including synchronous rectifier, inductor, and PCB resistances; V

DROP2

is the sum of the parasitic voltage drops in the charging path, including the high-side switch, inductor, and PCB resistances; and tONis the on-time calculated by the MAX17031.

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

16 ______________________________________________________________________________________

()

tVV V

×+

()

ON IN DROP DROP

OUT

OUT DROP

−

Modes of Operation

Forced-PWM Mode (V

SKIP

= 1.8V)

The low-noise forced-PWM mode (V

SKIP

= 1.8V) disables the zero-crossing comparator, which controls the low-side switch on-time. This forces the low-side gatedrive waveform to constantly be the complement of the high-side gate-drive waveform, so the inductor current reverses at light loads while DH maintains a duty factor of V

OUT/VIN

. The benefit of forced-PWM mode is to keep the switching frequency fairly constant. However, forced-PWM operation comes at a cost: the no-load 5V bias current remains between 20mA to 60mA depending on the switching frequency and MOSFET selection.

The MAX17031 automatically uses forced-PWM operation during shutdown regardless of the SKIP configuration.

Automatic Pulse-Skipping Mode (V

SKIP

> 2V)

In skip mode (V

SKIP

> 2V), an inherent automatic switchover to PFM takes place at light loads. This switchover is affected by a comparator that truncates the low-side switch on-time at the inductor current’s zero crossing. The zero-crossing comparator output is set by the differential voltage across LX and GND.

DC output-accuracy specifications refer to the integrated threshold of the error comparator. When the inductor is in continuous conduction, the MAX17031 regulates the valley of the output ripple and the internal integrator removes the actual DC output-voltage error caused by the output-ripple voltage and internal slope compensation. In discontinuous conduction (V

SKIP

> 2V and I

OUT

< I

LOAD(SKIP)

), the integrator cannot correct for the lowfrequency output ripple error, so the output voltage has a DC regulation level higher than the error comparator threshold by approximately 1.5% due to slope compensation and output ripple voltage.

Ultrasonic Mode (V

SKIP

= GND)

Shorting SKIP to ground activates a unique pulseskipping mode with a guaranteed minimum switching frequency of 20kHz. This ultrasonic pulse-skipping mode eliminates audio-frequency modulation that would otherwise be present when a lightly loaded controller automatically skips pulses. In ultrasonic mode, the controller automatically transitions to fixed-frequency PWM operation when the load reaches the same critical conduction point (I

LOAD(SKIP)

) that occurs when normally

pulse skipping.

An ultrasonic pulse occurs (Figure 4) when the controller detects that no switching has occurred within the last 37µs. Once triggered, the ultrasonic circuitry pulls DL high, turning on the low-side MOSFET to induce a negative inductor current. After the inductor current reaches the negative ultrasonic current threshold, the controller turns off the low-side MOSFET (DL pulled low) and triggers a constant on-time (DH driven high). When the on-time has expired, the controller reenables the low-side MOSFET until the inductor current drops below the zero-crossing threshold. Starting with a DL pulse greatly reduces the peak output voltage when compared to starting with a DH pulse.

The output voltage at the beginning of the ultrasonic pulse determines the negative ultrasonic current threshold, corresponding to:

where RCSis the current-sense resistance seen across LX to GND.

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 17

Figure 4. Ultrasonic Waveforms

VIR

NEG US L C

()

40µs (MAX)

INDUCTOR CURRENT

ZERO-CROSSING

DETECTION

SONIC

ON-TIME (tON)

MAX17031

Secondary Feedback (SKIP)

When the controller skips pulses (V

SKIP

> 2V), the long time between pulses (especially if the output is sinking current) allows the external charge-pump voltage or transformer secondary winding voltage to drop. Connecting a resistor-divider between the secondary output to SKIP to ground sets up a minimum refresh threshold. When the SKIP voltage drops below its 2V threshold, the MAX17031 enters forced-PWM mode. This forces the controller to begin switching, allowing the external unregulated charge pump (or transformer secondary winding) to be refreshed.

Valley Current-Limit Protection

The current-limit circuit employs a unique “valley” current-sensing algorithm that senses the inductor current through the low-side MOSFET—across LX to analog GND. If the current through the low-side MOSFET exceeds the valley current-limit threshold, the PWM controller is not allowed to initiate a new cycle. The actual peak current is greater than the valley currentlimit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the inductor value and battery voltage. When combined with the undervoltage protection circuit, this currentlimit method is effective in almost every circumstance.

In forced-PWM mode, the MAX17031 also implements a negative current limit to prevent excessive reverse inductor currents when V

OUT

is sinking current. The negative current-limit threshold is set to approximately 120% of the positive current limit.

POR, UVLO

When VCCrises above the power-on reset (POR) threshold, the MAX17031 clears the fault latches, forces the low-side MOSFET to turn on (DL high), and resets the soft-start circuit, preparing the controller for power-up. However, the VCCundervoltage lockout (UVLO) circuitry inhibits switching until VCCreaches 4.2V (typ). When VCCrises above 4.2V and the controller has been enabled (ON_ pulled high), the controller activates the enabled PWM controllers and initializes soft-start.

When VCCdrops below the UVLO threshold (falling edge), the controller stops switching, and DH and DL are pulled low. When the 2V POR falling-edge threshold is reached, the DL state no longer matters since there is not enough voltage to force the switching MOSFETs into a low on-resistance state, so the controller pulls DL high, allowing a soft discharge of the output capacitors (damped response). However, if the VCCrecovers

before reaching the falling POR threshold, DL remains low until the error comparator has been properly powered up and triggers an on-time.

Soft-Start and Soft-Shutdown

The MAX17031 includes voltage soft-start and softshutdown—slowly ramping up and down the target voltage. During startup, the slew-rate control softly slews the target voltage over a 1ms startup period. This long startup period reduces the inrush current during startup.

When ON1 or ON2 is pulled low or the output undervoltage fault latch is set, the respective output automatically enters soft-shutdown; the regulator enters PWM mode and ramps down its output voltage over a 1ms period. After the output voltage drops below 0.1V, the MAX17031 pulls DL high, clamping the output and LX switching node to ground, preventing leakage currents from pulling up the output and minimizing the negative output voltage undershoot during shutdown.

Output Voltage

DC output-accuracy specifications in the

Electrical

Characteristics

table refer to the error comparator’s threshold. When the inductor continuously conducts, the MAX17031 regulates the valley of the output ripple, so the actual DC output voltage is lower than the slope-compensated trip level by 50% of the output ripple voltage. For PWM operation (continuous conduction), the output voltage is accurately defined by the following equation:

where V

NOM

is the nominal feedback voltage, A

CCV

the integrator’s gain, and V

RIPPLE

is the output ripple

voltage (V

RIPPLE

= ESR x ∆I

INDUCTOR

, as described in

the

Output Capacitor Selection

section).

In discontinuous conduction (I

OUT

< I

LOAD(SKIP)

), the longer off-times allow the slope compensation to increase the threshold voltage by as much as 1%, so the output voltage regulates slightly higher than it would in PWM operation.

Internal Integrator

The internal integrator improves the output accuracy by removing any output accuracy errors caused by the slope compensation, output ripple voltage, and erroramplifier offset. Therefore, the DC accuracy (in forcedPWM mode) depends on the integrator’s gain, the integrator’s offset, and the accuracy of the integrator’s reference input.

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

18 ______________________________________________________________________________________

⎛

OUT PWM NOM

()

⎜ ⎝

RI PPLE

CCV

⎞ ⎟

⎠

Power-Good Outputs (PGOOD)

and Fault Protection

PGOOD is the open-drain output that continuously monitors both output voltages for undervoltage and overvoltage conditions. PGOOD is actively held low in shutdown (ON1 or ON2 = GND), during soft-start, and soft-shutdown. Approximately 20µs (typ) after the softstart terminates, PGOOD becomes high impedance as long as both output voltages exceed 85% of the nominal fixed-regulation voltage. PGOOD goes low if the output voltage drops 15% below the regulation voltage, or if the SMPS controller is shut down. For a logic-level PGOOD output voltage, connect an external pullup resistor between PGOOD and the logic power supply. A 100kΩ pullup resistor works well in most applications.

Overvoltage Protection (OVP)

When the output voltage rises 15% above the fixedregulation voltage, the controller immediately pulls PGOOD low, sets the overvoltage fault latch, and immediately pulls the respective DL_ high—clamping the output fault to GND. Toggle either ON1 or ON2 input, or cycle VCCpower below its POR threshold to clear the fault latch and restart the controller.

Undervoltage Protection (UVP)

When the output voltage drops 30% below the fixedregulation voltage, the controller immediately pulls the PGOOD low, sets the undervoltage fault latch, and begins the shutdown sequence. After the output voltage drops below 0.1V, the synchronous rectifier turns on, clamping the output to GND regardless of the output voltage. Toggle either ON1 or ON2 input, or cycle VCCpower below its POR threshold to clear the fault latch and restart the controller.

Thermal-Fault Protection (T

SHDN

)

The MAX17031 features a thermal-fault protection circuit. When the junction temperature rises above +160°C, a thermal sensor activates the fault latch, pulls PGOOD low, enables the 10Ω discharge circuit, and disables the controller—DH and DL pulled low. Toggle ONLDO or cycle IN power to reactivate the controller after the junction temperature cools by 15°C.

Design Procedure

Firmly establish the input-voltage range and maximum load current before choosing an inductor operating point (ripple-current ratio). The primary design goal is choosing a good inductor operating point, and the following three factors dictate the rest of the design:

• Input Voltage Range: The maximum value (V

IN(MAX)

) must accommodate the worst-case, high ACadapter voltage. The minimum value (V

IN(MIN)

) must account for the lowest battery voltage after drops due to connectors, fuses, and battery-selector switches. If there is a choice at all, lower input voltages result in better efficiency.

• Maximum Load Current: There are two values to consider. The peak load current (I

LOAD(MAX)

) determines the instantaneous component stresses and filtering requirements and thus drives output capacitor selection, inductor saturation rating, and the design of the current-limit circuit. The continuous load current (I

LOAD

) determines the thermal stresses and thus drives the selection of input capacitors, MOSFETs, and other critical heat-contributing components.

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 19

Table 3. Fault Protection and Shutdown Operation Table

MODE CONTROLLER STATE DRI VER STATE

Shutdown (ON_ = High to Low) Output UVP (Latched)

Output OVP (Latched)

UVLO (VCC Falling-Edge) Thermal Fault (Latched)

UVLO (VCC Ri s ing Edge)

VCC Below POR SMPS inactive, 10 output discharge active.

Voltage soft-shutdown initiated. Internal error-amplifier target slowly ramped down to GND and output activel y discharged (automaticall y enters forced-PWM mode).

Controller shuts down and EA target internally slewed down. Controller remains off until ON_ toggled or V power cycled.

SMPS controller disabled (assuming ON_ pulled high), 10 output discharge acti ve.

DL driven high and DH pulled low after soft-shutdown completed (output < 0.1V).

DL immediately DH pulled low.

DL and DH pulled low.

DL driven high, DH pulled low.

driven high,

MAX17031

Inductor Operating Point: This choice provides trade-offs between size vs. efficiency and transient response vs. output ripple. Low inductor values provide better transient response and smaller physical size, but also result in lower efficiency and higher output ripple due to increased ripple currents. The minimum practical inductor value is one that causes the circuit to operate at the edge of critical conduction (where the inductor current just touches zero with every cycle at maximum load). Inductor values lower than this grant no further size-reduction benefit. The optimum operating point is usually found between 20% and 50% value at which PFM/PWM switchover occurs.

Inductor Selection

The switching frequency and inductor operating point determine the inductor value as follows:

For example: I

LOAD(MAX)

= 4A, VIN= 12V, V

OUT2

2.5V, f

= 355kHz, 30% ripple current or LIR = 0.3:

Find a low-loss inductor having the lowest possible DC resistance that fits in the allotted dimensions. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 200kHz. The core must be large enough not to saturate at the peak inductor current (I

PEAK

Most inductor manufacturers provide inductors in standard values, such as 1.0µH, 1.5µH, 2.2µH, 3.3µH, etc. Also look for nonstandard values, which can provide a better compromise in LIR across the input voltage range. If using a swinging inductor (where the no-load inductance decreases linearly with increasing current), evaluate the LIR with properly scaled inductance values.

Transient Response

The inductor ripple current also impacts transientresponse performance, especially at low VIN- V

OUT

differentials. Low inductor values allow the inductor current to slew faster, replenishing charge removed from the output filter capacitors by a sudden load step. The amount of output sag is also a function of the maximum duty factor, which can be calculated from the ontime and minimum off-time:

where t

OFF(MIN)

is the minimum off-time (see the

Electrical Characteristics

table).

The amount of overshoot during a full-load to no-load transient due to stored inductor energy can be calculated as:

Setting the Current Limit

The minimum current-limit threshold must be great enough to support the maximum load current when the current limit is at the minimum tolerance value. The valley of the inductor current occurs at I

LOAD(MAX)

minus

half the ripple current; therefore:

where I

LIM(VAL)

equals the minimum valley current-limit threshold voltage divided by the current-sense resistance (R

SENSE

). When using a 100kΩ ILIM resistor, the

minimum valley current-limit threshold is 40mV.

Connect a resistor between ILIM_ and analog ground to set the adjustable current-limit threshold. The valley current-limit threshold is approximately 1/10 the ILIM voltage formed by the external resistance and internal 5µA current source. The 40kΩ to 400kΩ adjustment range corresponds to a 20mV to 200mV valley currentlimit threshold. When adjusting the current limit, use 1% tolerance resistors to prevent significant inaccuracy in the valley current-limit tolerance.

Output Capacitor Selection

The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load-transient requirements, yet have high enough ESR to satisfy stability requirements.

For processor core voltage converters and other applications where the output is subject to violent load transients, the output capacitor’s size depends on how much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag due to finite capacitance:

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

20 ______________________________________________________________________________________

12 355 4 0 3

VVV

OUT IN OUT

Vf I LIR

IN SW LOAD MAX

VVV

×−

25 12 25

()

VkHzA

×××

−

()

465

.µ

⎡

⎛

SAG

()

OUT OUT

() (2))

LOAD MAX

⎡

⎛

⎢

⎜ ⎝

⎢

⎣

SOA R

()

≈

⎢

⎜ ⎝

⎣

−

VV K

()

IN OUT

∆

LOAD MAX

OUT OUT

⎞

OUT

⎟

⎠

()

+∆

⎞ ⎟

⎠

LIM VAL LOAD MAX

>−

() ( )

ILIR

⎛

LOAD MAX

⎜ ⎝

()

OFF MIN

−2C V

OFF MMIN)

(

⎞ ⎟

⎠

⎤ ⎥ ⎦

⎤ ⎥ ⎥

⎦

PEAK LOAD MAX

LIR

⎛ ⎜

()

⎝

⎞

⎟ ⎠

In applications without large and fast load transients, the output capacitor’s size often depends on how much ESR is needed to maintain an acceptable level of output voltage ripple. The output ripple voltage of a stepdown controller equals the total inductor ripple current multiplied by the output capacitor’s ESR. Therefore, the maximum ESR required to meet ripple specifications is:

The actual capacitance value required relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value (this is true of tantalums, OS-CONs, polymers, and other electrolytics).

When using low-capacity filter capacitors, such as ceramic capacitors, size is usually determined by the capacity needed to prevent V

SAG

and V

SOAR

from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem (see the V

SAG

and V

SOAR

equa-

tions in the

Transient Response

section). However, lowcapacity filter capacitors typically have high ESR zeros that might affect the overall stability (see the

Output

Capacitor Stability Considerations

section).

Output Capacitor Stability Considerations

For Quick-PWM controllers, stability is determined by the value of the ESR zero relative to the switching frequency. The boundary of instability is given by the following equation:

where:

For a typical 300kHz application, the ESR zero frequency must be well below 95kHz, preferably below 50kHz. Tantalum and OS-CON capacitors in widespread use at the time of publication have typical ESR zero frequencies of 25kHz. In the design example used for inductor selection, the ESR needed to support 25mV

P-P

ripple is

25mV/1.2A = 20.8mΩ. One 220µF/4V SANYO polymer (TPE) capacitor provides 15mΩ (max) ESR. This results in a zero at 48kHz, well within the bounds of stability.

Do not put high-value ceramic capacitors directly on OUT1 and OUT2 pins to ensure stability. Large ceramic capacitors can have a high-ESR zero frequency and cause erratic, unstable operation. However, it is easy to add enough series resistance by placing the capacitors a couple of inches downstream from the feedback sense point, which should be as close as possible to the inductor.

Unstable operation manifests itself in two related but distinctly different ways: double-pulsing and fast-feedback loop instability. Double-pulsing occurs due to noise on the output or because the ESR is so low that there is not enough voltage ramp in the output-voltage signal. This “fools” the error comparator into triggering a new cycle immediately after the 400ns minimum offtime period has expired. Double-pulsing is more annoying than harmful, resulting in nothing worse than increased output ripple. However, it can indicate the possible presence of loop instability due to insufficient ESR. Loop instability results in oscillations at the output after line or load steps. Such perturbations are usually damped, but can cause the output voltage to rise above or fall below the tolerance limits.

The easiest method for checking stability is to apply a very fast zero-to-max load transient and carefully observe the output-voltage ripple envelope for overshoot and ringing. It can help to simultaneously monitor the inductor current with an AC current probe. Do not allow more than one cycle of ringing after the initial step-response under/overshoot.

Input Capacitor Selection

The input capacitor must meet the ripple current requirement (I

RMS

) imposed by the switching currents:

For most applications, nontantalum chemistries (ceramic, aluminum, or OS-CON) are preferred due to their resistance to power-up surge currents typical of systems with a mechanical switch or connector in series with the input. If the MAX17031 is operated as the second stage of a two-stage power conversion system, tantalum input capacitors are acceptable. In either configuration, choose a capacitor that has less than 10°C temperature rise at the RMS input current for optimal reliability and lifetime.

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 21

ESR

≤

ESR

STEP

≤

∆

ILIR

LOAD MAX

()

LOAD MAX

RI PPLE

()

ESR

2π

≤

ESR

××

ESR OUT

RMS LOAD

⎛

VVV

OUT IN OUT

⎜ ⎜

⎝

−

()

⎞ ⎟

⎟ ⎠

MAX17031

Power-MOSFET Selection

Most of the following MOSFET guidelines focus on the challenge of obtaining high load-current capability when using high-voltage (> 20V) AC adapters. Lowcurrent applications usually require less attention.

The high-side MOSFET (NH) must be able to dissipate the resistive losses plus the switching losses at both V

IN(MIN)

and V

IN(MAX)

. Ideally, the losses at V

IN(MIN)

should be roughly equal to the losses at V

IN(MAX)

, with

lower losses in between. If the losses at V

IN(MIN)

are significantly higher, consider increasing the size of NH. Conversely, if the losses at V

IN(MAX)

are significantly

higher, consider reducing the size of N

. If VINdoes not vary over a wide range, maximum efficiency is achieved by selecting a high-side MOSFET (N

) that

has conduction losses equal to the switching losses.

Choose a low-side MOSFET (N

) that has the lowest

possible on-resistance (R

DS(ON)

), comes in a moderate-sized package (i.e., 8-pin SO, DPAK, or D2PAK), and is reasonably priced. Ensure that the MAX17031 DL_ gate driver can supply sufficient current to support the gate charge and the current injected into the parasitic drain-to-gate capacitor caused by the high-side MOSFET turning on; otherwise, cross-conduction problems could occur. Switching losses are not an issue for the low-side MOSFET since it is a zero-voltage switched device when used in the step-down topology.

Power-MOSFET Dissipation

Worst-case conduction losses occur at the duty factor extremes. For the high-side MOSFET (NH), the worstcase power dissipation due to resistance occurs at minimum input voltage:

Generally, use a small high-side MOSFET to reduce switching losses at high input voltages. However, the R

DS(ON)

required to stay within package power-dissipation limits often limits how small the MOSFET can be. The optimum occurs when the switching losses equal the conduction (R

DS(ON)

) losses. High-side switching losses do not become an issue until the input is greater than approximately 15V.

Calculating the power dissipation in high-side MOSFETs (NH) due to switching losses is difficult, since it must allow for difficult-to-quantify factors that influence the turn-on and turn-off times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PCB layout

characteristics. The following switching loss calculation provides only a very rough estimate and is no substitute for breadboard evaluation, preferably including verification using a thermocouple mounted on NH:

where C

OSS

is the high-side MOSFET’s output capaci-

tance, Q

G(SW)

is the charge needed to turn on the high-

side MOSFET, and I

GATE

is the peak gate-drive

source/sink current (1A typ).

Switching losses in the high-side MOSFET can become a heat problem when maximum AC adapter voltages are applied due to the squared term in the switchingloss equation provided above. If the high-side MOSFET chosen for adequate R

DS(ON)

at low battery voltages becomes extraordinarily hot when subjected to V

IN(MAX)

, consider choosing another MOSFET with

lower parasitic capacitance.

For the low-side MOSFET (NL), the worst-case power dissipation always occurs at maximum battery voltage:

The absolute worst case for MOSFET power dissipation occurs under heavy overload conditions that are greater than I

LOAD(MAX)

but are not high enough to exceed the current limit and cause the fault latch to trip. To protect against this possibility, “overdesign” the circuit to tolerate:

where I

VALLEY(MAX)

is the maximum valley current allowed by the current-limit circuit, including threshold tolerance and sense-resistance variation. The MOSFETs must have a relatively large heatsink to handle the overload power dissipation.

Choose a Schottky diode (DL) with a forward voltage drop low enough to prevent the low-side MOSFET’s body diode from turning on during the dead time. As a general rule, select a diode with a DC current rating equal to 1/3 the load current. This diode is optional and can be removed if efficiency is not critical.

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

22 ______________________________________________________________________________________

PD (NH Resistive) =

⎛ ⎜

⎝

⎞

OUT

()

LOAD D

⎟

⎠

SSON()

PD (NH Switching) =

⎛

IN MAX LOAD SW G SW() ()

⎜ ⎝

⎛

NN OSS SW

⎜ ⎝

GGATE

⎞ ⎟

⎠

⎞ ⎟

⎠

VIfQ

⎡

PD (NL Resistive) = 1−

⎢ ⎢⎢

⎣

⎛

OUT

⎜

⎝

IN MAX()

⎤

⎞

⎥

()

⎟

LOAD DS ON2()

⎥

⎠

⎦

ILIR

LOAD VALLEY MAX

()

⎛

LOAD MAX

⎜ ⎝

()

⎞ ⎟

⎠

Applications Information

Step-Down Converter

Dropout Performance

The output voltage-adjustable range for continuousconduction operation is restricted by the nonadjustable minimum off-time one-shot. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on- and off-times. Manufacturing tolerances and internal propagation delays introduce an error to the TON K-factor. This error is greater at higher frequencies. Also, keep in mind that transient response performance of buck regulators operated too close to dropout is poor, and bulk output capacitance must often be added (see the V

SAG

equa-

tion in the

Design Procedure

section).

The absolute point of dropout is when the inductor current ramps down during the minimum off-time (∆I

DOWN

) as much as it ramps up during the on-time (∆IUP). The ratio h = ∆IUP/∆I

DOWN

indicates the controller’s ability to slew the inductor current higher in response to increased load, and must always be greater than 1. As h approaches 1, the absolute minimum dropout point, the inductor current cannot increase as much during each switching cycle, and V

SAG

greatly increases

unless additional output capacitance is used.

A reasonable minimum value for h is 1.5, but adjusting this up or down allows trade-offs between V

SAG

, output capacitance, and minimum operating voltage. For a given value of h, the minimum operating voltage can be calculated as:

where V

DROP2

is the parasitic voltage drop in the

charge path (see the

On-Time One-Shot

section),

OFF(MIN)

is from the

Electrical Characteristics

table, and K (1/fSW) is the switching period. The absolute minimum input voltage is calculated with h = 1.

If the calculated V

IN(MIN)

is greater than the required minimum input voltage, then operating frequency must be reduced or output capacitance added to obtain an acceptable V

SAG

. If operation near dropout is anticipated,

calculate V

SAG

to be sure of adequate transient response.

Dropout Design Example:

OUT2

= 2.5V

fSW= 355kHz

K = 3.0µs, worst-case K

MIN

= 3.3µs

OFF(MIN)

= 500ns

DROP2

= 100mV

h = 1.5:

Calculating again with h = 1 and the typical K-factor value (K = 3.3µs) gives the absolute limit of dropout:

Therefore, VINmust be greater than 3.06V, even with very large output capacitance, and a practical input voltage with reasonable output capacitance would be 3.47V.

PCB Layout Guidelines

Careful PCB layout is critical to achieving low switching losses and clean, stable operation. The switching power stage requires particular attention. If possible, mount all the power components on the top side of the board, with their ground terminals flush against one another. Follow these guidelines for good PCB layout:

• Keep the high-current paths short, especially at the ground terminals. This practice is essential for stable, jitter-free operation.

• Keep the power traces and load connections short. This practice is essential for high efficiency. Using thick copper PCBs (2oz vs. 1oz) can enhance fullload efficiency by 1% or more. Correctly routing PCB traces is a difficult task that must be approached in terms of fractions of centimeters, where a single milliohm of excess trace resistance causes a measurable efficiency penalty.

• Minimize current-sensing errors by connecting LX_ directly to the drain of the low-side MOSFET.

• When trade-offs in trace lengths must be made, it is preferable to allow the inductor charging path to be made longer than the discharge path. For example, it is better to allow some extra distance between the input capacitors and the high-side MOSFET than to allow distance between the inductor and the lowside MOSFET or between the inductor and the output filter capacitor.

• Route high-speed switching nodes (BST_, LX_, DH_, and DL_) away from sensitive analog areas (REF, and OUT_).

A sample layout is available in the MAX17031 Evaluation Kit data sheet.

MAX17031

Dual Quick-PWM Step-Down Controller with Low-

Power LDO and RTC Regulator for MAIN Supplies

______________________________________________________________________________________ 23

IN MIN

()

OUT DROP

⎛

−

⎜ ⎝

OFF MIN

()

⎞ ⎟

⎠

IN(MIN)

25 01

⎛

1 5 500

−

⎜ ⎝

25 01

⎛

1 500

−

⎜ ⎝

33.47V

⎞

⎟ ⎠

006V

=3.

⎞

⎟ ⎠

MAX17031

Dual Quick-PWM Step-Down Controller with LowPower LDO and RTC Regulator for MAIN Supplies

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

Layout Procedure

1) Place the power components first, with ground terminals adjacent (NL_source, CIN, C

OUT_

, and D

anode). If possible, make all these connections on the top layer with wide, copper-filled areas.

2) Mount the controller IC adjacent to the low-side MOSFET, preferably on the back side opposite N

and NH_in order to keep LX_, GND, DH_, and the DL_ gate-drive lines short and wide. The DL_ and DH_ gate traces must be short and wide (50 mils to 100 mils wide if the MOSFET is 1in from the controller IC) to keep the driver impedance low and for proper adaptive dead-time sensing.

3) Group the gate-drive components (BST_ capacitor, V

bypass capacitor) together near the controller IC.

4) Make the DC-DC controller ground connections as shown in Figure 1. This diagram can be viewed as having two separate ground planes: power ground, where all the high-power components go; and an analog ground plane for sensitive analog components. The analog ground plane and power ground plane must meet only at a single point directly at the IC.

5) Connect the output power planes directly to the output filter capacitor positive and negative terminals with multiple vias. Place the entire DC-DC converter circuit as close to the load as is practical.

Package Information

For the latest package outline information and land patterns, go to www.maxim-ic.com/packages

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

24 TQFN T2444-3

21-0139

Chip Information

TRANSISTOR COUNT: 12,197

PROCESS: BiCMOS

Rainbow Electronics MAX17031 User Manual

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Frequently Asked Questions

User Manual