Rainbow Electronics MAX16834 User Manual

General Description

The MAX16834 is a current-mode high-brightness LED
(HB LED) driver for boost, buck-boost, SEPIC, and high­side buck topologies. In addition to driving an n-channel
power MOSFET switch controlled by the switching con­troller, it also drives an n-channel PWM dimming switch to
achieve LED PWM dimming. The MAX16834 integrates
all the building blocks necessary to implement a fixed-fre­quency HB LED driver with wide-range dimming control.
The MAX16834 features constant-frequency peak cur­rent-mode control with programmable slope compensa­tion to control the duty cycle of the PWM controller.

A dimming driver designed to drive an external n-chan­nel MOSFET in series with the LED string provides
wide-range dimming control up to 20kHz. In addition to
PWM dimming, the MAX16834 provides analog dim­ming using a DC input at REFI. The programmable
switching frequency (100kHz to 1MHz) allows design
optimization for efficiency and board space reduction.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz while an
external clock signal at RT/SYNC disables the internal
oscillator and allows the MAX16834 to synchronize to
an external clock. The MAX16834’s integrated high­side current-sense amplifier eliminates the need for a
separate high-side LED current-sense amplifier in
buck-boost applications.

The MAX16834 operates over a wide supply range of

4.75V to 28V and includes a 3A sink/source gate driver
for driving a power MOSFET in high-power LED driver
applications. The MAX16834 is also suitable for DC-DC
converter applications such as boost or buck-boost.
Additional features include external enable/disable
input, an on-chip oscillator, fault indicator output (FLT)
for LED open/short or overtemperature conditions, and
an overvoltage protection sense input (OVP+) for true
overvoltage protection.

The MAX16834 is available in a thermally enhanced
4mm x 4mm, 20-pin TQFN-EP package and is specified
over the automotive -40°C to +125°C temperature range.

Applications

Single-String LED LCD Backlighting

Automotive Rear and Front Lighting

Projection System RGB LED Light Sources

Architectural and Decorative Lighting (MR16, M111)

Spot and Ambient Lights

DC-DC Boost/Buck-Boost Converters

Features

o Wide Input Operating Voltage Range (4.75V to

28V)

o 3000:1 PWM Dimming

o Analog Dimming

o Integrated PWM Dimming MOSFET Driver

o Integrated High-Side Current-Sense Amplifier for

LED Current Sense in Buck-Boost Converter

o 100kHz to 1MHz Programmable High-Frequency

Operation

o External Clock Synchronization Input

o Programmable UVLO

o Internal 7V Low-Dropout Regulator
o Fault Output (FLT) for Overvoltage, Overcurrent,

and Thermal Warning Faults

o Programmable True Differential Overvoltage

Protection

o 20-Pin TQFN-EP Package

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

________________________________________________________________

Maxim Integrated Products

1

Simplified Application Circuit

Ordering Information

19-4235; Rev 0; 8/08

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

+

Denotes a lead-free/RoHS-compliant package.

*

EP = Exposed pad.

PART TEMP RANGE

PIN-PACKAGE

MAX16834ATP+

-40°C to +125°C 20 TQFN-EP*

Pin Configuration appears at end of data sheet.

IN

MAX16834

PWMDIM

REFI

PGND

BOOST LED DRIVER

NDRV

CS

DIMOUT

SENSE+

LED+

LEDs

LED-

V

OFF

IN

ANALOG

ON

DIM

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VIN= VHV= 12V, V

UVEN

= 5V, VLV= V

PWMDIM

= SGND, C

VCC

= 4.7µF, C

LCV

= 100nF, C

REF

= 100nF, R

SENSE+

= 0.1Ω,

R

RT

= 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-

layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial

.

IN, HV, LV to SGND................................................-0.3V to +30V

OVP+, SENSE+, DIMOUT, CLV to SGND ..............-0.3V to +30V

SENSE+ to LV........................................................-0.3V to +0.3V

HV, IN to LV ............................................................-0.3V to +30V

OVP+, CLV, DIMOUT to LV ......................................-0.3V to +6V

PGND to SGND .....................................................-0.3V to +0.3V

V

CC

to SGND..........................................................-0.3V to +12V

NDRV to PGND...........................................-0.3V to (V

CC

+ 0.3V)

All Other Pins to SGND.............................................-0.3V to +6V

NDRV Continuous Current................................................±50mA

DIMOUT Continuous Current..............................................±2mA

V

CC

Short-Circuit Current to SGND Duration ...........................1s

Continuous Power Dissipation (T

A

= +70°C)
20-Pin TQFN 4mm x 4mm

(derate 25.6mW/°C* above +70°C) ............................2051mW

Junction-to-Ambient Thermal Resistance (

θ

JA

) (Note 1).....39°C/W

Junction-to-Case Thermal Resistance (

θ

JC

) (Note 1) ........6°C/W

Operating Temperature Range .........................-40°C to +125°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

*

As per JEDEC51 standard (multilayer board).

Input Voltage Range V

Quiescent Supply Current I

Shutdown Supply Current I

INTERNAL LINEAR REGULATOR (VCC)

Output Voltage V

Dropout Voltage V

Short-Circuit Current VCC = 0V, VIN = 12V 80 300 mA

LINEAR REGULATOR (CLV)

Output Voltage (V

Dropout Voltage V

Short-Circuit Current V

REFERENCE VOLTAGE (REF)

Output Voltage V

REF Short-Circuit Current V

UNDERVOLTAGE LOCKOUT/ENABLE INPUT (UVEN)

UVEN On Threshold Voltage V

UVEN Threshold Voltage
Hysteresis

Input Leakage Current I

PWMDIM

PWMDIM On Threshold Voltage V

PWMDIM Threshold Voltage
Hysteresis

Input Leakage Current V

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

IN

Q

SHDN

CC

OD

CLV - VLV

DO

REF

UVEN_THUP

LEAK

PWMDIM

Excluding I

V

UVEN

0 ≤ ICC ≤ 50mA, 9.5V ≤ VIN ≤ 28V 6.3 7 7.7 V

ICC = 35mA (Note 2) 0.65 1.8 V

0 ≤ I

CLV

)

6V ≤ V

I

= 2mA, 0 ≤ VLV ≤ 23.3V (Note 3) 0.5 V

CLV

= 12V, VIN = 12V, VHV = 24V 2.2 10 mA

CLV

0 ≤ I

REF

= 0 30 mA

REF

V

UVEN

PWMDIM

LED

= 0 30 60 µA

≤ 2mA, 6V ≤ VHV ≤ 28V,

≤ 22V

(HV-LV)

≤ 1mA, 4.75V ≤ VIN ≤ 28V 3.625 3.70 3.775 V

= 0 I1I µA

= 0 I1I µA

4.75 28 V

4.7 5 5.3 V

1.395 1.435 1.475 V

200 mV

1.395 1.435 1.475 V

200 mV

610mA

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VIN= VHV= 12V, V

UVEN

= 5V, VLV= V

PWMDIM

= SGND, C

VCC

= 4.7µF, C

LCV

= 100nF, C

REF

= 100nF, R

SENSE+

= 0.1Ω,

R

RT

= 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)

OSCILLATOR

Oscillator Frequency f

Oscillator Frequency Range (Note 4) 100 1000 kHz

External Sync Input Clock High
Threshold

External Sync Input Clock Low
Threshold

External Sync Input High Pulse
Width

Maximum External Sync Period 50 µs

SLOPE COMPENSATION (SC)

SC Pullup Current I

SC Discharge Resistance R

REFI

REFI Input Bias Current V

REFI Input Common-Mode Range (Note 4) 0 2 V

SENSE+

SENSE+ Input Bias Current (V

HIGH-SIDE LED CURRENT-SENSE AMPLIFIER (V

Input Offset Voltage VLV > 5V, (V

Voltage Gain A

3dB Bandwidth

LOW-SIDE LED CURRENT-SENSE AMPLIFIER

Input Offset Voltage VLV < 1V, (V

Voltage Gain A

3dB Bandwidth 600 kHz

CURRENT ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)

Transconductance g

Open-Loop DC Gain A

Input Offset Voltage -10 0 +10 mV

COMP Voltage Range V

PWM COMPARATOR

Input Offset Voltage 0.6 0.65 0.70 V

Propagation Delay t

Minimum On-Time t

Duty Cycle (Note 4) 90 99.5 %

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

OSC

SCPU

SCD

V

V

m

V

COMP

PD

ON(MIN)

R

RT/SYNC

R

RT/SYNC

(Note 4) 2 V

(Note 4) 0.4 V

(Note 4) 200 ns

VSC = 100mV 80 100 120 µA

VSC = 100mV 8 Ω

REFI

VLV > 5V, (V

(V

(V

VLV < 1V, (V

V

COMP

(Note 4) 0.4 2.5 V

50mV overdrive 40 ns

On-time includes blanking time 100 ns

= 5kΩ 0.9 1 1.1 MHz
= 25kΩ 180 200 220 kHz

= 1V I1I µA

- VLV) = 100mV 250 µA

SENSE+

- VLV)

SENSE+

- VLV) = 0.1V, no load 1.8 MHz

SENSE+

- VLV) = 0.02V, no load 600 kHz

SENSE+

= 2V, V

- VLV) = 5mV -2.4 0 +2.4 mV

SENSE+

- VLV) = 0.2V 9.7 9.9 10.1 V/V

SENSE+

- VLV) = 0V -2 0 +2 mV

SENSE+

- VLV) = 0.2V 9.7 9.9 10.1 V/V

SENSE+

= 5V 400 500 600 µS

PWMDIM

60 dB

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(VIN= VHV= 12V, V

UVEN

= 5V, VLV= V

PWMDIM

= SGND, C

VCC

= 4.7µF, C

LCV

= 100nF, C

REF

= 100nF, R

SENSE+

= 0.1Ω,

R

RT

= 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)

Note 2: Dropout voltage is defined as VIN- VCC, when VCCis 100mV below the value of VCCfor VIN= 9.5V.
Note 3: Dropout is defined as V

HV

- V

CLV

, when V

CLV

is 100mV below the value of V

CLV

for VHV= 8V.

Note 4: Not production tested. Guaranteed by design.

CURRENT PEAK LIMIT COMPARATOR

Trip Threshold Voltage 0.25 0.3 0.35 V

Propagation Delay 50mV overdrive with respect to NDRV 40 ns

OVERVOLTAGE PROTECTION INPUT (OVP+)

OVP+ On Threshold Voltage V

OVP+ Hysteresis 200 mV

OVP+ Input Leakage Current (V

HIGH-SIDE LED SHORT COMPARATOR

Off Threshold V

On Threshold V

Error Reject Blankout f

LOW-SIDE LED SHORT COMPARATOR

Off Threshold 0.27 0.30 0.33 V

Error Reject Blankout 5µs

HICCUP TIMER

Hiccup Time f

GATE-DRIVER OUTPUT (NDRV)

NDRV Peak Pullup Current VCC = 7V 3 A

NDRV Peak Pulldown Current VCC = 7V 3 A

p-Channel MOSFET R

n-Channel MOSFET R

DIMOUT

DIMOUT Peak Pullup Current (V

DIMOUT Peak Pulldown Current (V

p-Channel MOSFET R

n-Channel MOSFET R

PWMDIM to DIMOUT
Propagation Delay

FAULT FLAG (FLT)

FLT Pulldown Current V
FLT Leakage Current V

Thermal Warning On Threshold +140 °C

Thermal Warning Threshold
Hysteresis

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

DSON

DSON

DSON

DSON

OVP_ON

- VLV) = 1.235V -1 +1 µA

OVP

- V

CLV

LV

- V

CLV

LV

= 500kHz 256 µs

OSC

= 500kHz 8.2 ms

OSC

(VCC - V

V

(V

(V

NDRV

= 0.1V 0.9 1.7 Ω

NDRV

- VLV) = 5V 25 50 mA

CLV

- VLV) = 5V 25 50 mA

CLV

- V

CLV

DIMOUT

- VLV) = 0.1V 25 Ω

DIMOUT

= 0.2V 2 5 10 mA

FLT

= 1.0V I1I µA

FLT

) = 0.1V 1.2 1.9 Ω

) = 0.1V 31 Ω

1.375 1.435 1.495 V

4.0 4.3 4.6 V

4.1 4.4 4.7 V

200 ns

20 °C

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

_______________________________________________________________________________________ 5

V

(V)

Typical Operating Characteristics

(VIN= VHV= 12V, V

UVEN

= 5V, VLV= V

PWMDIM

= SGND, C

VCC

= 4.7µF, C

LCV

= 100nF, C

REF

= 100nF, R

SENSE+

= 0.1Ω,

R

RT

= 10kΩ, TA= +25°C, unless otherwise noted.)

V

vs. TEMPERATURE

REF

3.74

3.72

3.70

(V)

3.68

REF

V

3.66

3.64

3.62

3.60

-40 125
TEMPERATURE (°C)

VIN = 12V

11095-25 -10 5 35 50 6520 80

SUPPLY CURRENT

vs. SUPPLY VOLTAGE

20

18

16

14

12

10

8

6

SUPPLY CURRENT (mA)

4

2

0

428

SUPPLY VOLTAGE (V)

PWMDIM = 0

242016128

3.80

3.75

MAX16834 toc01

3.70

(V)

3.65

REF

V

3.60

3.55

3.50

10

9

MAX16834 toc04

8

7

6

5

4

3

SUPPLY CURRENT (mA)

2

1

0

V

REF

428

vs. TEMPERATURE

-40 125

vs. SUPPLY VOLTAGE

SUPPLY VOLTAGE (V)

SUPPLY CURRENT

VIN = 12V
PWMDIM = 0

TEMPERATURE (°C)

242016128

1109565 80-10 5 20 35 50-25

3.7020

3.7015

MAX16834 toc02

3.7010

3.7005

3.7000

REF

3.6995

3.6990

3.6985

3.6980

100

MAX16834 toc05

10

RT (kΩ)

1

V

vs. I

REF

010

I

(mA)

REF

RT vs. SWITCHING FREQUENCY

VIN = 12V

SWITCHING FREQUENCY (kHz)

REF

VIN = 12V

MAX16834 toc03

981 2 3 5 64 7

MAX16834 toc06

1000100

SWITCHING FREQUENCY

605
604
603
602
601
600
599
598
597
596
595
594

SWITCHING FREQUENCY (kHz)

593
592
591

590

-40 125

TEMPERATURE (°C)

VIN = 12V

1109565 80-10 5 20 35 50-25

MAX16834 toc07

7.16

7.14

7.12

7.10

7.08

7.06

(V)

7.04

CC

7.02

V

7.00

6.98

6.96

6.94

6.92

6.90
0100

vs. TEMPERATURE

VCC vs. I

ICC (mA)

CC

VIN = 12V

908060 7020 30 40 5010

MAX16834 toc08

7.2
TA = +125°C

7.1

(V)

7.0

TA = +25°C

CC

V

6.9

6.8

0100

VCC vs. I

TA = +100°C

TA = -40°C

ICC (mA)

CC

VIN = 12V

MAX16834 toc09

908070605040302010

Pin Description

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

6 _______________________________________________________________________________________

Typical Operating Characteristics (continued)

(VIN= VHV= 12V, V

UVEN

= 5V, VLV= V

PWMDIM

= SGND, C

VCC

= 4.7µF, C

LCV

= 100nF, C

REF

= 100nF, R

SENSE+

= 0.1Ω,

R

RT

= 10kΩ, TA= +25°C, unless otherwise noted.)

VCC vs. V

IN

MAX16834 toc10

VIN (V)

V

CC

(V)

26

22

1814

10

7.02

7.04

7.06

7.08

7.10

7.12

7.14

7.16

7.18

7.20

7.00
6

TA = +125°C

TA = +25°C

TA = -40°C

NDRV RISE/FALL TIME

vs. CAPACITANCE

MAX16834 toc11

CAPACITANCE (nF)

NDRV RISE TIME (ns)

987654321

10

20

30

40

50

0

010

VIN = 12V

RISE TIME

FALL TIME

V

CLV

vs. V

HV

MAX16834 toc13

VHV (V)

V

CLV

(V)

26

22

18

14

10

5.01

5.02

5.03

5.04

5.05

5.06

5.07

5.08

5.09

5.10

5.00
6

VIN = 12V

V

CLV

vs. I

CLV

MAX16834 toc12

I

CLV

(mA)

V

CLV

(V)

4.54.03.0 3.51.0 1.5 2.0 2.50.5

0.50

1.00

1.50

2.00

2.50

3.00

3.50

4.00

4.50

5.00

5.50

0

05.0

VIN = 12V

PIN NAME FUNCTION

LED-String Overvoltage Protection Input. Connect a resistive voltage-divider between the positive

1 OVP+

2 SGND Signal Ground

3 COMP

4 REF 3.7V Reference Output Voltage. Bypass REF to SGND with a 0.1µF to 0.22µF ceramic capacitor.

5 REFI

6SC

output, OVP+, and LV to set the overvoltage threshold. OVP+ has a 1.435V threshold voltage with a
200mV hysteresis.

Error-Amplifier Output. Connect an RC network from COMP to SGND for stable operation. See the
Feedback Compensation section.

Current Reference Input. V

provides a reference voltage for the current-sense amplifier to set the

REFI

LED current.

Current-Mode Slope Compensation Setting. Connect to an appropriate external capacitor from SC to
SGND to generate a ramp signal for stable operation.

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

_______________________________________________________________________________________ 7

Pin Description (continued)

Detailed Description

The MAX16834 is a current-mode, high-brightness LED
(HB LED) driver designed to control a single-string LED
current regulator with two external n-channel MOSFETs.

The MAX16834 integrates all the building blocks nec­essary to implement a fixed-frequency HB LED driver
with wide-range dimming control. The MAX16834
allows implementation of different converter topologies
such as SEPIC, boost, buck-boost, or high-side buck
current regulator.

The MAX16834 features a constant-frequency, peak-cur­rent-mode control with programmable slope compensa­tion to control the duty cycle of the PWM controller. A
dimming driver offers a wide-range dimming control for
the external n-channel MOSFET in series with the LED
string. In addition to PWM dimming, the MAX16834
allows for analog dimming of LED current.

The MAX16834 switching frequency (100kHz to 1MHz)
is adjustable using a single resistor from RT/SYNC. The

MAX16834 disables the internal oscillator and synchro­nizes if an external clock is applied to RT/SYNC. The
switching MOSFET driver sinks and sources up to 3A,
making it suitable for high-power MOSFETs driving in
HB LED applications, and the dimming control allows
for wide PWM dimming at frequencies up to 20kHz.

The MAX16834 is suitable for boost and buck-boost
LED drivers (Figures 2 and 3).

The MAX16834 operates over a wide 4.75V to 28V sup­ply range. Additional features include external
enable/disable input, an on-chip oscillator, fault indica­tor output (FLT) for LED open/short or overtemperature
conditions, and an overvoltage protection circuit for
true differential overvoltage protection (Figure 1).

The MAX16834 is also suitable for DC-DC converter
applications such as boost or buck-boost (Figures 6
and 7). Other applications include boost LED drivers
with automotive load dump protection (Figure 4) and
high-side buck LED drivers (Figure 5).

PIN NAME FUNCTION

7 FLT Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (

Resistor-Programmable Switching Frequency Setting/Sync Control Input. Connect a resistor from

8 RT/SYNC

9 UVEN

10 PWMDIM PWM Dimming Input. Connect to an external PWM signal for dimming operation.

11 CS

12 PGND Power Ground

13 NDRV External n-Channel Gate-Driver Output

14 V

15 IN Positive Power-Supply Input. Bypass to PGND with at least a 0.1µF ceramic capacitor.

16 HV High-Side Positive Supply Input Referred to LV. HV provides power to high-side linear regulator CLV.

17 CLV

18 DIMOUT External Dimming MOSFET Gate Driver. DIMOUT is capable of sinking/sourcing 50mA.

19 LV

20 SENSE+ LED Current-Sense Positive Input

—EP

CC

RT/SYNC to SGND to set the switching frequency. Drive RT/SYNC to synchronize the switching
frequency with an external clock.

Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable UVLO
threshold input with an enable feature. Connect UVEN to VIN through a resistive voltage-divider to
program the UVLO threshold. Observe the absolute maximum value for this pin.

Current-Sense Amplifier Positive Input. Connect a resistor from CS to PGND to set the inductor peak
current limit.

7V Low-Dropout Voltage Regulator. Bypass to PGND with at least a 1µF low-ESR ceramic capacitor.

provides power to the n-channel gate driver (NDRV).

V

CC

5V High-Side Regulator Output. CLV provides power to the dimming MOSFET driver. Connect a 0.1µF
to 1µF ceramic capacitor from CLV to LV for stable operation.

High-Side Reference Voltage Input. Connect to SGND for boost configuration. Connect to IN for buck­boost configuration.

Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power
dissipation. Do not use as the main IC ground connection. EP must be connected to SGND.

FLT

) section.

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

8 _______________________________________________________________________________________

Figure 1. Internal Block Diagram

IN

SGND

UVEN

RT/SYNC

SC

CS

REFI

SENSE+

1kΩ

V

LV

COMP

HV

V

LV

PWMDIM

OVP+

BG

V

REFERENCE

V

BG

OSC

5kΩ

BLANK

NDRVB

LPF

V

LV

HIGH-SIDE

REGULATOR

BG

V

AV = 9.9

5V

V

HV

REF

V

FLTA

7V

LDO

UVLO

RAMP

GENERATOR

CURRENT-LIMIT
COMPARATOR

0.3V

LED CURRENT­SENSE AMPLIFIERS

LV REFERENCE
SWITCH

BG

FLTB

ERROR

AMPLIFIER

g

REFHI

AND

TO
INTERNAL
CIRCUITRY

m

0.6V

V

LV

TEMPERATURE

PWM
COMP

4.3V

0.3V

SENSE+

SENSE

V

OR

PWMDIM

V

IN

REF

OT

OT

S

Q

R

AND

V

LV

128 TOSC

ERROR

REJECT

DELAY

5µs ERROR

REJECT

DELAY

AND

REFHI

FLTAFLTB

NDRVB

4096 TOSC

HICCUP

TIMER

AND

FLTB

REF

V

CC

NDRV

PGND

FLT

CLV

DIMOUT

V

BG

V

LV

V

LV

MAX16834

MAX16834

Undervoltage Lockout/Enable

The MAX16834 features an adjustable UVLO using the
enable input (UVEN). Connect UVEN to VINthrough a
resistive divider to set the UVLO threshold. The
MAX16834 is enabled when the V

UVEN

exceeds the

1.435V (typ) threshold. See the

Setting the UVLO

Threshold

section for more information.

UVEN also functions as an enable/disable input to the
device. Drive UVEN low to disable the output and high
to enable the output.

Reference Voltage (REF)

The MAX16834 features a 3.7V reference output, REF.
REF provides power to most of the internal circuit blocks
except for the output drivers and is capable of sourcing
1mA to external circuits. Connect a 0.1µF to 0.22µF
ceramic capacitor from REF to SGND. Connect REF to
REFI through a resistive divider to set the LED current.

Reference Input (REFI)

The output current is proportional to the voltage at
REFI. Applying an external DC voltage at REFI or using
a potentiometer from REF to SGND allows analog dim­ming of the output current.

High-Side Reference Voltage Input (LV)

LV is a reference input. Connect LV to SGND for boost
and SEPIC topologies. Connect LV to IN for buck-boost
and high-side buck topologies.

Dimming Driver Regulator

Input Voltage (HV)

The voltage at HV provides the input voltage for the
dimming driver regulator. For boost or SEPIC topology,
connect HV either to IN or to VCC. For buck-boost, con­nect HV to a voltage higher than IN. The voltage at HV
must not exceed 28V with respect to PGND. For the
high-side buck, connect HV to IN.

Dimming MOSFET Driver (DIMOUT)

The MAX16834 requires an external n-channel MOSFET
for PWM dimming. Connect the gate of the MOSFET to
the output of the dimming driver, DIMOUT, for normal
operation. The dimming driver is capable of sinking or
sourcing up to 50mA of current.

n-Channel MOSFET Switch Driver (NDRV)

The MAX16834 drives an external n-channel switching
MOSFET. NDRV swings between VCCand PGND.
NDRV is capable of sinking/sourcing 3A of peak current,
allowing the MAX16834 to switch MOSFETs in high­power applications. The average current demanded
from the supply to drive the external MOSFET depends
on the total gate charge (QG) and the operating

frequency of the converter, fSW. Use the following equa­tion to calculate the driver supply current I

NDRV

required for the switching MOSFET:

I

NDRV

= QGx f

SW

Pulse Dimming Inputs (PWMDIM)

The MAX16834 offers a dimming input (PWMDIM) for
pulse-width modulating the output current. PWM dim­ming can be achieved by driving PWMDIM with a pul­sating voltage source. When the voltage at PWMDIM is
greater than 1.435V, the PWM dimming MOSFET turns
on and when the voltage on PWMDIM is below 1.235V,
the PWM dimming MOSFET turns off.

High-Side Linear Regulator (V

CLV

)

The MAX16834’s 5V high-side regulator (CLV) powers
up the dimming MOSFET driver. V

CLV

is measured with
respect to LV and sources up to 2mA of current.
Bypass CLV to LV with a 0.1µF to 1µF low-ESR ceramic
capacitor. The maximum voltage on CLV with respect
to PGND must not exceed 28V. This limits the input volt­age for buck-boost topology.

Low-Side Linear Regulator (VCC)

The MAX16834’s 7V low-side linear regulator (VCC) pow­ers up the switching MOSFET driver with sourcing capa­bility of up to 50mA. Use at least a 1µF low-ESR ceramic
capacitor from VCCto PGND for stable operation.

LED Current-Sense Input (SENSE+)

The differential voltage from SENSE+ to LV is fed to an
internal current-sense amplifier. This amplified signal is
then connected to the negative input of the transcon­ductance error amplifier. The voltage gain factor of this
amplifier is 9.9 (typ).

Internal Transconductance Error Amplifier

The MAX16834 has a built-in transconductance amplifi­er used to amplify the error signal inside the feedback
loop. The amplified current-sense signal is connected
to the negative input of the gmamplifier with the current
reference connected to REFI. The output of the op amp
is controlled by the input at PWMDIM. When the signal
at PWMDIM is high, the output of the op amp connects
to COMP; when the signal at PWMDIM is low, the out­put of the op amp disconnects from COMP to preserve
the charge on the compensation capacitor. When the
voltage at PWMDIM goes high, the voltage on the com­pensation capacitor forces the converter into a steady
state. COMP is connected to the negative input of the
PWM comparator with CMOS inputs, which draw very
little current from the compensation capacitor at COMP
and thus prevent discharge of the compensation
capacitor when the PWMDIM input is low.

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

_______________________________________________________________________________________ 9

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

10 ______________________________________________________________________________________

Internal Oscillator

The internal oscillator of the MAX16834 is programma­ble from 100kHz to 1MHz using a single resistor at
RT/SYNC. Use the following formula to calculate the
switching frequency:

where RT is the resistor from RT/SYNC to SGND.

The MAX16834 synchronizes to an external clock signal
at RT/SYNC. The application of an external clock dis­ables the internal oscillator allowing the MAX16834 to
use the external clock for switching operation. The
internal oscillator is enabled if the external clock is
absent for more than 50µs. The synchronizing pulse
minimum width for proper synchronization is 200ns.

Switching MOSFET

Current-Sense Input (CS)

CS is part of the current-mode control loop. The switch­ing control uses the voltage on CS, set by R

CS

, to termi­nate the on pulse width of the switching cycle, thus
achieving peak current-mode control. Internal leading­edge blanking is provided to prevent premature turn-off
of the switching MOSFET in each switching cycle.

Slope Compensation (SC)

The MAX16834 uses an internal-ramp generator for
slope compensation. The ramp signal also resets at the
beginning of each cycle and slews at the rate pro­grammed by the external capacitor connected at SC.
The current source charging the capacitor is 100µA.

Overvoltage Protection (OVP+)

OVP+ sets the overvoltage threshold limit across the
LEDs. Use a resistive divider between output OVP+
and LV to set the overvoltage threshold limit. An internal
overvoltage protection comparator senses the differen­tial voltage across OVP+ and LV. If the differential volt­age is greater than 1.435V, NDRV is disabled and FLT
asserts. When the differential voltage drops by 200mV,
NDRV is enabled and FLT deasserts. The PWM dim-
ming MOSFET is still controlled by the PWMDIM input.

Fault Indicator (

FLT

)

The MAX16834 features an active-low, open-drain fault
indicator (FLT). FLT asserts when one of the following
occurs:

1) Overvoltage across the LED string

2) Short-circuit condition across the LED string, or

3) Overtemperature condition

When the output voltage drops below the overvoltage
set point minus the hysteresis, FLT deasserts. Similarly
during the short-circuit period, the fault signal
deasserts when the dimming MOSFET is on, which
happens every hiccup cycle during short circuit. During
overtemperature fault, the FLT signal is the inverse of
the PWM input.

Applications Information

Setting the UVLO Threshold

The UVLO threshold is set by resistors R1 and R2 (see
Figure 2). The MAX16834 turns on when the voltage
across R2 exceeds 1.435V, the UVLO threshold. Use
the following equation to set the desired UVLO thresh­old:

In a typical application, use a 10kΩ resistor for R2 and
then calculate R1 based on the desired UVLO threshold.

Setting the Overvoltage Threshold

The overvoltage threshold is set by resistors R4 and R9
(see Figure 2). The overvoltage circuit in the MAX16834
is activated when the voltage on OVP+ with respect to
LV exceeds 1.435V. Use the following equation to set
the desired overvoltage threshold:

Programming the LED Current

The LED current is programmed using the voltage on
REFI and the LED current-sense resistor R10 (see
Figure 2). The current is given by:

where V

REF

is 3.7V and the resistors R5, R6, and R10
are in ohms. The regulation voltage on the LED current­sense resistor must not exceed 0.3V to prevent activa­tion of the LED short-circuit protection circuit.

5000k

f (kHz)

OSC

=×

RT(k )

Ω

Ω

(kHz )

VVRRR

=+1 435 1 2 2.( )/

UVEN

VVRRR

=+1 435 4 9 9.( )/

OV

VR

×

I

=

LED

REF

RRR

×+×

10 6 5 9 9().

5

A

()

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 11

Boost Configuration

In the boost converter (Figure 2), the average inductor
current varies with the line voltage. The maximum aver­age current occurs at the lowest line voltage. For the
boost converter, the average inductor current is equal
to the input current.

Calculate maximum duty cycle using the below equation.

where V

LED

is the forward voltage of the LED string in
volts, VDis the forward drop of the rectifier diode D1 in
volts (approximately 0.6V), V

INMIN

is the minimum input

supply voltage in volts, and V

FET

is the average drain to
source voltage of the MOSFET Q1 in volts when it is on.
Use an approximate value of 0.2V initially to calculate
D

MAX

. A more accurate value of the maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.

Use the following equations to calculate the maximum
average inductor current IL

AVG

, peak-to-peak inductor
current ripple ∆IL, and the peak inductor current ILPin
amperes:

Figure 2. Boost LED Driver

V

IN

C1

L1

R1

LV

IN

UVEN

C2

R3

C5

R2

C4

R5

HV

MAX16834

SC

RT/SYNC

V

CC

REF

R6

REFI

SGND

FLT

NDRV

CS

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

Q1

ON

OFF

C7

R7

D1

C3

R4

Q2

R8

C6

R9

LED+

LEDs

LED-

R10

VVV

+−

LED D

=

VVV

+−

LEDDFET

D

MAX

INMIN

I

IL

AVG

LED

=

D

−1

MAX

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

12 ______________________________________________________________________________________

Allowing the peak-to-peak inductor ripple (∆IL) to be
±30% of the average inductor current:

and

The inductance value (L) of the inductor L1 in henries
(H) is calculated as:

where fSWis the switching frequency in hertz, V

INMIN

and V

FET

are in volts, and ∆IL is in amperes.

Choose an inductor that has a minimum inductance
greater than the calculated value. The current rating of
the inductor should be higher than ILPat the operating
temperature.

Buck-Boost Configuration

In the buck-boost LED driver (Figure 3), the average
inductor current is equal to the input current plus the
LED current.

Calculate maximum duty cycle using the below equation:

where V

LED

is the forward voltage of the LED string in
volts, VDis the forward drop of the rectifier diode D1
(approximately 0.6V) in volts, V

INMIN

is the minimum

input supply voltage in volts, and V

FET

is the average
drain to source voltage of the MOSFET Q1 in volts when
it is on. Use an approximate value of 0.2V initially to cal­culate D

MAX

. A more accurate value of maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.

Use the below equations to calculate the maximum
average inductor current IL

AVG

, peak-to-peak inductor
current ripple ∆IL, and the peak inductor current ILPin
amperes:

Allowing the peak-to-peak inductor ripple ∆I

L

to be

±30% of the average inductor current:

The inductance value (L) of the inductor L1 in henries is
calculated as:

where f

SW

is the switching frequency in hertz, V

INMIN

and V

FET

are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than
the calculated value.

Peak Current-Sense Resistor (R8)

The value of the switch current-sense resistor R8 for the
boost and buck-boost configurations is calculated as
follows:

where 0.25V is the minimum peak current-sense thresh­old, ILPis the peak inductor current in amperes, and
the factor 1.25 provides a 25% margin to account for
tolerances. The worst cycle-by-cycle current limiter trig­gers at 350mV (max). The I

SAT

of the inductor should

be higher than 0.35V/R8.

Output Capacitor

The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most applications, the output
ESR and ESL effects can be dramatically reduced by
using low-ESR ceramic capacitors. To reduce the ESL
and ESR effects, connect multiple ceramic capacitors
in parallel to achieve the required bulk capacitance. To
minimize audible noise generated by the ceramic
capacitors during PWM dimming, it may be necessary
to minimize the number of ceramic capacitors on the
output. In these cases an additional electrolytic or tan­talum capacitor provides most of the bulk capacitance.

Boost and buck-boost configurations: The calcula­tion of the output capacitance is the same for both
boost and buck-boost configurations. The output ripple
is caused by the ESR and the bulk capacitance of the
output capacitor if the ESL effect is considered negligi­ble. For simplicity, assume that the contributions from

∆∆IIL

∆IIL

=××03 2.

LAVG

=××

LAVG

IL IL

=+

PAVG

03 2

.

I

L

2

IL IL

=+

PAVG

I

∆

L

2

VVD

()

L

=

−×

INMIN FET MAX

fI

×

∆

SW L

VV

+

D

=

MAX

VVV V

LED D INMIN FET

IL

AVG

LED D

++ −

I

LED

=

D

−1

MAX

VVD

()

L

=

−×

INMIN FET MAX

fI

×

∆

SW L

025

R

8

.

=

IL

(.)

P

×

125

Ω

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 13

ESR and the bulk capacitance are equal, allowing 50%
of the ripple for the bulk capacitance. The capacitance
is given by:

where I

LED

is in amperes, C

OUT

is in farads, fSWis in

hertz, and ∆V

OUTRIPPLE

is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capaci­tor. Based on this, the ESR of the output capacitor is
given by:

where IL

P

is the peak inductor current in amperes.

Use the below equation to calculate the RMS current
rating of the output capacitor:

Input Capacitor

The input filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and the bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter.

For the boost configuration, the input current is the
same as the inductor current. For buck-boost

Figure 3. Buck-Boost LED Driver (V

LED+

< 28V)

V

IN

C1

R1

LV

IN

C2

R3

C5

R2

C4

R5

UVEN

SC

MAX16834

RT/SYNC

V

CC

REF

R6

REFI

FLT

SGND

NDRV

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

HV

Q1

CS

ON

OFF

C7

R7

L1

D1

C3

R8

C6

R4

Q2

R9

R10

LED+

LED-

V

LEDs

IN

ID

××

2

LED MAX

≥

Vf

∆

OUTRIPPLE SW

×

C

OUT

V

∆Ω()

<

OUTRIPPLE

IL

×

()2

P

ESR

COUT

(IL (1 D )) D

I

COUT(RMS)

=

AVG MAX

IL

+-(

VVG

A

××

××DD

MAX MAX

2

MAX

2

)( )

1-

MAX16834

configuration, the input current is the inductor current
minus the LED current. But for both configurations, the
ripple current that the input filter capacitor has to sup­ply is the same as the inductor ripple current with the
condition that the output filter capacitor should be con­nected to ground for buck-boost configuration. This
reduces the size of the input capacitor, as the inductor
current is continuous with maximum ±30% ripple.
Neglecting the effect of LED current ripple, the calcula­tion of the input capacitor for boost as well as buck­boost configurations is the same.

Neglecting the effect of the ESL, the ESR, and the bulk
capacitance at the input contributes to the input voltage
ripple. For simplicity, assume that the contribution from
the ESR and the bulk capacitance is equal. This allows
50% of the ripple for the bulk capacitance. The capaci­tance is given by:

where ∆ILis in amperes, CINis in farads, fSWis in hertz,
and ∆VINis in volts. The remaining 50% of allowable
ripple is for the ESR of the output capacitor. Based on
this, the ESR of the input capacitor is given by:

where ∆IL is in amperes, ESR

CIN

is in ohms, and ∆V

IN

is in volts.

Use the below equation to calculate the RMS current
rating of the input capacitor:

Slope Compensation

Slope compensation should be added to converters
with peak current-mode control operating in continuous
conduction mode with more than 50% duty cycle to
avoid current loop instability and subharmonic oscilla­tions. The minimum amount of slope added to the peak
inductor current to stabilize the current control loop is
half of the falling slope of the inductor.

In the MAX16834, the slope compensating ramp is
added to the current-sense signal before it is fed to the
PWM comparator. Connect a capacitor (C2 in the appli­cation circuit) from SC to ground for slope compensa­tion. This capacitor is charged with a 100µA current

source and discharged at the beginning of each switch­ing cycle to generate the slope compensation ramp.

The value of the slope compensation capacitor C2 is
calculated as shown below:

Boost configuration:

where C2 is in farads, L is the inductance of the induc­tor L1 in henries, 100µA is the pullup current from SC,
V

LED

and V

INMIN

are in volts, and R8 is the switch cur-

rent-sense resistor in ohms.

Buck-boost configuration:

where C2 is in farads, L is the inductance of the induc­tor L1 in henries, 100µA is the pullup current from SC,
V

LED

is in volts, and R8 is the switch current-sense

resistor in ohms.

Selection of Power Semiconductors

Switching MOSFET

The switching MOSFET (Q1) should have a voltage rat­ing sufficient to withstand the maximum output voltage
together with the diode drop of the rectifier diode D1
and any possible overshoot due to ringing caused by
parasitic inductances and capacitances. Use a
MOSFET with a drain-to-source voltage rating higher
than that calculated by the following equations:

Boost configuration:

where VDSis the drain-to-source voltage in volts and
VDis the forward drop of the rectifier diode D1. The fac­tor of 1.2 provides a 20% safety margin.

Buck-boost configuration:

where VDSis the drain-to-source voltage in volts and
VDis the forward drop of the rectifier diode D1. The fac­tor of 1.2 provides a 20% safety margin.

The continuous drain current rating of the selected
MOSFET, when the case temperature is at +70°C,
should be greater than the value calculated by the fol-

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

14 ______________________________________________________________________________________

I

∆

C

≥

IN

ESR

CIN

I

CIN(RMS)

L

Vf

××

∆4

IN SW

V

∆

IN

<

I

×

∆ 2

L

∆I

L

=

3

2

-

L

×× ×

C2

C2

3 100 10

=

(V - V ) R8

LED INMIN

L

×× ×

3 100 10

=

(V ) R8

××

LED

6

××

2

-

6

2

VVV

=+

()

DS LED D

×12.

VVV V

=+ +

()

DS LED INMAX D

×12.

lowing equation. The MOSFET must be mounted on a
board as per manufacturer specifications to dissipate
the heat.

The RMS current rating of the switching MOSFET Q1 is
calculated as follows for boost and buck-boost configu­rations:

where ID

RMS

is the MOSFET Q1’s drain RMS current in

amperes.

The MOSFET Q1 will dissipate power due to both
switching losses as well as conduction losses. The con­duction losses in the MOSFET is calculated as follows:

where R

DSON

is the on-resistance of Q1 in ohms with

an assumed junction temperature of +100°C, P

COND

is

in watts, and IL

AVG

is in amperes.

Use the following equations to calculate the switching
losses in the MOSFET:

Boost configuration:

Buck-boost configuration:

where IGONand IG

OFF

are the gate currents of the
MOSFET Q1 in amperes when it is turned on and
turned off, respectively, V

LED

and V

INMAX

are in volts,

IL

AVG

is in amperes, fSWis in hertz, and CGDis the

gate-to-drain MOSFET capacitance in farads.

Choose a MOSFET that has a higher power rating than
that calculated by the following equation when the
MOSFET case temperature is at +70°C:

Rectifier Diode

Use a Schottky diode as the rectifier (D1) for fast
switching and to reduce power dissipation. The select­ed Schottky diode must have a voltage rating 20%
above the maximum converter output voltage. The max­imum converter output voltage is V

LED

in boost configu-

ration and V

LED

+ V

INMAX

in buck-boost configuration.

The current rating of the diode should be greater than
I

D

in the following equation:

Dimming MOSFET

Select a dimming MOSFET (Q2) with continuous current
rating at +70°C, higher than the LED current by 30%.
The drain-to-source voltage rating of the dimming
MOSFET must be higher than V

LED

by 20%.

Feedback Compensation

The LED current control loop comprising of the switch­ing converter, the LED current amplifier, and the error
amplifier should be compensated for stable control of
the LED current. The switching converter small-signal
transfer function has a right half-plane (RHP) zero for
both boost and buck-boost configurations as the induc­tor current is in continuous conduction mode. The RHP
zero adds a 20dB/decade gain together with a 90°
phase lag, which is difficult to compensate. The easiest
way to avoid this zero is to roll off the loop gain to 0dB
at a frequency less than one-fifth of the RHP zero fre­quency with a -20dB/decade slope.

The worst-case RHP zero frequency (f

ZRHP

) is calculat-

ed as follows:

Boost configuration:

Buck-boost configuration:

where f

ZRHP

is in hertz, V

LED

is in volts, L is the induc-

tance value of L1 in henries (H), and I

LED

is in amperes.

The switching converter small-signal transfer function
also has an output pole for both boost and buck-boost
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as:

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 15

ID IL D

⎛

=

()

⎜

RMS AVG MAX

⎝

×

⎞

×213.

⎟
⎠

2

××

PILDR

=

COND AVG MAX DSON

()

⎛

IL V C f

P

SW

AVG LED GD SW

=

⎜
⎝

⎛

11

×+

⎜

IG IG

⎝

ON OF

⎛

IL V V C f

×+ ××

()

P

SW

AVG LED INMAX GD SW

=

⎜
⎝

⎛

1

×

⎜

GGIG

I

⎝

ON OFF

1

+

2

×××

2

⎞
⎟

⎠

⎞
⎟

⎠

FF

2

2

⎞
⎟

⎠

⎞
⎟

⎠

PWP WPW

() () ()=+

TOT COND SW

IIL D

=× ×().115-

DAVG MAX

f

ZRHP

V

=

()12-

LED

××

2LI

π

MAX

LED

D

×

D

×

f

=

ZRHP

2LI

V

π

()12-

LED

×× ×

MAX

LED

D

MAX

MAX16834

Boost configuration:

Buck-boost configuration:

where R

LED

is the dynamic impedance (rate of change
of voltage with current) of the LED string at the operat­ing current, R10 is the LED current-sense resistor in
ohms, V

LED

is in volts, and I

LED

is in amperes.

The output pole frequency for both boost and buck­boost configurations is calculated as follows:

where f

P2

is in hertz, C

OUT

is the output filter capaci-

tance in farads, R

OUT

is the effective output impedance

in ohms calculated above.

Compensation components R7 and C7 perform two
functions. C7 introduces a low-frequency pole that
introduces a -20dB/decade slope into the loop gain. R7
flattens the gain of the error amplifier for frequencies
above the zero formed by R7 and C7. For compensa­tion, this zero is placed at the output pole frequency f

P2

such that it provides a -20dB/decade slope for frequen­cies above f

P2

for the complete loop gain.

The value of R7 needed to fix the total loop gain at f

P2

such that the total loop gain crosses 0dB at

-20dB/decade at one-fifth of the RHP zero can be cal­culated as follows:

where R7 is the compensation resistor in ohms, f

ZRHP

and fP2are in hertz, R8 is the switch current-sense
resistor in ohms, R10 is the LED current-sense resistor
in ohms, factor 9.9 is the gain of the LED current ampli­fier, and GM

COMP

is the transconductance of the error

amplifier in Siemens.

The value of C7 can be calculated as:

where C7 is in farads, f

P2

is in hertz, and R7 is in ohms.

To minimize switching frequency noise, an additional
capacitor can be added in parallel with the series com­bination of R7 and C7. The pole from this capacitor and
R7 must be a decade higher than the loop crossover
frequency.

Short-Circuit Protection

Boost Configuration

In the boost configuration (Figure 2), if the LED string is
shorted then the excess current flowing in the LED cur­rent-sense resistor will cause NDRV to stop switching.
The input voltage will appear on the output capacitor,
and this causes very high peak currents to flow in the
LED current-sense resistor R10 because the dimming
MOSFET (Q2) is on. Once the voltage across the LED
current-sense resistor exceeds 300mV for more than
5µs, then the dimming MOSFET Q2 turns off and stays
off for 4096 switching clock cycles. At the same time,
NDRV is also off. The MAX16834 goes into the hiccup
mode and recovers from hiccup once the short has
been removed. The power dissipation in the dimming
MOSFET (Q2) is minimized during a short across the
LED string. During the same period, FLT only goes high
when the dimming MOSFET is on.

Buck-Boost Configuration

In the case of the buck-boost configuration (Figure 3),
once an LED string short occurs then the behavior is
different. A short across the LED string causes a high
current spike due to the external capacitors at the out­put. The regulation loop will cause NDRV to stop
switching. This causes the voltage on HV to drop if its
voltage is derived from LED+. The voltage on CLV will
drop, and this drop is detected after 128 clock cycles.
The dimming MOSFET and the switching MOSFET will
stop switching. It stays off for 4096 clock cycles, and
the cycle repeats itself. The short across the LED string
will cause the MAX16834 to go into a hiccup mode. At
the same time the FLT signal asserts itself for 4096
clock cycles every hiccup cycle. In the case where the
HV voltage is derived from a source different than
LED+, then the LED current will stay in regulation even
during a short across the LED string. In this case, FLT
does not assert itself during the short.

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

16 ______________________________________________________________________________________

R

OUT

()

=

RRIV

()

LED LED LED

10

LED LED

+×+

10

RRV

+×

R

=

OUT

RRID V

()

LED LED MAX

()

LED LED

+×× +

10

10

LLED

RRV

+×

1

××2π

CR

OUT OUT

=

f

P

2

fR

R

7

=

fDR GM

51 1099

××− × ××().

2

PMAX COMP

ZRHP

8

×

C

7

=

1

Rf

××2π

7

P

2

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 17

Figure 4. Boost LED Driver with Automotive Load Dump Protection

V

IN

C1

L1

R1

D2

24V

R2

Q3

C8

C2

R3

C5

C4

R5

LV

IN

UVEN

HV

MAX16834

SC

RT/SYNC

V

CC

REF

R6

REFI

SGND

FLT

NDRV

CS

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

Q1

ON

OFF

C7

R7

D1

C3

R4

Q2

R8

C6

R9

R10

LED+

LEDs

LED-

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

18 ______________________________________________________________________________________

Figure 5. High-Side Buck LED Driver

V

IN

LED+

C1

R1

R2

D1

LV

IN

UVEN

C2

SC

R3

C5

C4

R6

R5

RT/SYNC

V

CC

REF

REFI

FLT

SGND

MAX16834

NDRV

CS

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

HV

Q1

ON

OFF

C7

C6

R7

V

LV

C3

L1

V

LV

R4

Q2

R8

R9

R10

LED-

V

LEDs

LV

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 19

Figure 6. Boost DC-DC Converter

V

IN

C1

L1

V

R1

LV

FLT

D1

OUT

C2

R3

C5

R2

C4

R5

IN

UVEN

HV

MAX16834

SC

RT/SYNC

V

CC

REF

R6

REFI

SGND

NDRV

CS

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

V

REF

Q1

C3 R4

R10

C6

R8

R7

OPTIONAL

R9

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

20 ______________________________________________________________________________________

Figure 7. Buck-Boost DC-DC Converter

V

IN

C1

L1

R1

R2

LV

IN

C2

R3

C5

C4

R5

UVEN

SC

MAX16834

RT/SYNC

V

CC

REF

R6

REFI

FLT

SGND

HV

NDRV

CS

PWMDIM

DIMOUT

SENSE+

OVP+

CLV

COMP

PGND

Q1

V

REF

N.C.

C6

R7

D1

C3 R4

R9

R8

R11

R10

V

OUT

V

IN

MAX16834

High-Power LED Driver with Integrated High-Side LED

Current Sense and PWM Dimming MOSFET Driver

______________________________________________________________________________________ 21

Layout Recommendations

Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain cur­rent often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dv/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOS­FET power dissipation or shield it. Keep all PCB traces
carrying switching currents as short as possible to mini­mize current loops. Use ground planes for best results.

Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good
PCB layout:

1) Use a large contiguous copper plane under the

MAX16834 package. Ensure that all heat-dissipat­ing components have adequate cooling.

2) Isolate the power components and high-current

path from the sensitive analog circuitry.

3) Keep the high-current paths short, especially at the

ground terminals. This practice is essential for sta­ble, jitter-free operation. Keep switching loops short
such that:

a) The anode of D1 must be connected very close

to the drain of the MOSFET Q1.

b) The cathode of D1 must be connected very

close to C

OUT

.

c) C

OUT

and the current-sense resistor R8 must

be connected directly to the ground plane.

4) Connect PGND and SGND to a star-point configura­tion.

5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full­load efficiency.

6) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer
for the PGND and SGND plane as an EMI shield to
keep radiated noise away from the device, feed­back dividers, and analog bypass capacitors.

7) To prevent discharge of the compensation capaci­tors during the off-time of the dimming cycle,
ensure that the PCB area close to these compo­nents has extremely low leakage. Discharge of
these capacitors due to leakage results in reduced
performance of the dimming circuitry.

MAX16834

High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

22

____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

22

____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

Package Information

For the latest package outline information and land patterns, go
to www.maxim-ic.com/packages

.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

20-TQFN-EP T2044-3

21-0139

MAX16834

TQFN

TOP VIEW

19

20

18

17

7

6

8

SGND

REF

REFI

9

OVP+

V

CC

PGND

CS

IN

1+2

DIMOUT

45

15 14 12 11

LV

SENSE+

UVEN

RT/SYNC

FLT

SC

COMP

NDRV

3

13

CLV

16

*EP

*EP = EXPOSED PAD.

10

PWMDIM

HV

Pin Configuration

Chip Information

PROCESS: BiCMOS–DMOS