Rainbow Electronics MAX1637 User Manual

For free samples & the latest literature: http://www.maxim-ic.com, or phone [phone].
For small orders, phone [phone] ext. 3468.

General Description

The MAX1637 synchronous, buck, switch-mode power­supply controller generates the CPU supply voltage in
battery-powered systems. The MAX1637 is a stripped­down version of the MAX1636 in a smaller 16-pin QSOP
package. The MAX1637 is intended to be powered sep­arately from the battery by an external bias supply (typi­cally the +5V system supply) in applications where the
battery exceeds 5.5V. The MAX1637 achieves excellent
DC and AC output voltage accuracy. This device can
operate from a low input voltage (3.15V) and delivers the
excellent load-transient response needed by upcoming
generations of dynamic-clock CPUs.

Using synchronous rectification, the MAX1637 achieves
up to 95% efficiency. Efficiency is greater than 80%
over a 1000:1 load-current range, which extends bat­tery life in system-suspend or standby mode. Excellent
dynamic response corrects output load transients
caused by the latest dynamic-clock CPUs within five
300kHz clock cycles. Powerful 1A on-board gate driv­ersensure fast external N-channel MOSFET switching.

The MAX1637 features a logic-controlled and synchro­nizable, fixed-frequency, pulse-width-modulation
(PWM) operating mode. This reduces noise and RF
interference in sensitive mobile-communications and
pen-entry applications. Asserting the SKIP pin enables
fixed-frequency mode, for lowest noise under all load
conditions. For a stand-alone device that includes a
+5V VL linear regulator and low-dropout capabilities,
refer to the MAX1636 data sheet.

________________________Applications

Notebook Computers Subnotebook Computers
Handy-Terminals, PDAs

____________________________Features

♦ ±2% DC Accuracy
♦ 0.1% (typ) DC Load Regulation
♦ Adjustable Switching Frequency to 350kHz
♦ Idle Mode™ Pulse-Skipping Operation
♦ 1.10V to 5.5V Adjustable Output Voltage
♦ 3.15V Minimum IC Supply Voltage (at V

CC

pin)

♦ Internal Digital Soft-Start
♦ 1.1V ±2% Reference Output
♦ 1µA Total Shutdown Current
♦ Output Overvoltage Crowbar Protection
♦ Output Undervoltage Shutdown (foldback)
♦ Tiny 16-Pin QSOP Package

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

________________________________________________________________

Maxim Integrated Products

1

16
15
14
13
12
11
10

9

1
2
3
4
5
6
7
8

CSH SKIP

LX
DH
BST
PGND
DL
V

GG

V

CC

TOP VIEW

MAX1637

QSOP

CSL

FB

SHDN

CC

REF

SYNC

GND

__________________Pin Configuration

__________Typical Operating Circuit

19-1321; Rev 1; 2/98

PART

MAX1637EEE -40°C to +85°C

TEMP. RANGE PIN-PACKAGE

16 QSOP

EVALUATION KIT

AVAILABLE

______________Ordering Information

Idle Mode is a trademark of Maxim Integrated Products.

MAX1637

SHDN

GND

V

BATT

V

BIAS

DL

PGND

LX

DH

V

CC

V

GG

BST

CSH
CSL

FB

SKIP

SYNC

REF

CC

OUTPUT

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 1, VCC= VGG= 5V, SYNC = VCC, I

REF

= 0mA, TA= 0°C to +85°C, unless otherwise noted. Typical values are at

T

A

= +25°C.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

GND to PGND .............................................................+2V to -2V

LX, BST to GND......................................................-0.3V to +36V

BST, DH to LX...........................................................-0.3V to +6V

V

CC

, VGG, CSL, CSH, SHDN to GND.......................-0.3V to +6V

DL to GND..................................................-0.3V to (V

GG

+ 0.3V)

REF, SKIP, SYNC, CC to GND ...................-0.3V to (V

CC

+ 0.3V)

REF Output Current.............................................................20mA

REF Short-Circuit to GND..............................................Indefinite

Operating Temperature Range ...........................-40°C to +85°C

Continuous Power Dissipation (T

A

= +70°C)

QSOP (derate 8.3mW/°C above +70°C)......................667mW

Storage Temperature Range.............................-65°C to +160°C

Junction Temperature......................................................+150°C

Lead Temperature (soldering, 10sec).............................+300°C

SYNC = GND

SYNC = V

CC

FB tied to V

OUT

, 0mV < (CSH - CSL) < 80mV,

includes line and load regulation

VCC= 3.15V to 5.5V

REF load = 0µA to 50µA

VCC, V

GG

REF load = 0µA

Rising edge, hysteresis = 15mV

Rising edge, hysteresis = 15mV

SHDN = GND, VCC= V

GG

CSH - CSL = 0mV to CSH - CSL = 100mV

CSH - CSL

VCC= 5V
VCC= 3.3V

SHDN to full current limit, four levels

VFB= V

REF

CONDITIONS

170 200 230

Oscillator Frequency kHz

270 300 330

mV3REF Line Regulation

mV10REF Load Regulation

V1.080 1.100 1.120REF Output Voltage

V2.80 3.05VGGUndervoltage Lockout Threshold

V2.80 3.05VCCUndervoltage Lockout Threshold

%2AC Load Regulation

mV20 30 40Idle-Mode Switchover Threshold

clocks512Soft-Start Ramp Time

nA-50 50FB Input Current

V3.15 5.5Input Voltage Range

µA0.5 3Shutdown Supply Current

V1.080 1.100 1.120Output Voltage

V

REF

5.5

Output Adjustment Range V

V

REF

3.6

UNITSMIN TYP MAXPARAMETER

CSH > CSL
CSH < CSL

80 100 120

Current-Limit Threshold mV

-145 -100 -55

Output not switching

1.5 2.5

Power Consumption mW

1 1.75

SYNC = GND

SYNC = V

CC

ns200SYNC Input Pulse Width High

%

93 96

89 92

Maximum Duty Factor

(Note 1)

kHz240 340SYNC Input Frequency Range

ns200SYNC Input Rise/Fall Time

ns200SYNC Input Pulse Width Low

VCC= VGG= 5V
VCC= VGG= 3.3V

SMPS CONTROLLER

INTERNAL REFERENCE

OSCILLATOR

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, VCC= VGG= 5V, SYNC = VCC, I

REF

= 0mA, TA= 0°C to +85°C, unless otherwise noted. Typical values are at

T

A

= +25°C.)

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 1, VCC= VGG= 5V, SYNC = VCC, I

REF

= 0mA, TA= -40°C to +85°C, unless otherwise noted.) (Note 2)

High or low, DH or DL

DH or DL forced to 2V

CSH = CSL = 5V, VCC= VGG= GND,
either CSH or CSL input

FB to DL delay, 22mV overdrive, C

GATE

= 2000pF

FB, with respect to regulation point

From shutdown or power-on-reset state

Pin at GND or V

CC

SHDN, SKIP, SYNC

% of nominal output

SHDN, SKIP, SYNC

CONDITIONS

Ω7Gate Driver On-Resistance

A1Gate Driver Sink/Source Current

µA10Current-Sense Input Leakage Current

µA-1 1Logic Input Bias Current

V0.8Logic Input Voltage Low

V2.4Logic Input Voltage High

µs1.25Overvoltage Fault Propagation Delay

%4 7 10Overvoltage Trip Threshold

clocks6144Output Undervoltage Lockout Delay

%60 70 80Output Undervoltage Lockout Threshold

UNITSMIN TYP MAXPARAMETER

FB tied to V

OUT

, 0mV < (CSH - CSL) < 80mV,

includes line and load regulation

SYNC = GND

VCC, V

GG

SYNC = V

CC

Rising edge, hysteresis = 15mV

Rising edge, hysteresis = 15mV

VCC= 3.3V

VCC= VGG= 3.3V, output not switching

VCC= 5V

VCC= VGG= 5V, output not switching

CSH > CSL

CONDITIONS

kHz240 340SYNC Input Frequency Range

ns200SYNC Input Rise/Fall Time

ns200SYNC Input Pulse Width Low

ns200SYNC Input Pulse Width High

kHz

170 230

Oscillator Frequency

262 338

V2.80 3.05VGGUndervoltage Lockout Threshold

V2.80 3.05VCCUndervoltage Lockout Threshold

mW1.75

Power Consumption

mW2.5

mV70 130Current-Limit Threshold

V1.080 1.120Output Voltage

V3.15 5.5Input Voltage Range

V

REF

3.6

Output Adjustment Range V

V

REF

5.5

UNITSMIN TYP MAXPARAMETER

OVERVOLTAGE PROTECTION

INPUTS AND OUTPUTS

SMPS CONTROLLER

INTERNAL REFERENCE

OSCILLATOR

0

10

5

15

20

0 3 41 2 5 6 87 9

SUPPLY CURRENT

vs. LOAD CURRENT

MAX1637-07

LOAD CURRENT (A)

V

CC

+ V

GG

SUPPLY CURRENT (mA)

SYNC = HIGH

SYNC = LOW

SKIP = LOW

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, VCC= VGG= 5V, SYNC = VCC, I

REF

= 0mA, TA= -40°C to +85°C, unless otherwise noted.) (Note 2)

SHDN, SKIP, SYNC

SHDN, SKIP, SYNC

% of nominal output

FB, with respect to regulation point

CONDITIONS

V0.8Logic Input Voltage Low

V2.4Logic Input Voltage High

%60 80Output Undervoltage Lockout Threshold

%4.0 10Overvoltage Trip Threshold

UNITSMIN TYP MAXPARAMETER

Note 1: Guaranteed by design, not production tested.
Note 2: Specifications from -40°C to 0°C are guaranteed by design and not production tested.

__________________________________________Typical Operating Characteristics

(V

OUT

= 3.3V, TA = +25°C, unless otherwise noted.)

100

50

0.01 1010.1

EFFICIENCY vs. LOAD CURRENT

(1.7V/7A CIRCUIT)

70

60

90

80

MAX1637-01

LOAD CURRENT (A)

EFFICIENCY (%)

SKIP = LOW

V

BATT

= 7V

V

BATT

= 15V

V

BATT

= 22V

100

0

0.001 1010.01 0.1

EFFICIENCY vs. LOAD CURRENT

(2.5V/3A CIRCUIT)

40
30
20
10

80
70

90

60
50

MAX1637-02

LOAD CURRENT (A)

EFFICIENCY (%)

V

BATT

= 15V

SKIP = LOW

V

BATT

= 7V

V

BATT

= 22V

100

50

0.01 1010.1

EFFICIENCY vs. LOAD CURRENT

(2.5V/2A CIRCUIT)

70

60

90

80

MAX1637-03

LOAD CURRENT (A)

EFFICIENCY (%)

V

BATT

= 7V

SKIP = LOW

V

BATT

= 15V

V

BATT

= 22V

100

50

0.01 1010.1

EFFICIENCY vs. LOAD CURRENT

(3.3V/3A CIRCUIT)

70

60

90

80

MAX1637-04

LOAD CURRENT (A)

EFFICIENCY (%)

V

BATT

= 5V

V

BATT

= 30V

V

BATT

= 15V

SKIP = LOW

0

10

5

15

20

3.0 4.03.5 4.5 5.0

5.5

6.0

SUPPLY CURRENT

vs. SUPPLY VOLTAGE

MAX1637-06

SUPPLY VOLTAGE (V)

V

CC

+ V

GG

SUPPLY CURRENT (mA)

I

LOAD

= 1A

V

OUT

= 3.3V

SKIP = HIGH

SKIP = LOW

OVERVOLTAGE PROTECTION

INPUTS AND OUTPUTS

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

_______________________________________________________________________________________

5

10

-10

0.01 1010.1

LOAD REGULATION

vs. LOAD CURRENT

-2

-6

-8

6

8

4

0

-4

2

MAX1637-08

LOAD CURRENT (A)

LOAD REGULATION ∆V

OUT

(mV)

0

0.3

0.2

0.1

0.4

0.5

0.6

0 403010 20 50 60 70 80 90 100

REF LOAD-REGULATION ERROR

vs. REF LOAD CURRENT

MAX1637-09

REF LOAD CURRENT (µA)

REF LOAD REGULATION ∆V (mV)

900

0

0.01 1010.1

DROPOUT VOLTAGE

vs. LOAD CURRENT

300
200

100

700

800

600

400

500

MAX1637-10

LOAD CURRENT (A)

DROPOUT VOLTAGE (mV)

V

OUT

FORCED TO 3.27V

SYNC = V

CC

V

OUT

20mV/div

V

LX

INDUCTOR
CURRENT

1A

0V

5V

0A

SWITCHING WAVEFORMS

(PWM MODE)

MAX1637-13

1µs/div

V

OUT

50mV/div

LOAD
CURRENT

0A

2A

4A

LOAD-TRANSIENT RESPONSE

(3.3V/3A, PWM MODE)

MAX1637 TOC11

100µs/div

V

OUT

50mV/div

5A LOAD CURRENT
0A

10A

LOAD-TRANSIENT RESPONSE

(1.8V, PWM MODE)

MAX1637 TOC12

100µs/div

V

OUT

50mV/div

V

LX

INDUCTOR
CURRENT

1A

0V

5V

0A

SWITCHING WAVEFORMS

(PFM MODE)

MAX1637-14

20µs/div

V

OUT

= 1.7V

1µs/div

SWITCHING WAVEFORMS

DROPOUT OPERATION

MAX1637-15

V

OUT

10mV/div

V

LX

2V/div

INDUCTOR
CURRENT

1A
0A

V

OUT

FORCED TO 3.27V

SYNC = V

CC

____________________________________Typical Operating Characteristics (continued)

(V

OUT

= 3.3V, TA = +25°C, unless otherwise noted.)

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

6 _______________________________________________________________________________________

____________________________________Typical Operating Characteristics (continued)

(V

OUT

= 3.3V, TA = +25°C, unless otherwise noted.)

500µs/div

TIME EXITING SHUTDOWN
(V

OUT

= 3.3V, I

LOAD

= 7A)

MAX1637-16

V

OUT

1V/div

V

SHDN

5V/div

V

OUT

100mV/div

V

DL

INDUCTOR
CURRENT

-5A

0A

0V

5V

-10A

OVERVOLTAGE-PROTECTION WAVEFORMS

(V

IN

SHORTED TO V

OUT

THROUGH A 0.5Ω RESISTOR)

MAX1637-17

10µs/div

______________________________________________________________Pin Description

PIN

High-Side Current-Sense InputCSH1

FUNCTIONNAME

Low-Side Current-Sense InputCSL2

Compensation Pin. Connect a small capacitor to GND to set the integration time constant.CC4

Feedback Input. Connect to center of resistor divider.FB3

Shutdown Control Input. Turns off entire IC. When low, reduces supply current below 0.5µA (typ). Drive with
logic input or connect to RC network between GND and VCCfor automatic start-up.

SHDN

6

Analog GroundGND8

Oscillator Frequency Select and Synchronization Input. Tie to VCCfor 300kHz operation; tie to GND for
200kHz operation.

SYNC7

1.100V Reference Output. Capable of sourcing 50µA for external loads. Bypass with 0.22µF minimum.REF5

Gate-Drive and Boost-Circuit Power Supply. Can be driven from a supply other than VCC. If the same supply
is used by both VCCand VGG, isolate VCCfrom VGGwith a 20Ω resistor. Bypass to PGND with a 4.7µF
capacitor. VGGcurrent = (QG1+ QG2) x f, where QGis the MOSFET gate charge at VGS= VGG.

V

GG

10

Power GroundPGND12

Low-Side Gate-Driver OutputDL11

High-Side Gate-Driver OutputDH14

Low-Noise Mode Control. Forces fixed-frequency PWM operation when high.

SKIP

16

Inductor ConnectionLX15

Boost Capacitor ConnectionBST13

Main Analog Supply-Voltage Input to the Chip. VCCpowers the PWM controller, logic, and reference. Input
range is 3.15V to 5.5V. Bypass to GND with a 0.1µF capacitor close to the pin.

V

CC

9

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

_______________________________________________________________________________________ 7

MAX1637

0.1µF

V

BIAS

+5V

NOMINAL

0.1µF

1µF

470pF

C1

Q1

CMPSH-3

Q2

C2

L1

*

R1

R2

R3

OUTPUT

4.7µF

*SEE

RECTIFIER CLAMP DIODE

SECTION

**OPTIONAL RC NETWORK FOR POWER-ON-RESET

DL

PGND

LX

DH

BST

V

GG

V

CC

V

BATT

CSH

CSL

FB

1M

**

ON/OFF

CC

GND

SHDN

REF

SYNC

20Ω

SKIP

0.01µF**

Figure 1. Standard Application Circuit

______Standard Application Circuit

The basic MAX1637 buck converter (Figure 1) is easily
adapted to meet a wide range of applications where a
5V or lower supply is available. The components listed
in Table 1 represent a good set of trade-offs among
cost, size, and efficiency, while staying within the worst­case specification limits for stress-related parameters
such as capacitor ripple current. Do not change the cir­cuit’s switching frequency without first recalculating
component values (particularly inductance value at
maximum battery voltage).

The power Schottky diode across the synchronous rec­tifier is optional because the MOSFETs chosen incorpo­rate a high-speed silicon diode. However, installing the
Schottky will generally improve efficiency by about 1%.
If used, the Schottky diode DC current must be rated to
at least one-third of the maximum load current.

_______________Detailed Description

The MAX1637 is a BiCMOS, switch-mode power-supply
(SMPS) controller designed primarily for buck-topology
regulators in battery-powered applications where high
efficiency and low quiescent supply current are critical.
Light-load efficiency is enhanced by automatic idle­mode operation—a variable-frequency, pulse-skipping
mode that reduces transition and gate-charge losses.
The step-down, power-switching circuit consists of two
N-channel MOSFETs, a rectifier, and an LC output filter.
Output voltage for this device is the average AC volt­age at the switching node, which is regulated by
changing the duty cycle of the MOSFET switches. The
gate-drive signal to the high-side N-channel MOSFET,
which must exceed the battery voltage, is provided by
a flying-capacitor boost circuit that uses a 100nF
capacitor between BST and LX. Figure 2 shows the
major circuit blocks.

LOAD CURRENT

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

8 _______________________________________________________________________________________

Table 1. Component Selection for Standard Applications

Table 2. Component Suppliers

2.5VOutput Voltage Range

300kHzFrequency

Chipset SupplyApplication

1/2 Si4902DY or
1/2 MMDF3NO3HD

Q1 High-Side
MOSFET

7V to 22VInput Voltage Range

[phone](81) 7-2070-1174Sanyo

Tokin [phone]

[phone]

(1) 408-434-0375

(1) 847-390-4428TDK

Sprague

[phone](81) 3-3607-5144Sumida

[phone]

[phone]

[phone]

(1) 714-373-7183Panasonic

(1) 603-224-1430

(1) 408-970-3950Siliconix

[phone](1) 602-994-6430Motorola

[phone]

COMPANY

Matsuo [phone](1) 714-960-6492

USA PHONE

FACTORY FAX

(COUNTRY CODE)

(1) 847-696-9278

Marcon/United

Chemi-Con

COMPANY

Central
Semiconductor

Fairchild [phone](1) 408-721-1635

[phone](1) 512-992-3377IRC

Dale

[phone](1) 310-322-3332

International
Rectifier (IR)

[phone]

[phone]
[phone]

(1) 847-639-1469Coilcraft

(1) 605-665-1627

(1) 561-241-9339Coiltronics

USA PHONE

[phone]

[phone]

FACTORY FAX

(COUNTRY CODE)

(1) 516-435-1824

(1) 803-626-3123AVX

2.5V

300kHz

Chipset Supply

International Rectifier
IRF7403 or
Siliconix Si4412

7V to 22V

3.3V1.7V

300kHz300kHz

General PurposeCPU Core

International Rectifier
IRF7403 or
Siliconix Si4412

Fairchild FDS9412 or
International Rectifier
IRF7403

4.75V to 30V7V to 22V

10µF, 25V ceramic
Tokin C34Y5U1E106Z
or Marcon/United
Chemicon
THCR40E1E106ZT

10µF, 25V ceramic
Tokin C34Y5U1E106Z
or Marcon/United
Chemicon
THCR40E1E106ZT

C1 Input Capacitor

0.020Ω, 1% (2010)
Dale WSL-2010-R020F

0.033Ω, 1% (2010)
Dale WSL-2010-R033F

R1 Resistor

470µF, 6.3V tantalum
Kemet
T510X477(1)006AS or
470µF, 4V tantalum
Sprague
594D477X0004R2T

220µF, 6.3V tantalum
Sprague
595D227X96R3C2

C2 Output Capacitor

10µH
Sumida CDRH125-100

10µH
Coilcraft
DO3316P-103 or
Coiltronics UP2-100

L1 Inductor

International Rectifier
IRF7413 or
Siliconix Si4410DY

1/2 Si4902DY or
1/2 MMDF3NO3HD

Q2 Low-Side MOSFET

10µF, 30V
Sanyo OS-CON

4 x 10µF, 25V ceramic
Tokin C34Y5U1E106Z
or Marcon/United
Chemicon
THCR40E1E106ZT

0.020Ω, 1% (2010)
Dale WSL-2010-R020F

0.010Ω, 1% (2512)
Dale WSL-2512-R010F

470µF, 6.3V tantalum
Kemet
T510X477(1)006AS or
470µF, 4V tantalum
Sprague
594D477X0004R2T

3 x 470µF, 6.3V tantalum
Kemet
T510X477(1)006AS or
470µF, 4V tantalum
Sprague
594D477X0004R2T

10µH Sumida
CDRH125-100

2.2µH
Panasonic P1F2R0HL or
Coiltronics UP4-2R2 or
Coilcraft
DO5022P-222HC

International Rectifier
IRF7413 or
Siliconix Si4410DY

Fairchild FDS6680 or
Siliconix Si4420DY

COMPONENT

3A (EV KIT)2A 3A7A (EV KIT)

LOAD CURRENT

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

_______________________________________________________________________________________ 9

The pulse-width-modulation (PWM) controller consists
of a multi-input PWM comparator, high-side and low­side gate drivers, and logic. It uses a 200kHz/300kHz
synchronizable oscillator. The MAX1637 contains fault­protection circuits that monitor the PWM output for
undervoltage and overvoltage. It includes a 1.100V pre-

cision reference. The circuit blocks are powered from
an internal IC power rail that receives power from VCC.
VGGprovides direct power to the synchronous-switch
gate driver, but provides indirect power to the high­side-switch gate driver via an external diode-capacitor
boost circuit.

REF

IC

POWER

200kHz

TO

300kHz

OSC

PWM

LOGIC

3.15V TO 5.5V

V

BATT

V

OUT

REF

SHDN

SYNC

V

CC

DL

PGND

LX

DH

BST

V

GG

SKIP

CSH
CSL

FB

CC

REF

GND

V

REF

+7%

V

REF

-30%

+

60kHz

LP FILTER

SHUTDOWN

CONTROL

1.1V
REF.

ERROR
INTEGRATOR

+

-

+

-

+

-

+

-

+

MAX1637

gm

OVERVOLTAGE

FAULT

UNDER­VOLTAGE
FAULT

OFF

SLOPE
COMPENSATION

V

BIAS

Figure 2. Functional Diagram

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

10 ______________________________________________________________________________________

PWM Controller Block

The heart of the current-mode PWM controller is a
multi-input, open-loop comparator that sums four sig­nals: the output voltage error signal with respect to
the reference voltage, the current-sense signal, the
integrated voltage-feedback signal, and the slope-

compensation ramp (Figure 3). The PWM controller is
a direct-summing type, lacking a traditional error
amplifier and the phase shift associated with it. This
direct-summing configuration approaches ideal
cycle-by-cycle control over the output voltage.

SHOOT­THROUGH
CONTROL

R

Q

30mV

R Q

LEVEL
SHIFT

1X

gm

2X

OSC

LEVEL
SHIFT

REF

CURRENT
LIMIT

SYNCHRONOUS

RECTIFIER CONTROL

SHDN

CK

-100mV

CSH

CSL

CC

REF

FB

BST

DH

LX

V

GG

DL

PGND

S

S

SLOPE
COMPENSATION

SKIP

COUNTER

DAC

SOFT-START

Figure 3. PWM Controller Functional Diagram

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

______________________________________________________________________________________ 11

Idle Mode

When SKIP is low, idle-mode circuitry automatically
optimizes efficiency throughout the load-current range.
Idle mode dramatically improves light-load efficiency
by reducing the effective frequency, subsequently
reducing switching losses. It forces the peak inductor
current to ramp to 30% of the full current limit, deliver­ing extra energy to the output and allowing subsequent
cycles to be skipped. Idle mode transitions seamlessly
to fixed-frequency PWM operation as load current
increases (Table 3).

Fixed-Frequency Mode

When SKIP is high, the controller always operates in
fixed-frequency PWM mode for lowest noise. Each pulse
from the oscillator sets the main PWM latch that turns on
the high-side switch for a period determined by the duty
factor (approximately V

OUT

/ VIN). As the high-side switch
turns off, the synchronous rectifier latch is set; 60ns later,
the low-side switch turns on. The low-side switch stays on
until the beginning of the next clock cycle.

In PWM mode, the controller operates as a fixed-fre­quency, current-mode controller in which the duty fac­tor is set by the input/output voltage ratio. PWM mode
(SKIP = high) forces two changes on the PWM con-
troller. First, it disables the minimum-current compara­tor, ensuring fixed-frequency operation. Second, it
changes the detection threshold for reverse-current
limit from 0mV to -100mV, allowing the inductor current
to reverse at light loads. This results in fixed-frequency
operation and continuous inductor-current flow. PWM
mode eliminates discontinuous-mode inductor ringing
and improves cross-regulation of transformer-coupled,
multiple-output supplies.

The current-mode feedback system regulates the peak
inductor-current value as a function of the output volt­age error signal. In continuous-conduction mode, the
average inductor current is nearly the same as the
peak current, so the circuit acts as a switch-mode
transconductance amplifier. This pushes the second
output LC filter pole, normally found in a duty-factor­controlled (voltage-mode) PWM, to a higher frequency.
To preserve inner-loop stability and eliminate regenera­tive inductor-current “staircasing,” a slope-compensa­tion ramp is summed into the main PWM comparator to
make the apparent duty factor less than 50%.

The relative gains of the voltage-sense and current­sense inputs are weighted by the values of the current
sources that bias four differential input stages in the
main PWM comparator (Figure 4). The voltage sense
into the PWM has been conditioned by an integrated
component of the feedback voltage, yielding excellent
DC output voltage accuracy. See the

Output Voltage

Accuracy

section for details.

Constant frequency PWM,
continuous inductor current

HeavyLow

Constant frequency PWM,
continuous inductor current

HeavyHigh

Constant frequency PWM,
continuous inductor current

LightHigh

SKIP

Pulse-skipping, discontin­uous inductor current

LightLow

DESCRIPTION

LOAD

CURRENT

Table 3. SKIP PWM Table

PWM

PWM

PWM

Idle

MODE

Figure 4. Main PWM Comparator Functional Diagram

V

CC

CC

I1

REF

CSH

CSL

FB

I2

SLOPE COMPENSATION

R1

I3 I4

R2

UNCOMPENSATED
HIGH-SPEED
LEVEL TRANSLATOR
AND BUFFER

V

BIAS

TO PWM
LOGIC

OUTPUT DRIVER

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

12 ______________________________________________________________________________________

REF, VCC, and VGGSupplies

The 1.100V reference (REF) is accurate to ±2% over
temperature, making REF useful as a precision system
reference. Bypass REF to GND with a 0.22µF (min)
capacitor. REF can supply up to 50µA for external
loads. Loading REF reduces the main output voltage
slightly because of the reference load-regulation error.

The MAX1637 has two independent supply pins, V

CC

and VGG. VCCpowers the sensitive analog circuitry of
the SMPS, while VGGpowers the high-current MOSFET
drivers. No protection diodes or sequencing require­ments exist between the two supplies. Isolate VGGfrom
VCCwith a 20Ω resistor if they are powered from the
same supply. Bypass VCCto GND with a 0.1µF capaci­tor located directly adjacent to the pin. Use only small­signal diodes for the boost circuit (10mA to 100mA
Schottky or 1N4148 diodes are preferred), and bypass
VGGto PGND with a 4.7µF capacitor directly at the
package pins. The VCCand VGGinput range is 3.15V
to 5.5V.

High-Side Boost Gate Drive (BST)

Gate-drive voltage for the high-side N-channel switch is
generated by a flying-capacitor boost circuit (Figure 2).
The capacitor between BST and LX is alternately
charged from the VGGsupply and placed parallel to
the high-side MOSFET’s gate-source terminals.

On start-up, the synchronous rectifier (low-side
MOSFET) forces LX to 0V and charges the boost
capacitor to VGG. On the second half-cycle, the SMPS
turns on the high-side MOSFET by closing an internal
switch between BST and DH. This provides the neces­sary enhancement voltage to turn on the high-side
switch, an action that boosts the gate-drive signal
above the battery voltage.

Ringing at the high-side MOSFET gate (DH) in discon­tinuous-conduction mode (light loads) is a natural oper­ating condition. It is caused by residual energy in the
tank circuit, formed by the inductor and stray capaci­tance at the switching node, LX. The gate-drive nega­tive rail is referred to LX, so any ringing there is directly
coupled to the gate-drive output.

Synchronous-Rectifier Driver (DL)

Synchronous rectification reduces conduction losses in
the rectifier by shunting the normal Schottky catch
diode with a low-resistance MOSFET switch. Also, the
synchronous rectifier ensures proper start-up of the
boost gate-driver circuit. If the synchronous power
MOSFET is omitted for cost or other reasons, replace it
with a small-signal MOSFET, such as a 2N7002.

If the circuit is operating in continuous-conduction
mode, the DL drive waveform is simply the complement
of the DH high-side-drive waveform (with controlled
dead time to prevent cross-conduction or “shoot­through”). In discontinuous (light-load) mode, the syn­chronous switch is turned off as the inductor current
falls through zero.

Shutdown Mode and Power-On Reset

SHDN is a logic input with a threshold of about 1.5V
that, when held low, places the IC in its 0.5µA shut­down mode. The MAX1637 has no power-on-reset cir­cuitry, and the state of the device is not known on initial
power-up. In applications that use logic to drive SHDN,
it may be necessary to toggle SHDN to initialize the
part once VCCis stable. In applications that require
automatic start-up, drive SHDN through an external RC
network (Figure 5). The network will hold SHDN low
until VCCstabilizes. Typical values for R and C are 1MΩ
and 0.01µF. For slow-rising VCC, use a larger capacitor.
When cycling VCC, VCCmust stay low long enough to
discharge the 0.01µF capacitor, otherwise the circuit
may not start. A diode may be added in parallel with
the resistor to speed up the discharge.

Current-Limiting and Current-

Sense Inputs (CSH and CSL)

The current-limit circuit resets the main PWM latch and
turns off the high-side MOSFET switch whenever the
voltage difference between CSH and CSL exceeds
100mV. This limiting is effective for both current flow
directions, putting the threshold limit at ±100mV. The
tolerance on the positive current limit is ±20%, so the
external low-value sense resistor (R1) must be sized for
80mV / I

PEAK

, where I

PEAK

is the peak inductor current
required to support the full load current. Components
must be designed to withstand continuous current
stresses of 120mV / R1.

MAX1637

SHDN

R = 1MΩ
C = 0.01µF

V

IN

V

GG

C

R

V

CC

Figure 5. Power-On Reset RC Network for Automatic Start-Up

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

______________________________________________________________________________________ 13

For prototyping or for very high-current applications, it
may be useful to wire the current-sense inputs with a
twisted pair rather than PC traces (two pieces of
wrapped wire twisted together are sufficient). This
reduces the noise picked up at CSH and CSL, which can
cause unstable switching and reduced output current.

Oscillator Frequency

and Synchronization (SYNC)

The SYNC input controls the oscillator frequency as fol­lows: low selects 200kHz, high selects 300kHz. SYNC
can also be used to synchronize with an external 5V
CMOS or TTL clock generator. It has a guaranteed
240kHz to 340kHz capture range. A high-to-low transi­tion on SYNC initiates a new cycle.

Operation at 300kHz optimizes the application circuit
for component size and cost. Operation at 200kHz
increases efficiency, reduces dropout, and improves
load-transient response at low input-output voltage dif­ferences (see the

Low-Voltage Operation

section).

Output Voltage Accuracy (CC)

Output voltage error is guaranteed to be within ±2%
over all conditions of line, load, and temperature. The
MAX1637’s DC load regulation is typically better than

0.1%, due to its integrator amplifier. The device opti­mizes transient response by providing a feedback sig­nal with a direct path from the output to the main
summing PWM comparator. The integrated feedback
signal from the CC transconductance amplifier is also

summed into the PWM comparator, with the gain
weighted so that the signal has only enough gain to
correct the DC inaccuracies. The integrator’s response
time is determined by the time constant set by the
capacitor placed on the CC pin. The time constant
should neither be so fast that the integrator responds to
the normal V

OUT

ripple, nor too slow to negate the inte­grator’s effect. A 470pF to 1500pF CC capacitor is suf­ficient for 200kHz to 300kHz frequencies.

Figure 6 shows the output voltage response to a 0A to
3A load transient with and without the integrator. With
the integrator, the output voltage returns to within 0.1%
of its no-load value with only a small AC excursion.
Without the integrator, load regulation is degraded
(Figure 6b). Asymmetrical clamping at the integrator
output prevents worsening of load transients during
pulse-skipping mode.

Output Undervoltage Lockout

The output undervoltage-lockout circuit protects
against heavy overloads and short-circuits at the main
SMPS output. This scheme employs a timer rather than
a foldback current limit. The SMPS has an undervolt­age-protection circuit, which is activated 6144 clock
cycles after the SMPS is enabled. If the SMPS output is
under 70% of the nominal value, it is latched off and
does not restart until SHDN is toggled. Applications
that use the recommended RC power-on-reset circuit
will also clear the fault condition when VCCfalls below

0.5V (typical). Note that undervoltage protection can

0

2

4

-50

50

I

OUT

(A)

V

OUT

(mV)

(100µs/div)

CC = 470pF
V

OUT

= 3.3V

INTEGRATOR

ACTIVE

Figure 6a. Load-Transient Response with Integrator Active

0

2

4

-50

50

I

OUT

(A)

V

OUT

(mV)

(100µs/div)

CC = REF
V

OUT

= 3.3V

INTEGRATOR

DEACTIVATED

Figure 6b. Load-Transient Response with Integrator
Deactivated

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

14 ______________________________________________________________________________________

make prototype troubleshooting difficult since only
20ms or 30ms elapse before the SMPS is latched off.
The overvoltage crowbar protection is disabled in out­put undervoltage mode.

Output Overvoltage Protection

The overvoltage crowbar-protection circuit is intended
to blow a fuse in series with the battery if the main
SMPS output rises significantly higher than its standard
level (Table 4). In normal operation, the output is com­pared to the internal precision reference voltage. If the
output goes 7% above nominal, the synchronous-recti­fier MOSFET turns on 100% (the high-side MOSFET is
simultaneously forced off) in order to draw massive
amounts of battery current to blow the fuse. This safety
feature does not protect the system against a failure of
the controller IC itself, but is intended primarily to guard
against a short across the high-side MOSFET. A crow­bar event is latched and can only be reset by a rising
edge on SHDN (or by removal of the VCCsupply volt­age). The overvoltage-detection decision is made rela­tive to the regulation point.

Internal Digital Soft-Start Circuit

Soft-start allows a gradual increase of the internal cur­rent-limit level at start-up to reduce input surge cur­rents. The SMPS contains an internal digital soft-start
circuit controlled by a counter, a digital-to-analog con­verter (DAC), and a current-limit comparator. In shut­down, the soft-start counter is reset to zero. When the
SMPS is enabled, its counter starts counting oscillator
pulses, and the DAC begins incrementing the compari­son voltage applied to the current-limit comparator. The
DAC output increases from 0mV to 100mV in five equal
steps as the count increases to 512 clocks. As a result,
the main output capacitor charges up relatively slowly.

The exact time of the output rise depends on output
capacitance and load current, but it is typically 1ms
with a 300kHz oscillator.

Setting the Output Voltage

The output voltage is set via a resistor divider connect­ed to FB (Figure 1). Calculate the output voltage with
the following formula:

V

OUT

= V

REF

(1 + R2 / R3)

where V

REF

= 1.1V nominal.

Recommended normal values for R3 range from 5kΩ to
100kΩ. To achieve a 1.1V nominal output, connect FB
directly to CSL. Remote output voltage sensing is pos­sible by using the top of the external resistor divider as
the remote sense point.

__________________Design Procedure

The standard application circuit (Figure 1) contains a
ready-to-use solution for common application needs.
Use the following design procedure to optimize the
basic schematic for different voltage or current require­ments. But before beginning a design, firmly establish
the following:

• Maximum input (battery) voltage, V

IN(MAX)

. This
value should include the worst-case conditions, such
as no-load operation when a battery charger or AC
adapter is connected but no battery is installed.
V

IN(MAX)

must not exceed 30V.

• Minimum input (battery) voltage, V

IN(MIN)

. This value
should be taken at full load under the lowest battery
conditions. If the minimum input-output difference is
less than 1.5V, the filter capacitance required to
maintain good AC load regulation increases (see

Low-Voltage Operation

section).

Table 4. Operating Modes

All circuit blocks offLowShutdown

REF = off, DL = lowHigh

Output

Undervoltage

Lockout

REF = off, DL = highHigh

Overvoltage

(Crowbar)

—

V

OUT

below 70% of
nominal after 20ms to
30ms timeout expires

V

OUT

greater than 7%
above regulation point

V

OUT

in regulation

CONDITIONSMODE

All circuit blocks activeHighRun

STATUS

SHDN

Lowest current consumption

Rising edge on SHDN exits
UVLO

Rising edge on SHDN exits
crowbar

Normal operation

NOTES

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

______________________________________________________________________________________ 15

Inductor Value

The exact inductor value is not critical and can be
freely adjusted to allow trade-offs among size, cost,
and efficiency. Lower inductor values minimize size
and cost, but reduce efficiency due to higher peak­current levels. The smallest inductor value is obtained
by lowering the inductance until the circuit operates at
the border between continuous and discontinuous
mode. Further reducing the inductor value below this
crossover point results in discontinuous-conduction
operation, even at full load. This helps lower output filter
capacitance requirements, but efficiency suffers under
these conditions, due to high I2R losses. On the other
hand, higher inductor values produce greater efficien­cy, but also result in resistive losses due to extra wire
turns—a consequence that eventually overshadows the
benefits gained from lower peak current levels. High
inductor values can also affect load-transient response
(see the V

SAG

equation in the

Low-Voltage Operation

section). The equations in this section are for continu­ous-conduction operation.

Three key inductor parameters must be specified:
inductance value (L), peak current (I

PEAK

), and DC
resistance (RDC). The following equation includes a
constant, LIR, which is the ratio of inductor peak-to­peak AC current to DC load current. A higher LIR value
allows lower inductance, but results in higher losses
and ripple. A good compromise is a 30% ripple-current
to load-current ratio (LIR = 0.3), which corresponds to a
peak inductor current 1.15 times higher than the DC
load current.

L = V

OUT(VIN(MAX)

- V

OUT

) / (V

IN(MIN)

x ƒ x I

OUT

x

LIR)

where ƒ = switching frequency (normally 200kHz or
300kHz), and I

OUT

= maximum DC load current.

The peak current can be calculated as follows:

I

PEAK

= I

LOAD

+ [V

OUT(VIN(MAX)

- V

OUT

) / (2 x ƒ x L

x V

IN(MAX)

)]

The inductor’s DC resistance should be low enough
that RDCx I

PEAK

< 100mV, as it is a key parameter for
efficiency performance. If a standard, off-the-shelf
inductor is not available, choose a core with an LI2rat­ing greater than L x IPEAK2and wind it with the largest
diameter wire that fits the winding area. For 300kHz
applications, ferrite-core material is strongly preferred;
for 200kHz applications, Kool-Mu®(aluminum alloy) or
even powdered iron is acceptable. If light-load efficien­cy is unimportant (in desktop PC applications, for
example), then low-permeability iron-powder cores can

be acceptable, even at 300kHz. For high-current appli­cations, shielded-core geometries (such as toroidal or
pot core) help keep noise, EMI, and switching­waveform jitter low.

Current-Sense Resistor Value

The current-sense resistor value is calculated accord­ing to the worst-case, low-current limit threshold volt­age (from the

Electrical Characteristics

) and the peak

inductor current:

R

SENSE

= 80mV / I

PEAK

Use I

PEAK

from the second equation in the

Inductor

Value

section. Use the calculated value of R

SENSE

to
size the MOSFET switches and specify inductor satura­tion-current ratings according to the worst-case high­current-limit threshold voltage:

I

PEAK

= 120mV / R

SENSE

Low-inductance resistors, such as surface-mount metal
film, are recommended.

Input Capacitor Value

Connect low-ESR bulk capacitors directly to the drain
on the high-side MOSFET. The bulk input filter capaci­tor is usually selected according to input ripple current
requirements and voltage rating, rather than capacitor
value. Electrolytic capacitors with low enough equiva­lent series resistance (ESR) to meet the ripple-current
requirement invariably have sufficient capacitance val­ues. Aluminum electrolytic capacitors, such as Sanyo
OS-CON or Nichicon PL, are superior to tantalum
types, which risk power-up surge-current failure, espe­cially when connecting to robust AC adapters or low­impedance batteries. RMS input ripple current (I

RMS

) is
determined by the input voltage and load current, with
the worst case occurring at VIN= 2 x V

OUT

. Therefore,

when VINis 2 x V

OUT

:

I

RMS

= I

LOAD

/ 2

VCCand VGGshould be isolated from each other with a
20Ω resistor and bypassed to ground independently.
Place a 0.1µF capacitor between VCCand GND, as
close to the supply pin as possible. A 4.7µF capacitor
is recommended between VGGand PGND.

Output Filter Capacitor Value

The output filter capacitor values are generally deter­mined by the ESR and voltage-rating requirements,
rather than by actual capacitance requirements for loop
stability. In other words, the low-ESR electrolytic capac­itor that meets the ESR requirement usually has more
output capacitance than is required for AC stability.

Kool-Mu is a trademark of Magnetics, Inc.

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

16 ______________________________________________________________________________________

Use only specialized low-ESR capacitors intended for
switching-regulator applications, such as AVX TPS,
Sprague 595D, Sanyo OS-CON, or Nichicon PL series.
To ensure stability, the capacitor must meet both mini­mum capacitance and maximum ESR values as given
in the following equations:

COUT > V

REF

(1 + V

OUT

/ V

IN(MIN)

) / V

OUT

x R

SENSE

x ƒ

R

ESR

< R

SENSE

x V

OUT

/ V

REF

where R

ESR

can be multiplied by 1.5, as discussed

below.
These equations are worst case, with 45 degrees of

phase margin to ensure jitter-free, fixed-frequency
operation, and provide a nicely damped output
response for zero to full-load step changes. Some cost­conscious designers may wish to bend these rules with
less-expensive capacitors, particularly if the load lacks
large step changes. This practice is tolerable if some
bench testing over temperature is done to verify
acceptable noise and transient response.

No well-defined boundary exists between stable and
unstable operation. As phase margin is reduced, the first
symptom is timing jitter, which shows up as blurred edges
in the switching waveforms where the scope does not quite
sync up. Technically speaking, this jitter (usually harmless)
is unstable operation since the duty factor varies slightly.
As capacitors with higher ESRs are used, the jitter
becomes more pronounced, and the load-transient output
voltage waveform starts looking ragged at the edges.
Eventually, the load-transient waveform has enough ringing
on it that the peak noise levels exceed the allowable output
voltage tolerance. Note that even with zero phase margin
and gross instability, the output voltage seldom declines
beyond I

PEAK

x R

ESR

(under constant loads).

Designers of RF communicators or other noise-sensi­tive analog equipment should be conservative and stay
within the guidelines. Designers of notebook computers
and similar commercial-temperature-range digital sys­tems can multiply the R

ESR

value by a factor of 1.5

without affecting stability or transient response.
The output voltage ripple, which is usually dominated by

the filter capacitor’s ESR, can be approximated as
I

RIPPLE

x R

ESR

. There is also a capacitive term, so the
full equation for ripple in continuous-conduction mode is
V

RIPPLE(p-p)

= I

RIPPLE

x [R

ESR

+ 1 / (2πƒ x C

OUT

)]. In
idle mode, the inductor current becomes discontinuous,
with high peaks and widely spaced pulses, so the noise
can actually be higher at light load (compared to full
load). In idle mode, calculate the output ripple as follows:

V

RIPPLE(p-p)

= (0.02 x R

ESR

/ R

SENSE

) + [0.0003 x L x

(1 / V

OUT

+ 1 / (VIN- V

OUT

)) / R

SENSE

2

x CF]

Selecting Other Components

MOSFET Switches

The high-current N-channel MOSFETs must be logic­level types with guaranteed on-resistance specifications
at VGS= 4.5V. Lower gate-threshold specifications are
better (i.e., 2V max rather than 3V max). Drain-source
breakdown voltage ratings must at least equal the maxi­mum input voltage, preferably with a 20% margin. The
best MOSFETs have the lowest on-resistance per
nanocoulomb of gate charge. Multiplying R

DS(ON)

by
Qg provides a good figure of merit for comparing vari­ous MOSFETs. Newer MOSFET process technologies
with dense cell structures generally perform best. The
internal gate drivers tolerate >100nC total gate charge,
but 70nC is a more practical upper limit to maintain best
switching times.

In high-current applications, MOSFET package power
dissipation often becomes a dominant design factor.
I2R power losses are the greatest heat contributor for
both high-side and low-side MOSFETs. I2R losses are
distributed between Q1 and Q2 according to duty fac­tor, as shown in the following equations. Generally,
switching losses affect only the upper MOSFET since
the Schottky rectifier usually clamps the switching node
before the synchronous rectifier turns on. Gate-charge
losses are dissipated by the driver and do not heat the
MOSFET. Calculate the temperature rise according to
package thermal-resistance specifications to ensure
that both MOSFETs are within their maximum junction
temperature at high ambient temperature. The worst­case dissipation for the high-side MOSFET occurs at
both extremes of input voltage, and the worst-case dis­sipation for the low-side MOSFET occurs at maximum
input voltage.

Duty = (V

OUT

+ VQ2) / (VIN- VQ1)

P

D (UPPER FET)

= I

LOAD

2

x R

DS(ON)

x duty + VINx

I

LOAD

x ƒ x [(VINx C

RSS

) / I

GATE

+ 20ns]

P

D (LOWER FET)

= I

LOAD

2

x R

DS(ON)

x (1 - duty)

where VQ= the on-state voltage drop (I

LOAD

x

R

DS(ON)

), C

RSS

= the MOSFET reverse transfer capaci-

tance, I

GATE

= the DH driver peak output current capa­bility (1A typ), and the DH driver inherent rise/fall time is
20ns. The MAX1637’s output undervoltage shutdown
function protects the synchronous rectifier under output
short-circuit conditions. To reduce EMI, add a 0.1µF
ceramic capacitor from the high-side switch drain to
the low-side switch source.

Rectifier Clamp Diode

The rectifier is a clamp across the low-side MOSFET
that catches the negative inductor swing during the
60ns dead time between turning one MOSFET off and
turning each low-side MOSFET on. The latest genera­tions of MOSFETs incorporate a high-speed silicon
body diode, which serves as an adequate clamp diode
if efficiency is not of primary importance. A Schottky
diode can be placed in parallel with the body diode to
reduce the forward voltage drop, typically improving
efficiency 1% to 2%. Use a diode with a DC current rat­ing equal to one-third of the load current; for example,
use an MBR0530 (500mA-rated) type for loads up to

1.5A, a 1N5819 type for loads up to 3A, or a 1N5822
type for loads up to 10A. The rectifier’s rated reverse­breakdown voltage must be at least equal to the maxi­mum input voltage, preferably with a 20% margin.

Boost-Supply Diode D2

A signal diode such as a 1N4148 works well in most
applications. Do not use large power diodes, such as
1N5817 or 1N4001.

Low-Voltage Operation

Low input voltages and low input-output differential volt­ages each require extra care in their design. Low
VIN-V

OUT

differentials can cause the output voltage to
sag when the load current changes abruptly. The sag’s
amplitude is a function of inductor value and maximum
duty factor (D

MAX

, an

Electrical Characteristics

parame­ter, 93% guaranteed over temperature at f = 200kHz) as
follows:

V

SAG

= [(I

STEP

)2x L] / [2CFx (V

IN(MIN)

x D

MAX

-

V

OUT

)]

Table 5 is a low-voltage troubleshooting guide. The
cure for low-voltage sag is to increase the output
capacitor’s value. For example, at VIN= 5.5V, V

OUT

=

5V, L = 10µH, ƒ = 200kHz, and I

STEP

= 3A, a total
capacitance of 660µF keeps the sag below 200mV.
Note that only the capacitance requirement increases;
the ESR requirements do not change. Therefore, the

added capacitance can be supplied by a low-cost bulk
capacitor in parallel with the normal low-ESR capacitor.

__________Applications Information

Heavy-Load Efficiency Considerations

The major efficiency-loss mechanisms under loads are
as follows, in the usual order of importance:

• P(I2R) = I2R losses

• P(tran) = transition losses

• P(gate) = gate-charge losses

• P(diode) = diode-conduction losses

• P(cap) = capacitor ESR losses

• P(IC) = losses due to the IC’s operating supply current

Inductor core losses are fairly low at heavy loads
because the inductor’s AC current component is small.
Therefore, these losses are not considered in this
analysis. Ferrite cores are preferred, especially at
300kHz, but powdered cores, such as Kool-Mu, can
also work well.

Efficiency = P

OUT

/ PINx 100%

= P

OUT

/ (P

OUT

+ P

TOTAL

) x 100%

P

TOTAL

= P(I2R) + P(tran) + P(gate) + P(diode) +

P(cap) + P(IC)

P = (I2R) = I

LOAD

2

x (RDC+ R

DS(ON)+RSENSE

)

where RDCis the DC resistance of the coil, R

DS(ON)

is

the MOSFET on-resistance, and R

SENSE

is the current-

sense resistor value. The R

DS(ON)

term assumes iden­tical MOSFETs for the high-side and low-side switches
because they time-share the inductor current. If the
MOSFETs are not identical, their losses can be estimat­ed by averaging the losses according to duty factor.

PD(tran) = transition loss = VINx I

LOAD

x ƒ x

[(VINC

RSS

/ I

GATE

) + 20ns]

where C

RSS

is the reverse transfer capacitance of the

high-side MOSFET (a data sheet parameter), I

GATE

is
the DH gate-driver peak output current (1.5A typ), and
the rise/fall time of the DH driver is typically 20ns.

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

______________________________________________________________________________________ 17

Table 5. Low-Voltage Troubleshooting Guide

Low VIN-V

OUT

differential,

under 1V

Low VIN-V

OUT

differential,

under 1.5V

Dropout voltage is
too high

Sag or droop in V

OUT

under step-load change

SYMPTOM

Maximum duty-cycle limits
exceeded

Limited inductor-current
slew rate per cycle

ROOT CAUSECONDITION

Reduce operation to 200kHz.
Reduce MOSFET on-resistance
and coil DC resistance.

Increase bulk output capacitance
per formula (see

Low-Voltage

Operation

section). Reduce

inductor value.

SOLUTION

MAX1637

P(gate) = Qgx ƒ x V

GG

where Qgis the sum of the gate-charge values for low­side and high-side switches. For matched MOSFETs,
Qgis twice the data-sheet value of an individual
MOSFET. Efficiency can usually be optimized by con­necting VGGto the most efficient 5V source, such as
the system +5V supply.

P(diode) = diode conduction losses = I

LOAD

x V

FWD

x tDx ƒ

where tDis the diode conduction time (120ns typ), and
V

FWD

is the diode forward voltage. This power is dissi­pated in the MOSFET body diode if no external
Schottky diode is used.

P(cap) = input capacitor ESR loss = I

RMS

2

x R

ESR

where I

RMS

is the input ripple current as calculated in

the

Input Capacitor Value

section.

Light-Load Efficiency Considerations

Under light loads, the PWM operates in discontinuous
mode. The inductor current discharges to zero at some
point during the charging cycle. This makes the induc­tor current’s AC component high compared to the load
current, which increases core losses and I2R losses in
the input-output filter capacitors. For best light-load effi­ciency, use MOSFETs with moderate gate-charge lev­els and use ferrite MPP or other low-loss core material.
Avoid powdered-iron cores; even Kool-Mu (aluminum
alloy) is not as desirable as ferrite.

Low-Noise Operation

Noise-sensitive applications such as hi-fidelity multi­media-equipped systems, cellular phones, RF commu­nicating computers, and electromagnetic pen-entry
systems should operate the controller in PWM mode
(SKIP = high). This mode forces a constant switching
frequency, reducing interference due to switching
noise by concentrating the radiated EM fields at a
known frequency outside the system audio or IF bands.
Choose an oscillator frequency for which switching­frequency harmonics do not overlap a sensitive fre­quency band. If necessary, synchronize the oscillator
to a tight-tolerance external clock generator.

Powering From a Single

Low-Voltage Supply

The circuit of Figure 7 is powered from a single 3.3V to

5.5V source and delivers 4A at 2.5V. At input voltages
of 3.15V, this circuit typically achieves efficiencies of
90% at 3.5A load currents. When using a single supply
to power both V

BATT

and V

BIAS

, be sure that it does not

exceed the 5.5V rating (6V absolute maximum) for V

GG

and VCC. Also, heavy current surges from the input
may cause transient dips on VCC. To prevent this, the
decoupling capacitor on VCCmay need to be
increased to 2µF or greater. This circuit uses low­threshold (specified at VGS= 2.7V) IRF7401 MOSFETs
which allow a typical startup of 3.15V at above 4A. Low
input voltages demand the use of larger input capaci­tors. Sanyo OS-CONs are recommended for their high
capacity and low ESR.

PC Board Layout Considerations

Good PC board layout is required to achieve specified
noise, efficiency, and stable performance. The PC
board layout artist must be given explicit instructions,
preferably a pencil sketch showing the placement of
power-switching components and high-current routing.
See the PC board layout in the MAX1637 evaluation kit
manual for examples. A ground plane is essential for
optimum performance. In most applications, the circuit
will be located on a multi-layer board, and full use of
the four or more copper layers is recommended. Use
the top layer for high-current connections, the bottom
layer for quiet connections (REF, CC, GND), and the
inner layers for an uninterrupted ground plane. Use the
following step-by-step guide:

1) Place the high-power components (C1, C2, Q1, Q2,
D1, L1, and R1) first, with their grounds adjacent.

•

Minimize current-sense resistor trace lengths

and
ensure accurate current sensing with Kelvin con­nections (Figure 8).

•

Minimize ground trace lengths

in the high-current

paths.

•

Minimize other trace lengths

in the high-current

paths.
— Use >5mm-wide traces.
— CIN to high-side MOSFET drain: 10mm

max length
— Rectifier diode cathode to low side
— MOSFET: 5mm max length
— LX node (MOSFETs, rectifier cathode, induc-

tor): 15mm max length

Ideally, surface-mount power components are butted
up to one another with their ground terminals almost
touching. These high-current grounds are then con­nected to each other with a wide, filled zone of
top-layer copper so they do not go through vias. The
resulting top-layer subground plane is connected to the
normal inner-layer ground plane at the output ground
terminals, which ensures that the IC’s analog ground is

Miniature, Low-Voltage,
Precision Step-Down Controller

18 ______________________________________________________________________________________

sensing at the supply’s output terminals without interfer­ence from IR drops and ground noise. Other high-cur­rent paths should also be minimized, but focusing
primarily on short ground and current-sense connec­tions eliminates about 90% of all PC board layout prob­lems (see the PC board layouts in the MAX1637
evaluation kit manual for examples).

2) Place the IC and signal components. Keep the main
switching nodes (LX nodes) away from sensitive
analog components (current-sense traces and REF
capacitor). Place the IC and analog components on
the opposite side of the board from the power­switching node. Important: The IC must be no fur­ther than 10mm from the current-sense resistors.
Keep the gate-drive traces (DH, DL, and BST) short­er than 20mm and route them away from CSH, CSL,
and REF. Place ceramic bypass capacitors close to
the IC. The bulk capacitors can be placed further
away.

3) Use a single-point star ground where the input
ground trace, power ground (subground plane), and

normal ground plane meet at the supply's output
ground terminal. Connect both IC ground pins and
all IC bypass capacitors to the normal ground plane.

MAX1637

Miniature, Low-Voltage,

Precision Step-Down Controller

______________________________________________________________________________________ 19

MAX1637

SENSE RESISTOR

HIGH-CURRENT PATH

Figure 8. Kelvin Connections for the Current-Sense Resistors

MAX1637

0.1µF

1µF

IRF7401

CMPSH-3

IRF7401

1µF

470pF

MBRS130

470µF
LOW ESR
TANTALUM

4.7µF
TANTALUM

220µF
OS-CON

V

BIAS

10µH

CDHR125-100

20mΩ

1%

130k
1%

100k
1%

OUTPUT = 2.5V AT 4A

V

CC

20Ω

GND CC

DL

LX

DH

BST

V

GG

CSH

CSL

FB

SHDN

ON/OFF

SKIP

SYNC

REF

PGND

3.15V TO 5.5V

Figure 7. 3.15V to 5.5V Single-Supply Application Circuit

MAX1637

Miniature, Low-Voltage,
Precision Step-Down Controller

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

20

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© 1998 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

___________________Chip Information

________________________________________________________Package Information

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