Rainbow Electronics MAX1625 User Manual

_______________General Description

The MAX1624/MAX1625 are ultra-high-performance,
step-down DC-DC controllers for CPU power in high-end
computer systems. Designed for demanding applications
in which output voltage precision and good transient
response are critical for proper operation, they deliver
over 35A from 1.1V to 3.5V with ±1% total accuracy from
a +5V ±10% supply. Excellent dynamic response cor­rects output transients caused by the latest dynamically
clocked CPUs. These controllers achieve over 90% effi­ciency by using synchronous rectification. Flying-capaci­tor bootstrap circuitry drives inexpensive, external
N-channel MOSFETs.

The switching frequency is resistor programmable from
100kHz to 1MHz. High switching frequencies allow the
use of a small surface-mount inductor and decrease out­put filter capacitor requirements, reducing board area
and system cost.

The MAX1624 is available in a 24-pin SSOP and offers
additional features such as a digitally programmable out­put in 100mV increments; adjustable transient response;
selectable 0.5%, 1%, or 2% AC load regulation; and gate
drive for a current-boost MOSFET. The MAX1625 is resis­tor adjustable and comes in a 16-pin narrow SO pack­age. Other features in both controllers include internal
digital soft-start, a power-good output, and a 3.5V ±1%
reference output. For a similar controller compatible with
the latest Intel V

RM/VID

specification, see the MAX1638*

data sheet.

________________________Applications

Pentium Pro™, Pentium II™, PowerPC™, Alpha™,
and K6™ Systems

Desktop Computers
LAN Servers
Industrial Computers
GTL Bus Termination

____________________________Features

♦ Better than ±1% Output Accuracy Over

Line and Load

♦ 90% Efficiency
♦ Excellent Transient Response
♦ Resistor-Programmable Fixed Switching

Frequency from 100kHz to 1MHz

♦ Over 35A Output Current
♦ Digitally Programmable Output from 1.1V to 3.5V

in 100mV Increments (MAX1624)

♦ Resistor-Adjustable Output down to 1.1V

(MAX1625)

♦ Remote Sensing
♦ Adjustable AC Loop Gain (MAX1624)
♦ GlitchCatcher™ Circuit for Fast Load-Transient

Response (MAX1624)

♦ Power-Good (PWROK) Output
♦ Current-Mode Feedback
♦ Digital Soft-Start
♦ Strong 2A Gate Drivers
♦ Current-Limited Output

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

________________________________________________________________

Maxim Integrated Products

1

19-1227; Rev 1; 6/97

PART

MAX1624EAG
MAX1625ESE

-40°C to +85°C

-40°C to +85°C

TEMP. RANGE PIN-PACKAGE

24 SSOP
16 Narrow SO

______________Ordering Information

__________Typical Operating Circuit

Pin Configurations appear at end of data sheet.

*

Future product.
Pentium Pro and Pentium II are trademarks of Intel Corp.

PowerPC is a trademark of IBM Corp.
Alpha is a trademark of Digital Equipment Corp.
K6 is a trademark of Advanced Micro Devices.
GlitchCatcher is a trademark of Maxim Integrated Products.

EVALUATION KIT

AVAILABLE

V

CC

FREQ
CC2
CC1

REF

AGND

(SIMPLIFIED)

FB

DL

DH

PWROK

BST

CSL

CSH

LX

TO V

DD

TO AGND

PGND

OUTPUT

1.1V TO 4.5V

INPUT

+5V

V

DD

N

N

MAX1625

For free samples & the latest literature: http://www.maxim-ic.com, or phone [phone].
For small orders, phone [phone] ext. 3468.

kHz

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VDD= V

CC

= D4 = +5V, PGND = AGND = D0–D3 = 0V, R

FREQ

= 33.3kΩ, TA= 0°C to +85°C, unless otherwise noted.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

VDD, VCC, PWROK to AGND......................................-0.3V to 6V

PGND to AGND ..................................................................±0.3V

CSH, CSL to AGND....................................-0.3V to (VCC+ 0.3V)

NDRV, PDRV, DL to PGND.........................-0.3V to (VDD+ 0.3V)

REF, CC1, CC2, LG, D0–D4, FREQ,

FB to AGND................................................-0.3V to (V

CC

+ 0.3V)

BST to PGND............................................................-0.3V to 12V

BST to LX....................................................................-0.3V to 6V

DH to LX.............................................(LX - 0.3V) to (BST + 0.3V)

Continuous Power Dissipation (TA= ±70°C)

24 Pin SSOP (derate 8.00mW/°C above +70°C) ..........640mW

16 Pin Narrow SO (derate 8.70mW/°C above 70°C).....696mW

Operating Temperature Range

MAX162_E_ _.......................................................-40°C to +85°C

Storage Temperature Range.............................-65°C to +125°C

Lead Temperature (soldering, 10sec).............................+300°C

R

FREQ

= 33.3kΩ

R

FREQ

= 20kΩ

PWROK = 5.5V

VCC= VDD= 5.5V,
FB overdrive = 200mV

I

SINK

= 2mA, VCC= 4.5V

Falling FB, 1% hysteresis with respect to V

REF

VCCrising edge, 1% hysteresis

Rising FB, 1% hysteresis with respect to V

REF

VCC= V

DD

MAX1624, over line
and load (Note 1)

V

REF

= 0V

Rising edge, 1% hysteresis

0µA < I

LOAD

< 100µA

VCC= VDD= 5.5V, FB overdrive = 200mV,
operating or standby mode

MAX1625, over line
and load (Note 2)

No load

CONDITIONS

540 600 660

850 1000 1150

Switching Frequency

µA1PWROK Output Current High

V0.4PWROK Output Voltage Low

6.5 8 9.5

%

-7.5 -6 -4.5

PWROK Trip Level

1

2

1

%

0.5

AC Load Regulation
(Note 3)

±1.5

%

±1

FB Set Voltage

±1.5

2.5

VCCSupply Current

V4.0 4.2

V4.5 5.5Input Voltage Range

Input Undervoltage Lockout

%

±1

FB Accuracy

mA0.5 4.0Reference Short-Circuit Current

V2.7 3.0Reference Undervoltage Lockout

mV10Reference Load Regulation

mA

0.3
mA0.1VDDSupply Current

V3.465 3.5 3.535Reference Voltage

UNITSMIN TYP MAXPARAMETER

Operating mode
Standby mode

TA= +25°C to +85°C
TA= 0°C to +85°C
TA= +25°C to +85°C
TA= 0°C to +85°C

CSH - CSL =
0mV to 80mV

R

FREQ

= 200kΩ

kHz

85 100 115

LG = REF

LG = GND

LG = V

CC

MAX1624

MAX1625

LG = REF

CSH - CSL =
0mV to 80mV

LG = GND

LG = V

CC

0.1

MAX1624

MAX1625

0.2

0.1
%

0.05

DC Load Regulation
(Note 3)

ELECTRICAL CHARACTERISTICS (continued)

(VDD= V

CC

= D4 = +5V, PGND = AGND = D0–D3 = 0V, R

FREQ

= 33.3kΩ, TA= 0°C to +85°C, unless otherwise noted.)

MAX1624/MAX1625

_______________________________________________________________________________________

3

DH = DL = 2.5V

VDD= 4.5V

BST - LX = 4.5V

LG = GND (low)

100mV overdrive

R

FREQ

= 20kΩ

With respect to V

REF

,

FB going low

Minimum

MAX1625, CSH = CSL = 1.1V

D0–D4 = 0V, 5V

D0–D4; VCC= 4.5V

LG = REF (mid)

MAX1624, CSH = CSL = 1.3V,
D0–D3 = 5V, D4 = 0V

D0–D4; VCC= 5.5V

CONDITIONS

-2.75 -2 -1.25

LG = V

CC

(high)

ns0 30

FB = 1.1V

DH, DL Dead Time

A2DH, DL Source/Sink Current

Ω

0.7 2

Maximum

DH On-Resistance 0.7 2

%

-3 -1

µA100CC2 Source/Sink Current

4 V

CC

V

2.4 3.0

mmho1

kΩ10CC1 Output Resistance

µA±0.1

50

3.3 3.7

0.2

%85 90Maximum Duty Cycle

LG Input Voltage

µA

50

CSH, CSL Input Current

µA4LG Input Current

µA±1D0–D4 Input Current

V2.0Logic Input Voltage High

VCC- 0.2

V

0.8Logic Input Voltage Low

UNITSMIN TYP MAXPARAMETER

FB Input Current

CC2 Clamp Voltage

CC2 Transconductance

PDRV Trip Level

PDRV, NDRV Response Time FB overdrive = 5% ns75
PDRV, NDRV On-Resistance VDD= 4.5V Ω2 5
PDRV, NDRV Source/Sink Current PDRV = NDRV = 2.5V A0.5
PDRV, NDRV Minimum On-Time ns100
Current-Limit Trip Voltage mV85 100 115
Soft-Start Time To full current limit 1 / f

OSC

1536

BST Leakage Current BST = 12V, LX = 7V, REF = GND µA50

V

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

ΩDL On-Resistance

NDRV Trip Level

With respect to V

REF

,

FB going high

1.25 2 2.75
%

1 3

TA= +25°C
TA= 0°C to +85°C
TA= +25°C
TA= 0°C to +85°C

mA

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

4 _______________________________________________________________________________________

R

FREQ

= 33.3kΩ

R

FREQ

= 20kΩ

VCC= VDD= 5.5V,
FB overdrive = 200mV

Falling FB, 1% hysteresis with respect to V

REF

VCCrising edge, 1% hysteresis

Rising FB, 1% hysteresis with respect to V

REF

VCC= V

DD

R

FREQ

= 200kΩ

Operating mode

VCC= VDD= 5.5V, FB overdrive = 200mV,
operating or standby mode

MAX1624, over line and load

No load

CONDITIONS

510 600 690

Standby mode

800 1000 1200

Switching Frequency

6 8 10

kHz

80 100 120

%

-8 -6 -4

PWROK Trip Level

±2.5

3

VCCSupply Current

V3.9 4.3

V4.5 5.5Input Voltage Range

Input Undervoltage Lockout

%±2.5FB Accuracy

mA

0.4
mA0.2VDDSupply Current

V3.447 3.5 3.553Reference Voltage

UNITSMIN TYP MAXPARAMETER

MAX1625FB Set Voltage

BST - LX = 4.5V

R

FREQ

= 20kΩ

VDD= 4.5V

Ω0.7 2

%84 90

Ω0.7 2

Maximum Duty Cycle

DL On-Resistance

DH On-Resistance

Current-Limit Trip Voltage mV70 100 130

ELECTRICAL CHARACTERISTICS

(VDD= VCC= D4 = +5V, PGND = AGND = D0–D3= 0V, R

FREQ

= 33.3kΩ, TA= -40°C to +85°C, unless otherwise noted.) (Note 4)

Note 1: FB accuracy is 100% tested at FB = 3.5V (code 10000) with V

CC

= VDD= 4.5V to 5.5V and CSH - CSL = 0mV to 80mV. The

other DAC codes are tested at the major transition points with V

CC

= VDD= 5V and CSH - CSL = 0. FB accuracy at other

DAC codes over line and load is guaranteed by design.

Note 2: FB set voltage is 100% tested with VCC= VDD= 4.5V to 5.5V and CSH - CSL = 0mV to 80mV.
Note 3: AC load regulation sets the AC loop gain, to make tradeoffs between output filter capacitor size and transient response,

and has only a slight effect on DC accuracy or DC load-regulation error.

Note 4: Specifications from 0°C to -40°C are not production tested.

%

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

_______________________________________________________________________________________ 5

__________________________________________Typical Operating Characteristics

(TA = +25°C, using the MAX1624 evaluation kit, unless otherwise noted.)

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE DETAIL

(WITH GLITCHCATCHER)

(1.1V)

C

D

MAX1624/25 TOC01

A: PDRV, 5V/div
B: V

OUT

, 50mV/div, AC COUPLED
C: NDRV, 5V/div
D: LOAD CURRENT, 0A TO 10A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITH GLITCHCATCHER)

(1.1V )

C

MAX1624/25 TOC02

A: V

OUT

, 50mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 10A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITHOUT GLITCHCATCHER)

(1.1V)

C

MAX1624/25 TOC03

A: V

OUT

, 50mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 10A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITHOUT GLITCHCATCHER)

(3.5V)

C

MAX1624/25 TOC15

A: V

OUT

, 100mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 11A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITH GLITCHCATCHER)

(2.5V)

C

MAX1624/25 TOC17

A: V

OUT

, 50mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 10A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITHOUT GLITCHCATCHER)

(2.5V)

C

MAX1624/25 TOC18

A: V

OUT

, 50mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 10A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

10µs/div

MAX1624

LOAD-TRANSIENT RESPONSE

(WITH GLITCHCATCHER)

(3.5V)

C

MAX1624/25 TOC16

A: V

OUT

, 100mV/div, AC COUPLED
B: INDUCTOR CURRENT, 10A/div
C: LOAD CURRENT, 0A TO 11A, t

RISE

= t

FALL

= 100ns

B

A

LG = REF

1µs/div

MAX1624

SWITCHING WAVEFORMS

C

0

MAX1624/25 TOC10

VIN = 5V, V

OUT

= 2.5V, LOAD = 5A
A: LX, 5V/div
B: V

OUT

, 20mV/div, AC COUPLED

C: INDUCTOR CURRENT, 5A/div

B

A

1ms/div

MAX1624

STARTUP AND STANDBY RESPONSE

C

MAX1624/25 TOC11

VIN = 5V, V

OUT

= 2.5V, LOAD = 13.8A

A: V

OUT

, 1V/div
B: INDUCTOR CURRENT, 10A/div
C: STANDBY, D0–D4

B

A

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

6 _______________________________________________________________________________________

____________________________Typical Operating Characteristics (continued)

(TA = +25°C, using the MAX1624 evaluation kit, unless otherwise noted.)

100

0

0.1 1 10

MAX1624

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT

= 1.1V)

20
10

MAX1624/25 TOC04

OUTPUT CURRENT (A)

EFFICIENCY (%)

40
30

60

70

50

80

90

100

0

0.1 1 10

MAX1624

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT

= 2.5V)

20
10

MAX1624/25 TOC05

OUTPUT CURRENT (A)

EFFICIENCY (%)

40
30

60

70

50

80

90

100

0

0.1 1 10

MAX1624

EFFICIENCY vs. OUTPUT CURRENT

(V

OUT

= 3.5V)

20
10

MAX1624/25 TOC06

OUTPUT CURRENT (A)

EFFICIENCY (%)

40
30

60

70

50

80

90

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

_______________________________________________________________________________________ 7

____________________________Typical Operating Characteristics (continued)

(TA = +25°C, using the MAX1624 evaluation kit, unless otherwise noted.)

1.1020

1.1000

0.1 10.01 10

MAX1624

OUTPUT VOLTAGE vs. OUTPUT CURRENT

(V

OUT

= 1.1V)

1.1002

1.1006

1.1004

MAX1624/25 TOC07

OUTPUT CURRENT (A)

OUTPUT VOLTAGE (V)

1.1008

1.1010

1.1014

1.1012

1.1016

1.1018

LG = V

CC

LG = REF

LG = AGND

R9 AND R10 = 4.7Ω

2.500

2.490

0.1 10.01 10

MAX1624

OUTPUT VOLTAGE vs. OUTPUT CURRENT

(V

OUT

= 2.5V)

2.491

2.492

MAX1624/25 TOC08

OUTPUT CURRENT (A)

OUTPUT VOLTAGE (V)

2.493

2.494

2.496

2.495

2.497

2.498

2.499

LG = V

CC

LG = REF

LG = AGND

R9 AND R10 = 4.7Ω

3.500

3.480

0.1 10.01 10

MAX1624

OUTPUT VOLTAGE vs. OUTPUT CURRENT

(V

OUT

= 3.5V)

3.482

3.486

3.484

MAX1624/25 TOC09

OUTPUT CURRENT (A)

OUTPUT VOLTAGE (V)

3.488

3.494

3.490

3.492

3.496

3.498

LG = V

CC

LG = REF

LG = AGND

R9 AND R10 = 4.7Ω

5.094

1.094

0.001 0.1 10.01 10

REFERENCE VOLTAGE
vs. OUTPUT CURRENT

1.594

2.094

MAX1624/25 TOC12

OUTPUT CURRENT (mA)

REFERENCE VOLTAGE (V)

2.594

3.594

3.094

4.094

4.594

SOURCING

CURRENT

SINKING

CURRENT

50

55

0

MAXIMUM DUTY CYCLE

vs. SWITCHING FREQUENCY

65
60

70

MAX1624/25 tTOC13

SWITCHING FREQUENCY (kHz)

MAXIMUM DUTY CYCLE (%)

85

95
90

75

80

200 800 1000 1200

100

600400

10

-10

1.7 2.3 2.91.1 3.5

MAX1624

OUTPUT ERROR vs.

DAC OUTPUT VOLTAGE SETTING

-8

-4

-6

MAX1624/25 TOC19

DAC OUTPUT VOLTAGE SETTING (V)

OUTPUT ERROR (mV)

-2

4

0

2

6

8

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

8 _______________________________________________________________________________________

MAX1625MAX1624

PIN

High-Side Main MOSFET Switch Gate-Drive Output. DH is a floating driver output that
swings from LX to BST, riding on the LX switching-node voltage. See the section

BST

High-Side Gate-Driver Supply and MOSFET Drivers

.

DH1624

Switching Node. Connect LX to the high-side MOSFET source and inductor.LX1523

Power GroundPGND1422

DL

Low-Side Synchronous Rectifier Gate-Drive Output. DL swings between PGND and VDD.
See the section

BST High-Side Gate-Driver Supply and MOSFET Drivers

.

1321

V

DD

5V Power Input for MOSFET Drivers. Bypass VDDto PGND within 0.2 in. (5mm) of the
VDDpin using a 0.1µF capacitor and 4.7µF capacitor connected in parallel.

1220

PDRV GlitchCatcher P-Channel MOSFET Driver Output. PDRV swings between VDDand PGND. —19

NDRV

GlitchCatcher N-Channel MOSFET Driver Output. NDRV swings between VDDand
PGND.

—18

D4, D3 Digital Inputs for Programming the Output Voltage —16, 17

FREQ

Frequency-Programming Input. Attach a resistor within 0.2 in. (5mm) of FREQ to AGND to
set the switching frequency between 100kHz and 1MHz. The FREQ pin is normally 2V DC.

1115

CC2

Slow-Loop Compensation Capacitor Input. Connect a ceramic capacitor from CC2 to
AGND. See the section

Compensating the Feedback Loop.

1014

BST

Boost-Capacitor Bypass for High-Side MOSFET Gate Drive. Connect a 0.1µF capacitor
and low-leakage Schottky diode as a bootstrapped charge-pump circuit to derive a 5V
gate drive from V

DD

for DH.

11

NAME FUNCTION

______________________________________________________________Pin Description

CC1

Fast-Loop Compensation Capacitor Input. Connect a ceramic capacitor and resistor in
series from CC1 to AGND. See the section

Compensating the Feedback Loop

.

913

FB

Voltage-Feedback Input.
MAX1624: Connect FB to the CPU’s remote voltage-sense point. The voltage at this
input is regulated to a value determined by D0–D4.
MAX1625: Connect a feedback resistor voltage divider close to FB from the output to
AGND. FB is regulated to 1.1V.

812

PWROK

Open-Drain Logic Output. PWROK is high when the voltage on FB is within +8% and -6%
of its setpoint.

22

CSL

Current-Sense Amplifier’s Inverting Input. Place the current-sense resistor very close to
the controller IC, and use a Kelvin connection. Use an RC filter network at CSL (Figure 1).

33

CSH Current-Sense Amplifier’s Noninverting Input. Use an RC filter network at CSH (Figure 1).44

D2, D1,D0Digital Inputs for Programming the Output Voltage. D0–D4 are logic inputs that set the

output to a voltage between 1.1V and 3.5V in 100mV increments.

—5, 6, 7

LG

Loop Gain-Control Input. LG is a three-level input that is used to trade off loop gain vs.
AC load-regulation and load-transient response. Connect LG to VCC, REF, or AGND for
2%, 1%, or 0.5% AC load-regulation errors, respectively.

—8

V

CC

Analog Supply Input, 5V. Use an RC filter network, as shown in Figure 1. 59

REF

Reference Output, 3.5V. Bypass REF to AGND with 0.1µF (min). Sources up to 100µA for
external loads. Force REF below 2V to turn off the controller.

610

AGND Analog Ground711

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

_______________________________________________________________________________________ 9

N1

R1

N2

C2

D1
(OPTIONAL)

R9

(OPTIONAL)

R10

(OPTIONAL)

R8

39Ω

C12

4.7nF

C11

4.7nF

V

CC

V

DD

CSH

PWROK

CSL

BST

DH

LX

DL

PGND

FB

PDRV

NDRV

AGND

REF

CC1

CC2

CC2

CC1

RC1

TO

AGND

C6, 1.0µF
CERAMIC

R4, 40.1k

FOR 500kHz

R5

100k

C9

0.1µFC74.7µF

R6

100Ω

TO V

DD

FREQ

D0
D1
D2
D3
D4
LG

REF

C5

0.1µFC84.7µF

D2
CMPSH-3

C4

0.1µF

L1

V

IN

= 5V

C1

R7

39Ω

LOCAL
BYPASSING

MAX1624

P1

R11

V

OUT

= 1.1V

TO 3.5V

N3

LOAD

Figure 1. MAX1624 Standard Application Circuit

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

10 ______________________________________________________________________________________

LOCAL
BYPASSING

N1

R1

N2

C2

R3

100kΩ

R2

200kΩ

C10

(OPTIONAL)

D1
(OPTIONAL)

R9

(OPTIONAL)

R10

(OPTIONAL)

C12

4.7nF

C11

4.7nF

V

CC

V

DD

CSH

PWROK

CSL

BST

DH

LX

DL

PGND

FB

AGND

REF

CC1

CC2

CC2

CC1

RC1

TO

AGND

C6, 1.0µF

CERAMIC

R4, 40.1kΩ
FOR 500kHz

R5

100kΩ

C9

0.1µF

C7

4.7µF

R6

100Ω

TO V

DD

FREQ

C5

0.1µF

C8

4.7µF

D2
CMPSH-3

C4

0.1µF

V

IN

= 5V

C1

R7

39Ω

R8

39Ω

LOAD

V

OUT

L1

MAX1625

Figure 2. MAX1625 Standard Application Circuit

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 11

Table 1. Component List for Standard 3.3V Applications by Load Current*
(Output Voltage = 3.3V, Frequency = 500kHz)

*

MAX1624: LG = REF, D4–D0 = 10010.

C10 Capacitor
CC1 Capacitor

D2 Rectifier

L1 Inductor

R2 Resistor
R3 Resistor

Application Equipment

R11 Resistor (MAX1624)

C1 Input Capacitor

100µF, 10V Sanyo
OS-CON 10SL100M

Optional (see text)
680pF ceramic

1.5µH, 8A Coiltronics UP2-1R5

200kΩ, 1% resistor
100kΩ, 1% resistor

Power PC/Pentium/GTL
bus termination

3 x 100µF, 10V Sanyo
OS-CON 10SL100M

1000pF ceramic

0.5µH, 17A Coilcraft
DO5022P-501HC

N/A
N/A

Pentium Pro

500mΩ Dale WSL-2512-R500

C2 Output Capacitor

2 x 220µF, 4V Sanyo
OS-CON 4SP220M

3 x 220µF, 4V Sanyo
OS-CON 4SP220M

0.056µF ceramic 0.056µF ceramic CC2 Capacitor
Optional Schottky,

Nihon
NSQ03A02

Optional Schottky,
Nihon
NSQ03A02

D1 Rectifier

1kΩ, 5% resistor 1kΩ, 5% resistorRC1 Resistor

International Rectifier IRF7413

International Rectifier
IRL3103S, D2PAK

N1 High-Side MOSFET

International Rectifier IRF7413

International Rectifier
IRL3103S, D2PAK

International Rectifier IRF7107N3/P1 (MAX1624)

N2 Low-Side MOSFET

12mΩ Dale WSL-2512-R012-F

2 x 12mΩ in parallel,
Dale WSL-2512-R012-F

R1 Resistor

6A 12A

COMPONENT

3 x 2700µF, 6.3V
aluminum electrolytic,
Sanyo 6MV2700GX

1000pF ceramic

Central Semiconductor
CMPSH-3

0.5µH Coiltronics UP4-R47,
Coilcraft DO5022P-501HC

N/A
N/A

Pentium Pro

N/A

DESCRIPTION BY LOAD CURRENT

4 x 2700µF, 6.3V
aluminum electrolytic,
Sanyo 6MV2700GX

0.056µF ceramic
Optional Schottky,

Nihon
NSQ03A02

11A

(LOW-COST VRM)

1kΩ, 5% resistor

International Rectifier IRF7413

x 2

International Rectifier IRF7413

x 2

2 x 12mΩ in parallel
Dale WSL-2512-R012-F

Central Semiconductor
CMPSH-3

Central Semiconductor
CMPSH-3

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

12 ______________________________________________________________________________________

AVX [phone] [phone]

Coilcraft [phone] [phone]

Dale Inductors [phone] [phone]

Coiltronics [phone] [phone]

International
Rectifier

[phone] [phone]

Central
Semiconductor

[phone] [phone]

IRC [phone] [phone]
Matsuo [phone] [phone]
Motorola [phone] [phone]
Murata-Erie [phone] [phone]
Nichicon [phone] [phone]
NIEC [phone]* [81] 3-3494-7414
Sanyo [phone] [81] 7-2070-1174
Siliconix [phone] [phone]

SUPPLIER USA PHONE FACTORY FAX

Sprague [phone] [phone]
Sumida [phone] [81] 3-3607-5144

*

Distributor

†

See Table 4 for a complete listing.

D4 D2 D0

1 0 1

1 1 0

1 — —

1 1 1

1 0 0

0 0 0
0 0 1

0 — —

0 1 1
0 0 0
0 — —
0 1 0
0 1 1

D3

0

1

—

1

0

0
0

0

0
1
1
1
1

D1

0

1

—

1

0

0
0

—

1
0

—

1
1

OUTPUT

VOLTAGE

(V)

3.4

2.1

Decreases

in 100mV

increments

No CPU (OFF)

3.5

1.9

1.8

Decreases

in 100mV

increments

1.2

1.1

1.1

1.1

No CPU (OFF)

COMPATIBILITY

Intel-compatible

codes

Non-Intel

compatible codes

Table 2. Component Suppliers

Table 3. MAX1624 Output Voltage
Adjustment Settings (Abbreviated†)

_____Standard Application Circuits

The predesigned MAX1624/MAX1625 circuits shown in
Figures 1 and 2 meet a wide range of applications with
output currents up to 12A and higher. Use Table 1 to
select components appropriate for the desired output
current range, and adapt the evaluation kit PC board
layout as necessary. Table 2 lists suggested vendors.
These circuits represent a good set of trade-offs
between cost, size, and efficiency while staying within
the worst-case specification limits for stress-related
parameters, such as capacitor ripple current.

These MAX1624/MAX1625 circuits were designed for
the specified frequencies. Do not change the switching
frequency without first recalculating component val­ues—particularly the inductance, output filter capaci­tance, and RC1 resistance values. Recalculate the
voltage-feedback resistor and compensation-capacitor
values (CC1 and CC2) as necessary to reconfigure
them for different output voltages. Table 3 lists voltage
adjustment DAC codes for the MAX1624.

_______________Detailed Description

The MAX1624/MAX1625 are BiCMOS switch-mode,
power-supply controllers designed for buck-topology
regulators. They are optimized for powering the latest
high-performance CPUs—demanding applications
where output voltage precision, good transient
response, and high efficiency are critical for proper
operation. With appropriate external components, the
MAX1624/MAX1625 deliver over 15A between 1.1V and

3.5V with ±1% accuracy. The MAX1625 offers 1% typi­cal transient-load regulation from a +5V supply, while the
MAX1624 offers a selectable transient-load regulation of

0.5%, 1%, or 2%. Remote output sensing ensures volt­age precision by eliminating errors caused by PC board
trace impedance. These controllers achieve 90% effi­ciency by using synchronous rectification.

A typical application circuit consists of two N-channel
MOSFETs, a rectifier, and an LC output filter (Figure 1).
At each of the internal oscillator’s rising edges, the
high-side MOSFET switch (N1) is turned on and allows
current to ramp up through the inductor to the output
filter capacitor and load, storing energy in a magnetic
field. The current is monitored by reading the voltage

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 13

across the current-sense resistor (R1). When the induc­tor current ramps up to the current-sense threshold, the
MOSFET turns off and interrupts the flow of current from
the supply. This causes the magnetic field in the induc­tor to collapse, resulting in a voltage surge that forces
the rectifier diode (D1) or MOSFET body diode (N2) on
and keeps the inductor current flowing in the same
amplitude and direction. At this point, the synchronous
rectifier MOSFET turns on until the end of the cycle to
reduce conduction losses across the rectifier diode.
The current through the inductor ramps back down,

transferring the stored energy to the output filter capac­itor and load. The output filter capacitor stores energy
when inductor current is high and releases it when
inductor current is low, smoothing the voltage delivered
to the load.

The MAX1624/MAX1625 use a current-mode pulse­width-modulation (PWM) control scheme (Figures 3
and 4). The output voltage is regulated by switching at
a constant frequency and then modulating the peak
inductor current to change the energy transferred per
pulse and to adjust to changes in the load. The output

+

-

+

-

REF

REF4

AGND

REF3

REF2

REF

FB D0–D4 PWROK PDRV NDRV

CC2

CC1

REF1

5

N

10k

40k

WINDOW

CONTROL AND

DRIVE LOGIC

OSCILLATOR

SLOPE

COMPENSATION

REF

AGND

V

CC

FREQ

REF

REF1 REF2 REF3 REF4

CSL

CSH
LG

BST

DH

LX

V

DD

DL

RESET

Q

Q

SET

PGND

MAX1624

Figure 3. MAX1624 Simplified Block Diagram

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

14 ______________________________________________________________________________________

voltage is the average of the AC voltage at the switching
node, which is adjusted and regulated by changing the
duty cycle of the MOSFET switches. Slope compensa­tion is necessary to stabilize current-mode feedback
controllers with a duty cycle greater than 50%. Maximum
duty cycle is greater than 85% (see

Typical Operating

Characteristics

).

PWM Controller Block and Integrator

The heart of the current-mode PWM controller is a
multi-input open-loop comparator that sums three sig­nals: the buffered feedback signal, the current-sense

signal, and the slope-compensation ramp. This direct­summing configuration approaches ideal cycle-by­cycle control over the output voltage. The output
voltage error signal is generated by an error amplifier
that compares the amplified feedback voltage to an
internal reference.

Each pulse from the oscillator sets the main PWM latch
that turns on the high-side switch for a period deter­mined by the duty factor (approximately V

OUT

/ VIN).
The current-mode feedback system regulates the peak
inductor current as a function of the output voltage

REF

REF2

N

FB

REF

CC2

CC1

PWROK

REF1

10k

40k

WINDOW

CONTROL AND

DRIVE LOGIC

OSCILLATOR

SLOPE

COMPENSATION

REF

AGND

V

CC

FREQ

REF

REF1 REF2

CSL

CSH

BST

DH

LX

V

DD

RESET

Q

Q

SET

DL

PGND

MAX1625

Figure 4. MAX1625 Simplified Block Diagram

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 15

error signal. Since average inductor current is nearly
the same as peak current (assuming the inductor value
is set relatively high to minimize ripple current), the cir­cuit acts as a switch-mode transconductance amplifier.
It pushes the second output LC filter pole, normally
found in a duty-factor-controlled (voltage-mode) PWM,
to a higher frequency. To preserve inner-loop stability
and eliminate regenerative inductor current staircasing,
a slope-compensation ramp is summed into the main
PWM comparator. As the high-side switch turns off, the
synchronous rectifier latch is set. The low-side switch
turns on 30ns later and stays on until the beginning of
the next clock cycle. Under fault conditions where the
inductor current exceeds the maximum current-limit
threshold, the high-side latch resets, and the high-side
switch turns off.

Internal Reference

The internal 3.5V reference (REF) is accurate to ±1%
from 0°C to +85°C, making REF useful as a system ref­erence. Bypass REF to AGND with a 0.1µF (min)
ceramic capacitor. A larger value (such as 1µF) is rec­ommended for high-current applications. Load regula­tion is 10mV for loads up to 100µA. Loading REF
reduces the main output voltage slightly, according to
the reference-voltage load-regulation error (see

Typical

Operating Characteristics

). Reference undervoltage
lockout is between 2.7V and 3V. Short-circuit current is
less than 4mA.

Synchronous-Rectifier Driver

Synchronous rectification reduces conduction losses in
the rectifier by shunting the normal Schottky diode or
MOSFET body diode with a low-on-resistance MOSFET
switch. The synchronous rectifier also ensures proper
start-up by precharging the boost-charge pump used
for the high-side switch gate-drive circuit. Thus, if you
must omit the synchronous power MOSFET for cost or
other reasons, replace it with a small-signal MOSFET,
such as a 2N7002.

The DL drive waveform is simply the complement of the
DH high-side drive waveform (with typical controlled
dead time of 30ns to prevent cross-conduction or
shoot-through). The DL output’s on-resistance is 0.7Ω
(typ) and 2Ω (max).

BST High-Side Gate-Driver Supply

and MOSFET Drivers

Gate-drive voltage for the high-side N-channel switch is
generated using a flying-capacitor boost circuit (Fig­ure 5). The capacitor is alternately charged from the
+5V supply and placed in parallel with the high-side
MOSFET’s gate and source terminals.

On start-up, the synchronous rectifier (low-side
MOSFET) forces LX to 0V and precharges the BST
capacitor (C4) to 5V through a diode (D2). This pro­vides the necessary enhancement voltage to turn on
the high-side switch. On the next half-cycle, the PWM
control logic turns on the high-side MOSFET by closing
an internal switch between BST and DH. As the MOS­FET turns on, the LX node rises to the input voltage, an
action that boosts the 5V gate-drive signal above the
+5V supply. DH on-resistance is 0.7Ω (typical) and 2Ω
(max). Do not bias D2 with voltages greater than 5.5V,
as this will destroy the DH gate driver.

A 0.1µF minimum ceramic capacitor is recommended for
the boost supply. Use a low-power, SOT23 Schottky
diode to minimize reduction of the gate drive from the
diode’s forward voltage. Use a low-leakage Schottky
diode, such as a CMPSH-3 from Central Semiconductor
or a 1N4148, to prevent reverse leakage from discharg­ing the BST capacitor when the ambient temperature is
high. Place the BST capacitor and diode within 0.4 in.
(10mm) of the BST pin.

Gate-drive resistors (R9 and R10) can often be useful
to reduce jitter in the switching waveforms by slowing
down the fast-slewing LX node and reducing ground
bounce at the controller IC. Low-valued resistors from
around 1Ω to 5Ω are sufficient for many applications.

C4

D2

V

IN

= 5V

V

DD

N1

R10

DH

LEVEL

TRANSLATOR

CONTROL AND

DRIVE LOGIC

N2

R9

PGND

R9 AND R10
ARE OPTIONAL

LX

DL

BST

MAX1624
MAX1625

Figure 5. Boost Supply for Gate Drivers

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

16 ______________________________________________________________________________________

GlitchCatcher

Current-Boost Driver (MAX1624)

Drivers for an optional current-boost circuit are includ­ed in the MAX1624 to improve transient response.
Some dynamically clocked CPUs switch computational
blocks on and off as needed to reduce power con­sumption, and can generate load steps of several
amperes in a few tens of nanoseconds. The current­boost circuit is intended to improve transient response
to such load steps by bypassing the inductor’s lowpass
filter operation. When the output drops out of regulation
by more than ±1.5% to ±2.5%, the P-channel or
N-channel switches turn on and force the output back
into regulation. The MOSFET drivers’ response time is
typically 75ns, and their minimum on-time is typically
100ns.

Current Sense

and Overload Current Limiting

The current-sense circuit resets the main PWM latch
and turns off the high-side MOSFET switch whenever
the voltage difference between CSH and CSL from cur­rent through the sense resistor (R1) exceeds the peak
current limit (100mV typical).

Current-mode control offers a practical level of over­load protection in response to many fault conditions.
During normal operation, maximum output current is
enforced by the peak current limit. If the output is short­ed directly to GND through a low-resistance path, the
current-sense comparator may be unable to enforce a
current limit. Under such conditions, circuit parasitics
such as MOSFET R

DS(ON)

typically limit the short­circuit current to a value around the peak-current-limit
setting.

Attach a lowpass-filter network between the current­sense pins and resistor to reduce high-frequency com­mon-mode noise (Figure 6). The filter should be
designed with a time constant of around 200ns.
Resistors in the 20Ω to 100Ω range are recommended
for R7 and R8. Connect the filter capacitors C11 and
C12 from VCCto CSH and CSL, respectively.

Values of 39Ω and 4.7nF are suitable for many designs.
Place the current-sense filter network close to the IC,
within 0.1 in. (2.5mm) of the CSH and CSL pins.

Internal Soft-Start

Soft-start allows a gradual increase of the internal cur­rent limit at start-up to reduce input surge currents. In
the MAX1624/MAX1625, an internal DAC raises the cur­rent-limit threshold from 0V to 100mV in four steps
(25mV, 50mV, 75mV, and 100mV) over the span of
1536 oscillator cycles.

__________________Design Procedure

Setting the Output Voltage

MAX1624

Select the output voltage using the D0–D4 pins. The
MAX1624 uses an internal 5-bit DAC as a feedback­resistor voltage divider. The output voltage can be digi­tally set in 100mV increments from 1.1V to 3.5V using
the D0–D4 inputs (Table 4).

D0–D4 are logic inputs and accept both TTL and
CMOS voltage levels. The MAX1624 has both FB and
AGND inputs, allowing a Kelvin connection for remote
voltage and ground sensing to eliminate the effects of
trace resistance on the feedback voltage. (See

PC

Board Layout Considerations

for further details.) FB

input current is 0.1µA (max).
The MAX1624 DAC codes were designed for compati-

bility with Intel specifications for output voltages
between 3.5V and 2.1V. Codes 10000 through 11110
are compatible with Intel specifications, while codes
00000 through 01111 are not. Codes 11111 and 01111
turn off the buck controller, placing the IC in a low­current mode (0.2mA typical). For compatibility with
Intel codes for output voltages below 2.1V, see the
MAX1638/MAX1639 data sheet.

C12

4.7nF

R1

C11

4.7nF

R7

39Ω

R6

100Ω

R8

39Ω

CSH

V

CC

CSL

MAX1624
MAX1625

N1

C9

0.1µF

C7

4.7µF

C1

V

IN

Figure 6. Current-Sense Filter

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 17

MAX1625

Set the output voltage by connecting R2 and R3 (Fig­ure 7) to the FB pin from the output to AGND. R2 is
given by the following equation:

where VFB= 1.1V. Since the input bias current at FB
has a maximum value of ±0.1µA, values up to 100kΩ
can be used for R3 with no significant accuracy loss.

Values under 1kΩ are recommended to improve noise
immunity and minimize parasitic capacitance at the FB
node. Place R2 and R3 very close to the MAX1625,
within 0.2 in. (5mm) of the FB pin.

Selecting the Oscillator Frequency

Set the switching frequency between 100kHz and
1MHz by connecting a resistor from FREQ to AGND.
Select the resistor according to the following equation:

Low-frequency operation reduces controller IC quies­cent current and improves efficiency. High-frequency
operation reduces cost and PC board area by allowing
the use of smaller inductors and fewer and smaller out­put capacitors. Inductor energy-storage requirements
and output capacitor requirements at 1MHz are one­third those at 300kHz.

Choosing the

Error-Amplifier Gain (MAX1624)

Set the error-amplifier gain to match the voltage-preci­sion requirements of the CPU used. The MAX1624’s
loop-gain control input (LG) allows trade-offs in DC/AC
voltage accuracy versus output filter capacitor require­ments. AC load regulation can be set to 0.5%, 1%,
or 2% by connecting LG as shown in Table 5. The
MAX1625’s default AC regulation is 1%.

DC load regulation is typically 10 times better than AC
load regulation, and is determined by the gain set by
the LG pin.

Specifying the Inductor

Three key inductor parameters must be specified:
inductance value (L), peak current (I

PEAK

), and DC
resistance (RDC). The following equation includes a
constant LIR, which is the ratio of inductor peak-to­peak AC current to DC load current. A higher LIR
value allows for smaller inductors and better transient
response, but results in higher losses and output ripple.

R

x

f

OSC

4

2 10

=

10

R R x

V

V

OUT

FB

2 3 1 = −











AC LOAD-

REGULATION

ERROR

(%)

1

LG

CONNECTED

TO

DC LOAD-

REGULATION

ERROR

(%)

REF 0.1

GND 0.05

V

CC

0.2

0.5

2

TYPICAL

A

E

(V

GAIN

/

I

GAIN

)

8

2
4

Table 4. Output Voltage-Adjustment
Settings

Table 5. LG Pin Adjustment Settings

Non-Intel-

compatible

DAC codes

No CPU (off)11110

1.101110

1.1———10

1.110010

1.100010

1.211100

1.301100

1.410100

1.500100

1.611000

1.701000

1.810000

1.900000

Intel-compatible

DAC codes

No CPU (off)11111

2.101111

2.210111

COMPATIBILITY

2.3

2.4

2.5

2.6

2.7

2.8

2.9

3.0

3.5

3.1

3.3

3.2

3.4

OUTPUT

VOLTAGE (V)

0

1

1

0

0

1

1

0

0

0

1
1

0

D1

1

1

1

1

1

0

0

0

0

0

0
0

011

101

001

101

001

111

011

111

001

011

001
101

101

D0D2D40D3

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

18 ______________________________________________________________________________________

A good compromise between size and loss is a 45%
ripple current to load current ratio (LIR = 0.45), which
corresponds to a peak inductor current 1.23 times
higher than the DC load current.

where f is the switching frequency, between 100kHz
and 1MHz; I

OUT

is the maximum DC load current; and

LIR is the ratio of AC to DC inductor current (typically

0.45). The exact inductor value is not critical and can be
adjusted to make trade-offs among size, transient
response, cost, and efficiency. Although lower inductor
values minimize size and cost, they also reduce efficien­cy due to higher peak currents. In general, higher
inductor values increase efficiency, but at some point
resistive losses due to extra turns of wire exceed the
benefit gained from lower AC current levels. Load­transient response can be adversely affected by
high inductor values, especially at low (VIN- V

OUT

)

differentials.
The peak inductor current at full load is 1.23 x I

OUT

if
the previous equation is used; otherwise, the peak cur­rent can be calculated using the following equation:

The inductor’s DC resistance is a key parameter for effi­cient performance, and should be less than the current­sense resistor value.

Calculating the Current-Sense

Resistor Value

Calculate the current-sense resistor value according to
the worst-case minimum current-limit threshold voltage
(from the

Electrical Characteristics

) and the peak
inductor current required to service the maximum load.
Use I

PEAK

from the equation in the section

Specifying

the Inductor

.

The high inductance of standard wire-wound resistors
can degrade performance. Low-inductance resistors,
such as surface-mount power metal-strip resistors, are
preferred. The current-sense resistor’s power rating
should be higher than the following:

In high-current applications, connect several resistors
in parallel as necessary, to obtain the desired resis­tance and power rating.

Selecting the Output Filter Capacitor

Output filter capacitor values are generally determined
by effective series resistance (ESR) and voltage-rating
requirements, rather than by the actual capacitance
value required for loop stability. Due to the high switch­ing currents and demanding regulation requirements in
a typical MAX1624/MAX1625 application, use only spe­cialized low-ESR capacitors intended for switching­regulator applications, such as AVX TPS, Sprague
595D, Sanyo OS-CON, or Nichicon PL series. Do not
use standard aluminum-electrolytic capacitors, which
can cause high output ripple and instability due to high
ESR. The output voltage ripple is usually dominated by
the filter capacitor’s ESR, and can be approximated as
I

RIPPLE

x R

ESR

. To ensure stability, the capacitor must

meet

both

minimum capacitance and maximum ESR

values as given in the following equations:

C

V

V

V

V x R x f

R R

OUT

REF

OUT

IN MIN

OUT SENSE OSC

ESR SENSE

( )

>

+











<

1

R

mV

R

POWER RATING

SENSE

( )

=

115

2

R

mV

I

SENSE

PEAK

=

85

I I

V V V

f x L x V

PEAK OUT

OUT IN MAX OUT

OSC INMAX

( )

( )

= +

−

( )

2

L

V V V

V x f x I x LIR

OUT IN MAX OUT

IN MAX OSC OUT

( )

( )

=

−

( )

R3

PLACE VERY CLOSE

TO MAX1625

R2

FB

AGND

V

OUT

LOAD

MAX1625

Figure 7. MAX1625 Adjustable Output Operation

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 19

Compensating the Feedback Loop

The feedback loop needs proper compensation to pre­vent excessive output ripple and poor efficiency
caused by instability. Compensation cancels unwanted
poles and zeros in the DC-DC converter’s transfer func­tion that are due to the power-switching and filter ele­ments with corresponding zeros and poles in the
feedback network. These compensation zeros and
poles are set by the compensation components CC1,
CC2, and RC1. The objective of compensation is to
ensure stability by ensuring that the DC-DC converter’s
phase shift is less than 180° by a safe margin, at the
frequency where the loop gain falls below unity.

One simple method for ensuring adequate phase mar­gin is to place pole-zero pairs to approximate a single­pole response with a -20dB/decade slope all the way to
unity-gain crossover (Figure 8). (Other compensation
schemes are possible.) The order of undesired poles
and zeros may differ from that shown in Figure 8,
depending on the characteristics of the load, output
filter capacitor, switching frequency, and inductor.
These procedures are guidelines only, and empirical
experimentation is needed to select the compensation
components’ final values.

Canceling the Sampling Pole

and Output Filter ESR Zero

Compensate the fast-voltage feedback loop by con­necting a resistor and a capacitor in series from the
CC1 pin to AGND. The pole from CC1 can be set to
cancel the zero from the filter-capacitor ESR. Thus the
capacitor at CC1 should be as follows:

Resistor RC1 sets a zero that can be used to compen­sate for the sampling pole generated by the switching
frequency. Set RC1 to the following:

The CC1 pin’s output resistance is 10kΩ. In the sam­pling pole equation (Figure 8), D

MAX

is the maximum

duty cycle, or V

OUT

/ V

IN(MIN)

.

Setting the Dominant Pole

and Canceling the Load and Output Filter Pole

Compensate the slow-voltage feedback loop by adding
a ceramic capacitor from the CC2 pin to AGND. This is
an integrator loop used to cancel out the DC load­regulation error. Selection of capacitor CC2 sets the
dominant pole and a compensation zero. The zero is
typically used to cancel the unwanted pole generated
by the load and output filter capacitor at the maximum
load current. Select CC2 to place the zero close to or
slightly lower than the frequency of the unwanted pole,
as follows:

The transconductance of the integrator amplifier at CC2
is 1mmho. The voltage swing at CC2 is internally
clamped around 2.4V to 3V minimum and 4V to V

CC

maximum to improve transient response times. CC2
can source and sink up to 100µA.

CC

mmho x C

x

V

I

OUT OUT

OUT MAX

2

1

4

( )

=

RC

V

V

f x CC

OUT

IN

OSC

1

1

2 1

=

+











CC

C x R

k

OUT ESR

110

=Ω

FREQUENCY (LOG)

LOOP

GAIN

DOMINANT POLE
FROM INTEGRATOR

UNWANTED
POLE FROM
R

LOAD COUT

COMPENSATION ZERO
TO CANCEL POLE FROM
R

LOAD COUT

COMPENSATION
POLE TO CANCEL
ZERO FROM
C

OUT RESR

UNWANTED ZERO
FROM C

OUT RESR

UNWANTED
SAMPLING POLE

COMPENSATION
ZERO TO CANCEL
SAMPLING POLE

DESIRED RESPONSE

2π(RC1 x CC1)

2π(10kΩ x CC1)

1

1

(1 + D

MAX

) x π

f

OSC(MIN)

2πR

ESR

x C

OUT

1

2πR

LOAD(MAX)

x C

OUT

1

2π50kΩ x CC2

1

1mmho

2π x 4CC2

GAIN (dB LINEAR)

Figure 8. MAX1624/MAX1625 Bode Plot with Compensation
Poles and Zeros

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

20 ______________________________________________________________________________________

Calculating the Loop Gain (Optional)

The loop gain is an important parameter in alternative
compensation schemes:

where A

E

is the error-comparator relative gain, and
AI = 10 is the integrator gain. AEis 4 for the MAX1625,
but it is 2, 4, or 8 for LG pin settings of VCC, REF, or
AGND, respectively, for the MAX1624.

Feed-Forward Compensation (MAX1625)

An optional compensation capacitor, typically 220pF,
may be needed across the upper feedback resistor to
counter the effects of stray capacitance on the FB pin,
and to help ensure stable operation when high-value
feedback resistors are used (Figure 9). Empirically adjust
the feed-forward capacitor as needed.

Choosing the MOSFET Switches

The two high-current N-channel MOSFETs must be
logic-level types with guaranteed on-resistance specifi­cations at V

GS

= 4.5V. Lower gate-threshold specs are
better (i.e., 2V max rather than 3V max). Gate charge
should be less than 100nC to minimize switching losses
and reduce power dissipation.

I2R losses are the greatest heat contributor to MOSFET
power dissipation and are distributed between the high­and low-side MOSFETs according to duty factor, as follows:

Switching losses affect the upper MOSFET only, and are
insignificant at 5V input voltages. Gate-charge losses are
dissipated in the IC, and do not heat the MOSFETs.
Ensure that both MOSFETs are at a safe junction temper­ature by calculating the temperature rise according to
package thermal-resistance specifications. The high-side
MOSFET’s worst-case dissipation occurs at the maximum
output voltage and minimum input voltage. For the low­side MOSFET, the worst case is at the maximum input
voltage when the output is short-circuited (consider the
duty factor to be 100%).

Selecting the Rectifier Diode

The rectifier diode D1 is a clamp that catches the nega­tive inductor swing during the 30ns typical dead time
between turning off the high-side MOSFET and turning on
the low-side MOSFET synchronous rectifier. D1 must be a
Schottky diode, to prevent the MOSFET body diode from
conducting. It is acceptable to omit D1 and let the body
diode clamp the negative inductor swing, but efficiency
will drop 1% or 2% as a result. Use a 1N5819 diode for
loads up to 3A, or a 1N5822 for loads up to 10A.

Adding the BST Supply Diode

and Capacitor

A signal diode, such as a 1N4148, works well for D2 in
most applications, although a low-leakage Schottky diode
provides slightly improved efficiency. Do not use large
power diodes, such as the 1N4001 or 1N5817. Exercise
caution in the selection of Schottky diodes, since some
types exhibit high reverse leakage at high operating tem­peratures. Bypass BST to LX using a 0.1µF capacitor.

Selecting the Input Capacitors

Place a 0.1µF ceramic capacitor and 4.7µF capacitor
between VCCand AGND, as well as between VDDand
PGND, within 0.2 in. (5mm) of the VCCand VDDpins.

Select low-ESR input filter capacitors with a ripple­current rating exceeding the RMS input ripple current,
connecting several capacitors in parallel if necessary.
RMS input ripple current is determined by the input
voltage and load current, with the worst-possible case
occurring at VIN= 2 x V

OUT

:

I I

V V V

V

I I when V V

RMS LOAD MAX

OUT IN OUT

IN

RMS OUT IN OUT

( )

/

( )

=

−

= =2 2

P low side I x R x

V

V

P low side shorted I x R

where I mV R

D LOAD DS ON

OUT

IN

D LIMIT DS ON

LIMIT SENSE

( )

( , )

/ .

( )

( )

= −











=

=

2

1

115

2

P high side I x R x

V

V

D LOAD DS ON

OUT

IN

( )

( )

=

2

Loop Gain dB Log A

V

V

x

R

R

x A

Log A

V

mV

x

E

REF

OUT

LOAD

CS

I

E

REF

( )

=











=











20

20

85

10

R3

R2

OPTIONAL FEED-

FORWARD CAPACITOR

FB

AGND

OUTPUT

MAX1625

Figure 9. MAX1625 Optional Feed-Forward Compensation
Capacitors

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 21

Bypassing the Internal Reference

Bypass the internal 3.5V reference at the REF pin by
connecting a 0.1µF capacitor to AGND. Use a larger
value, such as 1µF, for high-current applications.

Choosing the GlitchCatcher MOSFETs

P-channel and N-channel switches and a series resistor
are required for the current-boost circuit (Figure 10).
Current through the MOSFETs and current-limiting
resistors must be sufficient to supply the load current,
with enough extra for prompt output regulation without
excessive overshoot. Design for boost-current values

1.5 times the maximum load current, and choose
MOSFETs and current-limiting resistors such that:

__________Applications Information

Efficiency Considerations

Refer to the MAX796–MAX799 data sheet for informa­tion on calculating losses and improving efficiency.

PC Board Layout Considerations

Good PC board layout and routing are

required

in high­current, high-frequency switching power supplies to
achieve good regulation, high efficiency, and stability.
The PC board layout artist must be provided with
explicit instructions concerning the placement of
power-switching components and high-current routing.

It is strongly recommended that the evaluation kit PC
board layouts be followed as closely as possible.
Contact Maxim’s Applications Department concerning
the availability of PC board examples for higher-current
circuits.

In most applications, the circuit is on a multilayer
board, and full use of the four or more copper layers is
recommended. Use the top layer for high-current
power and ground connections. Leave the extra cop­per on the board as a pseudo-ground plane. Use the
bottom layer for quiet connections (REF, FB, AGND),
and the inner layers for an uninterrupted ground plane.
A ground plane and pseudo-ground plane are essential
for reducing ground bounce and switching noise.

Follow these steps:

1) Place the high-power components (C1, R1, N1,
D1, N2, L1, and C2 in Figure 1) as close together
as possible, following these priorities:

• Minimize ground-trace lengths in high-current

paths. The surface-mount power components
should be butted up to one another with their
ground terminals almost touching. Connect their
ground terminals using a wide, filled zone of top­layer copper (the pseudo-ground plane), rather
than through the internal ground plane. At the out­put terminal, use vias to connect the top-layer
pseudo-ground plane to the normal inner-layer
ground plane at the output filter capacitor ground
terminals. This minimizes interference from IR
drops and ground noise, and ensures that the IC’s
AGND is sensing at the supply’s output terminals.

R R

V V

I

and
R R

V

I

DSON P MAX LIMIT

IN OUT

OUT MAX

DSON N MAX LIMIT

OUT

OUT MAX

, ( )

( )

, ( )

( )

.

.

+ ≈

−

+ ≈

1 5

1 5

R11

INPUT 5V

C3

C2

C1

OUTPUT

1.1V TO 3.5V

N3

NDRV

PDRV

LOAD

MAX1624

P1

Figure 10. GlitchCatcher Circuit

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

22 ______________________________________________________________________________________

• Minimize high-current path trace lengths. Use
very short and wide traces. From C1 to N1: 0.4 in.
(10mm) max length; D1 cathode to N2: 0.2 in.
(5mm) max length; LX node (N1 source, N2 drain,
D1 cathode, inductor L1): 0.6 in. (15mm) max
length.

2) Place the MAX1624/MAX1625 and supporting
components following these rules:

• Minimize trace lengths to the current-sense

resistor. The IC must be no farther than 0.4 in.

(10mm) from the current-sense resistor. Use a
Kelvin connection.

• Minimize ground trace lengths between the

MAX1624/MAX1625 and supporting compo­nents. Connect components for the REF, CC1,

CC2, and FREQ pin directly to AGND. Connect
AGND and PGND at the IC.

• Keep noisy nodes and components away from

sensitive analog nodes, such as the current­sense, voltage-feedback, REF, CC1, CC2, and

FREQ pins. Placing the IC and analog compo­nents on the opposite side of the board from the
power-switching node is desirable. Noisy nodes
include the main switching node (LX), inductor,
and gate-drive outputs.

• Place components for the FREQ, REF, CC1,

and CC2 pins as close to the IC as possible,

within 0.2 in. (5mm).

• Keep the gate-drive traces (DH, DL, and BST)
shorter than 20mm, and route them away from
CSH, CSL, REF, FB, etc.

• Filter the VCCsupply input to the IC. Bypass the
IC directly from VDDto PGND using a 0.1µF
ceramic capacitor and 4.7µF electrolytic capacitor
placed within 0.2 in. (5mm) of the IC.

• Place the voltage-feedback components close to
the FB pin of the MAX1625, within 0.2 in. (5mm).
Connect the voltage-feedback trace directly to the
CPU’s power input and route it to avoid noisy
traces.

MAX1624/MAX1625

High-Speed Step-Down Controllers with

Synchronous Rectification for CPU Power

______________________________________________________________________________________ 23

___________________Chip Information

TRANSISTOR COUNT: 2472
SUBSTRATE CONNECTED TO AGND

24
23
22
21
20
19
18
17

1
2
3
4
5
6
7
8

DH
LX
PGND
DLCSH

CSL

PWROK

BST

TOP VIEW

V

DD

PDRV
NDRV
D3LG

D0

D1

D2

16
15
14
13

9
10
11
12

D4
FREQ
CC2
CC1FB

AGND

REF

V

CC

SSOP

MAX1624

__________________________________________________________Pin Configurations

16
15
14
13
12
11
10

9

1
2
3
4
5
6
7
8

BST DH

LX
PGND
DL
V

DD

FREQ
CC2
CC1

MAX1625

SO

PWROK

CSL

REF

CSH

V

CC

AGND

FB

MAX1624/MAX1625

High-Speed Step-Down Controllers with
Synchronous Rectification for CPU Power

24 ______________________________________________________________________________________

________________________________________________________Package Information

SSOP.EPS