Rainbow Electronics MAX1585 User Manual

General Description

The MAX1584/MAX1585 provide a complete power­supply solution for slim digital cameras. They improve
performance, component count, and size compared to
conventional multichannel controllers in 2-cell AA, 1-cell
Li+, and dual-battery designs. On-chip MOSFETs pro­vide up to 95% efficiency for critical power supplies,
while additional channels operate with external FETs for
optimum design flexibility. This optimizes overall effi­ciency and cost, while also reducing board space.

The MAX1584/MAX1585 include 5 high-efficiency DC­DC conversion channels:

• Step-up DC-DC converter with on-chip FETs

• Step-down DC-DC converter with on-chip FETs

• Three PWM DC-DC controllers for CCD, LCD, LED,
or other functions

The step-down DC-DC converter can operate directly
from the battery or from the step-up output, providing
boost-buck capability with a compound efficiency of up
to 90%. Both devices include three PWM DC-DC con­trollers: the MAX1584 includes two step-up controllers
and one step-down controller, while the MAX1585
includes one step-up controller, one inverting controller,
and one step-down controller. All DC-DC channels
operate at one fixed frequency—settable from 100kHz
to 1MHz—to optimize size, cost, and efficiency. Other
features include soft-start, power-OK outputs, and over­load protection. The MAX1584/MAX1585 are available
in space-saving, 32-pin thin QFN packages. An evalua­tion kit is available to expedite designs.

Applications

Digital Cameras

PDAs

Features

♦ Step-Up DC-DC Converter, 95% Efficient

♦ Step-Down DC-DC Converter

Operate from Battery for 95% Efficient
Step-Down
90% Efficient Boost-Buck with Step-Up

♦ Three Auxiliary PWM DC-DC Controllers

♦ No Transformers (MAX1585)

♦ Up to 1MHz Operating Frequency

♦ 1mA Shutdown Mode

♦ Internal Soft-Start Control

♦ Overload Protection

♦ Compact 32-Pin Thin QFN Package (5mm x 5mm)

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

________________________________________________________________ Maxim Integrated Products 1

Ordering Information

MAX1585

INPUT

0.7V TO 5.5V

ONSU
ONSD
ON1
ON2
ON3

STEP-UP SYSTEM +5V

STEP-DOWN CORE +1.8V

AUX1 LCD, CCD, LED +15V

AUX2 CCD -7.5V

AUX3 LOGIC +3.3V

Typical Operating Circuit

19-2883; Rev 0; 7/03

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

Pin Configuration

PART TEMP RANGE

MAX1584ETJ -40°C to +85°C

MAX1585ETJ -40°C to +85°C

PIN-

PACKAGE

32 Thin QFN

5mm x 5mm

32 Thin QFN

5mm x 5mm

FB331CC330GND29DL128DL327DL226PV25INDL2

32

1CC1

FB1

2

PGSD

3

ON110ON211ON3

THIN QFN

5mm x 5mm

MAX1584
MAX1585

12

13

ONSU

REF

14

15

FBSU

LXSD

PVSD

ONSD

CCSD

FBSD

4

5

6

7

8

9

24 CC2

23

22

21

20

19

18

17

16

CCSU

AUX1OK

AUX

FUNCTIONS

2 step-up

1 step-down

1 step-up

1 step-down

1 inverting

FB2

PVSU

LXSU

PGSU

OSC

SCF

SDOK

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

PV, PVSU, PVSD, SDOK, AUX1OK, SCF, ON_, FB_ to

GND..........................................................................-0.3V to +6V

PGND to GND....................................................…-0.3V to +0.3V

INDL2, DL1, DL3 to GND.........................-0.3V to (PVSU + 0.3V)

DL2 to GND ............................................-0.3V to (INDL2 + 0.3V)

PV to PVSU ...........................................................-0.3V to + 0.3V

LXSU Current (Note 1) ..........................................................3.6A

LXSD Current (Note 1) ........................................................2.25A

REF, OSC, CC_ to GND...........................-0.3V to (PVSU + 0.3V)

Continuous Power Dissipation (T

A

= +70°C)

32-Pin Thin QFN (derate 22mW/°C above +70°C) ....1700mW

Operating Temperature Range ...........................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

ELECTRICAL CHARACTERISTICS

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= 0°C to +85°C, unless otherwise noted.)

Note 1: LXSU has internal clamp diodes to PVSU and PGND, and LXSD has internal clamp diodes to PVSD and PGND. Applications

that forward bias these diodes should take care not to exceed the device’s power dissipation limits.

GENERAL

PARAMETER CONDITIONS MIN TYP MAX UNITS

Input Voltage Range (Note 2) 0.7 5.5 V
Step-Up Minimum Startup

Voltage

I

< 1mA, TA = +25°C, startup voltage tempco is

LOAD

-2300ppm/°C (typ) (Note 3)

Shutdown Supply Current into PV PV = 3.6V 0.1 5 µA
Supply Current into PV with

Step-Up Enabled

Supply Current into PV with

Step-Up and Step-Down Enabled

ONSU = 3.6V, FBSU = 1.5V

(does not include switching losses)

ONSU = ONSD = 3.6V, FBSU = 1.5V, FBSD = 1.5V

(does not include switching losses)

0.9 1.1 V

300 450 µA

450 700 µA

Total Supply Current from PV and

PVSU with Step-Up and One AUX
Enabled

ONSU = ON1 = 3.6V, FBSU = 1.5V, FB2 = 1.5V

(does not include switching losses)

400 650 µA

REFERENCE
Reference Output Voltage I
Reference Load Regulation 10µA < I

= 20µA 1.23 1.25 1.27 V

REF

< 200µA 4.5 10 mV

REF

Reference Line Regulation 2.7 < PVSU < 5.5V 1.3 5 mV
OSCILLATOR
OSC Discharge Trip Level Rising edge 1.225 1.25 1.275 V
OSC Discharge Resistance OSC = 1.5V, I

= 3mA 52 80 Ω

OSC

OSC Discharge Pulse Width 150 ns
OSC Frequency R

= 47kΩ, C

OSC

=100pF 500 kHz

OSC

STEP-UP DC-DC CONVERTER
Step-Up Startup-to-Normal

Operating Threshold

Step-Up Startup-to-Normal

Operating Threshold Hysteresis

Rising edge or falling edge (Note 4) 2.30 2.5 2.65 V

80 mV

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= 0°C to +85°C, unless otherwise noted.)

Idle Mode is a trademark of Maxim Integrated Products, Inc.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Step-Up Voltage Adjust Range 3.0 5.5 V
Start Delay of ONSD, ON1, ON2,

ON3 after SU in Regulation

1024

OSC

cycles

FBSU Regulation Voltage 1.231 1.25 1.269 V
FBSU to CCSU

Transconductance

FBSU = CCSU 80 135 185 µS

FBSU Input Leakage Current FBSU = 1.25V -100 +1 +100 nA
Idle Mode

Current-Sense Amplifier

Transresistance

TM

Trip Level (Note 6) 150 mA

0.275 V/A

Step-Up Maximum Duty Cycle FBSU = 1V 80 85 90 %
PVSU Leakage Current V
LXSU Leakage Current V

Switch On-Resistance

= 0V, PVSU = 5.5V 0.1 5 µA

LX

= V

LXSU

= 5.5V 0.1 5 µA

OUT

N channel 95 150

P channel 150 250
N-Channel Current Limit 2.4 2.8 3.2 A
P-Channel Turn-Off Current 20 mA
Startup Current Limit PVSU = 1.8V (Note 5) 450 mA
Startup t

OFF

PVSU = 1.8V 700 ns
Startup Frequency PVSU = 1.8V 200 kHz
STEP-DOWN DC-DC CONVERTER
Step-Down Output Voltage

Adjust Range

PVSD must be greater than output (Note 7) 1.25 5.00 V

FBSD Regulation Voltage 1.231 1.25 1.269 V
FBSD to CCSD

Transconductance

FBSD = CCSD 80 135 185 µS

FBSD Input Leakage Current FBSD = 1.25V -100 +0.1 +100 nA
Idle Mode Trip Level (Note 6) 100 mA

Current-Sense Amplifier

Transresistance

LXSD Leakage Current V

Switch On-Resistance

0.5 V/A

= 0 to 3.6V, PVSU = 3.6V 0.1 5 µA

LXSD

N channel 95 150

P channel 150 250
P-Channel Current Limit 0.65 0.8 0.95 A
N-Channel Turn-Off Current 20 mA

OSC

Soft-Start Interval 2048

cycles

SDOK Output Low Voltage 0.1mA into SDOK 0.01 0.1 V
SDOK Leakage Current ONSU = GND 0.01 1 µA

mΩ

mΩ

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= 0°C to +85°C, unless otherwise noted.)

PARAMETER CONDITIONS MIN TYP MAX UNITS
AUX1, 2, 3 DC-DC CONTROLLERS
Maximum Duty Cycle FB_ = 1V 80 85 90 %
FB1 and FB3 Regulation Voltage FB_ = CC_ 1.231 1.25 1.269 V

FB2 (MAX1584) Regulation

Voltage

FB2 (MAX1585) (Inverter)

Regulation Voltage

FB_ = CC_ 1.231 1.25 1.269 V

FB_ = CC_ -0.01 0 +0.01 V

FB_ to CC_ Transconductance FB_ = CC_ 80 135 185 µS
FB_ Input Leakage Current FB_ = 1.25V -100 +1 +100 nA
DL_ Driver Resistance Output high or low 2.5 10 Ω
DL_ Drive Current Sourcing or sinking 0.5 A

Soft-Start Interval 4096

AUX1OK Output Low Voltage 0.1mA into AUX1OK 0.01 0.1 V
AUX1OK Leakage Current ONSU = GND 0.01 1 µA
OVERLOAD AND THERMAL PROTECTION

Overload-Protection Fault Delay 100,000

SCF Leakage Current ONSU = PVSU, FBSU = 1.5V 0.1 1 µA
SCF Output Low Voltage 0.1mA into SCF 0.01 0.1 V
Thermal Shutdown +160 °C
Thermal Hysteresis 20 °C
LOGIC INPUTS

ON_ Input Low Level

ON_ Input High Level

ON_ Impedance to GND ON_ = 3.35V 330 kΩ

1.1V < PVSU < 1.8V (ONSU only) 0.2

1.8V < PVSU< 5.5V 0.4
V

-

1.1V < PVSU < 1.8V (ONSU only)

PVSU

0.2

1.8V < PVSU < 5.5V 1.6

OSC

cycles

OSC

cycles

V

V

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

_______________________________________________________________________________________ 5

ELECTRICAL CHARACTERISTICS

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= -40°C to +85°C, unless otherwise noted.) (Note 8)

PARAMETER CONDITIONS MIN MAX UNITS
GENERAL
Input Voltage Range (Note 2) 0.7 5.5 V

Shutdown Supply Current

into PVSU

Supply Current into PV with

Step-Up Enabled

Supply Current into PV with

Step-Up and Step-Down Enabled

Total Supply Current from PV and

PVSU with Step-Up and One AUX
Enabled

PVSU = 3.6V 5 µA

ONSU = 3.6V, FBSU = 1.5V

(does not include switching losses)

ONSU = ONSD = 3.6V, FBSU = 1.5V, FBSD = 1.5V

(does not include switching losses)

ONSU = ON1 = 3.6V, FBSU = 1.5V, FB2 = 1.5V

(does not include switching losses)

450 µA

700 µA

650 µA

REFERENCE
Reference Output Voltage I
Reference Load Regulation 10µA < I

= 20µA 1.225 1.275 V

REF

< 200µA 10 mV

REF

Reference Line Regulation 2.7V < PVSU < 5.5V 5 mV
OSCILLATOR
OSC Discharge Trip Level Rising edge 1.225 1.275 V
OSC Discharge Resistance OSC = 1.5V, I

= 3mA 80 Ω

OSC

STEP-UP DC-DC CONVERTER
Step-Up Startup-to-Normal

Operating Threshold

Rising edge or falling edge (Note 4) 2.30 2.65 V

Step-Up Voltage Adjust Range 3.0 5.5 V
FBSU Regulation Voltage 1.225 1.275 V

FBSU to CCSU

Transconductance

FBSU = CCSU 80 185 µS

FBSU Input Leakage Current FBSU = 1.25V -100 +100 nA
Step-Up Maximum Duty Cycle FBSU = 1V 80 90 %
PVSU Leakage Current V
LXSU Leakage Current V

Switch On-Resistance

= 0V, PVSU = 5.5V 5 µA

LX

= V

LXSU

= 5.5V 5 µA

OUT

N channel 150
P channel 250

mΩ

N-Channel Current Limit 2.4 3.2 A
STEP-DOWN DC-DC CONVERTER

Step-Down Output Voltage Adjust

Range

PVSD must be greater than output (Note 7) 1.25 5.00 V

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

6 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= -40°C to +85°C, unless otherwise noted.) (Note 8)

Note 2: The MAX1584/MAX1585 are powered from the step-up output (PVSU). An internal low-voltage startup oscillator drives the

step-up starting at about 0.9V until PVSU reaches approximately 2.5V. When PVSU reaches 2.5V, the main control circuitry
takes over. Once the step-up is up and running, it can maintain operation with very low input voltages; however, output cur­rent is limited.

Note 3: Since the device is powered from PVSU, a Schottky rectifier, connected from the input battery to PVSU, is required for low-

voltage startup, or if PVSD is connected to V

IN

instead of PVSU.

Note 4: The step-up regulator is in startup mode until this voltage is reached. Do not apply full load current during startup. A power-

OK output can be used with an external PFET to gate the load until the step-up is in regulation. See the Applications
Information section.

PARAMETER CONDITIONS MIN MAX UNITS
FBSD Regulation Voltage 1.225 1.275 V
FBSD to CCSD

Transconductance

FBSD Input Leakage Current FBSD = 1.25V -100 +100 nA
LXSD Leakage Current V

Switch On-Resistance

P-Channel Current Limit 0.65 0.95 A
SDOK Output Low Voltage 0.1mA into SDOK 0.1 V
SDOK Leakage Current ONSU = GND 1 µA
AUX1, 2, 3 DC-DC CONTROLLERS
Maximum Duty Cycle FB_ = 1V 80 90 %
FB1 and FB3 Regulation Voltage FB_ = CC_ 1.225 1.275 V

FB2 (MAX1584) Regulation

Voltage

FB2 (MAX1585) (Inverter)

Regulation Voltage

FB_ to CC_ Transconductance FB_ = CC_ 80 185 µS
FB_ Input Leakage Current FB_ = 1.25V -100 +100 nA
DL_ Driver Resistance Output high or low 10 Ω
AUX1OK Output Low Voltage 0.1mA into AUX1OK 0.1 V
AUX1OK Leakage Current ONSU = GND 1 µA
OVERLOAD AND THERMAL PROTECTION
SCF Leakage Current ONSU = PVSU, FBSU = 1.5V 1 µA
SCF Output Low Voltage 0.1mA into SCF 0.1 V
LOGIC INPUTS

ON_ Input Low Level

ON_ Input High Level

FBSD = CCSD 80 185 µS

= 0 to 3.6V, PVSU = 3.6V 5 µA

LXSD

N channel 150
P channel 250

FB_ = CC_ 1.225 1.275 V

FB_ = CC_ -0.01 +0.01 V

1.1V < PVSU < 1.8V (ONSU only) 0.2

1.8V < PVSU < 5.5V 0.4

1.1V < PVSU < 1.8V (ONSU only) V

1.8V < PVSU < 5.5V 1.6

PVSU

- 0.2

mΩ

V

V

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

_______________________________________________________________________________________ 7

Typical Operating Characteristics

(Circuit of Figure 1, TA= +25°C, unless otherwise noted.)

Note 5: The step-up current limit in startup refers to the LXSU switch current limit, not an output current limit.
Note 6: The idle mode current threshold is the transition point between fixed-frequency PWM operation and idle mode operation

(where switching rate varies with load). The specification is given in terms of inductor current. In terms of output current, the
idle mode transition varies with input-output voltage ratio and inductor value. For the step-up, the transition output current is
approximately 1/3 the inductor current when stepping from 2V to 3.3V. For the step-down, the transition current in terms of
output current is approximately 3/4 the inductor current when stepping down from 3.3V to 1.8V.

Note 7: Operation in dropout (100% duty cycle) can only be maintained for 100,000 OSC cycles before the output is considered

faulted, triggering global shutdown.

Note 8: Specifications to -40°C are guaranteed by design, not production tested.

ELECTRICAL CHARACTERISTICS (continued)

(V

PVSU

= VPV= V

PVSD

= V

INDL2

= 3.6V, TA= -40°C to +85°C, unless otherwise noted.) (Note 8)

STEP-UP EFFICIENCY

vs. LOAD CURRENT

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

VIN = 4.5V

= 4.2V

V

IN

= 3.8V

V

IN

= 3.0V

V

IN

V

= 5V

0

11000

LOAD CURRENT (mA)

OUT

10010

MAX1584/85 toc01

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

STEP-DOWN EFFICIENCY

vs. LOAD CURRENT

100

90

80

MAX1584/85 toc02

VIN = 3.0V

= 3.8V

V

IN

= 4.2V

V

IN

= 4.5V

V

IN

PVSD CONNECTED TO BATTERY

= 1.5V

V

OUT

DOES NOT INCLUDE CURRENT USED
BY THE STEP-UP TO POWER THE IC

11000

LOAD CURRENT (mA)

10010

70

60

50

EFFICIENCY (%)

40

30

20

10

COMBINED BOOST-BUCK

EFFICIENCY vs. LOAD CURRENT

VIN = 4.5V
V

IN

V

IN

V

IN

1 1000

LOAD CURRENT (mA)

= 4.2V
= 3.8V
= 3.0V

10010

V
V

OUT3
OUTSU

= 3.3V

= 5.0V

MAX1584/85 toc03

EFFICIENCY vs. INPUT VOLTAGE

100

95

90

85

EFFICIENCY (%)

80

SU = 5V, 300mA
SD = 1.5V, 250mA
SU + AUX3 = 3.3V, 300mA

75

AUX1 = 15V, 40mA
AUX2 = -7.5V, 40mA

70

3.0 4.5
INPUT VOLTAGE (V)

4.03.5

MAX1584/85 toc04

AUX1 EFFICIENCY vs. LOAD CURRENT

MAX1585 AUX2 EFFICIENCY

100

90

80

70

60

EFFICIENCY (%)

50

40

30

11000

VIN = 4.5V

= 4.2V

V

IN

= 3.8V

V

IN

= 3.0V

V

IN

10010

LOAD CURRENT (mA)

V

= 15V

OUT1

MAX5184/85 toc05

90

80

70

60

EFFICIENCY (%)

50

40

30

1 1000

vs. LOAD CURRENT

VIN = 3.0V

= 3.8V

V

IN

= 4.2V

V

IN

= 4.5V

V

IN

10010

LOAD CURRENT (mA)

V

OUT2

MAX5184/85 toc06

= -7.5V

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

8 _______________________________________________________________________________________

Typical Operating Characteristics (continued)

(Circuit of Figure 1, TA= +25°C, unless otherwise noted.)

NO-LOAD INPUT CURRENT

vs. INPUT VOLTAGE (SWITCHING)

9

8

7

6

5

4

3

INPUT CURRENT (mA)

2

VSU = 5.0V

1

0

05.0

BOOST-BUCK (SU + AUX3)

= 5.0V, OUT3 = 3.33V

V

SU

INPUT VOLTAGE (V)

REFERENCE VOLTAGE

vs. TEMPERATURE

1.254

MAX1584/85 toc07

MINIMUM STARTUP VOLTAGE (V)

4.54.03.0 3.51.0 1.5 2.0 2.50.5

1.250

MINIMUM STARTUP VOLTAGE

vs. LOAD CURRENT (V

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0

0 1000

vs. REFERENCE LOAD CURRENT

SCHOTTKY DIODE CONNECTED
FROM IN TO V

LOAD CURRENT (mA)

SU

800600400200

REFERENCE VOLTAGE

)

SU

MAX5184/85 toc08

1.251

1.248

REFERENCE VOLTAGE (V)

1.246

1.243

-50 100
TEMPERATURE (°C)

OSCILLATOR FREQUENCY vs. R

1100

900

700

500

300

OSCILLATOR FREQUENCY (kHz)

100

-100
11000

C

= 470pF

OSC

C

= 330pF

OSC

C

= 220pF

OSC

C

OSC

C

OSC

10010

R

(kΩ)

OSC

MAX1584/85 toc09

7550250-25

OSC

MAX1584/85 toc11

= 100pF

= 47pF

1.249

1.248

1.247

1.246

REFERENCE VOLTAGE (V)

1.245

1.244
0300

REFERENCE LOAD CURRENT (µA)

MAX1584/85 toc10

25020015010050

SWITCHING FREQUENCY

vs. TEMPERATURE

510

509

508

507

506

505

504

SWITCHING FREQUENCY (kHz)

503

502

501

-50 100
TEMPERATURE (°C)

R

OSC

C

OSC

= 51kΩ
= 100pF

7550250-25

MAX1584/85 toc12

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

_______________________________________________________________________________________ 9

Typical Operating Characteristics (continued)

(Circuit of Figure 1, TA= +25°C, unless otherwise noted.)

MAXIMUM DUTY CYCLE (%)

AUX MAXIMUM DUTY CYCLE

vs. FREQUENCY

88

WHEN THIS DUTY CYCLE IS EXCEEDED
FOR 100,000 CLOCK CYCLES,

87

THE MAX1584/MAX1585 SHUT DOWN

86

85

84

83

82

81

80

0 1000

FREQUENCY (kHz)

STEP-DOWN STARTUP RESPONSE

0V

0V

0A

4ms/div

C

OSC

MAX1584/85 toc15

VIN = 3.5V

= 330pF

900800100 200 300 500 600400 700

MAX1584/85 toc13

ONSD
5V/div

OUTSD
5V/div

I

OUTSD

200mA/div

STEP-UP STARTUP RESPONSE

0V

0V

0A

0A

AUX1 STARTUP RESPONSE

0V

0V

0A

200µs/div

2ms/div

MAX1584/85 toc14

VIN = 3.5V

MAX1584/85 toc16

VIN = 3.5V

ONSU
5V/div

OUTSU
5V/div
I

OUTSU

200mA/div

I

IN

1.0A/div

ON1
5V/div

OUT1
10V/div

I

OUT1

100mA/div

STEP-DOWN LOAD-

STEP-UP LOAD-TRANSIENT RESPONSE

0V

0A

V

= 5.0V

OUTSU

= 3.5V

V

IN

400µs/div

MAX1584/85 toc17

V

OUTSU

AC-COUPLED
500mV/div

I

OUT_SU

200mA/div

0V

0A

TRANSIENT RESPONSE

VIN = 3.5V

= 1.5V

V

OUT_SD

400µs/div

MAX1584/85 toc18

V

OUTSD

AC-COUPLED
100mV/div

I

OUT_SD

100mA/div

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

10 ______________________________________________________________________________________

Pin Description

PIN NAME FUNCTION

AUX1 Controller Compensation Node. Connect a series resistor-capacitor from CC1 to GND to

1 CC1

compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and
thermal limit. See the AUX Compensation section.

2 FB1

3 PGSD

4 LXSD

AUX1 Controller Feedback Input. The feedback threshold is 1.25V. This pin is high impedance in

shutdown.

Step-Down Power Ground. Connect all PG_ pins together and to GND with short traces as close as

possible to the IC.

Step-Down Converter Switching Node. Connect to the inductor of the step-down converter. LXSD is high

impedance in shutdown.

Step-Down Converter Input. PVSD can connect to PVSU, effectively making OUTSD a boost-buck output

from the battery. Bypass to GND with a 1µF ceramic capacitor if connected to PVSU. PVSD can also be

5 PVSD

6 ONSD

connected to the battery but should not exceed PVSU by more than a Schottky diode forward voltage.
Bypass PVSD with a 10µF ceramic capacitor when connecting to the battery input. A 10kΩ internal
resistance connects PVSU and PVSD.

Step-Down Converter On/Off Control Input. Logic high = on; however, turn-on is locked out until the step-

up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND.

Step-Up Converter Compensation Node. Connect a series resistor-capacitor from CCSD to GND to

7 CCSD

8 FBSD

9 ON1

10 ON2

11 ON3

compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and
thermal limit. See the Step-Down Compensation section.

Step-Down Converter Feedback Input. Connect a resistive voltage-divider from OUTSD to FBSD to GND.

The FBSD feedback threshold is 1.25V. This pin is high impedance in shutdown.

AUX1 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after

the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND.

AUX2 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after

the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND.

AUX3 Controller On/Off Input. Logic high = on; however, turn-on is locked out until 1024 OSC cycles after

the step-up has reached regulation. This pin has an internal 330kΩ pulldown resistance to GND.

Step-Up Converter On/Off Control. Logic high = on. All other ON_ pins are locked out until 1024 OSC

12 ONSU

cycles after the step-up DC-DC converter output has reached its final value. This pin has an internal
330kΩ pulldown resistance to GND.

13 REF

14 FBSU

Reference Output. Bypass REF to GND with a 0.1µF or greater capacitor. The maximum allowed load on

REF is 200µA. REF is actively pulled to GND when all converters are shut down.

Step-Up Converter Feedback Input. Connect a resistive voltage-divider from PVSU to FBSU to GND. The

FBSU feedback threshold is 1.25V. This pin is high impedance in shutdown.

Step-Up Converter Compensation Node. Connect a series resistor-capacitor from CCSU to GND to

15 CCSU

compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and
thermal limit. See the Step-Up Compensation section.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 11

Pin Description (continued)

PIN NAME FUNCTION

16 AUX1OK

17 SDOK

Open-Drain Power-OK Signal for AUX1 Controller. AUX1OK is low when the AUX1 controller has

successfully completed soft-start. This pin is high impedance in shutdown, overload, and thermal limit.

Open-Drain Power-OK Signal for Step-Down Converter. SDOK is low when the step-down has successfully

completed soft-start. This pin is high impedance in shutdown, overload, and thermal limit.

Short-Circuit Flag, Active-Low, Open-Drain Output. SCF is high impedance when overload protection

18 SCF

occurs and during startup. SCF can drive high-side PFET switches connected to one or more outputs to
completely disconnect the load when the channel turns off in response to a logic command or an
overload. See the Status Outputs (

SDOK, AUX1OK

, SCF) section.

Oscillator Control. Connect a timing capacitor from OSC to GND and a timing resistor from OSC to PVSU

19 OSC

20 PGSU

21 LXSU

22 PVSU

(or other DC voltage) to set the oscillator frequency between 100kHz and 1MHz. See the Setting the
Switching Frequency section. This pin is high impedance in shutdown.

Step-Up Power Ground. Connect all PG_ pins together and to GND with short traces as close to the IC as

possible.

Step-Up Converter Switching Node. Connect to the inductor of the step-up converter. LXSU is high

impedance in shutdown.

Power Output of the Step-Up DC-DC Converter. Connect the output filter capacitor from PVSU to PGSU.

PVSU can also power other converter channels. Connect PVSU to PV at the IC.

MAX1585 (AUX2 inverter): The FB2 feedback threshold is 0V.

Connect a resistive voltage-divider from the output voltage to FB2 to
REF to set the output voltage.

MAX1584 (AUX2 step-up): The FB2 feedback threshold is 1.25V.

Connect a resistive voltage-divider from the output voltage to FB2 to
GND to set the output voltage.

23 FB2

AUX2 Controller Feedback Input.

This pin is high impedance in
shutdown.

AUX2 Controller Compensation Node. Connect a series resistor-capacitor from CC2 to GND to

24 CC2

compensate the control loop. CC2 is actively driven to GND in shutdown and thermal limit. See the AUX
Compensation section.

MAX1585 (AUX2 inverter): Connect INDL2 to the external P channel

MOSFET source (typically the battery) to ensure the P channel is
completely off when D2 swings high.

MAX1584 (AUX2 step-up): Connect INDL2 to PVSU for optimum

N-channel gate drive.

25 INDL2

Voltage Input for the AUX2 Gate

Driver. The voltage at INDL2 sets
the high gate-drive voltage.

26 PV IC Power Input. Connect PVSU and PV together.

MAX1585: DL2 drives a PFET in an inverter configuration. In

shutdown, overload, and thermal limit, DL2 is driven high.

MAX1584: DL2 drives an N-channel FET in a boost/flyback

configuration. In shutdown, overload, and thermal limit, DL2 is driven
low.

27 DL2

AUX2 Controller Gate-Drive

Output. DL2 drives between
INDL2 and GND.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

12 ______________________________________________________________________________________

Detailed Description

The MAX1584/MAX1585 are complete power-conver­sion ICs for slim digital still cameras. They can accept
input from a variety of sources, including single-cell Li+
batteries and 2-cell alkaline or NiMH batteries, as well
as systems designed to accept both battery types. The
MAX1584/MAX1585 include five DC-DC converter
channels to generate all required voltages (Figure 2
shows a functional diagram):

• Synchronous-rectified step-up DC-DC converter with
on-chip MOSFETs—Typically supplies 3.3V for main
system power or 5V to power other DC-DC convert­ers for boost-buck designs.

• Synchronous-rectified step-down DC-DC converter
with on-chip MOSFETs—Typically supplies 1.8V for
the DSP core. Powering the step-down from the
step-up output provides efficient (up to 90%) boost­buck functionality that supplies a regulated output
when the battery voltage is above or below the out­put voltage. The step-down can also be powered
from the battery if there is sufficient headroom.

• AUX1 step-up controller—Typically used for 15V to
bias one or more of the LCD, CCD, and LED back­lights.

• AUX2 step-up controller (MAX1584)—Typically sup­plies remaining bias voltages with either a multi-out­put flyback transformer or a boost converter with
charge-pump inverter. Alternately, can power white
LEDs for LCD backlighting.

• AUX2 inverter controller (MAX1585)—Typically sup­plies negative CCD bias when high current is need­ed for large pixel-count CCDs.

• AUX3 step-down controller—Typically steps 5V gen­erated at PVSU down to 3.3V for system logic in
boost-buck designs.

Step-Up DC-DC Converter

The step-up DC-DC switching converter is typically used
to generate a 5V output voltage from a 1.5V to 4.5V bat­tery input, but any voltage from VINto 5V can be set. An
internal NFET switch and a PFET synchronous rectifier
allow conversion efficiencies as high as 95%. Under
moderate to heavy loading, the converter operates in a
low-noise PWM mode with constant frequency and modu­lated pulse width. Switching harmonics generated by
fixed-frequency operation are consistent and easily fil­tered. Efficiency is enhanced under light (<75mA typ)
loading, by an idle mode that switches the step-up only
as needed to service the load. In this mode, the maxi­mum inductor current is 250mA for each pulse.

Pin Description (continued)

PIN NAME FUNCTION

28 DL3

29 DL1

AUX3 Step-Down Controller Gate-Drive Output. Connect to the gate of a P-channel MOSFET. DL3 swings

from GND to PVSU and supplies up to 500mA. DL3 is driven to PVSU in shutdown and thermal limit.

AUX1 Step-Up Controller Gate-Drive Output. Connect to the gate of an N-channel MOSFET. DL1 swings

from GND to PVSU and supplies up to 500mA. DL1 is driven to GND in shutdown and thermal limit.

30 GND Analog Ground. Connect to all PG_ pins as close to the IC as possible.

AUX3 Step-Down Controller Compensation Node. Connect a series resistor-capacitor from CC3 to FB3 to

31 CC3

compensate the converter control loop. This pin is actively driven to GND in shutdown, overload, and
thermal limit. See the AUX Compensation section.

PWM Step-Up Controller 3 Feedback Input. Connect a resistive voltage-divider from the output voltage to

32 FB3

FB3 to GND to set the output voltage. The FB3 feedback threshold is 1.25V. This pin is high impedance in
shutdown.

Exposed Underside Metal Pad. This pad must be soldered to the PC board to achieve package thermal

PAD EP

and mechanical ratings. There is no internal metal or bond wire physically connecting the exposed pad to
the GND pin(s). Connecting the exposed pad to ground does not remove the requirement for a good
ground connection to the appropriate IC pins.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 13

Figure 1. MAX1584/MAX1585 Typical Application for 2-Cell AA or 1-Cell Li+ Battery

47k

0.01µF

C1

R1

Ω

25k

4700pF

C2

R2

Ω

-7.5V

-CDD
BIAS

526k

93.1k

20k

1500pF

1.5V TO 4.2V

R8

Ω

R9

Ω

R3

Ω

10k

C3

1500pF

V

IN

10µF

C8

4.7µF

TO PVSU

R5

Ω

C5

C24

D2

61.9k

470pF

0.1µF

20k

330pF

R4

Ω

C4

P1

L6

3.6µH

Ω

TO FB3

INDL2

DL2

FB2

REF

1.25V

OSC

CCSU
CCSD
CC1
CC2
CC3

ONSU
ONSD
ON1
ON2
ON3

AUX1OK

REF

MAX1585

AUX2

V-MODE

INV

PWM

CURRENT-

MODE

STEP-UP

CURRENT-

MODE
STEP­DOWN

AUX1

V-MODE

STEP-UP

PWM

AUX3 V-MODE

STEP-DOWN

PWM

DL1

FB1

DL3

FB3

PVSU

LXSU

PGSU

FBSU

PVSD

LXSD

PGSD

FBSD

L3
2µH

N1

P2

D3

L1

5µH

C19
10µF

D4

OR PVSU

TO V

IN

L2

22µH

30µF

PV

C9

D1

10µH

TO V

L4

C23
10µF

C25

47µF

IN

C11
47µF

C6

4.7µF

TO PVSU
OR V

R10

18.2k

IN

R15

18.2k

R12
274k

R13

90.9k

R6

1M

Ω

R14

30.1k

Ω

5V 1A
MAIN SYSTEM

Ω

Ω

+1.5V
250mA
CORE

Ω

+15V, 80mA
+CCD
LCD
LED

R7

90.9k

Ω

C20

560pF

Ω

3.3V
250mA
LOGIC

R22

1.2k

Ω

SCF

SDOK

GND

R11

90.9k

Ω

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

14 ______________________________________________________________________________________

Figure 2. MAX1584/MAX1585 Functional Diagram

PVSU

2.35V

ONSU

V

ONSU

OSC

REF

1V

INTERNAL

POWER-OK

REFOK

REF

CCSU

FBSU

STEP-UP

SOFT-START

DONE (SUSSD)

CCSD

SOFT-START

RAMP

GENERATOR

100,000

CLOCK CYCLE

FAULT TIMER

150ns

ONE-SHOT

TO
V

NORMAL

MODE

OVER

TEMP

FAULT

IN

CLK

REF

FLTALL

FAULT

CURRENT-MODE

STEP-UP

FAULT

DC-DC

STARTUP

OSCILLATOR

TO INTERNAL

POWER

1.25V

REFERENCE

FLTALL

MAX1584/

MAX1585

SCF

PV

REF

GND

PVSU

LXSU

PGSU

ONSU

PVSD

FBSD

ONSD

SOFT-START

RAMP GENERATOR

SUSSD

FLTALL

CURRENT-MODE

DC-DC

STEP-DOWN

TO
V

REF

TO AUX_

CHANNELS

(SEE FIGURE 3)

LXSD

PGSD

SDOK

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 15

Step-Down DC-DC Converter

The step-down DC-DC converter is optimized for gen­erating low output voltages (down to 1.25V) at high effi­ciency. Output voltages lower than 1V can be set by
adding an additional resistor (see the Applications
Information section). The step-down runs from the volt­age at PVSD. This pin can be connected directly to the
battery if sufficient headroom exists to avoid dropout;
otherwise, PVSD can be powered from the output of
another converter. The step-down can also operate
with the step-up for boost-buck operation.

Under moderate to heavy loading, the converter oper­ates in a low-noise PWM mode with constant frequency
and modulated pulse width. Efficiency is enhanced
under light (<75mA typ) loading by assuming an idle
mode during which the step-down switches only as
needed to service the load. In this mode, the maximum
inductor current is 100mA for each pulse. The step­down DC-DC is inactive until the step-up DC-DC is in
regulation.

The step-down also features an open-drain SDOK out-
put that goes low when the step-down output is in regu­lation. SDOK can be used to drive an external MOSFET
switch that gates 3.3V power to the processor after the
core voltage is in regulation. This connection is shown
in Figure 13.

Boost-Buck Operation

The step-down input can be powered from the output
of the step-up. By cascading these two channels, the
step-down output can maintain regulation even as the
battery voltage falls below the step-down output volt­age. This is especially useful when trying to generate

3.3V from 1-cell Li+ inputs, or 2.5V from 2-cell alkaline
or NiMH inputs, or when designing a power supply that
must operate from both Li+ and alkaline/NiMH inputs.
Compound efficiencies of up to 90% can be achieved
when the step-up and step-down are operated in
series.

Note that the step-up output supplies both the step-up
load and the step-down input current when the step­down is powered from the step-up. The step-down
input current reduces the available step-up output cur­rent for other loads.

Direct Battery Step-Down Operation

The step-down converter can also be operated directly
from the battery as long as the voltage at PVSD does
not exceed PVSU by more than a Schottky diode for­ward voltage. When using this connection, connect an
external Schottky diode from the battery input to PVSU.
On the MAX1584/MAX1585, there is an internal 10kΩ
resistance from PVSU to PVSD. This adds a small addi-

tional current drain (of approximately (V

PVSU

- V

PVSD

) /
10kΩ) from PVSU when PVSD is not connected directly
to PVSU.

Step-down direct battery operation improves efficiency
for the step-down output (up to 95%), but restricts the
upper limit of the output voltage to 200mV less than the
minimum battery voltage. In 1-cell Li+ designs (with a

2.7V min), the output can be set up to 2.5V. In 2-cell
alkaline or NiMH designs, the output can be limited to

1.5V or 1.8V, depending on the minimum-allowed cell
voltage.

The step-down can only be briefly operated in dropout
since the MAX1584/MAX1585 fault protection detects
the out-of-regulation condition and activates after
100,000 OSC cycles (200ms at f

OSC

= 500kHz). At that

point, all MAX1584/MAX1585 channels shut down.

AUX1, AUX2, and AUX3 DC-DC Controllers

The three auxiliary controllers operate as fixed-frequen­cy voltage-mode PWM controllers. They do not have
internal MOSFETs, so output power is determined by
external components. The controllers regulate output
voltage by modulating the pulse width of the DL_ drive
signal to an external MOSFET switch. The MAX1584
contains two step-up/flyback controllers (AUX1 and
AUX2) and one step-down controller (AUX3). The
MAX1585 contains one step-up controller (AUX1), one
inverting controller (AUX2), and one step-down con­troller (AUX3).

Figure 3 shows a functional diagram of the AUX con­trollers. The inverting and step-down controllers differ
from the step-up controllers only in the gate-drive logic
and FB polarity and threshold. The sawtooth oscillator
signal at OSC governs timing. At the start of each
cycle, DL_ turns on the external MOSFET switch. For
step-up controllers, DL_ goes high, while for inverting
and step-down controllers, DL_ goes low (to turn on
PFETs). The external MOSFET then turns off when the
internally level-shifted sawtooth rises above CC_ or
when the maximum duty cycle is exceeded. The switch
remains off until the start of the next cycle. A transcon­ductance error amplifier forms an integrator at CC_ so
that high DC loop gain and accuracy can be main­tained. In step-up and step-down controllers, the FB_
threshold is 1.25V, and higher FB_ voltages reduce the
MOSFET duty cycle. In inverting controllers, the FB_
threshold is 0V, and lower (more negative) FB_ volt­ages reduce the MOSFET duty cycle.

Auxiliary controllers do not start until the step-up DC-DC
output is in regulation. If the step-up, step-down, or any
of the auxiliary controllers remains faulted for 100,000

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

16 ______________________________________________________________________________________

OSC cycles, then all MAX1584/MAX1585 channels latch
off.

Maximum Duty Cycle

The MAX1584/MAX1585 auxiliary PWM controllers have
a guaranteed maximum duty cycle of 80%. In boost
designs that employ continuous current, the maximum
duty cycle limits the boost ratio so that:

1 - VIN/ V

OUT

≤ 80%

With discontinuous inductor current, no such limit exists
for the input/output ratio since the inductor has time to
fully discharge before the next cycle begins.

AUX1

AUX1 can be used for conventional DC-DC boost and
flyback designs (Figure 5). Its output (DL1) is designed

Figure 3. AUX Controller Functional Diagrams

FB_

CC_

LEVEL
SHIFT

0.85
REF

REFI

(REFI RAMPS FROM

0V TO REF IN 1024 OSC

REF

OSC

MAX1584 AUX1 AND

AUX2, MAX1585 AUX1

STEP-UP CONTROLLER

SOFT-START

CYCLES)

CLK

R

S

FAULT

PROTECTION

FB3

CC3

REF

Q

ENABLE

0.85
REF

REFI

0V TO REF IN 1024 OSC

FB2

CC2

DL_

REF

OSC

MAX1584 AND MAX1585

LEVEL
SHIFT

SOFT-START

(REFI RAMPS FROM

CYCLES)

REFI

0.85
REF

AUX3 STEP-DOWN

CONTROLLER

R

S

LEVEL

SHIFT

SOFT-START

(REFI RAMPS FROM

REF TO 0V IN 1024 OSC

CYCLES)

Q

MAX1585 AUX2

INVERTING CONTROLLER

R

S

CLK

FAULT

PROTECTION

DL3

Q

DL2

ENABLE

CLK

OSC

FAULT

PROTECTION

ENABLE

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 17

to drive an N-channel MOSFET. Its feedback (FB1)
threshold is 1.25V.

AUX2

In the MAX1584, AUX2 is identical to AUX1.

In the MAX1585, AUX2 is an inverting controller that
generates a regulated negative output voltage, typically
for CCD and LCD bias. This is handy in height-limited
designs where transformers might not be desired.

The AUX2 MOSFET driver (DL2) in the MAX1585 is
designed to drive P-channel MOSFETs. DL2 swings
from GND to PVSU. See Figure 8 for a typical inverter
configuration.

AUX3 DC-DC Step-Down Controller

AUX3 can be used for conventional DC-DC step-down
(buck) designs (Figure 1). Its output (DL3) is designed to
drive a P-channel MOSFET and swings from GND to
PVSU. Its feedback (FB3) threshold is 1.25V.

Master/Slave Configurations

The MAX1584/MAX1585 support the MAX1801 slave
PWM controllers that obtain input power, a voltage ref­erence, and an oscillator signal directly from the
MAX1584/MAX1585 master. The master/slave configu­ration allows channels to be easily added and mini­mizes system cost by eliminating redundant circuitry.
The slaves also control the harmonic content of noise
since their operating frequency is synchronized to that
of the MAX1584/MAX1585 master converter. A
MAX1801 connection to the MAX1584/MAX1585 is
shown in Figure 12.

Status Outputs (

SDOK, AUX1OK

, SCF)

The MAX1584/MAX1585 include three versatile status
outputs that can provide information to the system. All
are open-drain outputs and can directly drive MOSFET
switches to facilitate sequencing, disconnect loads
during overloads, or perform other hardware-based
functions.

SDOK pulls low when the step-down has successfully
completed soft-start. SDOK goes high impedance in
shutdown, overload, and thermal limit. A typical use for
SDOK is to enable 3.3V power to the CPU I/O after the
CPU core is powered up (Figure 13), thus providing safe
sequencing in hardware without system intervention.

AUX1OK pulls low when the AUX1 controller has suc­cessfully completed soft-start. AUX1OK goes high
impedance in shutdown, overload, and thermal limit. A
typical use for AUX1OK is to drive a P-channel MOSFET
that gates 5V power to the CCD until the +15V CCD bias
(generated by AUX1) is powered up (Figure 14).

SCF goes high (high impedance, open drain) when
overload protection occurs. Under normal operation,
SCF pulls low. SCF can drive a high-side P-channel
MOSFET switch that can disconnect a load during
power-up or when a channel turns off in response to a
logic command or an overload. Several connections
are possible for SCF. One is shown in Figure 15, where
SCF provides load disconnect for the step-up on fault
and power-up.

Soft-Start

The MAX1584/MAX1585 channels feature a soft-start
function that limits inrush current and prevents exces­sive battery loading at startup by ramping the output
voltage of each channel up to the regulation voltage.
This is accomplished by ramping the internal reference
inputs to each channel error amplifier from 0V to the

1.25V reference voltage over a period of 4096 oscillator
cycles (16ms at 500kHz) when initial power is applied
or when a channel is enabled. Soft-start is not included
in the step-up converter in order to avoid limiting start­up capability with loading.

The step-down soft-start ramp takes half the time (2048
clock cycles) of the other channel ramps. This allows
the step-down and AUX3 output (when set to 3.3V) to
track each other and rise at nearly the same dV/dt rate
on power-up. Once the step-down output reaches its
regulation point (1.5V or 1.8V typ), the AUX3 output
(3.3V typ) continues to rise at the same ramp rate.

Fault Protection

The MAX1584/MAX1585 have robust fault and overload
protection. After power-up, the device is set to detect
an out-of-regulation state that could be caused by an
overload or short. If any DC-DC converter channel
(step-up, step-down, or any of the auxiliary controllers)
remains faulted for 100,000 clock cycles (200ms at
500kHz), then all outputs latch off until the step-up DC­DC converter is reinitialized by the ONSU pin or by
cycling the input power. The fault-detection circuitry for
any channel is disabled during its initial turn-on soft­start sequence.

An exception to the standard fault behavior is that there
is no 100,000 clock-cycle delay in entering the fault
state if the step-up output (PVSU) is dragged below its

2.5V UVLO threshold or is shorted. The step-up UVLO
immediately triggers and shuts down all channels. The
step-up then continues to attempt to start. If the step-up
output short remains, these attempts do not succeed
since PVSU remains near ground.

If a soft-short or overload remains on PVSU, the startup
oscillator switches the internal N-channel MOSFET, but
fault is retriggered if regulation is not achieved by the

MAX1584/MAX1585

end of the soft-start interval. If PVSU is dragged below
the input, the overload is supplied by the body diode of
the internal synchronous rectifier or by a Schottky diode
connected from the battery to PVSU. If desired, this
overload current can be interrupted by a P-channel
MOSFET controlled by SCF, as shown in Figure 15.

Reference

The MAX1584/MAX1585 have internal 1.250V refer­ences. Connect a 0.1µF ceramic bypass capacitor from
REF to GND within 0.2in (5mm) of the REF pin. REF can
source up to 200µA and is enabled when ONSU is high
and PVSU is above 2.5V. The auxiliary controllers and
MAX1801 slave controllers (if connected) each sink up
to 30µA REF current during startup. If the application
requires that REF be loaded beyond 200µA, buffer REF
with a unity-gain amplifier or op amp.

Oscillator

All MAX1584/MAX1585 DC-DC converter channels
employ fixed-frequency PWM operation. The operating
frequency is set by an RC network at the OSC pin. The
range of usable settings is 100kHz to 1MHz. When
MAX1801 slave controllers are added, they operate at
the frequency set by OSC.

The oscillator uses a comparator, a 150ns one-shot,
and an internal NFET switch in conjunction with an
external timing resistor and capacitor (Figure 4). When
the switch is open, the capacitor voltage exponentially
approaches the step-up output voltage from zero with a
time constant given by the product of R

OSC

and C

OSC

.
The comparator output switches high when the capaci­tor voltage reaches V

REF

(1.25V). In turn, the one-shot

activates the internal MOSFET switch to discharge the
capacitor within a 150ns interval, and the cycle
repeats. The oscillation frequency changes as the main
output voltage ramps upward following startup. The
oscillation frequency is then constant once the main
output is in regulation.

Low-Voltage Startup Oscillator

The MAX1584/MAX1585 internal control and reference­voltage circuitry receive power from PVSU and do not
function when PVSU is less than 2.5V. To ensure low­voltage startup, the step-up employs a low-voltage
startup oscillator that activates at 0.9V if a Schottky rec­tifier is connected from V

BATT

to PVSU (1.1V with no
Schottky rectifier). The startup oscillator drives the inter­nal N-channel MOSFET at LXSU until PVSU reaches

2.5V, at which point voltage control is passed to the
current-mode PWM circuitry.

Once in regulation, the MAX1584/MAX1585 operate
with inputs as low as 0.7V since internal power for the
IC is supplied by PVSU. At low input voltages, the step­up can have difficulty starting into heavy loads (see the
Minimum Startup Voltage vs. Load Current graph in the
Typical Operating Characteristics section); however,
this can be remedied by connecting an external P­channel load switch driven by SCF so the load is not
connected until the PVSU is in regulation (Figure 15).

ON_ Control Inputs

The step-up converter activates with a high input at
ONSU. The step-down and auxiliary DC-DC converters
1, 2, and 3 activate with a high input at ONSD, ON1,
ON2, and ON3, respectively. The step-down and auxil-

5-Channel Slim DSC Power Supplies

18 ______________________________________________________________________________________

Figure 4. Oscillator Functional Diagram

Figure 5. +15V LCD Bias with Basic Boost Topology

V

OSC

SU

R

OSC

C

OSC

V

REF

(1.25V)

150ns

ONE-SHOT

MAX1584
MAX1585
(PARTIAL)

AUX

PWM

PVSU

DL_

TO

V

BATT

+15V

Q1

D6

50mA
LCD

MAX1584
MAX1585

FB_

NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1 OR
AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585

iary converters and cannot be activated until PVSU is in
regulation. For automatic startup, connect ON_ to PVSU
or a logic level greater than 1.6V.

Design Procedure

Setting the Switching Frequency

Choose a switching frequency to optimize external
component size or circuit efficiency for the particular
application. Typically, switching frequencies between
400kHz and 500kHz offer a good balance between
component size and circuit efficiency—higher frequen­cies generally allow smaller components, and lower fre­quencies give better conversion efficiency. The
switching frequency is set with an external timing resistor
(R

OSC

) and capacitor (C

OSC

). At the beginning of a
cycle, the timing capacitor charges through the resistor
until it reaches V

REF

. The charge time, t1, is as follows:

t1 = -R

OSC

x C

OSC

x ln(1 - 1.25 / V

PVSU

)

The capacitor voltage then decays to zero over time t

2

= 150ns. The oscillator frequency is as follows:

f

OSC

= 1 / (t1+ t2)

f

OSC

can be set from 100kHz to 1MHz. Choose C

OSC

between 22pF and 470pF. Determine R

OSC

:

R

OSC

= (150ns - 1 / f

OSC

) / (C

OSC

x ln[1 - 1.25

V

PVSU

])

See the Typical Operating Characteristics section for
f

OSC

vs. R

OSC

using different values of C

OSC

.

Setting Output Voltages

The MAX1584/MAX1585 step-up and step-down con­verters and the AUX1 controllers have resistor­adjustable output voltages. When setting the voltage for
all channels except AUX2 on the MAX1585, connect a
resistive voltage-divider from the output voltage to the
corresponding FB_ input. The FB_ input bias current is
less than 100nA, so choose the low-side (FB_-to-GND)
resistor (RL) to be 100kΩ or less. Then calculate the
high-side (output-to-FB_) resistor (RH):

RH= RL[(V

OUT

/ 1.25) - 1]

AUX2 is an inverter on the MAX1585, so the FB2
threshold on the MAX1585 is 0V. To set the MAX1585
AUX2 negative output voltage, connect a resistive volt­age-divider from the negative output to the FB2 input,
and then to REF. The FB2 input bias current is less than
100nA, so choose the REF-side (FB2-to-REF) resistor
(R

REF

) to be 100kΩ or less. Then calculate the top-side

(negative output-to-FB2) resistor:

R

TOP

= R

REF

(-V

OUT(AUX2)

/ 1.25)

General Filter-Capacitor Selection

The input capacitor in a DC-DC converter reduces cur­rent peaks drawn from the battery or other input power
source and reduces switching noise in the controller.
The impedance of the input capacitor at the switching
frequency should be less than that of the input source
so high-frequency switching currents do not pass
through the input source.

The output capacitor keeps output ripple small and
ensures control-loop stability. The output capacitor
must also have low impedance at the switching fre­quency. Ceramic, polymer, and tantalum capacitors
are suitable, with ceramic exhibiting the lowest ESR
and high-frequency impedance.

Output ripple with a ceramic output capacitor is
approximately:

V

RIPPLE

= I

L(PEAK)

[1 / (2π x f

OSC

x C

OUT

)]

If the capacitor has significant ESR, the output ripple
component due to capacitor ESR is:

V

RIPPLE(ESR)

= I

L(PEAK)

x ESR

Output capacitor specifics are also discussed in each
converter’s Compensation section.

Step-Up Component Selection

The external components required for the step-up are
an inductor, an input and output filter capacitor, and a
compensation RC.

The inductor is typically selected to operate with contin­uous current for best efficiency. An exception might be
if the step-up ratio, (V

OUT

/ VIN), is greater than 1 / (1 -

D

MAX

), where D

MAX

is the maximum PWM duty factor

of 80%.

When using the step-up channel to boost from a low
input voltage, loaded startup is aided by connecting a
Schottky diode from the battery to PVSU. See the
Minimum Startup Voltage vs. Load Current graph in the
Typical Operating Characteristics section.

Step-Up Inductor

In most step-up designs, a reasonable inductor value
(L

IDEAL

) can be derived from the following equation,
which sets continuous peak-to-peak inductor current at
half the DC inductor current:

L

IDEAL

= [2V

IN(MAX)

x D(1 - D)] / (I

OUT

x f

OSC

)

where D is the duty factor given by:

D = 1 - (V

IN

/ V

OUT

)

Given L

IDEAL

, the consistent peak-to-peak inductor cur-

rent is 0.5 x I

OUT

/ (1 - D). The peak inductor current is

as follows:

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 19

MAX1584/MAX1585

I

IND(PK)

= 1.25 x I

OUT

/ (1 - D)

Inductance values smaller than L

IDEAL

can be used to
reduce inductor size; however, if much smaller values are
used, inductor current rises and a larger output capaci­tance might be required to suppress output ripple.

Step-Up Compensation

The inductor and output capacitor are usually chosen
first in consideration of performance, size, and cost.
The compensation resistor and capacitor are then cho­sen to optimize control-loop stability. In some cases, it
helps to readjust the inductor or output capacitor value
to get optimum results. For typical designs, the compo­nent values in the circuit of Figure 1 yield good results.

The step-up converter employs current-mode control,
thereby simplifying the control-loop compensation.
When the converter operates with continuous inductor
current (typically the case), a right-half-plane zero
appears in the loop-gain frequency response. To
ensure stability, the control-loop gain should cross over
(drop below unity gain) at a frequency (fC) much less
than that of the right-half-plane zero.

The relevant characteristics for step-up channel com­pensation are as follows:

• Transconductance (from FBSU to CCSU), g

MEA

(135µS)

• Current-sense amplifier transresistance, R

CS

(0.3V/A)

• Feedback regulation voltage, VFB(1.25V)

• Step-up output voltage, VSU, in V

• Output load equivalent resistance, R

LOAD

, in

Ω = V

SUOUT

/ I

LOAD

The key steps for step-up compensation are as follows:

1) Place fCsufficiently below the right-half-plane zero

(RHPZ) and calculate CC.

2) Select RCbased on the allowed load-step transient.

RCsets a voltage delta on the CCpin that corre­sponds to load-current step.

3) Calculate the output-filter capacitor (C

OUT

) required

to allow the RCand CCselected.

4) Determine if CPis required (if calculated to be >10pF).

For continuous conduction, the right-half-plane zero fre­quency (f

RHPZ

) is given by the following:

f

RHPZ

= V

SUOUT

(1 - D)2 / (2π x L x I

LOAD

)

where D = the duty cycle = 1 - (V

IN

/ V

OUT

), L is the

inductor value, and I

LOAD

is the maximum output cur­rent. Typically, target crossover (fC) for 1/6 of the
RHPZ. For example, if we assume f

OSC

= 500kHz, V

IN

= 2.5V, V

OUT

= 5V, and I

OUT

= 0.5A, then R

LOAD

=

10Ω. If we select L = 4.7µH, then:

f

RHPZ

= 5 (2.5 / 5)2 / (2π x 4.7 x 10-6x 0.5) = 84.65kHz

Choose fC= 14kHz. Calculate CC:

CC= (V

FB

/ V

OUT

)(R

LOAD

/ RCS)(gM/ 2π x fC)(1 - D)
= (1.25 / 5)(10 / 0.3) x (135µS / (6.28 x 14kHz) (2/5)
= 6.4nF

Choose 6.8nF.

Now select RCso transient-droop requirements are
met. As an example, if 4% transient droop is allowed,
the input to the error amplifier moves 0.04 x 1.25V, or
50mV. The error-amp output drives 50mV x 135µS, or

6.75µA across RCto provide transient gain. Since the
current-sense transresistance is 0.3V/A, the value of R

C

that allows the required load step swing is as follows:

RC= 0.3 I

IND(PK)

/ 6.75µA

In a step-up DC-DC converter, if L

IDEAL

is used, output

current relates to inductor current by:

I

IND(PK)

= 1.25 x I

OUT

/ (1 - D) = 1.25 x I

OUT

x V

OUT

/

V

IN

So for a 500mA output load step with VIN= 2.5V and
V

OUT

= 5V:

RC= [1.25(0.3 x 0.5 x 5) / 2)] / 6.75µA = 69.4kΩ

Note that the inductor does not limit the response in this
case since it can ramp at 2.5V / 4.7µH, or 530mA/µs.

The output filter capacitor is then chosen so the C

OUT

R

LOAD

pole cancels the RCCCzero:

C

OUT

x R

LOAD

= RCx C

C

For the example:

C

OUT

= 68kΩ x 6.8nF / 10Ω = 46µF

Choose 47µF for C

OUT

. If the available C

OUT

is sub­stantially different from the calculated value, insert the
available C

OUT

value into the above equation and

recalculate RC. Higher substituted C

OUT

values allow a
higher RC, which provides higher transient gain and
consequently less transient droop.

If the output filter capacitor has significant ESR, a zero
occurs at the following:

Z

ESR

= 1 / (2π x C

OUT

x R

ESR

)

5-Channel Slim DSC Power Supplies

20 ______________________________________________________________________________________

If Z

ESR

> fC, it can be ignored, as is typically the case

with ceramic output capacitors. If Z

ESR

is less than fC,

it should be cancelled with a pole set by capacitor C

P

connected from CCSU to GND:

CP= C

OUT

x R

ESR

/ R

C

If CPis calculated to be <10pF, it can be omitted.

Step-Down Component Selection

Step-Down Inductor

The external components required for the step-down
are an inductor, input and output filter capacitors, and
a compensation RC network.

The MAX1585/1585 step-down converter provides best
efficiency with continuous inductor current. A reason­able inductor value (L

IDEAL

) can be derived from the

following:

L

IDEAL

= [2(VIN) x D(1 - D)] / I

OUT

x f

OSC

which sets the peak-to-peak inductor current at half the
DC inductor current. D is the duty cycle:

D = V

OUT

/ V

IN

Given L

IDEAL

, the peak-to-peak inductor current is 0.5 x

I

OUT

. The absolute peak inductor current is 1.25 x I

OUT

.

Inductance values smaller than L

IDEAL

can be used to
reduce inductor size; however, if much smaller values
are used, inductor current rises and a larger output
capacitance may be required to suppress output ripple.
Larger values than L

IDEAL

can be used to obtain higher

output current, but with typically larger inductor size.

Step-Down Compensation

The relevant characteristics for step-down compensa­tion are as follows:

• Transconductance (from FBSD to CCSD), g

MEA

(135µS)

• Current-sense amplifier transresistance, RCS(0.6V/A)

• Feedback regulation voltage, VFB(1.25V)

• Step-down output voltage, VSD, in V

• Output load equivalent resistance, R

LOAD

, in

Ω = V

SD

/ I

LOAD

The key steps for step-down compensation are as fol­lows:

1) Set the compensation RC zero to cancel the R

LOAD

C

OUT

pole.

2) Set the loop crossover below 1/10 the switching fre-

quency.

If we assume VIN= 3.5V, V

OUT

= 1.5V, and I

OUT

=

250mA, then R

LOAD

= 6Ω.

If we select f

OSC

= 500kHz and L = 22µH,

choose fC= 24kHz and calculate CC:

CC= (V

FB

/ V

OUT

)(R

LOAD

/ RCS)(gM/ 2π x fC)

= (1.25 / 1.5)(6 / 0.6) x (135µS / (6.28 x 40kHz))

= 4.5nF

Choose 4.7nF.

Now select RCso transient-droop requirements are met.
For example, if 4% transient droop is allowed, the input
to the error amplifier moves 0.04 x 1.25V, or 50mV. The
error-amp output drives 50mV x 135µS, or 6.75µA across
RCto provide transient gain. Since the current-sense
transresistance is 0.6V/A, the value of RCthat allows the
required load step swing is as follows:

RC= 0.6 x I

IND(PK)

/ 6.75µA

In a step-down DC-DC converter, If L

IDEAL

is used, out-

put current relates to inductor current by the following:

I

IND(PK)

= 1.25 x I

OUT

So for a 250mA output load step with VIN= 3.5V and
V

OUT

= 1.5V:

RC= (1.25 x 0.6 x 0.25) / 6.75µA = 27.8kΩ

Choose 27kΩ.

The inductor does somewhat limit the response in this
case since it ramps at (V

IN

- V

OUT

) / 22µH, or (3.5 - 1.5)

/ 22µH = 90mA/µs.

The output filter capacitor is then chosen so the C

OUT

R

LOAD

pole cancels the RCCCzero:

C

OUT

x R

LOAD

= RCx C

C

For the example:

C

OUT

= 27kΩ x 4.7nF / 6Ω = 21µF

Choose 22µF or greater.

If the output filter capacitor has significant ESR, a zero
occurs at:

Z

ESR

= 1 / (2π x C

OUT

x R

ESR

)

If Z

ESR

> fC, it can be ignored, as is typically the case

with ceramic output capacitors. If Z

ESR

is less than fC,

it should be cancelled with a pole set by capacitor C

P

connected from CCSD to GND:

CP= C

OUT

x R

ESR

/ R

C

If CPis calculated to be <10pF, it can be omitted.

AUX Controller Component Selection

External MOSFET

MAX1584/MAX1585 AUX1(step-up) controllers drive
external logic-level N-channel MOSFETs. AUX3 (step­down) controllers drive P-channel MOSFETs. AUX2
(step-up) on the MAX1584 drives an N channel, while
AUX2 (inverting) on the MAX1585 drives a P channel.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 21

MAX1584/MAX1585

Significant MOSFET selection parameters are as fol­lows:

• On-resistance (R

DS(ON)

)

• Maximum drain-to-source voltage (V

DS(MAX)

)

• Total gate charge (QG)

• Reverse transfer capacitance (C

RSS

)

DL1 and DL3 swing between PVSU and GND. DL2
swings between INDL2 and GND. Use a MOSFET with
on-resistance specified at or below the DL_ drive volt­age. The gate charge, Q

G

, includes all capacitance
associated with charging the gate and helps to predict
MOSFET transition time between on and off states.
MOSFET power dissipation is a combination of on­resistance and transition losses. The on-resistance loss
is as follows:

P

RDSON

= D x I

L

2

x R

DS(ON)

where D is the duty cycle, ILis the average inductor
current, and R

DS(ON)

is the MOSFET on-resistance. The

transition loss is approximately:

P

TRANS

= (V

OUT

x ILx f

OSC

x tT) / 3

where V

OUT

is the output voltage, ILis the average

inductor current, f

OSC

is the switching frequency, and
tTis the transition time. The transition time is approxi­mately QG/ IG, where QGis the total gate charge, and
IGis the gate-drive current (0.5A typ). The total power
dissipation in the MOSFET is as follows:

P

MOSFET

= P

RDSON

+ P

TRANS

Diode

For most AUX applications, a Schottky diode rectifies
the output voltage. Schottky low forward voltage and
fast recovery time provide the best performance in
most applications. Silicon signal diodes (such as
1N4148) are sometimes adequate in low-current
(<10mA), high-voltage (>10V) output circuits where the
output voltage is large compared to the diode forward
voltage.

AUX Compensation

The auxiliary controllers employ voltage-mode control
to regulate their output voltage. Optimum compensa­tion depends on whether the design uses continuous or
discontinuous inductor current.

AUX Step-Up, Discontinuous Inductor Current

When the inductor current falls to zero on each switch­ing cycle, it is described as discontinuous. The inductor
is not utilized as efficiently as with continuous current,
but in light-load applications, this often has little nega­tive impact since the coil losses may already be low
compared to other losses. A benefit of discontinuous

inductor current is more flexible loop compensation, and
no maximum duty-cycle restriction on boost ratio.

To ensure discontinuous operation, the inductor must
have a sufficiently low inductance to fully discharge on
each cycle. This occurs when:

L < [V

IN

2

(V

OUT

- VIN) / V

OUT

3

] [R

LOAD

/ (2f

OSC

)]

A discontinuous current boost has a single pole at the
following:

FP= (2V

OUT

- VIN) / (2π x R

LOAD

x C

OUT

x V

OUT

)

Choose the integrator cap so the unity-gain crossover,
fC, occurs at f

OSC

/ 10 or lower. For many AUX circuits,
such as those powering motors, LEDs, or other loads
that do not require fast transient response, it is often
acceptable to overcompensate by setting fCat f

OSC

/

20 or lower.

CC is then determined by the following:

CC= [2V

OUT

x VIN/ ((2V

OUT

- VIN) x V

RAMP

)] [V

OUT

/

(K(V

OUT

- VIN))]

1/2

[(VFB/ V

OUT

)(gM/ (2π x fC))]

where:

K = 2L x f

OSC

/ R

LOAD

and V

RAMP

is the internal voltage ramp of 1.25V.

The CC RCzero is then used to cancel the fPpole, so:

RC= R

LOAD

x C

OUT

x V

OUT

/ [(2V

OUT

- VIN) x CC]

AUX Step-Up, Continuous Inductor Current

Continuous inductor current can sometimes improve
boost efficiency by lowering the ratio between peak
inductor current and output current. It does this at the
expense of a larger inductance value that requires larg­er size for a given current rating. With continuous
inductor-current boost operation, there is a right-half­plane zero, Z

RHP

, at the following:

Z

RHP

= (1 - D)2R

LOAD

/ (2π x L)

where (1 - D) = V

IN

/ V

OUT

(in a boost converter)

There is a complex pole pair at the following:

f0= V

OUT

/ [2π x VIN(L x C

OUT

)

1/2

]

If the zero due to the output capacitor capacitance and
ESR is less than 1/10 the right-half-plane zero:

Z

COUT

= 1 / (2π x C

OUT

x R

ESR

) < Z

RHP

/ 10

Then choose CCso the crossover frequency fC occurs
at Z

COUT

. The ESR zero provides a phase boost at

crossover:

CC = (V

IN

/ V

RAMP

)(V

FB

/ V

OUT

)(gM/ (2π x Z

COUT

))

Choose RCto place the integrator zero, 1 / (2π x RC x
CC), at f0 to cancel one of the pole pairs:

RC= VIN(L x C

OUT

)

1/2

/ (V

OUT

x CC)

5-Channel Slim DSC Power Supplies

22 ______________________________________________________________________________________

If Z

COUT

is not less than Z

RHP

/ 10 (as is typical with
ceramic output capacitors) and continuous conduction
is required, then cross the loop over before Z

RHP

and f0:

fC< f

0SC

/ 10, and fC< Z

RHP

/ 10

In that case:

CC = (V

IN

/ V

RAMP

)(V

FB

/ V

OUT

)(gM/ (2π x fC))

Place:

1 / (2π x RC x CC) = 1 / (2π x R

LOAD

x C

OUT

), so that

RC= R

LOAD

x C

OUT

/ C

C

Or, reduce the inductor value for discontinuous operation.

AUX3 Step-Down Compensation

It is expected that most AUX3 step-down applications
employ continuous inductor current to optimize induc­tor size and efficiency. To ensure stability, the control­loop gain should cross over (drop below unity gain) at
a frequency (fC) much less than that of the switching
frequency.

The relevant characteristics for voltage-mode step­down compensation are as follows:

• Transconductance (from FB3 to CC3), g

MEA

(135µS)

• Oscillator ramp voltage, V

RAMP

(1.25V)

• Feedback regulation voltage, VFB(1.25V)

• Output voltage, V

OUT3

, in V

• Output load equivalent resistance, R

LOAD

, in Ω =

V

OUT3

/ I

LOAD

• Characteristic impedance of the LC output filter, R

O

= (L / C)

1/2

The key steps for AUX3 step-down compensation are
as follows:

1) Place fCsufficiently below the switching frequency

(f

OSC

/ 10).

2) Calculate C

OUT

.

3) Calculate the complex pole pair due to the output

LC filter.

4) Add two zeros to cancel the complex pole pair.

5) Add two high-frequency poles to optimize gain and

phase margin.

If we assume VIN= 5V, V

OUT

= 3.3V, and I

OUT

=

300mA, then R

LOAD

= 11Ω. If we select f

OSC

= 500kHz
and L = 10µH, select the crossover frequency to be
1/10 the OSC frequency:

fC= f

OSC

/ 10 = 50kHz

For 3.3V output, select R14 = 30.1kΩ and R15 =

18.2kΩ. See the Setting Output Voltages section.

Calculate the equivalent impedance, REQ:

R

EQ

= R

SOURCE

+ RL+ ESR + R

DS(ON)

where R

SOURCE

is the output impedance of the source
(this is the output impedance of the step-up converter
when the AUX3 step-down is powered from the step­up), RL is the inductor DC resistance, ESR is the filter­capacitor equivalent resistance, and R

DS(ON)

is the

on-resistance of the external MOSFET.

The output impedance of the step-up converter
(R

SOURCE

) is approximately 1Ω at f0. Since the sum of

RL+ ESR + R

DS(ON)

is small compared to 1Ω, assume

R

EQ

= 1Ω. Choose C

OUT

so ROis less than R

EQ

/ 2:

C

OUT

> L / [(R

EQ

/ 2)2] = 10µH / 0.25 = 40µF

Choose C

OUT

= 47µF:

C4 = (VIN/ V

RAMP

)(1 / [2π x R14 x fC])

= (5 / 1.25)(1/ [2π x 30.1k x 50kHz) = 423pF

Choose C4 = 470pF.

Cancel one pole of the complex pole pair by placing
the R4 C4 zero at 0.75 f0. The complex pole pair is at
the following:

f0= 1 / [2π(L x C

OUT

)

1/2

]

= 1 / [2π(10µH x 47µF)

1/2

] = 7.345kHz

Choose R4 = 1 / (2π x C4 x 0.75 x f0)

= 1 / (2π x 470pF x 0.75 x 7.345kHz)

z

Choose R4 = 61.9kΩ (standard 1% value). Ensure that
R4 > 2 / g

MEA

= 14.8kΩ. If it is not greater, reselect

R14 and R15.

Cancel the second pole of the complex pole pair by
placing the R14 C20 zero at 1.25 x f

0

.

C20 = 1 / (2π x R14 x 1.25 x f0)

= 1 / (2π x 30.1k x 1.25 x 7.345kHz) = 576pF

Choose C20 = 560pF.

Roll off the gain below the switching frequency by plac­ing a pole at f

OSC

/ 2:

R22 = 1 / (2π x C20 [f

OSC

/ 2])

= 1 / (2π x 560pF x 250kHz) = 1.137kΩ

Choose R22 = 1.2kΩ.

If the output filter capacitor has significant ESR, a zero
occurs at the following:

Z

ESR

= 1 / (2π x C

OUT

x R

ESR

)

Use the R4 C22 pole to cancel the ESR zero:

C22 = C

OUT

x R

ESR

/ R4

If C22 is calculated to be <10pF, it can be omitted.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 23

MAX1584/MAX1585

MAX1585 AUX2 Inverter Compensation,

Discontinuous Inductor Current

If the load current is very low (40mA or less), discontin­uous current is preferred for simple loop compensation
and freedom from duty-cycle restrictions on the inverter
input-output ratio. To ensure discontinuous operation,
the inductor must have a sufficiently low inductance to
fully discharge on each cycle. This occurs when:

L < [V

IN

/ (|V

OUT

| + VIN)]2R

LOAD

/ (2f

OSC

)

A discontinuous current inverter has a single pole at:

fP= 2 / (2π x R

LOAD

x C

OUT

)

Choose the integrator cap so the unity-gain crossover,
fC, occurs at f

OSC

/ 10 or lower. Note that for many AUX
circuits that do not require fast transient response, it is
often acceptable to overcompensate by setting fCat
f

OSC

/ 20 or lower.

CC is then determined by the following:

CC= [V

IN

/ (K

1/2

x V

RAMP

][V

REF

/ (V

OUT

+ V

REF

)] [gM/

(2π x fC)]

where:

K = 2L x f

OSC

/ R

LOAD,

and V

RAMP

is the internal volt-

age ramp of 1.25V.

The CC RCzero then is used to cancel the fPpole, so:

RC= (R

LOAD

x C

OUT

) / (2 CC)

MAX1585 AUX2 Inverter Compensation,

Continuous Inductor Current

Continuous inductor current may be more suitable for
larger load currents (50mA or more). It improves effi­ciency by lowering the ratio between peak inductor cur­rent and output current. It does this at the expense of a
larger inductance value that requires larger size for a
given current rating. With continuous inductor-current
inverter operation, there is a right-half-plane zero,
Z

RHP

, at:

Z

RHP

= [(1 - D)2 / D] x R

LOAD

/ (2π x L)

where D = |V

OUT

| / (|V

OUT

| + VIN) (in an inverter).

There is a complex pole pair at:

f0= (1 - D) / (2π(L x C)

1/2

)

If the zero due to the output-capacitor capacitance and
ESR is less than 1/10 the right-half-plane zero:

Z

COUT

= 1 / (2π x C

OUT

x R

ESR

) < Z

RHP

/ 10

Then choose CCso the crossover frequency, f

C,

occurs

at Z

COUT

. The ESR zero provides a phase boost at

crossover.

CC = (V

IN

/ V

RAMP

)[V

REF

/ (V

REF

+ |V

OUT

|)][gM/ (2π x

Z

COUT

)]

Choose R

C

to place the integrator zero, 1 / (2π x RC x

CC), at f0 to cancel one of the pole pairs:

R

C

= (L x C

OUT

)

1/2

/ [(1 - D) x CC]

If Z

COUT

is not less than Z

RHP

/ 10 (as is typical with
ceramic output capacitors) and continuous conduction
is required, then cross the loop over before Z

RHP

and f0:

fC< f0 / 10, and fC< Z

RHP

/ 10

In that case:

CC = (V

IN

/ V

RAMP

)[V

REF

/ (V

REF

+ |V

OUT

|)][gM/ (2π x fC)]

Place:

1 / (2π x RC x CC) = 1 / (2π x R

LOAD

x C

OUT

), so that

RC= R

LOAD

x C

OUT

/ C

C

Or, reduce the inductor value for discontinuous operation.

Applications Information

LED, LCD, and Other Boost Applications

Any AUX channel can be used for a wide variety of
step-up applications. These include generating 5V or
some other voltage for motor or actuator drive, generat­ing 15V or a similar voltage for LCD bias, or generating
a step-up current source to efficiently drive a series
array of white LEDs to display backlighting. Figures 5
and 6 show examples of these applications.

Multiple-Output Flyback Circuits

Some applications require multiple voltages from a sin­gle converter channel. This is often the case when gen­erating voltages for CCD bias or LCD power. Figure 7
shows a two-output flyback configuration with AUX_.
The controller drives an external MOSFET that switches
the transformer primary. Two transformer secondaries
generate the output voltages. Only one positive output
voltage can be fed back, so the other voltages are set
by the turns ratio of the transformer secondaries. The
load stability of the other secondary voltages depends
on transformer leakage, inductance, and winding resis­tance. Voltage regulation is best when the load on the
secondary that is not fed back is small compared to the
load on the one that is fed back. Regulation also
improves if the load current range is limited. Consult
the transformer manufacturer for the proper design for
a given application.

5-Channel Slim DSC Power Supplies

24 ______________________________________________________________________________________

Transformerless Inverter for Negative CCD

Bias (AUX2, MAX1585)

On the MAX1585, AUX2 is set up to drive an external P­channel MOSFET in an inverting configuration. DL2 dri­ves low to turn on the MOSFET, and FB2 has inverted
polarity and a 0V threshold. This is useful for generating
negative CCD bias without a transformer, particularly
with high pixel-count cameras that have a greater neg­ative CCD load current. Figures 1 and 8 show such a
configuration for the MAX1585.

Boost with Charge Pump for Positive and

Negative Outputs

Another method of producing bipolar output voltages
without a transformer is with an AUX controller and a
charge-pump circuit as shown in Figure 9. When MOS­FET Q1 turns off, the voltage at its drain rises to supply
current to V

OUT+

. At the same time, C1 charges to the

voltage V

OUT+

through D1. When the MOSFET turns on,

C1 discharges through D3, thereby charging C3 to V

OUT-

minus the drop across D3 to create roughly the same
voltage as V

OUT+

at V

OUT-

, but with inverted polarity.

If different magnitudes are required for the positive and
negative voltages, a linear regulator can be used at one
of the outputs to achieve the desired voltages. One such

connection is shown in Figure 10. This circuit is some­what unique in that a positive-output linear regulator is
able to regulate a negative voltage output. It does this by
controlling the charge current flowing to the flying
capacitor rather than directly regulating at the output.

SEPIC Boost-Buck

The MAX1584/MAX1585s’ internal switch step-up and
step-down can be cascaded to make a high-efficiency
boost-buck converter, but it is sometimes desirable to
build a second boost-buck converter with an AUX_
controller.

One type of step-up/step-down converter is the SEPIC,
shown in Figure 11. Inductors L1 and L2 can be sepa­rate inductors or can be wound on a single core and
coupled like a transformer. Typically, a coupled inductor
improves efficiency since some power is transferred
through the coupling so less power passes through the
coupling capacitor (C2). Likewise, C2 should have low
ESR to improve efficiency. The ripple-current rating must
be greater than the larger of the input and output cur­rents. The MOSFET (Q1) drain-source voltage rating and
the rectifier (D1) reverse-voltage rating must exceed the
sum of the input and output voltages. Other types of
step-up/step-down circuits are a flyback converter and a
step-up converter followed by a linear regulator.

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 25

Figure 6. AUX_ Channel Powering a White LED Step-Up
Current Source

Figure 7. +15V and -7.5V CCD Bias with Transformer

TO

V

BATT

1µF

PVSU

1µF

WHITE

LEDS

62

Ω

(FOR 20mA)

NOTE: THIS CIRCUIT CAN
OPERATE WITH AUX1 OR
AUX2 ON THE MAX1584, AND WITH
AUX1 ON THE MAX1585.

DL_

FB_

MAX1585
(PARTIAL)

AUX_
PWM

TO

V

BATT

MAX1584
MAX1585

(PARTIAL)

AUX

PWM

PVSU

NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1
OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585.

DL_

FB_

Q1

D2

+15V
50mA
CCD+

-7.5V
30mA
CCD-

MAX1584/MAX1585

Adding a MAX1801 Slave

The MAX1801 is a 6-pin SOT slave DC-DC controller
that can be connected to generate additional output
voltages. It does not generate its own reference or
oscillator. Instead, it uses the reference and oscillator
of the MAX1584/MAX1585 (Figure 12).

5-Channel Slim DSC Power Supplies

26 ______________________________________________________________________________________

Figure 8. Regulated -7.5V Negative CCD Bias Provided by
Conventional Inverter (MAX1585 Only)

Figure 9. ±15V Output from AUX-Driven Boost with Charge­Pump Inversion

Figure 10. +15V and -7.5V CCD Bias Without Transformer from
AUX-Driven Boost and Charge Pump. A positive linear regula­tor (MAX1616) regulates the negative output of the charge
pump.

Figure 11. SEPIC Converter for Additional Boost-Buck Channel

MAX1585

TO V

BATT

(PARTIAL)

INDL2

DL2

AUX2

INVERTING

PWM

FB2

REF

R

TOP

R

REF

L1

TO V

BATT

AUX_
PWM

10µH

1µF

FB_

PVSU

DL_

MAX1584
MAX1585

(PARTIAL)

NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1
OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585.

D2

C2

1µF

C1

Q1

1µF

D1

R1
1M

R2

90.9k

D3

+15V
20mA

-7.5V
20mA

-7.5V
100mA

TO V

AUX_
PWM

BATT

FB_

PVSU

DL_

MAX1584/MAX1585

(PARTIAL)

SHDNIN

GND

OUT

+1.25V

V

OUT+

+15V
20mA

Ω

NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1
OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585.

MAX1616

FB_

Ω

V

OUT-

-15V

C3

10mA

1µF

INPUT

1-CELL

Li+

V

SU

L2

PV PVSU

PART OF
MAX1584
MAX1585

(PARTIAL)

DL_

FB_

L1

C2

Q1

D1

OUTPUT

3.3V

R1

NOTE: THIS CIRCUIT CAN OPERATE WITH AUX1
OR AUX2 ON THE MAX1584, AND WITH AUX1 ON THE MAX1585.

R2

The MAX1801 controller operation and design are simi­lar to that of the MAX1584/MAX1585 AUX controllers.
All comments in the AUX Controller Component

Selection section also apply to add-on MAX1801 slave
controllers. For more details, refer to the MAX1801 data
sheet.

Using

SDOK

and

AUX1OK

for Power Sequencing

The SDOK goes low when the step-down reaches regu­lation. Some microcontrollers with low-voltage cores
require the high-voltage (3.3V) I/O rail not be powered
up until the core has a valid supply. The circuit in
Figure 13 accomplishes this by driving the gate of a
PFET connected between the 3.3V output and the
processor I/O supply.

Figure 14 shows a similar application where AUX1OK
gates 5V power to the CCD only after the +15V output
is in regulation. Alternately, power sequencing can also
be implemented by connecting RC networks to delay
the appropriate converter ON_ inputs.

Using SCF for Full-Load Startup

The SCF output goes low only after the step-up reaches
regulation. It can be used to drive a P-channel MOSFET
switch that turns off the load of a selected supply in the

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

______________________________________________________________________________________ 27

Figure 12. Adding a PWM Channel with an External MAX1801
Slave Controller

Figure 13. Using

SDOK

to Gate 3.3V Power to CPU After the

Core Voltage Is in Regulation

Figure 14.

AUX1OK

drives an external PFET that switches 5V to

the CCD only after the +15V CCD bias supply is in regulation.

TO BATT

V

OUT

DL

FB

COMP

GND

MAX1801

IN

OSC

REF

DCON

PVSU

OSC

REF

MAX1584
MAX1585
(PARTIAL)

MAX1584/MAX1585

(PARTIAL)

AUX3 V-MODE

STEP-DOWN

PWM

DL3

TO PVSU

3.3V
LOGIC

MAX1584
MAX1585

(PARTIAL)

AUX1
PWM

CURRENT-

MODE

STEP-UP

PWM

PVSU

TO

V

BATT

AUX1OK

PGSU

DL1

FB1

PVSU

LXSU

FBSU

15V

D6

PV

TO

V

BATT

L2

100mA

GATED +5V
TO CCD

V

SU

+5V

FB3

TO V
OR PVSU

TO V

BATT

OR PVSU

BATT

+1.5V

CURRENT-

MODE

STEP-DOWN

PWM

ON3

SDOK

PVSD

LXSD

PGSD

MAX1584/MAX1585

event of an overload. Or, it can remove the load until
the supply reaches regulation, effectively allowing full­load startup. Figure 15 shows such a connection for the
step-up output.

Setting SDOUT Below 1.25V

The step-down feedback voltage is 1.25V. With a stan­dard two-resistor feedback network, the output voltage
can be set to values between 1.25V and the input volt­age. If a step-down output voltage less than 1.25V is
desired, it can be set by adding a third feedback resis­tor from FBSD to a voltage higher than 1.25V (the step­up output is a convenient voltage for this) as shown in
Figure 16.

The equation governing output voltage in Figure 16’s
circuit is as follows:

0 = [(V

SD

- V

FBSD

) / R1] + [(0 - V

FBSD

) / R2] + [(V

SU

-

V

FBSD

) / R3]

where VSDis the output voltage, V

FBSD

is 1.25V, and
VSUis the step-up output voltage. Any available volt­age that is higher than 1.25V can be used as the con­nection point for R3 in Figure 16, and for the VSDterm
in the equation. Since there are multiple solutions for
R1, R2, and R3, the above equation cannot be written
in terms of one resistor. The best method for determin­ing resistor values is to enter the above equation into a

spreadsheet and test estimated resistor values. A good
starting point is with 100kΩ at R2 and R3.

Designing a PC Board

Good PC board layout is important to achieve optimal
performance from the MAX1584/MAX1585. Poor design
can cause excessive conducted and/or radiated noise.

Conductors carrying discontinuous currents and any
high-current path should be made as short and wide as
possible. A separate low-noise ground plane contain­ing the reference and signal grounds should connect to
the power-ground plane at only one point to minimize
the effects of power-ground currents. Typically, the
ground planes are best joined right at the IC.

Keep the voltage-feedback network very close to the
IC, preferably within 0.2in (5mm) of the FB_ pin. Nodes
with high dV/dt (switching nodes) should be kept as
small as possible and should be routed away from
high-impedance nodes such as FB_. Refer to the
MAX1584/MAX1585 evaluation kit data sheet for a full
PC board example.

5-Channel Slim DSC Power Supplies

28 ______________________________________________________________________________________

Figure 15. SCF controls a PFET load switch to disconnect all
5V loads on fault. This also allows full-load startup.

Figure 16. Setting PVSD for Outputs Below 1.25V

Chip Information

TRANSISTOR COUNT: 8234

PROCESS: BiCMOS

V

MAX1584
MAX1585

(PARTIAL)

CURRENT-MODE

STEP-UP

PWM

OK

PWR-ON

OR FAULT

PVSU

LXSU

PGSU

FBSD

SCF

SU

3.3V

PV

TO

V

BATT

L2

V
+5V

SU

PVSU

PV

CURRENT-MODE

STEP-DOWN

FBSD

R3

100k

Ω

MAX1584
MAX1585

(PARTIAL)

V

FBSD

1.25V

R2
100k

Ω

56k

PVSD

10µF

LXSD

PGSD

R1

Ω

4.7µH

22µF

V

0.8V

SD

MAX1584/MAX1585

5-Channel Slim DSC Power Supplies

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 29

© 2003 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

Package Information

(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages

.)

PIN # 1
I.D.

D

C

0.15 C A

D/2

0.15

C B

E/2

E

0.10

C

A

0.08 C

A3

A1

(NE-1) X e

DETAIL A

L

D2

k

e

(ND-1) X e

L

e e

PROPRIETARY INFORMATION

TITLE:

PACKAGE OUTLINE
16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm

APPROVAL

C

L

D2/2

b

0.10 M

E2/2

L

DOCUMENT CONTROL NO.

21-0140

C A B

PIN # 1 I.D.

0.35x45

C

E2

L

k

CC

QFN THIN.EPS

L

L

REV.

1

C

2

COMMON DIMENSIONS

NOTES:

1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994.

2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES.

3. N IS THE TOTAL NUMBER OF TERMINALS.

4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1
SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE
ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE.

5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm
FROM TERMINAL TIP.

6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY.

7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION.

8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS.

9. DRAWING CONFORMS TO JEDEC MO220.

10. WARPAGE SHALL NOT EXCEED 0.10 mm.

EXPOSED PAD VARIATIONS

PROPRIETARY INFORMATION

TITLE:

PACKAGE OUTLINE

16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm

21-0140

REV.DOCUMENT CONTROL NO.APPROVAL

2

C

2