QPROX QT110-D, QT110-IS, QT110-S, QT110H-D, QT110H-IS Datasheet

APPLICATIONS -

!

Elevator buttons

!

Toys & games

!

Access systems

!

Pointing devices

!

Appliance control

!

Security systems

!

Light switches

!

Industrial panels

The QT110 / QT110H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch. It
will project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and most kinds of wood. It can also turn
small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with its ability to
self calibrate continuously can lead to entirely new product concepts.

It is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical
switch or button may be found; it may also be used for some material sensing and control applications provided that the presence
duration of objects does not exceed the recalibration timeout interval.

The IC requires only a common inexpensive capacitor in order to function. A bare piezo beeper can be connected to create a ‘tactile’
feedback clicking sound; the beeper itself then doubles as the required external capacitor, and it can also become the sensing
electrode. An LED can also be added to provide visual sensing indication. With a second inexpensive capacitor the device can
operated in 2-wire mode, where both power and signal traverse the same wire pair to a host. This mode allows the sensor to be wired
to a controller with only a twisted pair over long distances.

Power consumption is under 20µA in most applications, allowing operation from Lithium cells for many years. In most cases the power
supply need only be minimally regulated.

The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make the device
survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Even sensitivity is digitally determined and remains
constant in the face of large variations in sample capacitor C

S

and electrode CX. No external switches, opamps, or other analog

components aside from CS are usually required.
The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light

switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power
rail, permitting a simple 2-wire interface. The Quantum-pioneered HeartBeat™ signal is also included, allowing a host controller to
monitor the health of the QT110 continuously if desired. By using the charge transfer principle, the IC delivers a level of performance
clearly superior to older technologies in a highly cost-effective package.

Quantum Research Group Ltd

Copyright © 1999 Quantum Research Group Ltd

R1.01/0106

!!!!

Less expensive than many mechanical switches

!!!!

Projects a ‘touch button’ through any dielectric

!!!!

Turns small objects into intrinsic touch sensors

!!!!

100% autocal for life - no adjustments required

!!!!

Only one external part required - a 1¢ capacitor

!!!!

Piezo sounder direct drive for ‘tactile’ click feedback

!!!!

LED drive for visual feedback

!!!!

2.5 to 5V 20µµµµA single supply operation

!!!!

Toggle mode for on/off control (strap option)

!!!!

10s or 60s auto-recalibration timeout (strap option)

!!!!

Pulse output mode (strap option)

!!!!

Gain settings in 3 discrete levels

!!!!

Simple 2-wire operation possible

!!!!

HeartBeat™ health indicator on output

!!!!

Active Low (QT110), Active High (QT110H) versions

-QT110H-IS-400C to +850C

-QT110-IS-400C to +850C

QT110H-DQT110H-S00C to +700C

QT110-DQT110-S0

0

C to +700C

8-PIN DIPSOICT

A

AVAILABLE OPTIONS

QProx™™™™ QT110 / QT110H

C

HARGE-TRANSFER TOUCH SENSOR

Sns2

Vss

Sns1

GainOpt2

Opt1

Out

Vdd 1

2

3

45

6

7

8

QT110

1 - OVERVIEW

The QT110 is a digital burst mode charge-transfer (QT)
sensor designed specifically for touch controls; it includes all
hardware and signal processing functions necessary to
provide stable sensing under a wide variety of changing
conditions. Only a single low cost, non-critical capacitor is
required for operation.

Figure 1-1 shows the basic QT110 circuit using the device,
with a conventional output drive and power supply
connections. Figure 1-2 shows a second configuration using
a common power/signal rail which can be a long twisted pair
from a controller; this configuration uses the built-in pulse
mode to transmit output state to the host controller (QT110
only).

1.1 BASIC OPERATION

The QT110 employs short, ultra-low duty cycle bursts of
charge-transfer cycles to acquire its signal. Burst mode
permits power consumption in the low microamp range,
dramatically reduces RF emissions, lowers susceptibility to
EMI, and yet permits excellent response time. Internally the
signals are digitally processed to reject impulse noise, using
a 'consensus' filter which requires four consecutive
confirmations of a detection before the output is activated.

The QT switches and charge measurement hardware
functions are all internal to the QT110 (Figure 1-3). A 14-bit
single-slope switched capacitor ADC includes both the
required QT charge and transfer switches in a configuration
that provides direct ADC conversion. The ADC is designed to
dynamically optimize the QT burst length according to the
rate of charge buildup on Cs, which in turn depends on the
values of Cs, Cx, and Vdd. Vdd is used as the charge
reference voltage. Larger values of Cx cause the charge
transferred into Cs to rise more rapidly, reducing available
resolution; as a minimum resolution is required for proper
operation, this can result in dramatically reduced apparent
gain. Conversely, larger values of Cs reduce the rise of
differential voltage across it, increasing available resolution
by permitting longer QT bursts. The value of Cs can thus be
increased to allow larger values of Cx to be tolerated (Figures
4-1, 4-2, 4-3 in Specifications, rear).

The IC is highly tolerant of changes in Cs since it computes
the threshold level ratiometrically with respect to absolute
load, and does so dynamically at all times.

Cs is thus non-critical; as it drifts with temperature, the
threshold algorithm compensates for the drift automatically.

A simple circuit variation is to replace Cs with a bare piezo
sounder (Section 2), which is merely another type of
capacitor, albeit with a large thermal drift coefficient. If C

piezo

is in the proper range, no other external component is
required. If C

piezo

is too small, it can simply be ‘topped up’ with
an inexpensive ceramic capacitor connected in parallel with
it. The QT110 drives a 4kHz signal across SNS1 and SNS2
to make the piezo (if installed) sound a short tone for 75ms
immediately after detection, to act as an audible confirmation.

Option pins allow the selection or alteration of several special
features and sensitivity.

1.2 ELECTRODE DRIVE

The internal ADC treats Cs as a floating transfer capacitor; as
a direct result, the sense electrode can be connected to
either SNS1 or SNS2 with no performance difference. In both
cases the rule Cs >> Cx must be observed for proper
operation. The polarity of the charge buildup across Cs
during a burst is the same in either case.

It is possible to connect separate Cx and Cx’ loads to SNS1
and SNS2 simultaneously, although the result is no different
than if the loads were connected together at SNS1 (or
SNS2). It is important to limit the amount of stray capacitance
on both terminals, especially if the load Cx is already large,

for example by minimizing trace lengths
and widths so as not to exceed the Cx
load specification and to allow for a
larger sensing electrode size if so
desired.

The PCB traces, wiring, and any
components associated with or in contact
with SNS1 and SNS2 will become touch
sensitive and should be treated with
caution to limit the touch area to the
desired location. Multiple touch
electrodes can be used, for example to
create a control button on both sides of
an object, however it is impossible for the
sensor to distinguish between the two
touch areas.

- 2 -

Figure 1-1 Standard mode options

SENSING
ELECTRODE

C

s

10nF

3

46

5

1

+2.5 to 5

72

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

OUTPUT=D C
TIMEOUT=10 Secs
TOGGLE=OFF
GAIN=HIGH

C

x

8

Figure 1-2 2-wire operation, self-powered (QT110 only)

C

x

+

22µF10V AL

10nF

CMOS
GATE

+3V

2.2k Twisted
pair

C

s

8

S E NSING
ELECTRODE

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

3

46

5

1

72

1.3 ELECTRODE DESIGN

1.3.1 E

LECTRODE GEOMETRY AND SIZE

There is no restriction on the shape of
the electrode; in most cases common
sense and a little experimentation can
result in a good electrode design. The
QT110 will operate equally well with
long, thin electrodes as with round or
square ones; even random shapes are
acceptable. The electrode can also be
a 3-dimensional surface or object.
Sensitivity is related to electrode
surface area, orientation with respect
to the object being sensed, object
composition, and the ground coupling
quality of both the sensor circuit and
the sensed object.

If a relatively large electrode surface is
desired, and if tests show that the
electrode has more capacitance than
the QT110 can tolerate, the electrode
can be made into a sparse mesh (Figure 1-4) having lower
Cx than a solid plane. Sensitivity may even remain the same,
as the sensor will be operating in a lower region of the gain
curves.

1.3.2 K

IRCHOFF’S CURRENT LAW

Like all capacitance sensors, the QT110 relies on Kirchoff’s
Current Law (Figure 1-5) to detect the change in capacitance
of the electrode. This law as applied to capacitive sensing
requires that the sensor’s field current must complete a loop,
returning back to its source in order for capacitance to be
sensed. Although most designers relate to Kirchoff’s law with
regard to hardwired circuits, it applies equally to capacitive
field flows. By implication it requires that the signal ground
and the target object must both be coupled together in some
manner for a capacitive sensor to operate properly. Note that
there is no need to provide actual hardwired ground
connections; capacitive coupling to ground (Cx1) is always
sufficient, even if the coupling might seem very tenuous. For
example, powering the sensor via an isolated transformer will
provide ample ground coupling, since there is capacitance
between the windings and/or the transformer core, and from
the power wiring itself directly to 'local earth'. Even when
battery powered, just the physical size of the PCB and the
object into which the electronics is embedded will generally
be enough to couple a few picofarads back to local earth.

1.3.3 V

IRTUAL CAPACITIVE GROUNDS

When detecting human contact (e.g. a fingertip), grounding
of the person is never required. The human body naturally
has several hundred picofarads of ‘free space’ capacitance to
the local environment (Cx3 in Figure 1-5), which is more than
two orders of magnitude greater than that required to create
a return path to the QT110 via earth. The QT110's PCB
however can be physically quite small, so there may be little
‘free space’ coupling (Cx1 in Figure 1-5) between it and the
environment to complete the return path. If the QT110 circuit
ground cannot be earth grounded by wire, for example via
the supply connections, then a ‘virtual capacitive ground’ may
be required to increase return coupling.

A ‘virtual capacitive ground’ can be created by connecting the
QT110’s own circuit ground to:

(1) A nearby piece of metal or metallized housing;
(2) A floating conductive ground plane;
(3) A nail driven into a wall when used with small

electrodes;

(4) A larger electronic device (to which its output might be

connected anyway).

- 3 -

Figure 1-3 Internal Switching & Timing

C

s

C

x

SNS2

SNS1

ELECTRODE

Single -S lope 14-bit

Switched Capacitor ADC

Charge

Amp

Burst Controller

Result

Done

Start

Figure 1-4 Mesh Electrode Geometry

Figure 1-5 Kirchoff's Current Law

Sense Electrode

C

X2

Surrounding environment

C

X3

SENSOR

C

X1

Free-floating ground planes such as metal foils should
maximize exposed surface area in a flat plane if possible. A
square of metal foil will have little effect if it is rolled up or
crumpled into a ball. Virtual ground planes are more effective
and can be made smaller if they are physically bonded to
other surfaces, for example a wall or floor.

1.3.4 F

IELD SHAPING

The electrode can be prevented from sensing in undesired
directions with the assistance of metal shielding connected to
circuit ground (Figure 1-6). For example, on flat surfaces, the
field can spread laterally and create a larger touch area than
desired. To stop field spreading, it is only necessary to
surround the touch electrode on all sides with a ring of metal
connected to circuit ground; the ring can be on the same or
opposite side from the electrode. The ring will kill field
spreading from that point outwards.

If one side of the panel to which the electrode is fixed has
moving traffic near it, these objects can cause inadvertent
detections. This is called ‘walk-by’ and is caused by the fact
that the fields radiate from either surface of the electrode
equally well. Again, shielding in the form of a metal sheet or
foil connected to circuit ground will prevent walk-by; putting a
small air gap between the grounded shield and the electrode
will keep the value of Cx lower and is
encouraged. In the case of the QT110, the
sensitivity is low enough that 'walk-by' should not
be a concern if the product has more than a few
millimeters of internal air gap; if the product is
very thin and contact with the product's back is a
concern, then some form of rear shielding may be
required.

1.3.5 S

ENSITIVITY

The QT110 can be set for one of 3 gain levels
using option pin 5 (Table 1-1). If left open, the
gain setting is high. The sensitivity change is
made by altering the numerical threshold level
required for a detection. It is also a function of
other things: electrode size, shape, and
orientation, the composition and aspect of the

object to be sensed, the thickness and composition of any
overlaying panel material, and the degree of ground coupling
of both sensor and object are all influences.

1.3.5.1 Increasing Sensitivity

In some cases it may be desirable to increase sensitivity
further, for example when using the sensor with very thick
panels having a low dielectric constant.

Sensitivity can often be increased by using a bigger
electrode, reducing panel thickness, or altering panel
composition. Increasing electrode size can have diminishing
returns, as high values of Cx will reduce sensor gain (Figures
4-1 ~ 4-3). Also, increasing the electrode's surface area will
not substantially increase touch sensitivity if its diameter is
already much larger in surface area than the object being
detected. The panel or other intervening material can be
made thinner, but again there are diminishing rewards for
doing so. Panel material can also be changed to one having
a higher dielectric constant, which will help propagate the
field through to the front. Locally adding some conductive
material to the panel (conductive materials essentially have
an infinite dielectric constant) will also help dramatically; for
example, adding carbon or metal fibers to a plastic panel will
greatly increase frontal field strength, even if the fiber density
is too low to make the plastic bulk-conductive.

1.3.5.2 Decreasing Sensitivity

In some cases the QT110 may be too sensitive, even on low
gain. In this case gain can be lowered further by any of a
number of strategies: making the electrode smaller,
connecting a very small capacitor in series with the sense
lead, or making the electrode into a sparse mesh using a
high space-to-conductor ratio (Figure 1-4). A deliberately
added Cx capacitor can also be used to reduce sensitivity
according to the gain curves (see Section 4).

Intermediate levels of gain (e.g. between 'medium' and 'low'
can be obtained by a combination of jumper settings with one
or more of the above strategies.

- 4 -

Figure 1-6

Shielding Against Fringe Fields

Sense

wire

Sense

wire

Unshielded

Electrode

Shielded

Electrode

Figure 2-1 Drift Compensation

Threshold

Signal

Hysteresis

Reference

Output

Pin 7

Low

Pin 6

Medium

None

High

Tie Pin 5 to:Gain

Table 1-1 Gain Setting Strap Options

2 - QT110 SPECIFICS

2.1 SIGNAL PROCESSING

The QT110 processes all signals using 16 bit math, using a
number of algorithms pioneered by Quantum. The algorithms
are specifically designed to provide for high 'survivability' in
the face of all kinds of adverse environmental changes.

2.1.1 D

RIFT COMPENSATION ALGORITHM

Signal drift can occur because of changes in Cx and Cs over
time. It is crucial that drift be compensated for, otherwise
false detections, non-detections, and sensitivity shifts will
follow.

Drift compensation (Figure 2-1) is performed by making the
reference level track the raw signal at a slow rate, but only
while there is no detection in effect. The rate of adjustment
must be performed slowly, otherwise legitimate detections
could be ignored. The QT110 drift compensates using a
slew-rate limited change to the reference level; the threshold
and hysteresis values are slaved to this reference.

Once an object is sensed, the drift compensation mechanism
ceases since the signal is legitimately high, and therefore
should not cause the reference level to change.

The QT110's drift compensation is 'asymmetric': the
reference level drift-compensates in one direction faster than
it does in the other. Specifically, it compensates faster for
decreasing signals than for increasing signals. Increasing
signals should not be compensated for quickly, since an
approaching finger could be compensated for partially or
entirely before even touching the sense pad. However, an
obstruction over the sense pad, for which the sensor has
already made full allowance for, could suddenly be removed
leaving the sensor with an artificially elevated reference level
and thus become insensitive to touch. In this latter case, the
sensor will compensate for the object's removal very quickly,
usually in only a few seconds.

2.1.2 T

HRESHOLD CALCULATION

Sensitivity is dependent on the threshold level as well as
ADC gain; threshold in turn is based on the internal signal
reference level plus a small differential value. The threshold
value is established as a percentage of the absolute signal
level. Thus, sensitivity remains constant even if Cs is altered
dramatically, so long as electrode coupling to the user
remains constant. Furthermore, as Cx and Cs drift, the
threshold level is automatically recomputed in real time so
that it is never in error.

The QT110 employs a hysteresis dropout below the
threshold level of 50% of the delta between the reference and
threshold levels.

2.1.3 MAX ON-D

URATION

If an object or material obstructs the sense pad the signal
may rise enough to create a detection, preventing further
operation. To prevent this, the sensor includes a timer which
monitors detections. If a detection exceeds the timer setting,
the timer causes the sensor to perform a full recalibration.
This is known as the Max On-Duration feature.

After the Max On-Duration interval, the sensor will once again
function normally, even if partially or fully obstructed, to the
best of its ability given electrode conditions. There are two
timeout durations available via strap option: 10 and 60
seconds.

2.1.4 D

ETECTION INTEGRATOR

It is desirable to suppress detections generated by electrical
noise or from quick brushes with an object. To accomplish

this, the QT110 incorporates a detect integration counter that
increments with each detection until a limit is reached, after
which the output is activated. If no detection is sensed prior
to the final count, the counter is reset immediately to zero. In
the QT110, the required count is 4.

The Detection Integrator can also be viewed as a 'consensus'
filter, that requires four detections in four successive bursts to
create an output. As the basic burst spacing is 75ms, if this
spacing was maintained throughout all 4 counts the sensor
would react very slowly. In the QT110, after an initial
detection is sensed, the remaining three bursts are spaced
about 18ms apart, so that the slowest reaction time possible
is 75+18+18+18 or 129ms and the fastest possible is 54ms,
depending on where in the initial burst interval the contact
first occurred. The response time will thus average 92ms.

2.1.5 F

ORCED SENSOR RECALIBRATION

The QT110 has no recalibration pin; a forced recalibration is
accomplished only when the device is powered up. However,
supply drain is so low it is a simple matter to treat the entire
IC as a controllable load; simply driving the QT110's Vdd pin
directly from another logic gate or a microprocessor port
(Figure 2-2) will serve as both power and 'forced recal'. The
source resistance of most CMOS gates and microprocessors
is low enough to provide direct power without any problems.
Note that most 8051-based micros have only a weak pullup
drive capability and will require true CMOS buffering. Any
74HC or 74AC series gate can directly power the QT110, as
can most other microprocessors.

Option strap configurations are read by the QT110 only on
powerup. Configurations can only be changed by powering
the QT110 down and back up again; again, a microcontroller
can directly alter most of the configurations and cycle power
to put them in effect.

- 5 -

10sVddGnd

Pulse

10sGndGnd

Toggle

60sGndVdd

DC Out

10sVddVdd

DC Out

Max On-

Duration

Tie

Pin 4 to:

Tie

Pin 3 to:

Table 2-1 Output Mode Strap Options

Figure 2-2 Powering From a CMOS Port Pin

0.01µF

C MO S

microcontroller

OUT

PORT X.m

PORT X.n

Vdd

Vss

QT110

2.2 OUTPUT FEATURES

The QT110 / QT110H are designed for maximum flexibility
and can accommodate most popular sensing requirements.
These are selectable using strap options on pins OPT1 and
OPT2. All options are shown in Table 2-1.

2.2.1 DC M

ODE OUTPUT

The output of the device can respond in a DC mode, where
the output is active-low (QT110) or active-high (QT110H)
upon detection. The output will remain active for the duration
of the detection, or until the Max On-Duration expires,
whichever occurs first. If the latter occurs first, the sensor
performs a full recalibration and the output becomes inactive
until the next detection.

In this mode, two Max On-Duration timeouts are available: 10
and 60 seconds.

2.2.2 T

OGGLE MODE OUTPUT

This makes the sensor respond in an on/off mode like a flip
flop. It is most useful for controlling power loads, for example
in kitchen appliances, power tools, light switches, etc.

Max On-Duration in Toggle mode is fixed at 10 seconds.
When a timeout occurs, the sensor recalibrates but leaves
the output state unchanged.

2.2.3 P

ULSE MODE OUTPUT

This generates a pulse of 75ms duration (QT110 ­negative-going; QT110H - positive-going) with every new
detection. It is most useful for 2-wire operation, but can also
be used when bussing together several devices onto a
common output line with the help of steering diodes or logic
gates, in order to control a common load from several places.

Max On-Duration is fixed at 10 seconds if in Pulse output
mode.

2.2.4 H

EARTBEAT

™ O

UTPUT

The output has a full-time HeartBeat™ ‘health’ indicator
superimposed on it. This operates by taking 'Out' into a
3-state mode for 350µs once before every QT burst. This
output state can be used to determine that the sensor is
operating properly, or, it can be ignored using one of several
simple methods.

QT110: The HeartBeat indicator can be sampled by using a
pulldown resistor on Out, and feeding the resulting
negative-going pulse into a counter, flip flop, one-shot, or

other circuit. Since Out is normally high, a pulldown resistor
will create negative HeartBeat pulses (Figure 2-3) when the
sensor is not detecting an object; when detecting an object,
the output will remain active for the duration of the detection,
and no HeartBeat pulse will be evident.

QT110H: Same as QT110 but inverted logic (use a pull-down
resistor instead of a pull-up etc.)

If the sensor is wired to a microprocessor as shown in Figure
2-4, the microprocessor can reconfigure the load resistor to
either ground or Vcc depending on the output state of the
device, so that the pulses are evident in either state.

Electromechanical devices will usually ignore this short
pulse. The pulse also has too low a duty cycle to visibly
activate LED’s. It can be filtered completely if desired, by
adding an RC timeconstant to filter the output, or if interfacing
directly and only to a high-impedance CMOS input, by doing
nothing or at most adding a small non-critical capacitor from
Out to ground (Figure 2-5).

2.2.5 P

IEZO ACOUSTIC DRIVE

A piezo drive signal is generated for use with a bare piezo
sounder immediately after a detection is made; the tone lasts
for a nominal 75ms to create a reassuring ‘tactile feedback’
sound.

The sensor will drive most common bare piezo ‘beepers’
directly using an H-bridge drive configuration for the highest
possible sound level at all supply voltages; H-bridge drive
effectively doubles the supply voltage across the piezo. The
piezo is connected across pins SNS1 and SNS2. This drive
operates at a nominal 4kHz frequency, a common resonance
point for enclosed piezo sounders. Other frequencies can be
obtained upon special request.

If desired a bare piezo sounder can be directly adhered to
the rear of a control panel, provided that an acoustically
resonant cavity is also incorporated to give the desired sound
level.

Since piezo sounders are merely high-K ceramic capacitors,
the sounder will double as the Cs capacitor, and the piezo's
metal disc will act as the sensing electrode. Piezo transducer
capacitances typically range from 6nF to 30nF (0.006µF to

0.03µF) in value; at the lower end of this range an additional
capacitor should be added to bring the total Cs across SNS1
and SNS2 to at least 10nF, or more if Cx is large.

- 6 -

Figure 2-4

Using a micro to obtain HB pulses in either output state

(QT110 or QT110H)

Figure 2-3

Getting HB pulses with a pull-down resistor (QT110 shown;

use pull-up resistor with QT110H)

3

46

5

1

+2.5 to 5

72

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

8

Ro

HeartBeat™ Pulses

Microprocesso r

PORT_M.x

PORT_M.y

3

46

5

72

OUT

OPT1

OPT2

GAIN

SNS 1

SNS 2

R

o

The burst acquisition process induces a small but audible
voltage step across the piezo resonator, which occurs when
SNS1 and SNS2 rapidly discharge residual voltage stored on
the resonator. The resulting slight clicking sound can be used
to provide an audible confirmation of functionality if desired,
or, it can be suppressed by placing a non-critical 1M to 2M
ohm bleed resistor in parallel with the resonator. The resistor
acts to slowly discharge the resonator, preempting the
occurrence of the harmonic-rich step (Figure 2-6).

With the resistor in place, an almost inaudible clicking sound
may still be heard, which is caused by the small charge
buildup across the piezo device during each burst.

2.2.6 O

UTPUT DRIVE

The QT110’s `output is active low (QT110) or active high
(QT110H) and can source 1mA or sink 5mA of non-inductive
current. If an inductive load is used, such as a small relay,
the load should be diode clamped to prevent damage.

Care should be taken when the IC and the load are both
powered from the same supply, and the supply is minimally
regulated. The device derives its internal references from the
power supply, and sensitivity shifts can occur with changes in
Vdd, as happens when loads are switched on. This can
induce detection ‘cycling’, whereby an object is detected, the
load is turned on, the supply sags, the detection is no longer
sensed, the load is turned off, the supply rises and the object
is reacquired, ad infinitum. To prevent this occurrence, the
output should only be lightly loaded if the device is operated
from an unregulated supply, e.g. batteries. Detection
‘stiction’, the opposite effect, can occur if a load is shed when
Out is active.

QT110: The output of the QT110 can directly drive a
resistively limited LED. The LED should be connected with its
cathode to the output and its anode towards Vcc, so that it
lights when the sensor is active-low. If desired the LED can
be connected from Out to ground, and driven on when the
sensor is inactive, but only with less drive current (1mA).

QT110H: This part is active-high, so it works in reverse to
that described above.

3 - CIRCUIT GUIDELINES

3.1 SAMPLE CAPACITOR

Charge sampler Cs can be virtually any plastic film or high-K
ceramic capacitor. Since the acceptable Cs range is
anywhere from 10nF to 30nF, the tolerance of Cs can be the
lowest grade obtainable so long as its value is guaranteed to
remain in the acceptable range under expected temperature

conditions. Only if very fast, radical temperature swings are
expected will a higher quality capacitor be required, for
example polycarbonate, PPS film, or NPO/C0G ceramic.

3.2 PIEZO SOUNDER

The use of a piezo sounder in place of Cs is described in the
previous section. Piezo sounders have very high,
uncharacterized thermal coefficients and should not be used
if fast temperature swings are anticipated.

3.3 OPTION STRAPPING

The option pins Opt1 and Opt2 should never be left floating.
If they are floated, the device will draw excess power and the
options will not be properly read on powerup. Intentionally,
there are no pullup resistors on these lines, since pullup
resistors add to power drain if tied low.

The Gain input is designed to be floated for sensing one of
the three gain settings. It should never be connected to a
pullup resistor or tied to anything other than Sns1 or Sns2.

Table 2-1 shows the option strap configurations available.

3.4 POWER SUPPLY, PCB LAYOUT

The power supply can range from 2.5 to 5.0 volts. At 3 volts
current drain averages less than 20µA in most cases, but can
be higher if Cs is large. Interestingly, large Cx values will
actually decrease power drain. Operation can be from
batteries, but be cautious about loads causing supply droop
(see Output Drive, previous section).

As battery voltage sags with use or fluctuates slowly with
temperature, the IC will track and compensate for these
changes automatically with only minor changes in sensitivity.

If the power supply is shared with another electronic system,
care should be taken to assure that the supply is free of
digital spikes, sags, and surges which can adversely affect
the device. The IC will track slow changes in Vdd, but it can
be affected by rapid voltage steps.

if desired, the supply can be regulated using a conventional
low current regulator, for example CMOS regulators that have
nanoamp quiescent currents. Care should be taken that the
regulator does not have a minimum load specification, which
almost certainly will be violated by the QT110's low current
requirement.

- 7 -

Figure 2-6 Damping Piezo Clicks with R

x

SENSING
ELECTRO DE

Piezo Sounde r

10-30nF

3

46

5

1

+2.5 to 5

72

OUT

OPT2

GAIN

SNS2

SNS1

Vss

Vdd

C

x

8

OPT1

R

x

Figure 2-5 Eliminating HB Pulses

3

46

5

72

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

CMOS

100pF

C

o

GATE OR

MICRO INPUT

Since the IC operates in a burst mode, almost
all the power is consumed during the course of
each burst. During the time between bursts the
sensor is quiescent.

3.4.1 M

EASURING SUPPLY CURRENT

Measuring average power consumption is a
fairly difficult task, due to the burst nature of
the device’s operation. Even a good quality
RMS DMM will have difficulty tracking the
relatively slow burst rate.

The simplest method for measuring average
current is to replace the power supply with a
large value low-leakage electrolytic capacitor,
for example 2,700µF. 'Soak' the capacitor by
connecting it to a bench supply at the desired
operating voltage for 24 hours to form the
electrolyte and reduce leakage to a minimum.
Connect the capacitor to the circuit at T=0,
making sure there will be no detections during
the measurement interval; at T=30 seconds measure the
capacitor's voltage with a DMM. Repeat the test without a
load to measure the capacitor's internal leakage, and
subtract the internal leakage result from the voltage droop
measured during the QT110 load test. Be sure the DMM is
connected only at the end of each test, to prevent the DMM's
impedance from contributing to the capacitor's discharge.

Supply drain can be calculated from the adjusted voltage
droop using the basic charge equation:

i =

✁VC

t

where C is the large supply cap value, t is the elapsed
measurement time in seconds, and ∆V is the adjusted
voltage droop on C.

3.4.2 ESD

PROTECTION

In cases where the electrode is placed behind a dielectric
panel, the IC will be protected from direct static discharge.
However, even with a panel, transients can still flow into the
electrode via induction, or in extreme cases, via dielectric
breakdown. Porous materials may allow a spark to tunnel
right through the material; partially conducting materials like
'pink poly' will conduct the ESD right to the electrode. Testing
is required to reveal any problems. The device does have
diode protection on its terminals which can absorb and
protect the device from most induced discharges, up to
20mA; the usefulness of the internal clamping will depending
on the dielectric properties, panel thickness, and rise time of
the ESD transients.

ESD dissipation can be aided further with an added diode
protection network as shown in Figure 2-7, in extreme cases.
Because the charge and transfer times of the QT110 are
relatively long, the circuit can tolerate very large values of Re,
more than 100k ohms in most cases where electrode Cx is

small. The added diodes shown (1N4150, BAV99 or
equivalent low-C diodes) will shunt the ESD transients away
from the part, and Re1 will current limit the rest into the
QT110's own internal clamp diodes. C1 should be around
10µF if it is to absorb positive transients from a human body
model standpoint without rising in value by more than 1 volt.
If desired C1 can be replaced with an appropriate zener
diode.

Directly placing semiconductor transient protection
devices or MOV's on the sense lead is not advised; these
devices have extremely large amounts of parasitic C which will
swamp the IC.

Re1 should be as large as possible given the load value of
Cx and the diode capacitances of D1 and D2. Re1 should be
low enough to permit at least 6 timeconstants of RC to occur
during the charge and transfer phases.

Re2 functions to isolate the transient from the Vdd pin;
values of around 1K ohms are reasonable.

As with all ESD protection networks, it is crucial that the
transients be led away from the circuit. PCB ground layout is
crucial; the ground connections to D1, D2, and C1 should all
go back to the power supply ground or preferably, if
available, a chassis ground connected to earth. The currents
should not be allowed to traverse the area directly under the
IC.

If the device is connected to an external circuit via a cable or
long twisted pair, it is possible for ground-bounce to cause
damage to the Out pin; even though the transients are led
away from the IC itself, the connected signal or power ground
line will act as an inductor, causing a high differential voltage
to build up on the Out wire with respect to ground. If this is a
possibility, the Out pin should have a resistance Re3 in
series with it to limit current; this resistor should be as large
as can be tolerated by the load.

- 8 -

Figure 2-7 ESD Protection

3

46

5

1

+2.5 to 5

72

OUT

OPT1

OPT2

GAIN

SNS1

SNS2

Vss

Vdd

8

R

C

D

D

R

R

e3

s

2

1

e2

e1

SENSING
ELECTRODE

10µF

+

C1

4.1 ABSOLUTE MAXIMUM SPECIFICATIONS

Operating temp ................................................................ as designated by suffix

Storage temp ....................................................................... -55

O

C to +125OC

V

DD

...................................................................................-0.5 to +6.5V

Max continuous pin current, any control or drive pin ...............................................±20mA

Short circuit duration to ground, any pin ......................................................... infinite

Short circuit duration to V

DD

, any pin ............................................................ infinite

Voltage forced onto any pin ....................................................-0.6V to (Vdd + 0.6) Volts

4.2 RECOMMENDED OPERATING CONDITIONS

VDD................................................................................... +2.5 to 5.5V

Supply ripple+noise ................................................................... 20mV p-p max

Load capacitance ..........................................................................0 to 20pF

Cs value .............................................................................. 10nF to 30nF

4.3 AC SPECIFICATIONS Vdd = 3.0, Ta = recommended operating range

µs300Heartbeat pulse widthT

HB

ms75Pulse output width on OutT

PO

ms75Piezo drive durationT

P

kHz4Piezo drive frequencyF

P

ms129Response timeT

R

ms70.5Burst lengthT

BL

ms75Burst spacing intervalT

BS

µs2Transfer durationT

PT

µs2Charge durationT

PC

ms550Recalibration timeT

RC

NotesUnitsMaxTypMinDescriptionParameter

4.4 SIGNAL PROCESSING

Note 3secs6010Post-detection recalibration timer duration

ms/level75Negative drift compensation rate

ms/level750Positive drift compensation rate

samples4Consensus filter length

Note 2%50Hysteresis

Note 1%6.25Threshold differential, low gain

Note 1%4.7Threshold differential, medium gain

Note 1%3.1Threshold differential, high gain

NotesUnitsMaxTypMinDescription

Note 1: Of absolute full scale signal
Note 2: Of signal threshold
Note 3: Strap option.

- 9 -

4.5 DC SPECIFICATIONS

Vdd = 3.0V, Cs = 10nF, Cx = 5pF, TA = recommended range, unless otherwise noted

Refer to Figures 4-1 through 4-3pF3Sensitivity - low gainS[3]

Refer to Figures 4-1 through 4-3pF1.5Sensitivity - medium gainS[2]

Refer to Figures 4-1 through 4-3pF1Sensitivity - high gainS[1]

bits14Acquisition resolutionA

R

Resistance from SNS1 to SNS2

✡

500KMin shunt resistanceI

X

pF300Load capacitance rangeC

X

OPT1, OPT2µA±1Input leakage currentI

IL

OUT, 1mA sourceVVdd-0.7High output voltageV

OH

OUT, 4mA sinkV0.6Low output voltageV

OL

OPT1, OPT2V2.2High input logic levelV

HL

OPT1, OPT2V0.8Low input logic levelV

IL

Required for proper startupV/s100Supply turn-on slopeV

DDS

µA20Supply currentI

DD

V5.252.45Supply voltageV

DD

NotesUnitsMaxTypMinDescriptionParameter

Preliminary Data: All specifications subject to change.

- 10 -

Figure 4-1 High Gain Sensitivity Figure 4-2 Medium Gain Sensitivity

and Range @ Vdd = 3V and Range @ Vdd = 3V

Cx=30pF Cx=30pF

25pF 25pF
20pF 20pF
10pF 10pF

5pF 5pF
0pF 0pF

Valid operating range Valid operating range

10 20

Cs, nF

0.5

1.0

2.0

3.0

Sensitivity, pF

30

1.5

2.5

10 20

Cs, nF

1.0

2.0

3.0

Sensitivity, pF

30

4.0

Figure 4-3 Low Gain Sensitivity

and Range @ Vdd = 3V

Cx=30pF

25pF

20pF
10pF

5pF
0pF

Valid operating range

10 20

Cs, nF

2.0

4.0

6.0

S ens it iv ity, p F

30

8.0

0.0150.0080.3810.203Y

0.390.329.9068.128x

BSC0.30.3BSC7.0627.62Aa

0.16-4.064-S1

0.140.123.5563.048S

-0.015-0.381r

0.150.123.813.048R

Typical0.1020.098Typical2.5912.489F

0.0650.0551.6511.397L1

0.0220.0140.5590.355L

-0.01-0.254P

-0.035-0.889Q

BSC0.30.3BSC7.627.62m

Typical0.430.355Typical10.9229.017M

0.3250.38.2557.62A

0.280.247.112 6.096a

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin Dual-In-Line

8º0º8º0ºØ

0.030.2290.7620.381ß

0.010.0070.2490.19e

0.040.021.0160.508E

0.0190.0140.4830.355L

BSC0.050.050BSC1.271.27D

0.010.0040.7620.101h

0.0680.0541.7281.371H

0.1570.153.9883.81Aa

0.2440.2296.1985.816W

0.1960.1894.9794.800M

NotesMaxMinNotesMaxMin

InchesMillimeters

SYMBOL

Package type: 8pin SOIC

- 11 -

5 - ORDERING INFORMATION

QT1 + AISOIC-8-40 - 85CQT110H-IS

QT1 + ASOIC-80 - 70CQT110H-S

QT1 +10HPDIP0 - 70CQT110H-D

QT1 + ISOIC-8-40 - 85CQT110-IS

QT1SOIC-80 - 70CQT110-S

QT1 + 10PDIP0 - 70CQT110-D

MARKINGPACKAGETEMP RANGEPART

Quantum Research Group Ltd

©1999 QRG Ltd.

Patented and patents pending

651 Holiday Drive Bldg. 5 / 300

Pittsburgh, PA 15220 USA

Tel: [phone] Fax: [phone]

[e-mail]

http://www.qprox.com

In the United Kingdom

Enterprise House, Southampton, Hants SO14 3XB

Tel: [phone] Fax: [phone]

Notice: This device expressly not for use in any medical or human safety related application without the express written consent of an officer

of the company.