POWER INTEGRATIONS LNK 501GN Datasheet

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Page 1

LNK501

LinkSwitch

™

Family

Energy Efficient, CV/CC Switcher for Very Low Cost Chargers and Adapters

Product Highlights

Cost Effective Linear/RCC Replacement

• Lowest cost, component count, constant voltage, constant current (CV/CC) solution

• Extremely simple circuit configuration

• Up to 75% lighter power supply reduces shipping cost

• Primary based CV/CC solution eliminates 10 to 20 secondary components for low system cost

• Combined primary clamp, feedback, IC supply, and loop compensation functions–minimizes external components

• Fully integrated auto-restart for short circuit and open loop fault protection–saves external component costs

• 42 kHz operation simplifies EMI filter design

•3 W output with EE13 core for low cost and small size

Much Higher Performance Over Linear/RCC

• Universal input range allows worldwide operation

• Up to 70% reduction in power dissipation–reduces enclosure size significantly

• CV/CC output characteristic without secondary feedback

• System level thermal and current limit protection

•Meets all single point failure requirements with only one additional clamp capacitor

• Controlled current in CC region provides inherent soft-start

• Optional opto feedback improves output voltage accuracy

*Estimated tolerance achievable in high volume production including transformer and other component tolerances.

**See Optional Secondary Feedback section.

Figure 1. Typical Application – not a Simplified Circuit (a) and

LinkSwitch

Wide Range HV DC Input

(a)

±5%

±10%

±20%*

For Circuit

Shown Above

Output Characteristic Tolerance Envelopes (b).

(b)

Secondary Feedback**

Output

±20%*

With Optional

PI-2776-091302

)

EcoSmart

- Extremely Energy Efficient

• Consumes <300 mW at 265 VAC input with no load

•Meets Blue Angel, Energy Star, and EC requirements

• No current sense resistors–maximizes efficiency

Applications

• Linear transformer replacement in all ≤ 3 W applications

• Chargers for cell phones, cordless phones, PDAs, digital cameras, MP3/portable audio devices, shavers, etc.

• Home appliances, white goods and consumer electronics

• TV standby and other auxilliary supplies

Description

LinkSwitch is specifically designed to replace all linear transformer/ RCC chargers and adapters in the ≤ 3 W universal range at equal or lower system cost with much higher performance and energy efficiency. LinkSwitch introduces a revolutionary topology for the design of low power switching power supplies that rivals the simplicity and low cost of linear adapters, and enables a much

OUTPUT POWER TABLE

PRODUCT

230 VAC ±15%

85-265 VAC

LNK501 P or G 4 W 3 W

Table 1. Notes: 1. Typical output power for a design achieving <300 mW no load consumption in an enclosed adapter measured at 50 °C ambient. Higher reflected voltage will extend power capability with increased no load consumption. See Key Application Considerations. 2. See Part Ordering Information.

smaller, lighter, and attractive package when compared with the traditional "brick". With efficiency of up to 75% at 3 W output and < 300 mW no-load consumption, a LinkSwitch solution can save

the end user enough energy over a linear design to completely pay for the power supply cost in less than one year. LinkSwitch

integrates a 700 V power MOSFET, PWM control, high voltage start-up, current limit, and thermal shutdown circuitry, onto a monolithic IC.

September 2002

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LNK501

CONTROL

DCS

SHUNT REGULATOR/

ERROR AMPLIFIER

OSCILLATOR

MAX

CLOCK

SAW

INTERNAL SUPPLY

SHUTDOWN/

AUTO-RESTART

5.7 V

5.7

4.7

COMPARATOR

HYSTERETIC

THERMAL

SHUTDOWN

LOW

FREQUENCY

OPERATION

÷ 8

SRQ

CURRENT

LIMIT

ADJUST

CURRENT LIMIT

COMPARATOR

LEADING

EDGE

BLANKING

DRAIN

Figure 2. Block Diagram.

Pin Functional Description

DRAIN (D) Pin:

Power MOSFET drain connection. Provides internal operating current for start-up. Internal current limit sense point for drain current.

CONTROL (C) Pin:

Error amplifier and feedback current input pin for duty cycle and current limit control. Internal shunt regulator connection to provide internal bias current during normal operation. It is also used as the connection point for the supply bypass and autorestart/compensation capacitor.

SOURCE (S) Pin:

Output MOSFET source connection for high voltage power return. Primary side control circuit common and reference point.

P Package (DIP-8B)

G Package (SMD-8B)

Figure 3. Pin Configuration.

LNK501

SOURCE

PI-2777-071802

PI-2711-073107

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LinkSwitch

Functional Description

The duty cycle, current limit and operating frequency relationships with CONTROL pin current are shown in Figure 4. Figure 5 shows a typical power supply outline schematic which is used below to describe the LinkSwitch operation.

Power Up

During power up, as VIN is first applied (Figure 5), the CONTROL pin capacitor C1 is charged through a switched high voltage current source connected internally between the DRAIN and CONTROL pins (see Figure 2). When the CONTROL pin voltage reaches approximately 5.7 V relative to the SOURCE pin, the high voltage current source is turned off, the internal control circuitry is activated and the high voltage internal MOSFET starts to switch. At this point, the charge stored on C1 is used to supply the internal consumption of the chip.

Constant Current (CC) Operation

As the output voltage, and therefore the reflected voltage across the primary transformer winding ramp up, the feedback CONTROL current I internal current limit increases with IC and reaches I is equal to I

. The internal current limit vs. IC characteristic

DCT

increases. As shown in Figure 4, the

LIM

when I

is designed to provide an approximately constant power supply output current as the power supply output voltage rises.

Constant Voltage (CV) Operation

When IC exceeds I

, typically 2 mA (Figure 4), the maximum

DCS

duty cycle is reduced. At a value of IC that depends on power supply input voltage, the duty cycle control limits LinkSwitch peak current below the internal current limit value. At this point the power supply transitions from CC to CV operation. With minimum input voltage in a typical universal input design, this transition occurs at approximately 30% duty cycle. R1 (Figure 5) is therefore initially selected to conduct a value of I approximately equal to I

when V

DCT

is at the desired value

OUT

at the minimum power supply input voltage. The final choice of R1 is made when the rest of the circuit design is complete. When the duty cycle drops below approximately 4%, the frequency is reduced, which reduces energy consumption under light load conditions.

Internal Current Limit

Auto-restart

LIM

CONTROL Current I

Duty Cycle

Auto-restart

77%

30%

3.8%

Frequency

OSC

OSC(low)

CONTROL Current I

Auto-restart

CONTROL Current I

Figure 4. CONTROL Characteristics.

CD1

LinkSwitc

DCS

DCT

PI-2799-112101

Auto-Restart Operation

When a fault condition, such as an output short circuit or open loop, prevents flow of an external current into the CONTROL pin, the capacitor C1 discharges towards 4.7 V. At 4.7 V, autorestart is activated, which turns the MOSFET off and puts the control circuitry in a low current standby mode. In auto-restart, LinkSwitch periodically restarts the power supply so that normal power supply operation can be restored when the fault is removed.

Figure 5. Power Supply Outline Schematic.

OUT

PI-2715-100801

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The characteristics described above provide an approximate CV/CC power supply output without the need for secondary side voltage or current feedback. The output voltage regulation is influenced by how well the voltage across C2 tracks the reflected output voltage. This tracking is influenced by the value of the transformer leakage inductance which introduces an error. Resistor R2 and capacitor C2 partially filter the leakage inductance voltage spike reducing this error. This circuitry, used with standard transformer construction techniques provides much better output load regulation than a linear transformer, making this an ideal power supply solution in many low power applications. If tighter load regulation is required, an optocoupler configuration can be used while still employing the constant output current characteristics provided by LinkSwitch.

Optional Secondary Feedback

Figure 6 shows a typical power supply outline schematic using LinkSwitch with optocoupler feedback to improve output voltage regulation. On the primary side, the schematic differs from

LinkSwitch

LNK501

85-265

VAC

R2 D1

Figure 5 by the addition of R3, C3 and optocoupler U1. R3 forms a potential divider with R1 to limit the U1 collector emitter voltage.

On the secondary side, the addition of voltage sense circuit components R4, VR1 and U1 LED provide the voltage feedback signal. In the example shown, a simple Zener (VR1) reference is used though more accurate references may be employed for improved output voltage tolerancing and cable drop compensation, if required. R4 provides biasing for VR1. The regulated output voltage is equal to the sum of the VR1 Zener voltage plus the forward voltage drop of the U1 LED. R5 is an optional low value resistor to limit U1 LED peak current due to output ripple. Manufacturer’s specifications for U1 current and VR1 slope resistance should be consulted to determine whether R5 is required.

U1 is arranged with collector connected to primary ground and emitter to the anode of D1. This connection keeps the opto in an electrically “quiet” position in the circuit. If the opto was

VR1

OUT

RTN

PI-2787-092002

Figure 6. Power Supply Outline Schematic with Optocoupler Feedback.

Output V oltage

Inherent

CC to CV

transition

Voltage

feedback

threshold

Load variation during battery

charging

point

Output Current

Figure 7. Influence of the Optocoupler on the Power Supply Output Characteristic.

9/02

Tolerance envelope

without optocoupler

Typical inherent

characteristic without

optocoupler

Characteristic with

optocoupler

PI-2788-092101

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LNK501

instead placed on the cathode side of D1, it would become a switching node, generating additional common mode EMI currents through its internal parasitic capacitance.

The feedback configuration in Figure 6 is simply a resistive divider made up of R1 and R3 with D1, R2, C1 and C2 rectifying, filtering and smoothing the primary winding voltage signal. The optocoupler therefore effectively adjusts the resistor divider ratio to control the DC voltage across R1 and therefore, the feedback current received by the LinkSwitch CONTROL pin.

When the power supply operates in the constant current (CC) region, for example when charging a battery, the output voltage is below the voltage feedback threshold defined by U1 and VR1 and the optocoupler is fully off. In this region, the circuit behaves exactly as previously described with reference to Figure 5 where the reflected voltage increases with increasing output voltage and the LinkSwitch internal current limit is adjusted to provide an approximate CC output characteristic. Note that for similar output characteristics in the CC region, the value of R1 in Figure 5 will be equal to the value of R1+R3 in Figure 6.

When the output reaches the voltage feedback threshold set by U1 and VR1, the optocoupler turns on. Any further increase in the power supply output voltage results in the U1 transistor current increasing, which increases the percentage of the reflected voltage appearing across R1. The resulting increase in the LinkSwitch CONTROL current reduces the duty cycle according to Figure 4 and therefore, maintains the output voltage regulation.

Normally, R1 and R3 are chosen to be equal in value. However, increasing R3 (while reducing R1 to keep R1 + R3 constant) increases loop gain in the CV region, improving load regulation. The extent to which R3 can be increased is limited by opto

transistor voltage and dissipation ratings and should be fully tested before finalizing a design. The values of C2 and C3 are less important other than to make sure they are large enough to have very little influence on the impedance of the voltage division circuit set up by R1, R3 and U1 at the switching frequency. Normally, the values of C2 and C3 in Figure 6 are chosen equal to the value of C2 in Figure 5, though the voltage rating may be reduced depending on the relative values of R1 and R2 discussed above. See Applications section for typical values of components.

Figure 7 shows the influence of optocoupler feedback on the output characteristic. The envelope defined by the dashed lines represent the worst case power supply DC output voltage and current tolerances (unit-to-unit and over the input voltage range) if an optocoupler is not used. A typical example of an inherent (without optocoupler) output characteristic is shown dotted. This is the characteristic that would result if U1, R4 and VR1 were removed. The optocoupler feedback results in the characteristic shown by the solid line. The load variation arrow in Figure 7 represents the locus of the output characteristic normally seen during a battery charging cycle. The two characteristics are identical as the output voltage rises but then separate as shown when the voltage feedback threshold is reached. This is the characteristic seen if the voltage feedback threshold is above the output voltage at the inherent CC to CV transition point also indicated in Figure 7.

Figure 8 shows a case where the voltage feedback threshold is set below the voltage at the inherent CC to CV transition point. In this case, as the output voltage rises, the secondary feedback circuit takes control before the inherent CC to CV transition occurs. In an actual battery charging application, this simply limits the output voltage to a lower value.

Output V oltage

Voltage

feedback

threshold

Load variation during battery

charging

Figure 8. Output Characteristic with Optocoupler Regulation (Reduced Voltage Feedback Threshold).

Inherent

CC to CV

transition

point

O(MAX)

Characteristic observed with

load variation often applied during

laboratory bench testing

Output Current

Tolerance envelope

without optocoupler

Typical inherent

characteristic without

optocoupler

Characteristic with

optocoupler

Power Supply peak

output power curve

PI-2790-092101

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However, in laboratory bench tests, it is often more convenient to test the power supply output characteristic starting from a low output current and gradually increasing the load. In this case, the optocoupler feedback regulates the output voltage until the peak output power curve is reached as shown in Figure 8. Under these conditions, the output current will continue to rise until the peak power point is reached and the optocoupler turns off. Once the optocoupler is off, the CONTROL pin feedback current is determined only by R1 and R3 and the output current therefore folds back to the inherent CC characteristic as shown. Since this type of load transition does not normally occur in a battery charger, the output current never overshoots the inherent constant current value in the actual application.

In some applications it may be necessary to avoid any output current overshoot, independent of the direction of load variation. To achieve this goal, the minimum voltage feedback threshold should be set at V

. This will ensure that the voltage at the

O(MAX)

CC to CV transition point of the inherent characteristic will always occur below the voltage feedback threshold. However, the output voltage tolerance is then increased, since the inherent CV characteristic tolerance below V

is added to the

O(MAX)

tolerance of the optocoupler feedback circuit.

Applications Example

The circuit shown in Figure 9 shows a typical implementation of an approximate constant voltage / constant current (CV/CC) charger using LinkSwitch. This design delivers 2.75 W with a nominal peak power point voltage of 5.5 V and a current of 500 mA. Efficiency is greater than 70% over an input range of 85 VAC to 265 VAC.

The bridge rectifier, BR1, rectifies the AC input. Resistor, RF1 is a fusible type providing protection from primary side short circuits. The rectified AC is smoothed by C1 and C2 with inductor L1 forming a pi-filter in conjunction with C1 and C2 to filter conducted EMI. The switching frequency of 42 kHz allows such a simple EMI filter to be used without the need for a Y capacitor while still meeting international EMI standards.

When power is applied, high voltage DC appears at the DRAIN pin of LinkSwitch (U1). The CONTROL pin capacitor C3 is then charged through a switched high voltage current source connected internally between the DRAIN and CONTROL pins. When the CONTROL pin reaches approximately 5.7 V relative to the SOURCE pin, the internal current source is turned off. The internal control circuitry is activated and the high voltage MOSFET starts to switch, using the energy in C3 to power the IC.

When the MOSFET is on, the high voltage DC bus is connected to one end of the transformer primary, the other end being connected to primary return. As the current ramps in the primary of flyback transformer T1, energy is stored. This energy is delivered to the output when the MOSFET turns off each switching cycle.

The secondary of the transformer is rectified and filtered by D6 and C5 to provide the DC output to the load.

LinkSwitch dramatically simplifies the secondary side by controlling both the constant voltage and constant current regions entirely from the primary side. This is achieved by monitoring the primary-side VOR (voltage output reflected).

Diode D5 and capacitor C4 form the primary clamp network. This both limits the peak drain voltage due to leakage inductance and provides a voltage across C4, which is equal to the VOR plus an error due to the parasitic leakage inductance. Resistor R2 filters the leakage inductance spike and reduces the error in the value of the VOR. Resistor R1 converts this voltage into a current that is fed into the CONTROL pin to regulate the output.

During CV operation the output is regulated through control of the duty cycle. As the current into the CONTROL pin exceeds approximately 2 mA, the duty cycle begins to reduce, reaching 30% at a CONTROL pin current of 2.3 mA.

Under light or no-load conditions, when the duty cycle reaches approximately 4%, the switching frequency is reduced to lower energy consumption.

If the output load is increased beyond the peak power point (defined by 0.5·L·I2·f), the output voltage and VOR falls. The reduced CONTROL pin current will lower the internal LinkSwitch current limit (current limit control) providing an approximately constant current output characteristic. If the load is increased and the CONTROL pin current falls below approximately 1 mA, the CONTROL pin capacitor C3 will discharge and the supply enters auto-restart.

Current limit control removes the need for any secondary side current sensing components (sense resistor, transistor, opto coupler and associated components). Removing the secondary sense circuit dramatically improves efficiency, giving the associated benefit of reduced enclosure size.

Key Application Considerations

Design Output Power

Table 1 (front page) shows the maximum continuous output power that can be obtained under the following conditions:

1. The minimum DC input bus voltage is 90 V or higher. This corresponds to a filter capacitor of 3 µF/W for universal input and 1 µF/W for 230 VAC or 115 VAC input with doubler input stage.

2. Design is a discontinuous mode flyback converter, with nominal primary inductance value and a VOR in the range 40-60 V. Note: The simple LinkSwitch circuit configuration is designed specifically for discontinuous mode operation.

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RF1 10 Ω 1 W Fusible

85-265

VAC

BR1

1 A, 600 V

L1 1 mH

4.7 µF 400 V

LNK501

LinkSwitch

4.7 µF 400 V

20.5 kΩ 1%

D5 1N4937

100 Ω

0.22 µF 50 V

0.1 µF 100 V

116 T #34 AWG

LNK501

EE13

= 2.55 mH

15 T #30 AWG TIW

D6 11DQ06

C5 470 µF 10 V

PERFORMANCE SUMMARY Output Power: 2.75 W

Efficiency: ≥72% No Load Consumption: 260 mW, 230 VAC 200 mW, 115 VAC

5.5 V , 500 mA

RTN

PI-2904-083002

Figure 9. 2.75 W Constant Voltage/Constant Current (CV/CC) Charger using LinkSwitch.

OUT

Output Voltage (V

1000 200 300 400 500 600 700

Output Current (I

OUT

mA)

= 85 VAC

= 115 VAC

= 185 VAC

= 265 VAC

PI-2964-072202

Figure 10. Measured Output Characteristic of the Circuit in Figure 9.

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LNK501

Continuous mode designs can result in loop instability and are therefore not recommended.

3. A secondary output of 5 V with a Schottky rectifier diode.

4. Assumed efficiency of 70%

5. The part is board mounted with SOURCE pins soldered to sufficient area of copper to keep the die temperature at or below 100 °C.

In addition to the thermal environment (sealed enclosure, ventilated, open frame, etc), the maximum power capability of LinkSwitch in a given application depends on transformer core size, efficiency, primary inductance tolerance, minimum specified input voltage, input storage capacitance, output voltage output diode forward drop etc., and can be different from the values shown in Table 1.

In designs not required to meet 300 mW no-load consumption, the transformer can be designed with higher VOR to extend power capability as noted in the following section.

Transformer Design

To provide an approximately CV/CC output, the transformer should be designed to be discontinuous; all the energy stored in the transformer is transferred to the secondary during the MOSFET off time. Energy transfer in discontinuous mode is independent of line voltage.

The peak power point prior to entering constant current operation is defined by the maximum power transferred by the transformer. The power transferred is given by the expression P = 0.5·L·I2·f, where L is the primary inductance, I2 is the primary peak current squared and f is the switching frequency.

To simplify analysis, the data sheet parameter table specifies an I2f coefficient. This is the product of current limit squared and switching frequency normalized to the feedback parameter I

. This provides a single term that specifies the variation of

DCT

the peak power point in the power supply due to LinkSwitch.

As primary inductance tolerance is part of the expression that determines the peak output power point (start of the CC characteristic) this parameter should be well controlled. For an estimated overall output peak power tolerance of ±20% the primary inductance tolerance should be ±10% or better. This is achievable using standard low cost, center leg gapping techniques where the gap size is typically 0.08 mm or larger. Smaller gap sizes are possible but require non standard, tighter ferrite A tolerances.

Other gapping techniques such as film gapping allow tighter tolerances (±7% or better) with associated improvements in the tolerance of the peak power point. Please consult your transformer vendor for guidance.

Core gaps should be uniform. Uneven core gapping, especially with small gap sizes, may cause variation in the primary inductance with flux density (partial saturation) and make the constant current region non-linear. To verify uniform gapping it is recommended that the primary current wave-shape be examined while feeding the supply from a DC source. The gradient is defined as di/dt = V/L and should remain constant throughout the MOSFET on time. Any change in gradient of the current ramp is an indication of uneven gapping.

Measurements made using a LCR bridge should not be solely relied upon; typically these instruments only measure at currents of a few milliamps. This is insufficient to generate high enough flux densities in the core to show uneven gapping.

For a typical EE13 core using center leg gapping, a 0.08 mm gap (ALG of 190 nH/t2) allows a primary inductance tolerance of ±10% to be maintained in standard high volume production. This allows the EE13 to be used in designs up to 2.75 W. If film gapping is used then this increases to 3 W with less than 300 mW no-load consumption. Moving to a larger core, EE16 for example, allows a 3 W output with center leg gapping.

The transformer turns ratio should be selected to give a V (output voltage reflected through secondary to primary turns ratio) of 40 - 60 V. In designs not required to meet 300 mW noload consumption targets, the transformer can be designed with higher VOR as long as discontinuous mode operation is maintained. This increases the output power capability. For example, a 230 VAC input design using an EE19 transformer core with VOR >70 V, is capable of delivering up to 5 W typical output power. Note: the linearity of the CC region of the power supply output characteristic is influenced by VOR. If this is an important aspect of the application, the output characteristic should be checked before finalizing the design.

Output Characteristic Variation

Both the device tolerance and external circuit govern the overall tolerance of the LinkSwitch output characteristic. Estimated peak power point tolerances for a 2.75 W design are ±10% for voltage and ±20% for current limit for overall variation in high volume manufacturing. This includes device and transformer tolerances and line variation. Lower power designs may have poorer constant current linearity.

As the output load reduces from the peak power point, the output voltage will tend to rise due to tracking errors compared to the load terminals. Sources of these errors include the output

cable drop, output diode forward voltage and leakage inductance, which is the dominant cause. As the load reduces, the primary operating peak current reduces, together with the leakage inductance energy, which reduces the peak charging of the clamp capacitor. With a primary leakage inductance of 50 µH, the output voltage typically rises 30% over a 100% to 5% load change.

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At very light or no-load, typically less than 2 mA of output current, the output voltage rises due to leakage inductance peak charging of the secondary. This voltage rise can be reduced with a small preload with little change to no-load power consumption.

The output voltage load variation can be improved to be ±5% across the whole load range by adding an optocoupler and secondary reference (Figure 6). The secondary reference is designed to only provide feedback above the normal peak power point voltage to maintain the correct constant current characteristic.

Component Selection

The schematic shown in Figure 5 outlines the key components needed for a LinkSwitch Supply.

Clamp diode – D1

Diode D1 should be either a fast (trr <250 ns) or ultra-fast type (trr <50 ns), with a voltage rating of 600 V or higher. Fast recovery types are preferred, being typically lower cost. Slow diodes are not recommended; they can allow excessive DRAIN ringing and the LinkSwitch to be reverse biased.

Clamp Capacitor – C2

Capacitor C2 should be a 0.1 µF, 100 V capacitor. Low cost metallized plastic film types are recommended. The tolerance of this part has a very minor effect on the output characteristic so any of the standard ±5%, ±10% or ±20% tolerances are acceptable. Ceramic capacitors are not recommended. The common dielectrics used such as Y5U or Z5U are not stable with voltage or temperature and may cause output instability. Ceramic capacitors with high stability dielectrics may be used but are expensive compared to metallized film types.

voltage plus the input voltage transformed through the turns ratio (a typical VOR of 50 V requires a diode PIV of 50 V). Slow recovery diodes are not recommended (1N400X types).

Output Capacitor – C4

Capacitor C4 should be selected such that its voltage and ripple current specifications are not exceeded.

LinkSwitch

Layout considerations

Primary Side Connections

As the SOURCE pins in a LinkSwitch supply are switching nodes, the copper area connected to SOURCE together with C1, C2 and R1 (Figure 5) should be minimized, within the thermal contraints of the design, to reduce EMI coupling.

The CONTROL pin capacitor C1 should be located as close as possible to the SOURCE and CONTROL pins.

To minimize EMI coupling from the switching nodes on the primary to both the secondary and AC input, the LinkSwitch should be positioned away from the secondary of the transformer and AC input.

Routing the primary return trace from the transformer primary around LinkSwitch and associated components further reduces coupling.

Y capacitor

If a Y capacitor is required, it should be connected close to the transformer secondary output return pin(s) and the primary bulk capacitor negative return. Such placement will maximize the EMI benefit of the Y capacitor and avoid problems in common-mode surge testing.

Control Pin Capacitor – C1

Capacitor C1 is used during start-up to power LinkSwitch and sets the Auto-Restart frequency. For designs that have a battery load this component should have a value of 0.22 µF and for resistive loads a value of 1 µF. This ensures there is sufficient time during start-up for the output voltage to reach regulation. Any capacitor type is acceptable with a voltage rating of 10 V or above.

Feedback Resistor – R1

The value of R1 is selected to give a feedback current into the CONTROL pin of approximately 2.3 mA at the peak output power point of the supply. The actual value depends on the V

selected during design. Any 1%, 0.25 W resistor is suitable.

Output Diode – D2

Either PN fast, PN ultra fast or Schottky diodes can be used depending on the efficiency target for the supply, Schottky diodes giving higher efficiency then PN diodes. The diode voltage rating should be sufficient to withstand the output

Quick Design Checklist

As with any power supply design, all LinkSwitch designs should be verified on the bench to make sure that component specifications are not exceeded under worst case conditions. Note: In a LinkSwitch circuit, the SOURCE is a switching node. This should be taken into consideration during testing. Oscilloscope measurements should be made with probe grounded to DC voltages such as primary return or DC rail but not to SOURCE. Power supply input voltage should always be supplied using an isolation transformer. The following minimum set of tests is strongly recommended:

1. Maximum drain voltage – Verify that V

does not exceed

DSS

675 V at highest input voltage and peak output power.

2. Maximum drain current – At maximum ambient temperature, maximum input voltage and peak output power, verify drain current waveforms at start-up for any signs of transformer saturation and excessive leading edge current spikes.

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LinkSwitch has a minimum leading edge blanking time of 200 ns to prevent premature termination of the on-cycle. Verify that the leading edge current spike event is below current limit at the end of the 200 ns blanking period.

3. Thermal check – At peak output power, minimum input voltage and maximum ambient temperature, verify that the temperature specifications are not exceeded for LinkSwitch, transformer, output diode and output capacitors. Enough thermal margin should be allowed for part-to-part variation of the R

of LinkSwitch as specified in the data sheet.

DS(ON)

Under low line, peak power, a maximum LinkSwitch SOURCE pin temperature of 100 °C is recommended to allow for these variations.

4. Centered output characteristic – Using a transformer with nominal primary inductance and at an input voltage midway between low and high line, verify that the peak power point occurs at the desired nominal output current, with the correct output voltage. If this does not occur then the design should be refined to ensure the overall tolerance limits are met.

Design Tools

Up to date information on design tools can be found at the Power Integrations Web site: www.powerint.com.

Input Filter

Capacitor

LinkSwitch

HV DC

Input

Transformer

Y1-

Capacitor

DC Out

Output

Capacitor

Figure 11. Recommended Circuit Board Layout for LinkSwitch using P Package.

SSSS

PI-2900-070202

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ABSOLUTE MAXIMUM RATINGS

DRAIN Voltage ........................................ -0.3 V to 700 V

DRAIN Peak Current .............................................400 mA

CONTROL Voltage ..................................... - 0.3 V to 9 V

CONTROL Current (not to exceed 9 V)................100 mA

Storage Temperature ................................-65 °C to 150 °C

Operating Junction Temperature Lead Temperature

(3)

................................................ 260 °C

(2)

...........-40 °C to 150 °C

THERMAL IMPEDANCE

Thermal Impedance: P/G Package:

(θJA) .............. 70 °C/W

(1)

(θJC)

................................. 11 °C/W

(2)

, 55 °C/W

Conditions

Parameter

Symbol

(Unless Otherwise Specified)

SOURCE = 0 V; TJ = -40 °C to 125 °C

See Figure 12

(1,4)

Notes:

1. All voltages referenced to SOURCE, T

= 25 °C.

2. Normally limited by internal circuitry.

3. 1/16" from case for 5 seconds.

4. Maximum ratings specified may be applied, one at a time, without causing permanent damage to the product. Exposure to Absolute Maximum Rating conditions for extended periods of time may affect product reliability.

Notes:

(3)

1. Measured on the SOURCE pin close to plastic interface.

2. Soldered to 0.36 sq. inch (232 mm2), 2 oz. (610 gm/m2) copper clad.

3. Soldered to 1 sq. inch (645 mm2). 2 oz. (610 gm/m2) copper

clad.

Min Typ Max

Units

CONTROL FUNCTIONS

Switching Frequency

Low Switching Frequency

OSC(LOW)

Duty Cycle at Low Switching

Frequency Low Frequency

Duty Cycle Range Maximum

Duty Cycle PWM

Gain CONTROL Pin

Current at 30% Duty Cycle

CONTROL Pin Voltage

OSC

(RANGE)

MAX

REG

DCT

C(IDCT)

IC = I

Duty Cycle = DC

, TJ = 25 °C

DCT

TJ = 25 °C

Frequency Switching from f

OSC(LOW)

Frequency = f

, TJ = 25 °C

OSC(LOW)

IC = 1.5 mA

IC = I

, TJ = 25 °C

DCT

TJ = 25 °C

See Figure 4

IC = I

DCT

OSC

38 42 46

26 30 34

2.4 3.8 5.2

1.8 3.15 4.5

74 77 80

-0.45 -0.35 -0.25

2.24 2.30 2.36

5.5 5.75 6

kHz

%/µA

Dynamic Impedance

IC = I

, TJ = 25 °C

DCT

60 90 120

9/02

Ω

Page 12

LNK501

Parameter

SHUTDOWN/AUTO-RESTART

Symbol

SOURCE = 0 V; TJ = -40 °C to 125 °C

Conditions

(Unless Otherwise Specified)

See Figure 12

Min Typ Max

Units

CONTROL Pin Charging Current

Control/Supply/ Discharge Current

Auto-restart Threshold Voltage

Auto-restart Hysteresis Voltage

Auto-restart Duty Cycle

Auto-restart Frequency

CIRCUIT PROTECTION

Self-protection Current Limit

I2 f Coefficient Current Limit at

Auto-restart Power Up Reset

Threshold Voltage

C(CH)

CD1

CD2

C(AR)

C(AR)hyst

(AR)

LIM

LIM(AR)

C(RESET)

VC = 0 V

TJ = 25 °C

VC = 5.15 V

Output MOSFET Enabled

Output MOSFET Disabled

Short circuit applied at

power supply output

Short circuit applied at power supply

output, C1 = 0.22 µF (See Figure 12)

TJ = 25 °C

di/dt = 90 mA/µs

See Note C

TJ = 25 °C

di/dt = 90 mA/µs

See Notes C, D

TJ = 25 °C

IC = I

CD1

-4.5 -3.25 -2

-2.3 -1.3 -0.3

0.95 1.06 1.14

0.7 0.9 1.1

5.6

0.9

300

241 254 267

2547 2710 2873

158

1.5 2.75 4.0

A2Hz

Leading Edge Blanking Time

Current Limit Delay

Thermal Shutdown Temperature

Thermal Shutdown Hysteresis

9/02

LEB

IL(D)

TJ = 25 °C

IC = I

DCT

TJ = 25 °C

IC = I

DCT

200 300

100

125 135

°C

Page 13

Parameter

OUTPUT

Symbol

Conditions

(Unless Otherwise Specified)

SOURCE = 0 V

; T

= -40 °C to 125 °C

See Figure 12

LNK501

Min Typ Max

Units

ON-State Resistance

OFF-State Current

Breakdown Voltage

DRAIN Supply Voltage

DS(ON)

DSS

ID = 25 mA

TJ = 25 °C

See Note B, VC = 6.2 V

VD = 560 V, TA = 125 °C

See Note B

VC = 6.2 V, TA = 25 °C

See Note E

= 100 °C

700

36 50

28 32 42 48

Ω

µA

NOTES:

A. For specifications with negative values, a negative temperature coefficient corresponds to an increase in magnitude with

increasing temperature, and a positive temperature coefficient corresponds to a decrease in magnitude with increasing temperature.

B. Breakdown voltage may be checked against minimum BV

exceeding minimum BV

C. IC is increased gradually to obtain maximum current limit at di/dt of 90 mA/µs. Increasing IC further would terminate the cycle

through duty cycle control. D. This parameter is normalized to I E. It is possible to start up and operate

current is reduced, which affects start-up time, auto-restart frequency, and auto-restart duty cycle. Refer to the characteristic

graph on CONTROL pin charge current (IC) vs. DRAIN voltage (Figure 13) for low voltage operation characteristics.

DSS

to correlate to power supply output current (it is multiplied by I

DCT

LinkSwitch

at DRAIN voltages well below 36 V. However, the CONTROL pin charging

specification by ramping the DRAIN pin voltage up to but not

DSS

(nominal)/I

DCT

9/02

Page 14

LNK501

2.15 2.25 2.35 2.45 2.55 2.65

Control Pin Current (mA)

Duty Cycle (%)

PI-2902-071602

20 10

40 30

40 V

Figure 12. LinkSwitch General Test Circuit.

VC = 5.15 V

1.6

1.2

40 V

750 Ω

10 kΩ

PI-2901-071602

0.22 µF

LinkSwitch

S C

90%

S S

PI-2894-062802

90%

0.8

CONTROL Pin

0.4

Charging Current (mA)

0 20 40 60 80 100

DRAIN Voltage (V)

Figure 13. IC vs. DRAIN Voltage.

120

100

Control Pin Current (mA)

0.0 2.0 4.0 8.06.0 10.0 12.0 14.0

Control Pin Voltage (V)

Figure 15. CONTROL Pin I-V Characteristic.

F 9/02

PI-2895-062802

DRAIN

VOLTAGE

10%

0 V

Figure 14. Duty Cycle Measurement.

Figure 16. Duty Cycle vs. CONTROL Pin Current.

D =

PI-2048-050798

Page 15

Typical Performance Characteristics

1.2

0.8

0.4

0.6

0.2

-50 0 50 100 150

Temperature (°C)

PI-2899-062802

PWM Gain (Normalized for 25 °C)

1.2

0.8

1.0

0.0

-50 -25 0 25 7550 100 150125

Junction Temperature (°C)

f Coefficient

(Normalized for 25

°C)

PI-2910-071602

0.2

0.6

0.4

LNK501

1.1

1.0

Breakdown Voltage

(Normalized to 25 °C)

0.9

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

Figure 17. Breakdown Voltage vs. Temperature.

1.200

1.000

°C)

0.800

0.600

PI-2213-012301

PI-2897-062802

1.200

1.000

°C)

0.800

0.600

0.400

Switching Frequency

(Normalized for 25

0.200

0.000

-50 0 50 100 150

Junction Temperature (°C)

Figure 18. Switching Frequency vs. Temperature.

PI-2896-062802

0.400

Current Limit

(Normalized for 25

0.200

0.000

-50 -25 0 25 50 75 100 125 150

Junction Temperature (°C)

Figure 19. Current Limit vs. Temperature.

1.200

1.000

0.800

0.600

0.400

(Normalized for 25 °C)

0.200

DCT

0.000

-50 0 50 100 150

Junction Temperature (°C)

Figure 21. I

vs. Temperature.

DCT

PI-2898-062802

Figure 20. I

f Coefficient vs. Temperature.

Figure 22. PWM Gain vs. Temperature.

9/02

Page 16

LNK501

Typical Performance Characteristics (cont.)

300

=25 °C

CASE

=100 °C

250

200

150

100

Drain Current (mA)

0246810

Figure 23. Output Characteristics (DRAIN Current vs.

DRAIN Voltage.

CASE

Drain Voltage (V)

PI-2221-031401

F 9/02

Page 17

PART ORDERING INFORMATION

LNK 501 G - TL

D S

.004 (.10)

⊕

-E-

.245 (6.22) .255 (6.48)

Pin 1

-D-

.128 (3.25) .132 (3.35)

-TSEATING PLANE

.100 (2.54) BSC

.375 (9.53) .385 (9.78)

.014 (.36) .022 (.56)

T E D S

⊕

.048 (1.22) .053 (1.35)

.010 (.25) M

LinkSwitch

Product Family Series Number Package Identifier

G Plastic Surface Mount DIP P Plastic DIP

Package/Lead Options

Blank Standard Configuration

TL Tape & reel in 1000 pc multiples, G package only

DIP-8B

Notes:

1. Package dimensions conform to JEDEC specification MS-001-AB (Issue B 7/85) for standard dual-in-line (DIP) package with .300 inch row spacing.

2. Controlling dimensions are inches. Millimeter sizes are shown in parentheses.

3. Dimensions shown do not include mold flash or other protrusions. Mold flash or protrusions shall not exceed .006 (.15) on any side.

4. Pin locations start with Pin 1, and continue counter-clock wise to Pin 8 when viewed from the top. The notch and/or dimple are aids in locating Pin 1. Pin 6 is omitted.

5. Minimum metal to metal spacing at the package body for the omitted lead location is .137 inch (3.48 mm).

6. Lead width measured at package body.

.057 (1.45) .063 (1.60)

(NOTE 6) .015 (.38)

MINIMUM

.125 (3.18) .135 (3.43)

7. Lead spacing measured with the leads constrained to be perpendicular to plane T.

.010 (.25) .015 (.38)

.300 (7.62) BSC

(NOTE 7)

.300 (7.62) .390 (9.91)

LNK501

P08B

PI-2551-070302

9/02

Page 18

LNK501

SMD-8B

-E-

.245 (6.22) .255 (6.48)

Pin 1

-D-

.128 (3.25) .132 (3.35)

.032 (.81) .037 (.94)

D S

.004 (.10)

⊕

.100 (2.54) (BSC)

.375 (9.53) .385 (9.78)

.048 (1.22) .053 (1.35)

.372 (9.45) .388 (9.86)

E S

⊕

.057 (1.45) .063 (1.60)

(NOTE 5)

.009 (.23)

.010 (.25)

.046

.060

Pin 1

.086

.186

.286

Solder Pad Dimensions

.004 (.10)

.004 (.10) .012 (.30)

.036 (0.91) .044 (1.12)

.060

.046

.080

Notes:

1. Controlling dimensions are inches. Millimeter sizes are shown in parentheses.

2. Dimensions shown do not include mold flash or other protrusions. Mold flash or

.420

protrusions shall not exceed .006 (.15) on any side.

3. Pin locations start with Pin 1, and continue counter-clock Pin 8 when viewed from the top. Pin 6 is omitted.

4. Minimum metal to metal spacing at the package body for the omitted lead location is .137 inch (3.48 mm).

5. Lead width measured at package body.

6. D and E are referenced datums on the package body.

0 -

G08B

PI-2546-081601

F 9/02

Page 19

LNK501

9/02

Page 20

LNK501

Revision

D E

Notes

1) Final release data sheet

1) Enhanced tolerance with optocoupler designs

Date 7/02

8/02

2) Updated P and G packages thermal impedance

1) Corrected minor errors in text and figures

9/02

2) Updated Figure 6 and text description

LIFE SUPPORT POLICY

POWER INTEGRATIONS' PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS, INC. As used herein:

1. Life support devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user.

2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness.

For the latest updates, visit our Web site: www.powerint.com Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power Integrations does not assume any liability arising from the use of any device or circuit described herein, nor does it convey any license under its patent rights or the rights of others.

The products and applications illustrated herein may be covered by one or more U.S. and foreign patents or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrations’ patents may be found at www.powerint.com.

The PI Logo,

PI Expert

WORLD HEADQUARTERS AMERICAS

Power Integrations, Inc. 5245 Hellyer Avenue San Jose, CA 95138 USA Main: +1 408-414-9200 Customer Service: Phone: +1 408-414-9665 Fax: +1 408-414-9765

e-mail: [email protected]

CHINA

Power Integrations International Holdings, Inc. Rm# 1705, Bao Hua Bldg. 1016 Hua Qiang Bei Lu Shenzhen Guangdong, 518031 China Phone: +86-755-8367-5143 Fax: +86-755-8377-9610

e-mail: [email protected]

TOPSwitch, TinySwitch

and

EcoSmart

are registered trademarks of Power Integrations, Inc.

EUROPE & AFRICA

Power Integrations (Europe) Ltd. Centennial Court Easthampstead Road Bracknell Berkshire, RG12 1YQ United Kingdom Phone: +44-1344-462-300 Fax: +44-1344-311-732

e-mail: [email protected]

KOREA

Power Integrations International Holdings, Inc. 8th Floor, dongSung Building, 17-8 Yoido-dong, Youngdeungpo-gu, Seoul, 150-874, Korea Phone: +82-2-782-2840 Fax: +82-2-782-4427

e-mail: [email protected]

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e-mail: [email protected]

JAPAN

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e-mail: [email protected]

TAIWAN

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e-mail: [email protected]

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e-mail: [email protected]

APPLICATIONS HOTLINE APPLICATIONS FAX

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9/02