Noty an61fa Linear Technology

Application Note 61

August 1994

Practical Circuitry for Measurement and Control Problems

Circuits Designed for a Cruel and Unyielding World

Jim Williams

INTRODUCTION

This collection of circuits was worked out between June
1991 and July of 1994. Most were designed at customer
request or are derivatives of such efforts. All represent
substantial effort and, as such, are disseminated here

1

for wider study and (hopefully) use.

The examples are
roughly arranged in categories including power conver­sion, transducer signal conditioning, amplifiers and signal
generators. As always, reader comment and questions
concerning variants of the circuits shown may be addressed
directly to the author.

Clock Synchronized Switching Regulator

Gated oscillator type switching regulators permit high
efficiency over extended ranges of output current. These
regulators achieve this desirable characteristic by using
a gated oscillator architecture instead of a clocked pulse
width modulator. This eliminates the “housekeeping”

V

IN

2V TO 3.2V

(2 CELLS)

currents associated with the continuous operation of fixed
frequency designs. Gated oscillator regulators simply
self-clock at whatever frequency is required to maintain
the output voltage. Typically, loop oscillation frequency
ranges from a few hertz into the kilohertz region, depend­ing upon the load.

In most cases this asynchronous, variable frequency opera­tion does not create problems. Some systems, however, are
sensitive to this characteristic. Figure 1 slightly modifies
a gated oscillator type switching regulator by synchroniz­ing its loop oscillation frequency to the systems clock. In
this fashion the oscillation frequency and its attendant
switching noise, albeit variable, become coherent with
system operation.

Note 1: “Study” is certainly a noble pursuit but we never fail to
emphasize use.

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.

SW2

47Ω

I

LIM

SW1

SET

GND

PRE1

CLR1

Q1

CLK1

74HC74

FLIP-FLOP

D1

PRE2CLR2Q1

V

CC

D2

Q2

CLK2

GND

47k

100kHz CLOCK

POWERED FROM

5V OUTPUT

100k

LT1107

A

OUT

V

1.2V

FB

REF

–

AUXILIARY

AMP

+

+

COMP

OSCILLATOR

–

L1 = COILTRONICS CTX-20-2
*= 1% METAL FILM RESISTOR

V

IN

V

REF

Figure 1. A Synchronizing Flip-Flop Forces Switching Regulator Noise to Be Coherent with the Clock

L1
22μH

1N5817

221k*

82.5k*

100k*

5V

OUT

+

100μF

AN61 F01

an61fa

AN61-1

Application Note 61

Circuit operation is best understood by temporarily ignor-

®

ing the flip-flop and assuming the LT

and FB pins are connected. When the output voltage

A

OUT

decays the set pin drops below V

1107 regulator’s

, causing A

REF

OUT

to
fall. This causes the internal comparator to switch high,
biasing the oscillator and output transistor into conduc­tion. L1 receives pulsed drive, and its flyback events are
deposited into the 100μF capacitor via the diode, restoring
output voltage. This overdrives the set pin, causing the IC
to switch off until another cycle is required. The frequency
of this oscillatory cycle is load dependent and variable. If,
as shown, a flip-flop is interposed in the A

-FB pin path,

OUT

synchronization to a system clock results. When the output
decays far enough (trace A, Figure 2) the A

pin (trace B)

OUT

goes low. At the next clock pulse (trace C) the flip-flop Q2
output (traceD) sets low, biasing the comparator-oscillator.
This turns on the power switch (V

pin is trace E), which

SW

pulses L1. L1 responds in flyback fashion, depositing its
energy into the output capacitor to maintain output volt­age. This operation is similar to the previously described
case, except that the sequence is forced to synchronize
with the system clock by the flip-flops action. Although the
resulting loops oscillation frequency is variable it, and all
attendant switching noise, is synchronous and coherent
with the system clock.

A start-up sequence is required because this circuit’s
clock is powered from its output. The start-up circuitry
was developed by Sean Gold and Steve Pietkiewicz of
LTC. The flip-flop’s remaining section is connected as a

A = 50mV/DIV

AC-COUPLED

B = 5V/DIV

C = 5V/DIV

D = 5V/DIV

E = 5V/DIV

20μs/DIV

Figure 2. Waveforms for the Clock Synchronized
Switching Regulator. Regulator Only Switches (TraceE)
on Clock Transitions (Trace C), Resulting in Clock
Coherent Output Noise (Trace A)

AN61 F02

buffer. The CLR1-CLK1 line monitors output voltage via the
resistor string. When power is applied Q1 sets CLR2 low.
This permits the LT1107 to switch, raising output voltage.
When the output goes high enough Q1 sets CLR2 high
and normal loop operation commences.

The circuit shown is a step-up type, although any switch­ing regulator configuration can utilize this synchronous
technique.

High Power 1.5V to 5V Converter

Some 1.5V powered systems (survival 2-way radios,
remote, transducer-fed data acquisition systems, etc.)
require much more power than stand-alone IC regulators
can provide. Figure 3’s design supplies a 5V output with
200mA capacity.

The circuit is essentially a flyback regulator. The LT1170
switching regulator’s low saturation losses and ease of
use permit high power operation and design simplicity.
Unfortunately this device has a 3V minimum supply require­ment. Bootstrapping its supply pin from the 5V output is
possible, but requires some form of start-up mechanism.

1.5V

IN

+

47μF

L1
25μH

LT1170

1k

6.8μF

V

GND

SW

V

FB

IN

I

LIM

V

IN

V

C

+

L1 = PULSE ENGINEERING #PE-92100
* = 1% METAL FILM RESISTOR

Figure 3. 200mA Output 1.5V to 5V Converter. Lower
Voltage LT1073 Provides Bootstrap Start-Up for LT1170
High Power Switching Regulator

SW2

SW1

LT1073

1N5823

SENSE

GND

3.74k*

+

1k*

240Ω*

AN61 F03

5V

OUT

200mA MAX

470μF

AN61-2

an61fa

Application Note 61

The 1.5V powered LT1073 switching regulator forms a
start-up loop. When power is applied the LT1073 runs,
causing its V

pin to periodically pull current through L1.

SW

L1 responds with high voltage flyback events. These events
are rectified and stored in the 470μF capacitor, producing
the circuits DC output. The output divider string is set up
so the LT1073 turns off when circuit output crosses about

4.5V. Under these conditions the LT1073 obviously can no
longer drive L1, but the LT1170 can. When the start-up
circuit goes off, the LT1170 V

pin has adequate supply

IN

voltage and can operate. There is some overlap between
start-up loop turn-off and LT1170 turn-on, but it has no
detrimental effect.

The start-up loop must function over a wide range of
loads and battery voltages. Start-up currents approach
1A, necessitating attention to the LT1073’s saturation and
drive characteristics. The worst case is a nearly depleted
battery and heavy output loading.

Figure 4 plots input-output characteristics for the circuit.
Note that the circuit will start into all loads with V

BAT

=

1.2V. Start-up is possible down to 1.0V at reduced loads.
Once the circuit has started, the plot shows it will drive full
200mA loads down to V
sible down to V

= 0.6V (a very dead battery)! Figure5

BAT

= 1.0V. Reduced drive is pos-

BAT

graphs efficiency at two supply voltages over a range of
output currents. Performance is attractive, although at
lower currents circuit quiescent power degrades efficiency.
Fixed junction saturation losses are responsible for lower
overall efficiency at the lower supply voltage.

1.5

= 5V

1.4

OUT

1.3

1.2

1.1

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1
0

MINIMUM INPUT VOLTAGE TO MAINTAIN V

START

RUN

120 140

100

0

60 80

20

40

OUTPUT CURRENT (mA)

160 180

200

AN61 F04

100

V

= 5V

OUT

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0 20 40 60 80 100 120

OUTPUT CURRENT (mA)

VIN = 1.5V

VIN = 1.2V

140 160 180 200

AN61 F05

Figure 5. Efficiency vs Operating Point for the 1.5V to
5V Converter. Efficiency Suffers at Low Power Because
of Relatively High Quiescent Currents

Low Power 1.5V to 5V Converter

Figure 6, essentially the same approach as the preced­ing circuit, was developed by Steve Pietkiewicz of LTC.
It is limited to about 150mA output with commensurate
restrictions on start-up current. It’s advantage, good ef­ficiency at relatively low output currents, derives from its
low quiescent power consumption.

The LT1073 provides circuit start-up. When output voltage,
sensed by the LT1073’s “set” input via the resistor divider,
rises high enough Q1 turns on, enabling the LT1302. This
device sees adequate operating voltage and responds by
driving the output to 5V, satisfying its feedback node. The
5V output also causes enough overdrive at the LT1073
feedback pin to shut the device down.

Figure 7 shows maximum permissible load currents for
start-up and running conditions. Performance is quite
good, although the circuit clearly cannot compete with the
previous design. The fundamental difference between the
two circuits is the LT1170’s (Figure 3) much larger power
switch, which is responsible for the higher available power.
Figure 8, however, reveals another difference. The curves
show that Figure 6 is significantly more efficient than the
LT1170 based approach at output currents below 100mA.
This highly desirable characteristic is due to the LT1302’s
much lower quiescent operating currents.

Figure 4. Input-Output Data for the 1.5V to 5V
Converter Shows Extremely Wide Start-Up and
Running Range into Full Load

an61fa

AN61-3

Application Note 61

220Ω

1.5V
CELL

C1 = AVX TPSD476M016R0150
C2 = AVX TPSE227M010R0100

SET

FB

I

LIM

GND

LT1073

V

SW2

IN

SW1

A

L1

3.3μH

+

C1
47μF

O

L1 = COILCRAFT DO3316-332
D1 = MOTOROLA MBR3130LT3

D1

100k

220μF

10Ω

NC

+

C2

0.1μF

Figure 6. Single-Cell to 5V Converter Delivers 150mA
with Good Efficiency at Lower Currents

SW

I

LIM

V

IN

PGND

100k

LT1302

SHDN

GND

Q1
2N3906

100k

FB

V

C

0.01μF

100pF

20k

R1
301k
1%

56.2k
1%

4.99k
1%

36.5k
1%

5V

OUT

AN61 F06

1000

100

10

MAXIMUM LOAD CURRENT (mA )

1

RUN

START

0.6

1.0 1.2 1.4 1.81.6

0.8 2.0
INPUT VOLTAGE (V)

AN61 F07

Figure 7. Maximum Permissible Loads for Start-Up
and Running Conditions. Allowable Load Current
During Start-Up Is Substantially Less Than Maximum
Running Current.

72
70
68
66
64
62
60
58

EFFICIENCY (%)

56
54
52
50
48

VIN = 1.5V

VIN = 1.2V

1

10 100 1000

LOAD CURRENT (mA)

AN61 F08

Figure 8. Efficiency Plot for Figure 6. Performance
Is Better Than the Previous Circuit at Lower
Currents, Although Poorer at High Power

AN61-4

an61fa

Application Note 61

Low Power, Low Voltage Cold Cathode Fluorescent
Lamp Power Supply

Most Cold Cathode Fluorescent Lamp (CCFL) circuits
require an input supply of 5V to 30V and are optimized
for bulb currents of 5mA or more. This precludes lower
power operation from 2- or 3-cell batteries often used in
palmtop computers and portable apparatus. A CCFL power
supply that operates from 2V to 6V is detailed in Figure 9.
This circuit, contributed by Steve Pietkiewicz of LTC, can
drive a small CCFL over a 100μA to 2mA range.

The circuit uses an LT1301 micropower DC/DC converter IC
in conjunction with a current driven Royer class converter
comprised of T1, Q1 and Q2. When power and intensity
adjust voltage are applied the LT1301’s I

pin is driven

LIM

slightly positive, causing maximum switching current
through the IC’s internal switch pin (SW). Current flows
from T1’s center tap, through the transistors, into L1. L1’s
current is deposited in switched fashion to ground by the
regulator’s action.

The Royer converter oscillates at a frequency primarily
set by T1’s characteristics (including its load) and the

0.068μF capacitor. LT1301 driven L1 sets the magnitude of
the Q1-Q2 tail current, hence T1’s drive level. The 1N5817
diode maintains L1’s current flow when the LT1301’s
switch is off. The 0.068μF capacitor combines with T1’s
characteristics to produce sine wave voltage drive at the
Q1 and Q2 collectors. T1 furnishes voltage step-up and
about 1400Vp-p appears at its secondary. Alternating cur­rent flows through the 22pF capacitor into the lamp. On
positive half-cycles the lamp’s current is steered to ground
via D1. On negative half-cycles the lamp’s current flows
through Q3’s collector and is filtered by C1. The LT1301’s

pin acts as a 0V summing point with about 25μA

I

LIM

bias current flowing out of the pin into C1. The LT1301
regulates L1’s current to equalize Q3’s average collector
current, representing 1/2 the lamp current, and R1’s cur­rent, represented by V
to DC. When V

is set to zero, the I

A

/R1. C1 smooths all current flow

A

pin’s bias current

LIM

forces about 100μA bulb current.

97

V

2V TO 6V

IN

NC

0.1μF

SHUTDOWN

T1 = COILTRONICS CTX110654-1
L1 = COILCRAFT D03316-473

0.68μF = WIMA MKP-20

V

IN

SENSE

SHDN

GND

1Ω

LT1301

SELECT

SW

I

LIM

PGND

1N5817

L1

47μH

+

C1
1μF

INTENSITY ADJUST

100μA TO 2mA BULB CURRENT

R1

7.5K
1%

V

0V TO 5VDC

120Ω

Q3

2N3904

A

IN

15TI4

Q1

ZTX849

D1
1N4148

32

0.68μF

+

10μF

Q2
ZTX849

AN61 F09

Figure 9. Low Power Cold Cathode Fluorescent Lamp Supply Is Optimized for Low Voltage Inputs and Small Lamps

22pF
3kV

CCFL

an61fa

AN61-5

Application Note 61

Circuit efficiency ranges from 80% to 88% at full load, de­pending on line voltage. Current mode operation combined
with the Royer’s consistent waveshape vs input results
in excellent line rejection. The circuit has none of the line
rejection problems attributable to the hysteretic voltage
control loops typically found in low voltage micropower
DC/DC converters. This is an especially desirable charac­teristic for CCFL control, where lamp intensity must remain
constant with shifts in line voltage. Interaction between
the Royer converter, the lamp and the regulation loop is
far more complex than might be supposed, and subject
to a variety of considerations. For detailed discussion see
Reference 3.

Low Voltage Powered LCD Contrast Supply

Figure 10, a companion to the CCFL power supply previ­ously described, is a contrast supply for LCD panels. It
was designed by Steve Pietkiewicz of LTC. The circuit is
noteworthy because it operates from a 1.8V to 6V input,
significantly lower than most designs. In operation the
LT1300/LT1301 switching regulator drives T1 in flyback

fashion, causing negative biased step-up at T1’s sec­ondary. D1 provides rectification, and C1 smooths the
output to DC. The resistively divided output is compared
to a command input, which may be DC or PWM, by the
IC’s “I
the I

” pin. The IC, forcing the loop to maintain 0V at

LIM

pin, regulates circuit output in proportion to the

LIM

command input.
Efficiency ranges from 77% to 83% as supply voltage

varies from 1.8V to 3V. At the same supply limits, available
output current increases from 12mA to 25mA.

HeNe Laser Power Supply

Helium-Neon lasers, used for a variety of tasks, are dif­ficult loads for a power supply. They typically need almost
10kV to start conduction, although they require only about
1500V to maintain conduction at their specified operating
currents. Powering a laser usually involves some form
of start-up circuitry to generate the initial breakdown
voltage and a separate supply for sustaining conduction.
Figure11’s circuit considerably simplifies driving the laser.

V

1.8V TO 6V

IN

+

100μF

T1 = DALE LPE-5047-AO45

NC

NC

V

IN

SENSE

SELECT

PGND

LT1300

OR

LT1301

1

10

SW

SHDN

I

LIM

GND

T1

SHUTDOWN

4

7
3

8
2

9

1N5819

D1

COMMAND INPUT

PWM OR DC
0% TO 100%
OR 0V TO 5V

150k

12k

12k

CONTRAST OUTPUT
V

OUT

C1
22μF

+

35V

+

2.2μF

–4V TO –29V

AN61 F10

AN61-6

Figure 10. Liquid Crystal Display Contrast Supply Operates from 1.8V to 6V with –4V to –29V Output Range

an61fa

Application Note 61

The start-up and sustaining functions have been combined
into a single, closed-loop current source with over 10kV
of compliance. The circuit is recognizable as a reworked

2

CCFL power supply with a voltage tripled DC output.
When power is applied, the laser does not conduct and

the voltage across the 190Ω resistor is zero. The LT1170
switching regulator FB pin sees no feedback voltage, and
its switch pin (V

) provides full duty cycle pulse width

SW

modulation to L2. Current flows from L1’s center tap
through Q1 and Q2 into L2 and the LT1170. This current
flow causes Q1 and Q2 to switch, alternately driving L1.
The 0.47μF capacitor resonates with L1, providing boosted
sine wave drive. L1 provides substantial step-up, causing

0.01μF
5kV

about 3500V to appear at its secondary. The capacitors and
diodes associated with L1’s secondary form a voltage tripler,
producing over 10kV across the laser. The laser breaks
down and current begins to flow through it. The 47k resistor
limits current and isolates the laser’s load characteristic.
The current flow causes a voltage to appear across the
190Ω resistor. A filtered version of this voltage appears
at the LT1170 FB pin, closing a control loop. The LT1170
adjusts pulse width drive to L2 to maintain the FB pin at

1.23V, regardless of changes in operating conditions. In
this fashion, the laser sees constant current drive, in this

Note 2: See References 2 and 3 and this text’s Figure 9.

1800pF

10kV

1800pF

10kV

47k
5W

V

IN

9V TO 35V

HV DIODES =

0.47μF =
Q1, Q2 =

LASER =

L1

5

+

SEMTECH-FM-50
WIMA 3w 0.15μF TYPE MKP-20
ZETEX ZTX849

L1 =

COILTRONICS CTX02-11128-2

L2 =

PULSE ENGINEERING PE-92105
HUGHES 3121H-P

4

150Ω

MUR405

2.2μF

10μF

V

1

+

IN

V

11

Q1

V

LT1170

C

SW

0.47μF

GND

2

L2
145μH

FB

8

Q2

3

HV DIODES

+

2.2μF

10k

0.1μF

V

IN

10k

1N4002
(ALL)

LASER

190Ω
1%

AN61 F11

Figure 11. LASER Power Supply Is Essentially A 10,000V Compliance Current Source

an61fa

AN61-7

Application Note 61

case 6.5mA. Other currents are obtainable by varying the
190Ω value. The 1N4002 diode string clamps excessive
voltages when laser conduction first begins, protecting
the LT1170. The 10μF capacitor at the V

pin frequency

C

compensates the loop and the MUR405 maintains L1’s
current flow when the LT1170 V

pin is not conducting.

SW

The circuit will start and run the laser over a 9V to 35V
input range with an electrical efficiency of about 80%.

Compact Electroluminescent Panel Power Supply

Electroluminescent (EL) panel LCD backlighting presents an
attractive alternative to fluorescent tube (CCFL) backlighting
in some portable systems. EL panels are thin, lightweight,
lower power, require no diffuser and work at lower voltage
than CCFLs. Unfortunately, most EL DC/AC inverters use a

0.1μF
100V

L1

V

SW2

100μH

+

IN

SW1

2.26M

FB

30.1k

R1

25k

V

2V TO 12V

33pF

IN

+

= MOTOROLA MURS120T3
L1 = COILCRAFT DO3316-104

47Ω

I

LIM

U1

LT1108CS8

GND

large transformer to generate the 400Hz 95V square wave
required to drive the panel. Figure 12’s circuit, developed
by Steve Pietkiewicz of LTC, eliminates the transformer
by employing an LT1108 micropower DC/DC converter
IC. The device generates a 95VDC potential via L1 and the
diode-capacitor doubler network. The transistors switch
the EL panel between 95V and ground. C1 blocks DC and
R1 allows intensity adjustment. The 400Hz square wave
drive signal can be supplied by the microprocessor or a
simple multivibrator. When compared to conventional EL
panel supplies, this circuit is noteworthy because it can
be built in a square inch with a 0.5 inch height restriction.
Additionally, all components are surface mount types, and
the usual large and heavy 400Hz transformer is eliminated.

95V

1M

MMBTA42

0.1μF
100V

0.47μF
200V

MMBTA42

10k

MMBTA42

400Hz DRIVE
SQUARE WAVE

C1

1μF

100V

MMBTA92

EL

PANEL

AN61 F12

AN61-8

Figure 12. Switch Mode EL Panel Driver Eliminates Large 400Hz Transformer

an61fa

Application Note 61

3.3V Powered Barometric Pressure Signal Conditioner

The move to 3.3V digital supply voltage creates problems
for analog signal conditioning. In particular, transducer
based circuits often require higher voltage for proper
transducer excitation. DC/DC converters in standard
configurations can address this issue but increase power
consumption. Figure 13’s circuit shows a way to provide
proper transducer excitation for a barometric pressure
sensor while minimizing power requirements.

The 6kΩ transducer T1 requires precisely 1.5mA of ex­citation, necessitating a relatively high voltage drive. A1
senses T1’s current by monitoring the voltage drop across
the resistor string in T1’s return path.

A = 10

–

1/2 LT1078

+

LT1034

1.2V

A1

≈10V DURING OPERATIONT1

3.3V

+

A3

1/2 LT1078

–

0.05μF

1μF

1N4148

10k

3.3V

PRESSURE

TRANSDUCER

5

10

6

BRIDGE

CURRENT

TRIM

BRIDGE
CURRENT
MONITOR
(0.1500V)

* = 1% FILM RESISTOR
** = 0.1% FILM RESISTOR
L1 = TOKO 262-LYF-0095K
T1 = NOVASENSOR (FREMONT, CA)
NPH-8-100AH

4

700Ω*

50Ω

100Ω**

100k

1μF

–

A2

LT1101

+

A1’s output biases the LT1172 switching regulator’s op­erating point, producing a stepped up DC voltage which
appears as T1’s drive and A2’s supply voltage. T1’s return
current out of pin 6 closes a loop back at A1 which is
slaved to the 1.2V reference. This arrangement provides
the required high voltage drive (≈10V) while minimizing
power consumption. This is so because the switching
regulator produces only enough voltage to satisfy T1’s
current requirements. Instrumentation amplifier A2 and A3

®

provide gain and LTC

1287 A/D converter gives a 12-bit
digital output. A2 is bootstrapped off the transducer supply,
enabling it to accept T1’s common-mode voltage. Circuit
current consumption is about 14mA. If the shutdown pin
is driven high the switching regulator turns off, reducing

2N3904

10k*

1M
CALIB

2.2μF

100k

+

3.3V

+

22μF

V

V

C

SHUTDOWN

1N752

5.6V

MUR110

IN

LT1172

TO PROCESSOR

CS

+IN

–IN

3.3V

L1
150μH

V

SW

FB

GNDE2E1

CLK D

LTC1287

GND

NC

OUT

V

REF

V

CC

1N4148

AN61 F13

3.3V

Figure 13. 3.3V Powered, Digital Output, Barometric Pressure Signal Conditioner

an61fa

AN61-9

Application Note 61

total power consumption to about 1mA. In shutdown the

3.3V powered A/D’s output data remains valid. In practice,
the circuit provides a 12-bit representation of ambient
barometric pressure after calibration. To calibrate, adjust
the “bridge current trim” for exactly 0.1500V at the indicated
point. This sets T1’s current to the manufacturers speci­fied point. Next, adjust A3’s trim so that the digital output
corresponds to the known ambient barometric pressure.
If a pressure standard is not available the transducer is
supplied with individual calibration data, permitting circuit
calibration.

Some applications may require operation over a wider
supply range and/or a calibrated analog output. Figure14’s
circuit is quite similar, except that the A/D converter is
eliminated and a 2.7V to 7V supply is acceptable. The
calibration procedure is identical, except that A3’s analog
output is monitored.

5

T1

10

6

4

–

LT1101

+

A = 10

A2

+

1/2 LT1078

–

Single Cell Barometers

It is possible to power these circuits from a single cell with­out sacrificing performance. Figure 15, a direct extension
of the above approaches, simply substitutes a switching
regulator that will run from a single 1.5V battery. In other
respects loop action is nearly identical.

Figure 16, also a 1.5V powered design, is related but
eliminates the instrumentation amplifier. As before, the
6kΩ transducer T1 requires precisely 1.5mA of excitation,
necessitating a relatively high voltage drive. A1’s positive
input senses T1’s current by monitoring the voltage drop
across the resistor string in T1’s return path. A1’s negative
input is fixed by the 1.2V LT1004 reference. A1’s output
biases the 1.5V powered LT1110 switching regulator. The
LT1110’s switching produces two outputs from L1. Pin4’s
rectified and filtered output powers A1 and T1. A1’s output,

A3

20k*

OUTPUT
0V TO 3.100V =
0 TO 31.00" Hg.

2.2μF

1N4148

+

3.3V

+

22μF

3.3V

MUR110

V

IN

V

C

V

LT1172

GNDE2E1

1.500mA

700Ω*

BRIDGE

CURRENT

* = 1% FILM RESISTOR
L1 = TOKO 262-LYF-0095K
T1 = NOVASENSOR NPH-8-100AH

50Ω

TRIM

100Ω

0.1%

100k

–

1/2 LT1078

+

10k

LT1034

1.2V

1μF

10k*

OUTPUT

A1

3.3V

TRIM

1k

Figure 14. Single Supply Barometric Pressure Signal Conditioner Operates Over a 2.7V to 7V Range

SW

L1
150μH

FB

NC

AN61 F13

an61fa

AN61-10

Application Note 61

T1

5

10

50Ω
TRIM FOR
150mV AT
POINT “A”

* = 1% FILM RESISTOR
L1 = COILTRONICS CTX50-1
T1 = NOVASENSOR NPH-8-100AH

4

6

700Ω
1%

A

100Ω

0.1%

–

A2

LT1101

+

100k

LT1004-1.2

A =10

+

A1

1/2 LT1078

–

100k

+

–

A3

LT1078

10k

OUTPUT
0 TO 3.100V=
0 TO 31.00"Hg

200k*

100k*

10k

CAL

1.5V
AA CELL

+

1μF

V

IN

FB

LT1110

SET

150

I

L

SW1

AONC NC

SW2GND

L1

50μH

1N5818

100Ω

+

150μF

AN61 F15

Figure 15. 1.5V Powered Barometric Pressure Signal Conditioner Uses
Instrumentation Amplifier and Voltage Boosted Current Loop

in turn, closes a feedback loop at the regulator. This loop
generates whatever voltage step-up is required to force
precisely 1.5mA through T1. This arrangement provides
the required high voltage drive while minimizing power
consumption. This occurs because the switching regula­tor produces only enough voltage to satisfy T1’s current
requirements.

L1 pins 1 and 2 source a boosted, fully floating voltage,
which is rectified and filtered. This potential powers A2.
Because A2 floats with respect to T1, it can look differ­entially across T1’s outputs, pins 10 and 4. In practice,
pin10 becomes “ground” and A2 measures pin 4’s output
with respect to this point. A2’s gain-scaled output is the
circuit’s output, conveniently scaled at 3.000V = 30.00"Hg.
A2’s floating drive eliminates the requirement for an
instrumentation amplifier, saving cost, power, space and
error contribution.

To calibrate the circuit, adjust R1 for 150mV across the
100Ω resistor in T1’s return path. This sets T1’s current
to the manufacturer’s specified calibration point. Next,
adjust R2 at a scale factor of 3.000V = 30.00"Hg. If R2
cannot capture the calibration, reselect the 200k resistor
in series with it. If a pressure standard is not available,
the transducer is supplied with individual calibration data,
permitting circuit calibration.

This circuit, compared to a high-order pressure standard,
maintained 0.01"Hg accuracy over months with widely
varying ambient pressure shifts. Changes in pressure,
particularly rapid ones, correlated quite nicely to changing
weather conditions. Additionally, because 0.01"Hg corre­sponds to about 10 feet of altitude at sea level, driving over
hills and freeway overpasses becomes quite interesting.

an61fa

AN61-11

Application Note 61

"Hg.

0V TO 3.100V =

0 TO 31.00

OUTPUT

200k*

0.1μF

A2

LT1077

+

–

R2

10k

1%

1k

1%

100Ω

+

1N4148

100μF

100Ω

1N5818

1

2

390μF

16V

NICHICON

PL

+

430k

4

L1

3

L

I

150Ω

21

IN

V

83

1μF

SW1

FB

LT1110

AN61 F16

39k

7

SET

SW2

54

GND

AO

6

L1 = COILTRONICS CTX50-1

68k

5

†

T1

NOVASENSOR

NPH-8-100AH

4

10

1μF

NON-POLAR

+

= LUCAS NOVASENSOR

FREMONT, CA [phone]

†

100k

100Ω

0.1%

AA CELL

A2

LT1004-1.2

Figure 16. 1.5V Powered Barometric Pressure Signal Conditioner Floats Bridge Drive to

Eliminate Instrumentation Amplifier. Voltage Boosted Current Loop Drives Transducer

* = NOMINAL VALUE. EACH SENSOR REQUIRES SELECTION

** = TRIM FOR 150mV ACROSS A1-A2

100k

R1**

1N4148

50Ω

A1

0.1μF

A1

LT1077

+

6

100k

698Ω

–

1%

AN61-12

an61fa