Noty an43f Linear Technology

Bridge Circuits

Marrying Gain and Balance

Jim Williams

Application Note 43

June 1990

Bridge circuits are among the most elemental and powerful
electrical tools. They are found in measurement, switch­ing, oscillator and transducer circuits. Additionally, bridge
techniques are broadband, serving from DC to bandwidths
well into the GHz range. The electrical analog of the me­chanical beam balance, they are also the progenitor of all
electrical differential techniques.

Resistance Bridges

Figure 1 shows a basic resistor bridge. The circuit is
usually credited to Charles Wheatstone, although S. H.
Christie, who demonstrated it in 1833, almost certainly

1

preceded him.

If all resistor values are equal (or the two
sides ratios are equal) the differential voltage is zero. The
excitation voltage does not alter this, as it affects both
sides equally. When the bridge is operating off null, the
excitation’s magnitude sets output sensitivity. The bridge
output is nonlinear for a single variable resistor. Similarly,
two variable arms (e.g., R

and RB both variable) produce

C

nonlinear output, although sensitivity doubles. Linear
outputs are possible by complementary resistance swings
in one or both sides of the bridge.

A great deal of attention has been directed towards this
circuit. An almost uncountable number of tricks and tech­niques have been applied to enhance linearity, sensitivity

R

EXCITATION

VOLTAGE

A

DIFFERENTIAL

OUTPUT

+

VOLTAGE

R

B

R

C

R

D

and stability of the basic configuration. In particular, trans­ducer manufacturers are quite adept at adapting the bridge
to their needs (see Appendix A, “Strain Gauge Bridges”).
Careful matching of the transducer’s mechanical charac­teristics to the bridge’s electrical response can provide a
trimmed, calibrated output. Similarly, circuit designers
have altered performance by adding active elements (e.g.,
amplifiers) to the bridge, excitation source or both.

Bridge Output Amplifiers

A primary concern is the accurate determination of the
differential output voltage. In bridges operating at null the
absolute scale factor of the readout device is normally
less important than its sensitivity and zero point stability.
An off-null bridge measurement usually requires a well
calibrated scale factor readout in addition to zero point
stability. Because of their importance, bridge readout
mechanisms have a long and glorious history (see Ap­pendix B, “Bridge Readout—Then and Now”). Today’s
investigator has a variety of powerful electronic techniques
available to obtain highly accurate bridge readouts. Bridge
amplifiers are designed to accurately extract the bridges
differential output from its common mode level. The
ability to reject common mode signal is quite critical. A
typical 10V powered strain gauge transducer produces
only 30mV of signal “riding” on 5V of common mode
level. 12-bit readout resolution calls for an LSB of only

7.3μV…..almost 120dB below the common mode signal!
Other significant error terms include offset voltage, and
its shift with temperature and time, bias current and gain
stability. Figure 2 shows an “Instrumentation Amplifier,”
which makes a very good bridge amplifier. These devices
are usually the first choice for bridge measurement,
and bring adequate performance to most applications.

AN43 F01

Figure 1. The Basic Wheatstone Bridge,
Invented by S. H. Christie

Note 1: Wheatstone had a better public relations agency, namely himself.

For fascinating details, see reference 19.

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.

an43f

AN43-1

Application Note 43

In general, instrumentation amps feature fully differential
inputs and internally determined stable gain. The absence
of a feedback network means the inputs are essentially pas­sive, and no significant bridge loading occurs. Instrumenta­tion amplifiers meet most bridge requirements. Figure 3
lists performance data for some specific instrumentation
amplifiers. Figure 4’s table summarizes some options
for DC bridge signal conditioning. Various approaches
are presented, with pertinent characteristics noted. The
constraints, freedoms and performance requirements of
any particular application define the best approach.

+

–

mNO FEEDBACK RESISTORS USED
mGAIN FIXED INTERNALLY (TYP 10 OR 100)

OR SOMETIMES RESISTOR PROGRAMMABLE
mBALANCED, PASSIVE INPUTS

Figure 2. Conceptual Instrumentation Amplifier

AN43 F02

DC Bridge Circuit Applications

Figure 5, a typical bridge application, details signal con­ditioning for a 350Ω transducer bridge. The specified
strain gauge pressure transducer produces 3mV output
per volt of bridge excitation (various types of strain-based
transducers are reviewed in Appendix A, “Strain Gauge

®

Bridges”). The LT

1021 reference, buffered by A1A and
A2, drives the bridge. This potential also supplies the
circuits ratio output, permitting ratiometric operation of
a monitoring A/D converter. Instrumentation amplifier
A3 extracts the bridge’s differential output at a gain of

100, with additional trimmed gain supplied by A1B. The
configuration shown may be adjusted for a precise 10V
output at full-scale pressure. The trim at the bridge sets
the zero pressure scale point. The RC combination at A1B’s
input filters noise. The time constant should be selected
for the system’s desired lowpass cutoff. “Noise” may
originate as residual RF/line pick-up or true transducer
responses to pressure variations. In cases where noise
is relatively high it may be desirable to filter ahead of A3.
T h i s p r e v e n t s a n y p o s s i b l e s i g n a l i n f i d e l i t y d u e t o n o n l i n e a r
A3 operation. Such undesirable outputs can be produced
by saturation, slew rate components, or rectification
effects. When filtering ahead of the circuits gain blocks
remember to allow for the effects of bias current induced
errors caused by the filter’s series resistance. This can be
a significant consideration because large value capacitors,
particularly electrolytics, are not practical. If bias current
induced errors rise to appreciable levels FET or MOS input
amplifiers may be required (see Figure 3).

To trim this circuit apply zero pressure to the transducer
and adjust the 10k potentiometer until the output just
comes off 0V. Next, apply full-scale pressure and trim the
1k adjustment. Repeat this procedure until both points
are fixed.

Common Mode Suppression Techniques

Figure 6 shows a way to reduce errors due to the bridges
common mode output voltage. A1 biases Q1 to servo the
bridges left mid-point to zero under all operating condi­tions. The 350Ω resistor ensures that A1 will find a stable
operating point with 10V of drive delivered to the bridge.
This allows A2 to take a single-ended measurement,

PARAMETER LTC1100 LT1101 LT1102

Offset
Offset Drift
Bias Current
Noise (0.1Hz to 10Hz)
Gain
Gain Error
Gain Drift
Gain Nonlinearity
CMRR
Power Supply
Supply Current
Slew Rate
Bandwidth

10μV
100nV/°C
50pA
2μV

P-P

100

0.03%
4ppm/°C
8ppm
104dB
Single or Dual, 16V Max

2.2mA

1.5V/μs
8kHz

Figure 3. Comparison of Some IC Instrumentation Amplifiers

160μV
2μV/°C
8nA

0.9μV
10,100

0.03%
4ppm/°C
8ppm
100dB
Single or Dual, 44V Max
105μA

0.07V/μs
33kHz

500μV

2.5μV/°C
50pA

2.8μV
10,100

0.05%
5ppm/°C
10ppm
100dB
Dual, 44V Max
5mA
25V/μs
220kHz

AN43-2

(USING LTC1050 AMPLIFIER)

LTC1043

0.5μV
50nV/°C
10pA

1.6μV
Resistor Programmable
Resistor Limited 0.001% Possible
Resistor Limited <1ppm/°C Possible
Resistor Limited 1ppm Possible
160dB
Single, Dual 18V Max
2mA
1mV/ms
10Hz

an43f

Application Note 43

CONFIGURATION ADVANTAGES DISADVANTAGES

+V

RATIO
OUT

Best general choice. Simple,
straightforward. CMRR typically
>110dB, drift 0.05μV/°C to 2μV/°C,
gain accuracy 0.03%, gain drift
4ppm/°C, noise 10nV√Hz – 1.5μV
for chopper-stabilized types. Direct
ratiometric output.

AN43 F04a

+

–

INSTRUMENTATION

AMPLIFIER

OUT

CMRR, drift and gain stability
may not be adequate in highest
precision applications. May require
second stage to trim gain.

+V

RATIO
OUT

CMRR > 120dB, drift 0.05μV/°C.
Gain accuracy 0.001% possible.
Gain drift 1ppm with appropriate
resistors. Noise 10nV√Hz – 1.5μV

Multi-package—moderately
complex. Limited bandwidth.
Requires feedback resistors to set
gain.

for chopper-stabilized types. Direct

+

OUT

–

ratiometric output. Simple gain
trim. Flying capacitor commutation
provides lowpass filtering. Good
choice for very high performance—
monolithic versions (LTC1043)
available.

OP AMP

AN43 F04b

CMRR > 160dB, drift 0.05μV/°C to

0.25μV/°C, gain accuracy 0.001%
possible, gain drift 1ppm/°C with
appropriate resistors plus floating

Requires floating supply. No direct
ratiometric output. Floating supply
drift is a gain term. Requires
feedback resistors to set gain.

supply error, simple gain trim,

+

OUT

+

Noise 1nV√Hz possible.

–

OP AMP

+V

AN43 F04c

CMRR ≈ 140dB, drift 0.05μV/°C to

0.25μV/°C, gain accuracy 0.001%
possible, gain drift 1ppm/°C with
appropriate resistors plus floating
supply error, simple gain trim,
noise 1nV√Hz possible.

+

OUT

–

No direct ratiometric output.
Zener supply is a gain and offset
term error generator. Requires
feedback resistors to set gain.
Low impedance bridges require
substantial current from shunt
regulator or circuitry which
simulates it. Usually poor choice if
precision is required.

–V

OP AMP

AN43 F04d

Figure 4. Some Signal Conditioning Methods for Bridges

an43f

AN43-3

Application Note 43

CONFIGURATION ADVANTAGES DISADVANTAGES

+V

+

–

OP AMP

RATIO
OUT

AN43 F04e

CMRR > 160dB, drift 0.05μV/°C to

0.25μV/°C, gain accuracy 0.001%
possible, gain drift 1ppm/°C with
appropriate resistors, simple gain
trim, ratiometric output, noise
1nV√Hz possible.

+

OUT

+

Requires precision analog level
shift, usually with isolation
amplifier. Requires feedback
resistors to set gain.

+V

+

–

–V

+

–

OP AMP

+V

+

–

OP AMP

AN43 F04f

RATIO
OUT

OUT

AN43 F04g

RATIO
OUT

OUT

CMRR ≈ 120dB to 140dB, drift

0.05μV/°C to 0.25μV/°C, gain
accuracy 0.001% possible, gain
drift 1ppm/°C with appropriate
resistors, simple gain trim, direct
ratiometric output, noise 1nV√Hz
possible.

CMRR = 160dB, drift 0.05μV/°C to

0.25μV/°C, gain accuracy 0.001%
possible, gain drift 1ppm/°C,
simple gain trim, direct ratiometric
output, noise 1nV√Hz possible.

Requires tracking supplies.
Assumes high degree of bridge
symmetry to achieve best CMRR.
Requires feedback resistors to set
gain.

Practical realization requires two
amplifiers plus various discrete
components. Negative supply
necessary.

Figure 4. Some Signal Conditioning Methods for Bridges (Continued)

eliminating all common mode voltage errors. This approach
works well, and is often a good choice in high precision
work. The amplifiers in this example, CMOS chopper-sta­bilized units, essentially eliminate offset drift with time and
temperature. Trade-offs compared to an instrumentation
amplifier approach include complexity and the require­ment for a negative supply. Figure 7 is similar, except that
low noise bipolar amplifiers are used. This circuit trades
slightly higher DC offset drift for lower noise and is a good
candidate for stable resolution of small, slowly varying
measurands. Figure 8 employs chopper-stabilized A1 to

AN43-4

reduce Figure 7’s already small offset error. A1 measures
the DC error at A2’s inputs and biases A1’s offset pins to
force offset to a few microvolts. The offset pin biasing at
A2 is arranged so A1 will always be able to find the servo
point. The 0.01μF capacitor rolls off A1 at low frequency,
with A2 handling high frequency signals. Returning A2’s
feedback string to the bridges mid-point eliminates A4’s
offset contribution. If this was not done A4 would require
a similar offset correction loop. Although complex, this
approach achieves less than 0.05μV/°C drift, 1nV√Hz noise
and CMRR exceeding 160dB.

an43f

10k

ZERO

301k*

15V 15V

A2

LT1010

350Ω STRAIN GAGE

PRESSURE TRANSDUCER

15V

+

A3

LT1101

A = 100

–

1/2 LT1078

100k

A1A

+

–

0.33

LT1021

+

A1B

1/2 LT1078

–

Application Note 43

15V

10V

10V RATIO
OUTPUT

OUTPUT
0V TO 10V =
0 TO 250 PSI

10k*

*1% FILM RESISTOR
PRESSURE TRANSDUCER =
BLH #DHF-350—3MV/VOLT GAIN FACTOR

Figure 5. A Practical Instrumentation Amplifier-Based Bridge Circuit

0.02

A1

LTC1150

*1% FILM RESISTOR

Figure 6. Servo Controlling Bridge Drive Eliminates Common Mode Voltage

3.65k*

1k – GAIN

AN43 F05

350Ω

15V

1/2W

10μF

+

OUTPUT

TRIM
100Ω

250* 100k*

RATIO
OUTPUT

–

350Ω

100k

–

+

1k

STRAIN
GAUGE

BRIDGE

3MV/V

TYPE

Q1
2N2905

–15V

LTC1150

+

A2

AN43 F06

OUTPUT
0V TO 10V

Single Supply Common Mode Suppression Circuits

The common mode suppression circuits shown require a
negative power supply. Often, such circuits must function
in systems where only a positive rail is available. Figure 9

®

shows a way to do this. A2 biases the LTC

1044 positive­to-negative converter. The LTC1044’s output pulls the
bridge’s output negative, causing A1’s input to balance at
0V. This local loop permits a single-ended amplifier (A2)

to extract the bridge’s output signal. The 100k-0.33μF RC
filters noise and A2’s gain is set to provide the desired
output scale factor. Because bridge drive is derived from
the LT1034 reference, A2’s output is not affected by supply
shifts. The LT1034’s output is available for ratio operation.
Although this circuit works nicely from a single 5V rail the
transducer sees only 2.4V of drive. This reduced drive

an43f

AN43-5

Application Note 43

15V

LT1021-5

350Ω

BRIDGE

3

5V

2

2

–

A3

LT1028

3

+

–15V

+

–

15V

7

A1

LT1007

4

–15V

7

6

4

330Ω

6

RATIO
REFERENCE
OUT

15V

330Ω

301k*

10k
ZERO
TRIM

*1% FILM RESISTOR

3

2

+

LT1028

–

A3

–15V

7

6

4

1μF

5k
GAIN
TRIM

Figure 7. Low Noise Bridge Amplifier with Common Mode Suppression

15V

0V TO 10V
OUTPUT

30.1k*

49.9Ω*

AN43 F07

LT1021-5

350Ω

BRIDGE

*1% FILM RESISTOR

3

+

5V

2

–

15V

2

–

A4

LT1028

3

+

–15V

A3

LT1007

7

4

–15V

7

4

330Ω

6

REFERENCE
OUT

+

301k*

10k
ZERO
TRIM

6

330Ω

A1

LTC1150

–

0.01

15V

130Ω100k 30k

1

7

+

LT1028

–

A2

–15V

8

4

5k
GAIN
TRIM

68Ω

15V

OUTPUT

30.1k*

(A = 1000)

49.9Ω*

AN43 F08

AN43-6

Figure 8. Low Noise, Chopper-Stabilized Bridge Amplifier with Common Mode Suppression

an43f

Application Note 43

1.2V REFERENCE OUTPUT
TO A/D CONVERTER
FOR RATIOMETRIC

OPERATION. 0.1mA MAXIMUM
10k
ZERO
TRIM

+

A2

1/2 LT1078

39k

0.1μF

–

A1

1/2 LT1078

5V

220

LT1034

1.2V

+

PRESSURE
TRANSDUCER
350Ω

D

E A

C

2.4V

301k

100k

0.33μF

–

100μF

8

+

V

2

+

CAP

+

3

LTC1044
CAP
GND

V

–

OUT

5

LV

100μF

+

64

*1% FILM RESISTOR
PRESSURE TRANSDUCER-BLH/DHF-350
CIRCLED LETTER IS PIN NUMBER

= 350Ω

Z

IN

0.047μF

Figure 9. Single Supply Bridge Amplifier with Common Mode Suppression

OUTPUT
0V TO 3.5V =
0 TO 350 PSI

2k
GAIN
TRIM

46k*

100Ω*

AN43 F09

40Ω

5V

10μF

+

5k

OUTPUT

TRIM

5k*

1M*

–

A2

1/2 LT1078

+

OUTPUT
0V TO 3V

AN43 F10

10μF

350Ω

3k

STRAIN

GAUGE

BRIDGE

3mV/V

TYPE

Q2
2N2222

100k

+

A1

1/2 LT1078

0.02

200k

1

FB/SD

2

+

+

CAP

LT1054

3

GND

4

–

CAP

V

*1% FILM RESISTOR

OUT

5V

8

+

V

+

5

–

100μF
SOLID
TANTALUM

100k

100μF

+

8V

10k

1μF

Figure 10. High Resolution Version of Figure 9. Bipolar Voltage Converter Gives Greater Bridge Drive, Increasing Output Signal

results in lower transducer outputs for a given measurand
value, effectively magnifying amplifier offset drift terms.
The limit on available bridge drive is set by the CMOS
LTC1044’s output impedance. Figure 10’s circuit employs
a bipolar positive-to-negative converter which has much

lower output impedance. The biasing used permits 8V to
appear across the bridge, requiring the 100mA capability
LT1054 to sink about 24mA. This increased drive results
in a more favorable transducer gain slope, increasing
signal-to-noise ratio.

an43f

AN43-7

Application Note 43

Switched-Capacitor Based Instrumentation Amplifiers

Switched-capacitor methods are another way to signal
condition bridge outputs. Figure 11 uses a flying capacitor
configuration in a very high precision-scale application. This
design, intended for weighing human subjects, will resolve

0.01 pound at 300.00 pounds full scale. The strain gauge
based transducer platform is excited at 10V by the LT1021
reference, A1 and A2. The LTC1043 switched-capacitor
building block combines with A3, forming a differential
input chopper-stabilized amplifier. The LTC1043 alternately
connects the 1μF flying capacitor between the strain gauge
bridge output and A3’s input. A second 1μF unit stores
the LTC1043 output, maintaining A3’s input at DC. The
LTC1043’s low charge injection maintains differential to
single-ended transfer accuracy of about 1ppm at DC and
low frequency. The commutation rate, set by the 0.01μF
capacitor, is about 400Hz. A3 takes scaled gain, providing

3.0000V for 300.00 pounds full-scale output.

15V

15V

LT1021

10V

+

–

7

13

A1

LT1012

LTC1043

15V

4

11

1μF

12

16 17

0.01

14

8

A2

LT1010

301k

1% FILM

1μF

5.8k*

2.5k
ZERO

80k*

15V

+

A3

LTC1150

–

0.68μF

10k
1% FILM

50k
GAIN

0.68/2μF = POLYSTYRENE
* = ULTRONIX 105A RESISTOR
STRAIN BRIDGE PLATFORM = NCI 3224

15k

25Ω

135k* 1k

1k*

100k

The extremely high resolution of this scale requires filtering
to produce useful results. Very slight body movement acting
on the platform can cause significant noise in A3’s output.
This is dramatically apparent in Figure 12’s tracings. The
total force on the platform is equal to gravity pulling on
the body (the “weight”) plus any additional accelerations
within or acting upon the body. Figure 12 (Trace B) clearly
shows that each time the heart pumps, the acceleration due
to the blood (mass) moving in the arteries shows up as
“weight”. To prove this, the subject gets off the scale and
runs in place for 15 seconds. When the subject returns to
the platform the heart should work harder. Trace A confirms
this nicely. The exercise causes the heart to work harder,
forcing a greater acceleration-per-stroke.

Note 2: Cardiology aficionados will recognize this as a form of
Ballistocardiograph (from the Greek “ballein”—to throw, hurl or eject
and “kardia,” heart). A significant amount of effort was expended in
attempts to reliably characterize heart conditions via acceleration detection
methods. These efforts were largely unsuccessful when compared against
the reliability of EKG produced data. See references for further discussion.

15V

–

A5A

1/2 LT1018

+

–

LT1012

+

15V

–

A5B

1/2 LT1018

+

RC FILTER

680k

39k 2μF

2k

2

10V RATIO
OUTPUT

HEARTBEAT
OUTPUT

A4

WEIGHT
OUTPUT
0V TO 3.0000V =
0LB TO 300.00LB

AN43 F11

AN43-8

= HEWLETT-PACKARD HSSR-8200

Figure 11. High Precision Scale for Human Subjects

an43f

A = 0.45LB/FULL SCALE

B = 0.45LB/FULL SCALE

Application Note 43

HORIZ = 1s/INCH

Figure 12. High Precision Scale’s Heartbeat Output. Trace B Shows Subject at Rest; Trace A After Exercise. Discontinuous Components
in Waveforms Leading Edges Are Due to XY Recorder Slew Limitations

Another source of noise is due to body motion. As the
body moves around, its mass doesn’t change but the
instantaneous accelerations are picked up by the platform
and read as “weight” shifts.

All this seems to make a 0.01 pound measurement mean­ingless. However, filtering the noise out gives a time aver­aged value. A simple RC lowpass will work, but requires
excessively long settling times to filter noise fundamentals
in the 1Hz region. Another approach is needed.

A4, A5 and associated components form a filter which
switches its time constant from short to long when the
output has nearly arrived at the final value. With no weight
on the platform A3’s output is zero. A4’s output is also
zero, A5B’s output is indeterminate and A5A’s output is
low. The MOSFET opto-couplers LED comes on, putting the
RC filter into short time constant mode. When someone
gets on the scale A3’s output rises rapidly. A5A goes high,
but A5B trips low, maintaining the RC filter in its short
time constant mode. The 2μF capacitor charges rapidly,

the 2μF capacitor, returning A4’s output rapidly to zero.
The bias string at A5A’s input maintains the scale in fast
time constant mode for weights below 0.50 pounds. This
permits rapid response when small objects (or persons)
are placed on the platform. To trim this circuit, adjust
the zero potentiometer for 0V out with no weight on the
platform. Next, set the gain adjustment for 3.0000V out
for a 300.00 pound platform weight. Repeat this procedure
until both points are fixed.

Optically Coupled Switched-Capacitor
Instrumentation Amplifier

Figure 13 also uses optical techniques for performance
enhancement. This switched-capacitor based instru­mentation amplifier is applicable to transducer signal
conditioning where high common mode voltages exist.
The circuit has the low offset and drift of the LTC1150
but also incorporates a novel switched-capacitor “front
end” to achieve some specifications not available in a
conventional instrumentation amplifier.

AN43 F12

and A4 quickly settles to final value ± body motion and
heartbeat noise. A5B’s negative input sees 1% attenuation
from A3; its positive input does not. This causes A5B to
switch high when A4’s output arrives within 1% of final
value. The opto-coupler goes off and the filter switches
into long time constant mode, eliminating noise in A4’s
output. The 39k resistor prevents overshoot, ensuring
monotonic A4 outputs. When the subject steps off the
scale A3 quickly returns to zero. A5A goes immediately

Common mode rejection ratio at DC for the front end
exceeds 160dB. The amplifier will operate over a ±200V
common mode range and gain accuracy and stability are
limited only by external resistors. A1, a chopper stabilized
unit, sets offset drift at 0.05μV/°C. The high common
mode voltage capability of the design allows it to with­stand transient and fault conditions often encountered in
industrial environments.

low, turning on the opto-coupler. This quickly discharges

an43f

AN43-9

Application Note 43

15V

ACQUIRE

0.05

10k

10k

0.05

15V

15V

Q

74C74

DCK

÷ 4

Q

2N3904

10k

10k

15V

+E BRIDGE

+

S1

C1

2k 2k

1μF

S2

–

= HEWLETT-PACKARD HSSR-8200

= 1/6 74C04

= 1/4 74C02

* = 1% FILM RESISTOR

Figure 13. Floating Input Bridge Instrumentation Amplifier with 200V Common Mode Range

The circuit’s inputs are fed to LED-driven optically-coupled
MOSFET switches, S1 and S2. Two similar switches, S3
and S4, are in series with S1 and S2. CMOS logic func­tions, clocked from A1’s internal oscillator, generate non­overlapping clock outputs which drive the switch’s LEDs.
When the “acquire pulse” is high, S1 and S2 are on and
C2 acquires the differential voltage at the bridge’s output.
During this interval, S3 and S4 are off. When the acquire
pulse falls, S1 and S2 begin to go off. After a delay to allow
S1 and S2 to fully open, the “read pulse” goes high, turn­ing on S3 and S4. Now C1 appears as a ground-referred
voltage source which is read by A1. C2 allows A1’s input
to retain C1’s value when the circuit returns to the acquire
mode. A1 provides the circuit’s output. Its gain is set in
normal fashion by feedback resistors. The 0.33μF feedback
capacitor sets roll-off. The differential-to-single-ended
transition performed by the switches and capacitors means
that A1 never sees the input’s common mode signal. The

READ

S3

15V

+

LTC1150

C2
1μF

S4

–

A1

–15V

100pF

CLK OUT

OUTPUT

100k*

0.33

100Ω*

AN43 F13

breakdown specification of the optically-driven MOSFET
switch allows the circuit to withstand and operate at com­mon mode levels of ±200V. In addition, the optical drive
to the MOSFETs eliminates the charge injection problems
common to FET switched-capacitive networks.

Platinum RTD Resistance Bridge Circuits

Platinum RTDs are frequently used in bridge configura­tions for temperature measurement. Figure 14’s circuit is
highly accurate and features a ground referred RTD. The
ground connection is highly desirable for noise rejection.
The bridges RTD leg is driven by a current source while
the opposing bridge branch is voltage biased. The current
drive allows the voltage across the RTD to vary directly with
its temperature induced resistance shift. The difference
between this potential and that of the opposing bridge leg
forms the bridges output.

an43f

AN43-10

15V

27k

10k*

LT1009

2.5V

2k

* = 1% FILM RESISTOR

= ROSEMOUNT 118MFRTD

R

P

15V

+

A1A

1/2 LT1078

–

0.1μF

LT1101

A = 10

Application Note 43

274k*

15V

88.7Ω*

R

P

100Ω AT
0°C RTD

50k

ZERO

8.25k*

–

LT1101

A = 10

+

A3

LINEARITY

+

A2

–

250k*

+

5k

A1B

1/2 LT1078

–

0V TO 10V
0°C TO 400°C ±0.05°C

2k
GAIN

13k*

10k*

AN43 F14

OUT

=

Figure 14. Linearized Platinum RTD Bridge. Feedback to Bridge from A3 Linearizes the Circuit

A1A and instrumentation amplifier A2 form a voltage-con­trolled current source. A1A, biased by the LT1009 refer­ence, drives current through the 88.7Ω resistor and the
RTD. A2, sensing differentially across the 88.7Ω resistor,
closes a loop back to A1A. the 2k-0.1μF combination sets
amplifier roll-off, and the configuration is stable. Because
A1A’s loop forces a fixed voltage across the 88.7Ω resistor,
the current through R

is constant. A1’s operating point is

P

primarily fixed by the 2.5V LT1009 voltage reference.
The RTD’s constant current forces the voltage across it

to vary with its resistance, which has a nearly linear posi­tive temperature coefficient. The nonlinearity could cause
several degrees of error over the circuit’s 0°C to 400°C
operating range. The bridges output is fed to instrumenta­tion amplifier A3, which provides differential gain while
simultaneously supplying nonlinearity correction. The
correction is implemented by feeding a portion of A3’s
output back to A1’s input via the 10k-250k divider. This
causes the current supplied to R

to slightly shift with

P

its operating point, compensating sensor nonlinearity to

within ±0.05°C. A1B, providing additional scaled gain,
furnishes the circuit output.

To calibrate this circuit, substitute a precision decade
box (e.g., General Radio 1432k) for R

. Set the box to

P

the 0°C value (100.00Ω) and adjust the offset trim for a

0.00V output. Next, set the decade box for a 140°C output
(154.26Ω) and adjust the gain trim for a 3.500V output
reading. Finally, set the box to 249.0Ω (400.00°C) and trim
the linearity adjustment for a 10.000V output. Repeat this
sequence until all three points are fixed. Total error over
the entire range will be within ±0.05°C. The resistance
values given are for a nominal 100.00Ω (0°C) sensor.
Sensors deviating from this nominal value can be used
by factoring in the deviation from 100.00Ω. This devia­tion, which is manufacturer specified for each individual
sensor, is an offset term due to winding tolerances during
fabrication of the RTD. The gain slope of the platinum is
primarily fixed by the purity of the material and has a very
small error term.

an43f

AN43-11

Application Note 43

Figure 15 is functionally identical to Figure 14, except that
A2 and A3 are replaced with an LTC1043 switched-capacitor
building block. The LTC1043 performs the differential­to-single-ended transitions in the current source and
bridge output amplifier. Value shifts in the current source
and output stage reflect the LTC1043’s lack of gain. The
primary trade-off between the two circuits is component
count versus cost.

Digitally Corrected Platinum Resistance Bridge

The previous examples rely on analog techniques to
achieve a precise, linear output from the platinum RTD
bridge. Figure 16 uses digital corrections to obtain simi­lar results. A processor is used to correct residual RTD

27k

15V

10k*

LT1009

2.5V

15V

+

1/2 LT1078

–

0.1μF

nonlinearities. The bridges inherent nonlinear output is
also accommodated by the processor.

The LT1027 drives the bridge with 5V. The bridge differential
output is extracted by instrumentation amplifier A1. A1’s
output, via gain scaling stage A2, is fed to the LTC1290
12-bit A/D. The LTC1290’s raw output codes reflect the
bridges nonlinear output versus temperature. The pro­cessor corrects the A/D output and presents linearized,
calibrated data out. RTD and resistor tolerances mandate
zero and full-scale trims, but no linearity correction is
necessary. A2’s analog output is available for feedback
control applications. The complete software code for the
68HC05 processor, developed by Guy M. Hoover, appears
in Figure 17.

250k*

15V

2k

1μF

15V

4

7

13

0.01μF

8

11

1μF

12

14

1716

887Ω*

R

P

100Ω
AT 0°C

274k*

50k

ZERO

8.25k*

4

1/2 LTC1043

5

2

3

15

* = 1% FILM RESISTOR
R

6

1μF

18

= ROSEMOUNT 118MFRTD

P

5

1μF

6

+

1/2 LT1078

–

GAIN

ADJUST

0°C TO 400°C ±0.05°C

7

2k

LINEARITY

13k*

619Ω*

AN43 F15

0V TO 10V

5k

OUT

=

Figure 15. Switched-Capacitor-Based Version of Figure 14

AN43-12

an43f

Application Note 43

Thermistor Bridge

Figure 18, another temperature measuring bridge, uses
a thermistor as a sensor. The LT1034 furnishes bridge
excitation. The 3.2k and 6250Ω resistors are supplied
with the thermistor sensor. The networks overall response
is linearly related to the thermistor’s sensed temperature.
The network forms one leg of a bridge with resistors fur­nishing the opposing leg. A trim in this opposing leg sets
bridge output to zero at 0°C. Instrumentation amplifier A1
takes gain with A2 providing additional trimmed gain to
furnish a calibrated output. Calibration is accomplished
in similar fashion to the platinum RTD circuits, with the
linearity trim deleted.

Low Power Bridge Circuits

Low power operation of bridge circuits is becoming increas­ingly common. Many bridge-based transducers are low
impedance devices, complicating low power design. The
most obvious way to minimize bridge power consumption
is to restrict drive to the bridge. Figure 19a is identical to

Figure 5, except that the bridge excitation has been re­duced to 1.2V. This cuts bridge current from nearly 30mA
to about 3.5mA. The remaining circuit elements consume
negligible power compared to this amount. The trade-off
is the sacrifice in bridge output signal. The reduced drive
causes commensurately lowered bridge outputs, making
the noise and drift floor a greater percentage of the signal.
More specifically, a 0.01% reading of a 10V powered 350Ω
strain gauge bridge requires 3μV of stable resolution. At

1.2V drive, this number shrinks to a scary 360nV.
Figure 19b is similar, although bridge current is reduced

below 700μA. This is accomplished by using a semi­conductor-based bridge transducer. These devices have
significantly higher input resistance, minimizing power
dissipation. Semiconductor-based pressure transducers
have major cost advantages over bonded strain gauge
types, although accuracy and stability are reduced. Ap­pendix A, “Strain Gauge Bridges,” discusses trade-offs
and theory of both technologies.

5V

12k*

1k*

OUT

12.5k*

500k
ZERO°C
TRIM

–

+

R

PLAT

*TRW-IRC MAR-6 RESISTOR—0.1%
**1% FILM RESISTOR

= 1kΩ AT 0°C—ROSEMOUNT #118MF

R

PLAT

15V

A1

LT1101

A = 10

+

LT1006

–

+

10μF

15V

V

A2

30.1k**

1μF

3.92M**

AN43 F16

LTC1290

500k
400°C
TRIM

REF

+V

15V

SERIAL OUT TO
68HC05 PROCESSOR

LT102715V

Figure 16. Digitally Linearized Platinum RTD Signal Conditioner

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AN43-13

Application Note 43

* PLATINUM RTD LINEARIZATION PROGRAM (0.0 TO 400.0 DEGREES C)
* WRITTEN BY GUY HOOVER LINEAR TECHNOLOGY CORPORATION
* 3/14/90
* N IS THE NUMBER OF SEGMENTS THAT RTD RESPONSE IS DIVIDED INTO
* TEMPERATURE (DEG. C*10)=M*X+B
* M IS SLOPE OF RTD RESPONSE FOR A GIVEN SEGMENT
* X IS A/D OUTPUT MINUS SEGMENT END POINT
* B IS SEGMENT START POINT IN DEGREES C *10.
*
*****************************************************************************************
* LOOK UP TABLES
*
ORG $1000
* TABLE FOR SEGMENT END POINTS IN DECIMAL
* X IS FORMED BY SUBTRACTING PROPER SEGMENT END POINT FROM A/D OUTPUT
FDB 60,296,527,753,976,1195,1410,1621,1829,2032
FDB 2233,2430,2623,2813,3000,3184,3365,3543,3718,3890
ORG $1030
* TABLE FOR M IN DECIMAL
* M IS SLOPE OF RTD OVER A GIVEN TEMPERATURE RANGE
FDB 3486,3535,3585,3685,3735,3784,3884,3934,3984,4083
FDB 4133,4232,4282,4382,4432,4531,4581,4681,4730,4830
ORG $1060
* TABLE FOR B IN DECIMAL
* B IS DEGREES C TIMES TEN
FDB 0,200,400,600,800,1000,1200,1400,1600,1800
FDB 2000,2200,2400,2600,2800,3000,3200,3400,3600,3800
ORG $10FF
FCB 39 (N*2)-1 IN DECIMAL
*
* END LOOK UP TABLES
*****************************************************************************************
* BEGIN MAIN PROGRAM
*
ORG $0100
LDA #$F7 CONFIGURATION DATA FOR PORT C DDR
STA $06 LOAD CONFIGURATION DATA INTO PORT C
BSET 0,$02 INITIALIZE B0 PORT C
MES90L NOP
LDA #$2F DIN WORD FOR 1290 CH4 WITH RESPECT
* TO CH5, MSB FIRST, UNIPOLAR, 16 BITS
STA $50 STORE DIN WORD IN DIN BUFFER
JSR READ90 CALL READ90 SUBROUTINE (DUMMY READ)
JSR READ90 CALL READ90 SUBROUTINE (MSBS IN $61 LSBS IN $62)
LDX $10FF LOAD SEGMENT COUNTER INTO X \ FOR N=20 TO 1
DOAGAIN LDA $1000,X LOAD LSBS OF SEGMENT N \
STA $55 STORE LSBS IN $55 \
DECX DECREMENT X \
LDA $1000,X LOAD MSBS OF SEGMENT N \
STA $54 STORE MSBS IN $54 \ FIND B
JSR SUBTRCT CALL SUBTRCT SUBROUTINE /
BPL SEGMENT IF RESULT IS PLUS GOTO SEGMENT /
JSR ADDB CALL ADDB SUBROUTINE /
DECX DECREMENT X /
JMP DOAGAIN GOTO CODE AT LABEL DOAGAIN / NEXT N

AN43-14

Figure 17. Software Code for 68HC05 Processor-Based RTD Linearization

an43f

Application Note 43

*
*
*
*
*
SEGMENT LDA $1030,X LOAD MSBS OF SLOPE \
STA $54 STORE MSBS IN $54 \
INCX INCREMENT X \ M*X
LDA $1030,X LOAD LSBS OF SLOPE /
STA $55 STORE LSBS IN $55 /
JSR TBMULT CALL TBMULT SUBROUTINE /
LDA $1060,X LOAD LSBS OF BASE TEMP \
STA $55 STORE LSBS IN $55 \
DECX DECREMENT X > B ADDED TO M*X
LDA $1060,X LOAD MSBS OF BASE TEMP /
STA $54 STORE MSBS IN $54 /
JSR ADDB CALL ADDB SUBROUTINE
* TEMPERATURE IN DEGREES C * 10 IS IN $61 AND $62
* END MAIN PROGRAM
*****************************************************************************************
*
*
JMP MES90L RUN MAIN PROGRAM IN CONTINUOUS LOOP
*
*****************************************************************************************
* SUBROUTINES BEGIN HERE
*
*****************************************************************************************
* READ90 READS THE LTC1290 AND STORES THE RESULT IN $61 AND $62
*
READ90 LDA #$50 CONFIGURATION DATA FOR SPCR \
STA $0A LOAD CONFIGURATION DATA > CONFIGURE PROCESSOR
LDA $50 LOAD DIN WORD INTO THE ACC /
BCLR 0,$02 BIT 0 PORT C GOES LOW (CS GOES LOW) \
STA $0C LOAD DIN INTO SPI DATA REG. START TRANSFER. |
BACK90 TST $0B TEST STATUS OF SPIF |
BPL BACK90 LOOP TO PREVIOUS INSTRUCTION IF NOT DONE |
LDA $0C LOAD CONTENTS OF SPI DATA REG. INTO ACC |
STA $0C START NEXT CYCLE |
STA $61 STORE MSBS IN $61 | XFER
BACK92 TST $0B TEST STATUS OF SPIF | DATA
BPL BACK92 LOOP TO PREVIOUS INSTRUCTION IF NOT DONE |
BSET 0,$02 SET BIT 0 PORT C (CS GOES HIGH) |
LDA $0C LOAD CONTENTS OF SPI DATA REG INTO ACC |
STA $62 STORE LSBS IN $62 /
LDA #$04 LOAD COUNTER WITH NUMBER OF SHIFTS \
SHIFT CLC CLEAR CARRY \
ROR $61 ROTATE MSBS RIGHT THROUGH CARRY \ RIGHT
ROR $62 ROTATE LSBS RIGHT THROUGH CARRY / JUSTIFY
DECA DECREMENT COUNTER / DATA
BNE SHIFT IF NOT DONE SHIFTING THEN REPEAT LOOP /
RTS RETURN TO MAIN PROGRAM
*
* END READ90
*****************************************************************************************

Figure 17. Software Code for 68HC05 Processor-Based RTD Linearization (Continued)

an43f

AN43-15