Noty an126fa Linear Technology

Application Note 126

October 2010

2-Wire Virtual Remote Sensing for Voltage Regulators

Clairvoyance Marries Remote Sensing

Jim Williams, Jesus Rosales, Kurk Mathews, Tom Hack

Introduction

Wires and connectors have resistance. This simple, un­avoidable truth dictates that a power source’s remote load
voltage will be less than the source’s output voltage. Figure 1
shows this, and implies that intended load voltage can
be maintained by raising regulator output. Unfortunately,
line resistance and load variations introduce uncertainties,
limiting achievable performance.

WIRING DROPS

POWER
SUPPLY

WIRING DROPS

Figure 1. Unavoidable Wiring Drops Cause Low Load
Voltage. Line and Load Resistance Variations Introduce
Additional Load Voltage Uncertainty, Mitigating Against
Compensation by Raising Supply Voltage

POWER
SUPPLY

LOAD

VOLTAGE

REGULATOR

Figure 2. Local Regulation Stabilizes
Load Voltage But is Ineffi cient

V

SENSE

POWER

SUPPLY

SENSE

V

OUT

OUT

VOLTAGE DROP R

+

+

–

VOLTAGE DROP R

–

WIRE

WIRE

Figure 3. Classical “4-Wire” Remote Sensing. V
Voltage Drops Are Compensated by Regulator Sensing at Load.
High Impedance Sense Inputs Negate Sense Wire Resistance.
Approach Requires Four Wires

AN125 F01

AN125 F02

OUT

LOAD

LOAD

LOAD

AN125 F03

Line

Figure 2 illustrates one compensatory approach. Locally
positioned regulation stabilizes load voltage against line
drops but is ineffi cient due to regulator losses. Figure 3,
the classical approach, utilizes “4-wire” remote sensing to
eliminate line drop effects. The power supply sense inputs
are fed from load referred sense wires. The sense inputs
are high impedance, negating sense line resistance effects.
This scheme works well, but requires dedicated sense
wires, a signifi cant disadvantage in many applications.

“Virtual” Remote Sensing

Figure 4 retains the advantages of classical 4-wire re­mote sensing while eliminating the sense leads. Here,
the LT4180 Virtual Remote Sense™ (VRS) IC alternates
output current between 95% and 105% of the nominal
required output current. The LT4180 forces the power
supply to provide a DC current plus a small square wave
current with peak-to-peak amplitude equal to 10% of the
DC current. Decoupling capacitor C

, normally required

LOAD

for low impedance under transient conditions in non-VRS
systems, takes an additional role by fi ltering out the VRS
square wave excursions.

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and Virtual
Remote Sense is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.

SENSE

+

V

POWER SUPPLY

OUT

–

CONTROL PIN

LT4180

V

= DC + SQUAREWAVE FROM WIRING VOLTAGE DROP

OUT

REMOVES SQUAREWAVE, SO VL CONTAINS ONLY DC.

C

LOAD

= DC + SQUAREWAVE

I

L

Figure 4. LT4180 2-Wire Virtual Remote Sense Estimates
Wiring Voltage Drops, Compensates by Adjusting Supply
Output Voltage. Wiring Loss Is Determined by Measuring
Small Signal Square Wave Carrier Induced Voltage Drop.
Load Capacitor Absorbs Square Wave; Load Is at DC

R

R

WIRE

WIRE

I

L

/2I

+

V

C

L

/2

LOAD

LOAD

–

AN125 F04

an126fa

AN126-1

Application Note 126

Because C is sized to produce an “AC short” at the square
wave frequency, a square wave voltage is produced at the
power supply equal to V

= 0.1 • IDC • R

OUTAC

WIREVP-P

. The
square wave voltage at the power supply has a peak-to­peak amplitude equal to one tenth the DC wiring drop. This
is a direct measurement of wiring drop, not an estimate,
accurate over all load currents. Signal processing produces
a DC voltage from this AC signal which is introduced into
the supply feedback loop to provide accurate load regula-

1

. Note that the “power supply” may be an IC linear or

tion
switching regulator, a module or any other power source
capable of variable output. Power supplies can be syn­chronized to the LT4180 and VRS operating frequency is
adjustable over more than three decades. Optional spread
spectrum operation provides partial immunity from single­tone interference and a 3V to 50V input range simplifi es
design. Because this technique is based on an estimate
of load voltage, not a direct measurement, the resultant
correction is an approximation, but a very good one.

Typical LT4180 load regulation is plotted in Figure 5. In
this example, load current increases from zero until it
produces a 2.5V wiring drop. Load voltage drops only
73mV at maximum current. A voltage drop equivalent to
50% of load voltage results in only a 1.5% shift in load
voltage value. Smaller wiring drops produce even better
results.

Note 1. Readers fi nding their intellectual prowess unsatiated by this

admittedly cursory description will fi nd more studious coverage in
Appendix A, “A Primer on LT4180 VRS Operation.”

5.00

4.99

4.98

4.97

4.96

(V)

LOAD

4.95

V

4.94

4.93

4.92

4.91
0

0.5 1.51 2 2.5 3

V

(V)

WIRING

AN126 F05

Applications

The following applications are all VRS augmented voltage
regulators of various descriptions. The power regulation
stages employed are, with one exception, generic LTC
designs and are spared exhaustive commentary, permit­ting emphasis on the LT4180 VRS role. Additionally, the
similarity of the VRS associated circuitry across the broad
array of applications shown should be noted, and is indica­tive of the relative ease of implementation. Surprisingly
little change is needed to use the VRS in the different
situations presented.

VRS Linear Regulators

Figure 6 adds a simple stage to the LT4180 to implement
a complete VRS aided linear regulator. The LT4180 senses
current via the 0.2 shunt and feedback controls Q1 with
Q2, completing a control loop. Cascoded Q2 permits the
ICs 5V capable open drain output to control a high voltage
at Q1’s gate. Components at the compensation pin furnish

2

loop stability, promoting good transient response
shows Figure 6’s load step waveforms. They include V
(trace A), V

LOAD

(B) and I

(C). Transient response is

LOAD

. Figure 7

SENSE

determined by loop compensation, load capacitance and
remote sense sample rate. Figure 8 shows response with

increased to 1100µF. Load voltage transient excur-

C

LOAD

sion reduces and duration increases.

Figure 9, employing a monolithic regulator, adds current
limiting and simplifi es loop compensation. Transient re­sponse approximates Figure 6’s. As before, the LT4180’s
low voltage drain pin requires a cascode transistor to
control the high voltage at the LT3080 set pin.

Note 2. Value selection procedure for LT1480 VRS circuits is detailed in

Appendix B, “Design Guidelines for LT4180 VRS Circuits.”

Figure 5. Typical LT4180 Virtual Remote Sense Performance
Shows 1.6% Regulation vs 0V → 2.5V Wiring Drop

AN126-2

an126fa

Application Note 126

V
20V

Q1

10µF
25V

IRLZ44

27k

10k

GUARD PINS NOT SHOWN

200k

4.7µF

5.36k
1%

Q2

INTV

VN2222

63.4k
1%

3.74k
1%

2.2k
1%

CC

330pF

RUN

FB

OV

COMP GNDDRAIN

IN

0.2
1%

1µF

SENSE

V

IN

CHOLD1 CHOLD2 CHOLD3 CHOLD4

WIRING DROP

LOAD RETURN
WIRING DROP

LT4180

470pF47nF

DIV2

100µF

33nF

LOAD VOLTAGE
12V, 500mA
8Ω TOTAL WIRING DROP

LOAD
RETURN

V

DIV0DIV1

PP

C

OSC

470pF470pF

INTV

SPREAD

R

OSC

CC

41.2k
1%

AN126 F06

INTV

CC

1µF

Figure 6. Virtual Remote Sense Controls Discrete Linear Regulator. Q2 Cascodes Drain Output,
Buffering High Voltage Q1 Gate Drive. COMP Pin Associated Components Stabilize Loop

A = 2V/DIV

B = 2V/DIV

AC COUPLED

C = 0.2A/DIV

ON 0.2A

DC LEVEL

5ms/DIV

AN126 F07

Figure 7. Figure 6’s Load Step Waveforms with 100μF Load
Capacitor Include V

(Trace A), V

SENSE

LOAD

(B) and I

LOAD

(C).
Transient Response is Determined by Loop Compensation,
Load Capacitance and Remote Sense Sample Rate

A = 2V/DIV

B = 2V/DIV

AC COUPLED

C = 0.2A/DIV

ON 0.2A

DC LEVEL

5ms/DIV

Figure 8. Same Conditions as Figure 7 with C
1100μF. V

Transient Excursion Reduces, Duration Extends

LOAD

LOAD

AN126 F08

Increased to

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AN126-3

Application Note 126

V
18V

IN

LT3080

IN OUT

SET

10µF
25V

100k

GUARD PINS NOT SHOWN

10k

4.7µF

INTV

VN2222

51k

60.4k
1%

3.57k
1%

1.78k
1%

5.36k
1%

CC

1500pF

RUN

FB

OV

COMP GNDDRAIN

0.2
1%

1µF

SENSE

V

IN

CHOLD1 CHOLD2 CHOLD3 CHOLD4

WIRING DROP

LOAD RETURN
WIRING DROP

DIV2

LT4180

470pF47nF

470µF

47nF

DIV0DIV1

LOAD VOLTAGE
12V, 500mA
4Ω TOTAL
WIRING DROP

LOAD
RETURN

V

PP

C

OSC

330pF470pF

INTV

SPREAD

R

OSC

CC

22.1k
1%

AN126 F06

INTV

CC

1µF

Figure 9. Figure 6’s Approach Utilizing IC Regulator Adds Current Limiting,
Simplifi es Loop Compensation. Transient Response Approximates Figure 6’s

VRS Equipped Switching Regulators

VRS based switching regulators are readily constructed.
Figure 10’s fl yback voltage boost confi guration has similar
architecture to the linear examples although output voltage
is above the input. In this case, the LT4180 open drain output
is directly compatible with the LT3581 boost regulator low
voltage V

pin––no cascode stage is necessary.

C

Step down (“Buck”) VRS equipped switching regulators are
similarly easily achieved. Figure 11’s scheme, reminiscent
of the previously described linear regulators, substitutes
an LT3685 step down regulator which is directly controlled
from the LT4180 open drain output. A single pole roll-off
stabilizes the loop and a 12V, 1.5A output is maintained
from a 22V to 36V input despite a 0Ω to 2.5Ω wiring drop
loss. Figure 11A is similar, except that it provides a 5V, 3A
output from a 12V to 36V input.

VRS Based Isolated Switching Supplies

The VRS approach is adaptable to isolated output supplies.
Figure 12’s 24V output converter utilizes an approach
similar to the previous examples except that it supplies
a fully isolated output. The virtual remote sense feature
accommodates a 10Ω wire resistance. The LT3825 and T1
form a transformer coupled power stage. Opto-coupled
feedback maintains output isolation.

Figure 13’s 48V → 3.3V, 3A design also has a fully isolated
output, facilitated by power delivery through a transformer
and optically coupled feedback loop closure. The LT3758
drives T1 via Q1. T1’s rectifi ed and fi ltered secondary
supplies output power which is corrected for line drops
by the LT4180. Isolation is maintained by transmitting the
feedback signal with an opto-isolator. The opto-isolators
output collector ties back to the LT3578 V

pin, closing

C

the control loop.

AN126-4

an126fa

OV

FB

DIV0DIV1

V

IN

INTV

CC

INTV

CC

V

PP

COMP GND

DRAIN

DIV2

CHOLD1 CHOLD2 CHOLD3 CHOLD4

47nF

AN126 F10

RUN

R

OSC

C

OSC

40.2Ω

1%

SENSE

SPREAD

LT4180

470pF470pF

470pF47nF

10nF

1µF

41.7k

1%

73.2Ω

1%

1.24k

1%

24.3k

1µF

0.2

1%

WIRING DROP

VIN5V

LOAD VOLTAGE

12V, 500mA

(100mA MIN.)

6Ω TOTAL WIRING DROP

LOAD

RETURN

10µF

25V

4.7µF

16V

L1

4.7µH

DFLS220

LOAD RETURN

WIRING DROP

100µF

191k

100k

10k

107Ω

1%

47pF

15k

LT3581

SW2SW2SW2SW1SW1SW1GATE

RT SSSYNC GND

V

CC

SHDN

FAULT

FB

VC

84.5k

0.1µF

L1 = VISHAY IHLPI525CZ-11

GUARD PINS NOT SHOWN

+

–

Application Note 126

Pin

C

Figure 10. Virtual Remote Sensed Voltage Boost Confi guration.

LT4180 Drain Output Controls Flyback Regulator via LT3581 V

an126fa

AN126-5

Application Note 126

12V, 1.5A

2.5Ω TOTAL

WIRING DROP

CC

INTV

LOAD

RETURN

1µF

AN126 F11

22.1k

R

C

1%

OSC

330pF470pF

OSC

CC

INTV

SPREAD

PP

V

470µF

WIRING DROP

LOAD RETURN

0.067 WIRING DROP

61.9k

1%

DIV0DIV1

DIV2

LT4180

SENSE

IN

V

1µF

RUN

22µF

2k

25V

1%

OV

DRAIN

5.36k

1%

3.65k

FB

1%

CHOLD1 CHOLD2 CHOLD3 CHOLD4

COMP GND

CC

INTV

47pF

47nF

470pF47nF

3.3nF

28k

Step-Down

IN

to 36V

IN

AN126-6

1µF

22µF

+

IN

V

22V TO 36V

50V

50V

BD

IN

V

100k

INTV

CC

0.47µF

BOOST

RUN/SD

0.1µF

LT3685

50V

30.1k

10µH

SW

FBRTSYNC

DFLS240

VC

GND

10k

68.1k

1%

CMDSH-3

1k

Regulator Maintains 12V Output Despite Wiring Losses

Figure 11. Remote Sense Corrected 22V

L1 - VISHAY IHLP2020CZ-11

GUARD PINS NOT SHOWN

an126fa

OUT

5V, 3A

0.4Ω TOTAL

WIRING DROP

V

C2

470µF

10V

+

C1

470µF

10V

+

–

WIRING DROP

1%

0.033Ω

CC

LOAD

RETURN

INTV

WIRING DROP

LOAD RETURN

Application Note 126

1µF

AN126 F11A

22.1k

CC

INTV

SPREAD

PP

V

DIV0

DIV1

DIV2

LT4180

SENSE

IN

V

1µF

OSC

R

OSC

C

CHOLD1 CHOLD2 CHOLD3 CHOLD4

1%

330pF

47nF

470pF

470pF47nF

4.7µF

50V

100k

BD

V

IN

21.5k

1%

BOOST

RUN/SD

1.87k

6.8µH

0.47µF

LT3693EDD

RUN

FB

2.15k

1%

47µF

10V

47µF

10V

MBRA340T3G

VC

SW

GND

FBRTSYNC

1%

OV

DRAIN

5.36k

1%

COMP GND

4.7nF

Step-Down Remote

OUT

47pF

CMDSH-3

CC

INTV

23.2k

1k

GUARD PINS NOT SHOWN

C1 = C2 = AVXTPSE477M010R0050

to 5V

IN

→ 36V

IN

Figure 11A. 12V

Sensed Regulator Has Similar Architecture to Figure 11

+

IN

V

8V TO 36V

22µF

50V

68.1k

1%

10k

1%

1%

an126fa

0.1µF

30.1k

CC

INTV

AN126-7