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LM6211
Low Noise, RRO Operational Amplifier with CMOS Input
and 24V Operation
LM6211 Low Noise, RRO Op-Amp with CMOS Input and 24V Operation
June 2006
General Description
The LM6211 is a wide bandwidth, low noise op amp with a
wide supply voltage range and a low input bias current. The
LM6211 operates with a single supply voltage of 5V to 24V,
is unity gain stable, has a ground-sensing CMOS input
stage, and offers rail-to-rail output swing.
The LM6211 is designed to provide optimal performance in
high voltage, low noise systems. The LM6211 has a unity
gain bandwidth of 20 MHz and an input referred voltage
noise density of 5.5 nV/
achieves these specifications with a low supply current of
only 1 mA. The LM6211 has a low input bias current of
2.3 pA, an output short circuit current of 25 mA and a slew
rate of 5.6 V/us. The LM6211 also features a low commonmode input capacitance of 5.5 pF which makes it ideal for
use in wide bandwidth and high gain circuits. The LM6211 is
well suited for low noise applications that require an op amp
with very low input bias currents and a large output voltage
swing, like active loop-filters for wide-band PLLs. A low total
harmonic distortion, 0.01% at 1 kHz with loads as high as
600Ω, also makes the LM6211 ideal for high fidelity audio
and microphone amplifiers.
The LM6211 is available in the small SOT package, allowing
the user to implement ultra-small and cost effective board
layouts.
at 10 kHz. The LM6211
Typical Application
Features
(Typical 24V supply unless otherwise noted)
n Supply voltage range 5V to 24V
n Input referred voltage noise 5.5 nV/
n Unity gain bandwidth 20 MHz
n 1/f corner frequency 400 Hz
n Slew rate 5.6 V/µs
n Supply current 1.05 mA
n Low input capacitance 5.5 pF
n Temperature range -40˚C to 125˚C
n Total harmonic distortion 0.01%
n Output short circuit current 25 mA
@
1 kHz, 600Ω
Applications
n PLL loop filters
n Low noise active filters
n Strain gauge amplifiers
n Low noise microphone amplifiers
20120303
20120304
© 2006 National Semiconductor Corporation DS201203 www.national.com
Absolute Maximum Ratings (Note 1)
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If Military/Aerospace specified devices are required,
LM6211
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Junction Temperature (Note 3) +150˚C
Soldering Information
Infrared or Convection (20 sec) 235˚C
Wave Soldering Lead Temp. (10 sec) 260˚C
ESD Tolerance (Note 2)
Human Body Model 2000V
Operating Ratings (Note 1)
Machine Model 200V
Differential
V
IN
Supply Voltage (V
=V+–V−) 25V
S
Voltage at Input/Output pins V
+
+0.3V, V−−0.3V
±
0.3V
Storage Temperature Range −65˚C to +150˚C
Temperature Range −40˚C to +125˚C
Supply Voltage (V
Package Thermal Resistance (θ
=V+–V−) 5Vto24V
S
(Note 3))
JA
5-Pin SOT23 178˚C/W
5V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA= 25˚C, V+= 5V, V−= 0V, VCM=VO=V+/2. Boldface limits apply
at the temperature extremes.
Symbol Parameter Conditions Min
(Note 6)
V
OS
TC V
I
B
I
OS
CMRR Common Mode Rejection
PSRR Power Supply Rejection Ratio V
Input Offset Voltage VCM= 0.5V 0.1
Input Offset Average Drift VCM= 0.5V (Note 7) 2 µV/C
OS
Input Bias Current VCM= 0.5V (Notes 8, 9) 0.5 5
Input Offset Current VCM= 0.5V 0.1 pA
Ratio
0V≤ V
0.4 V ≤ V
+
≤ 3V
CM
≤ 2.3 V
CM
= 5V to 24V, VCM= 0.5V 85
83
70
78
+
= 4.5V to 25V, VCM= 0.5V 80 95
V
CMVR Input Common-Mode Voltage
Range
A
VOL
Large Signal Voltage Gain VO= 0.35V to 4.65, RL=2kΩ to V+/2 82
CMRR ≥ 65 dB
CMRR ≥ 60 dB
0
0
80
= 0.25V to 4.75, RL=10kΩ to V+/2 85
V
O
82
V
O
Output Swing High RL=2kΩ to V+/2 50 150
=10kΩ to V+/2 20 85
R
L
Output Swing Low R
=2kΩ to V+/2 39 150
L
RL=10kΩ to V+/2 13 85
I
I
OUT
S
Output Short Circuit Current Sourcing to V+/2
Supply Current 0.96 1.10
SR Slew Rate A
= 100 mV (Note 10)
V
ID
Sinking to V
= −100 mV (Note 10)
V
ID
= +1, 10% to 90% (Note 11) 5.5 V/µs
V
+
/2
13
10
20
10
GBW Gain Bandwidth Product 17 MHz
e
n
Input-Referred Voltage Noise f = 10 kHz 5.5
f = 1 kHz 6.0
i
n
Input-Referred Current Noise f = 1 kHz 0.01 pA/
Typ
(Note 5)
98
98
110
110
16
30
Max
(Note 6)
±
±
165
170
1.25
2.5
2.8
10
3.3
2.4
90
90
Units
mV
pA
nA
dB
dB
V
dB
mV from
rail
mA
mA
nV/
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5V Electrical Characteristics (Note 4) (Continued)
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Unless otherwise specified, all limits are guaranteed for TA= 25˚C, V+= 5V, V−= 0V, VCM=VO=V+/2. Boldface limits apply
at the temperature extremes.
Symbol Parameter Conditions Min
(Note 6)
THD Total Harmonic Distortion A
=2,RL= 600Ω to V+/2 0.01 %
V
Typ
(Note 5)
Max
(Note 6)
Units
24V Electrical Characteristics (Note 4)
Unless otherwise specified, all limits are guaranteed for TA= 25˚C, V+= 24V, V−= 0V, VCM=VO=V+/2. Boldface limits apply
at the temperature extremes.
Symbol Parameter Conditions Min
(Note 6)
V
OS
TC V
I
B
I
OS
CMRR Common Mode Rejection
PSRR Power Supply Rejection Ratio V
Input Offset Voltage VCM= 0.5V 0.25
Input Offset Average Drift VCM= 0.5V (Note 7)
OS
Input Bias Current VCM= 0.5V (Notes 8, 9) 2 25
Input Offset Current VCM= 0.5V 0.1 pA
85
70
Ratio
0 ≤ V
0.4 ≤ V
≤ 21V
CM
≤ 20V
CM
+
= 5V to 24V, VCM= 0.5V 85
78
V+= 4.5V to 25V, VCM= 0.5V 80 98
CMVR Input Common-Mode Voltage
Range
A
VOL
Large Signal Voltage Gain VO= 1.5V to 22.5V, RL=2kΩ to V+/2 82
CMRR ≥ 65 dB
CMRR ≥ 60 dB
0
0
77
VO= 1V to 23V, RL=10kΩ to V+/2 85
82
V
O
Output Swing High RL=2kΩ to V+/2 212 400
RL=10kΩ to V+/2 48 150
Output Swing Low R
=2kΩ to V+/2 150 350
L
RL=10kΩ to V+/2 38 150
I
I
OUT
S
Output Short Circuit Current Sourcing to V+/2
= 100 mV (Note 10)
V
ID
Sinking to V
= −100 mV (Note 10)
V
ID
+
/2
Supply Current 1.05 1.25
SR Slew Rate AV= +1, VO=18V
PP
20
15
30
20
10% to 90% (Note 11)
GBW Gain Bandwidth Product 20 MHz
e
n
Input-Referred Voltage Noise f = 10 kHz 5.5
f = 1 kHz 6.0
i
n
Input-Referred Current Noise f = 1 kHz 0.01 pA/
THD Total Harmonic Distortion AV=2,RL=2kΩ to V+/2 0.01 %
Typ
(Note 5)
±
2 µV/C
Max
(Note 6)
±
2.7
±
3.0
10
105
98
21.5
20.5
120
120
520
165
420
170
25
38
1.40
5.6 V/µs
Units
mV
pA
nA
dB
dB
V
dB
mV from
rail
mA
mA
nV/
LM6211
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Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
查询"LM6211"供应商
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics Tables.
Note 2: Human Body Model is 1.5 kΩ in series with 100 pF. Machine Model is 0Ω in series with 200 pF.
LM6211
Note 3: The maximum power dissipation is a function of T
P
=(T
D
J(MAX)-TA
Note 4: Electrical table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating of
the device.
Note 5: Typical values represent the most likely parametric norm at the time of characterization.
Note 6: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlations using the Statistical Quality
Control (SQC) method.
Note 7: Offset voltage average drift is determined by dividing the change in V
Note 8: Positive current corresponds to current flowing into the device.
Note 9: Input bias current is guaranteed by design.
Note 10: The device is short circuit protected and can source or sink its limit currents continuously. However, care should be taken such that when the output is
driving short circuit currents, the inputs do not see more than
Note 11: Slew rate is the average of the rising and falling slew rates.
)/θJA. All numbers apply for packages soldered directly onto a PC board.
Connection Diagram
, θJA, and TA. The maximum allowable power dissipation at any ambient temperature is
J(MAX)
at the temperature extremes into the total temperature change.
OS
±
0.3V differential voltage.
5-Pin SOT23
Top View
20120301
Ordering Information
Package Part Number Package Marking Transport Media NSC Drawing
5-Pin SOT-23
LM6211MF
LM6211MFX 3k Units Tape and Reel
AT1A
1k Units Tape and Reel
MF05A
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LM6211
Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2.
V
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Supply Current vs. Supply Voltage V
20120318
VOSvs. V
CM
vs. Supply Voltage
OS
VOSvs. V
= 25˚C, VS= 24V, V+=VS,
A
20120319
CM
Input Bias Current vs. V
CM
20120320
20120350
Input Bias Current vs. V
20120321
CM
20120351
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Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2. (Continued)
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V
LM6211
= 25˚C, VS= 24V, V+=VS,
A
Input Bias Current vs. V
CM
20120353
Input Bias Current vs. V
CM
Sourcing Current vs. Supply Voltage Sinking Current vs. Supply Voltage
20120352
20120334
Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage
20120330
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20120333
20120332
LM6211
Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2. (Continued)
V
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Positive Output Swing vs. Supply Voltage Negative Output Swing vs. Supply Voltage
20120329
Sourcing Current vs. Output Voltage Sinking Current vs. Output Voltage
= 25˚C, VS= 24V, V+=VS,
A
20120331
20120328 20120327
Sourcing Current vs. Output Voltage Sinking Current vs. Output Voltage
20120326
20120325
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Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2. (Continued)
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V
LM6211
Open Loop Gain and Phase with Resistive Load Open Loop Gain and Phase with Capacitive Load
Input Referred Voltage Noise vs. Frequency THD+N vs. Frequency
= 25˚C, VS= 24V, V+=VS,
A
20120309 20120308
20120304
THD+N vs. Output Amplitude THD+N vs. Output Amplitude
20120316
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20120317
20120315
LM6211
Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2. (Continued)
V
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Slew Rate vs. Supply Voltage Overshoot and Undershoot vs. Capacitive Load
20120313 20120314
Small Signal Transient Response Large Signal Transient Response
= 25˚C, VS= 24V, V+=VS,
A
20120322
20120324
Phase Margin vs. Capacitive Load (Stability) Phase Margin vs. Capacitive Load (Stability)
20120310 20120311
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Typical Performance Characteristics Unless otherwise specified, T
−
=0V,VCM=VS/2. (Continued)
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V
LM6211
Closed Loop Output Impedance vs. Frequency PSRR vs. Frequency
CMRR vs. Frequency
= 25˚C, VS= 24V, V+=VS,
A
20120305 20120312
20120306
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Application Notes
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ADVANTAGES OF THE LM6211
High Supply Voltage, Low Power Operation
The LM6211 has performance guaranteed at supply voltages of 5V and 24V. The LM6211 is guaranteed to be
operational at all supply voltages between 5V and 24V. In
this large range of operation, the LM6211 draws a fairly
constant supply current of 1 mA, while providing a wide
bandwidth of 20 MHz. The wide operating range makes the
LM6211 a versatile choice for a variety of applications ranging from portable instrumentation to industrial control systems.
Low Input Referred Noise
The LM6211 has very low flatband input referred voltage
noise, 5.5 nV/
is about 400 Hz. The CMOS input stage allows for an
extremely low input current (2 pA) and a very low input
referred current noise (0.01 pA/
LM6211 to maintain signal fidelity and makes it ideal for
audio, wireless or sensor based applications.
Low Input Bias Current and High Input Impedance
The LM6211 has a CMOS input stage, which allows it to
have very high input impedance, very small input bias currents (2 pA) and extremely low input referred current noise
(0.01 pA/
amps used in sensor applications, which deal with extremely
low currents of the order of a few nanoamperes. In this case,
the op amp is being driven by a sensor, which typically has a
source impedance of tens of MΩ. This makes it essential for
the op amp to have a much higher impedance.
. The 1/f corner frequency, also very low,
). This allows the
). This level of performance is essential for op
should be taken to prevent the inputs from seeing more than
±
0.3V differential voltage, which is the absolute maximum
differential input voltage.
Small Size
The small footprint of the LM6211 package saves space on
printed circuit boards, and enables the design of smaller and
more compact electronic products. Long traces between the
signal source and the op amp make the signal path susceptible to noise. By using a physically smaller package, the
LM6211 can be placed closer to the signal source, reducing
noise pickup and enhancing signal integrity
STABILITY OF OP AMP CIRCUITS
Stability and Capacitive Loading
The LM6211 is designed to be unity gain stable for moderate
capacitive loads, around 100 pF. That is, if connected in a
unity gain buffer configuration, the LM6211 will resist oscillation unless the capacitive load is higher than about 100 pF.
For higher capacitive loads, the phase margin of the op amp
reduces significantly and it tends to oscillate. This is because
an op amp cannot be designed to be stable for high capacitive loads without either sacrificing bandwidth or supplying
higher current. Hence, for driving higher capacitive loads,
the LM6211 needs to be externally compensated.
LM6211
Low Input Capacitance
The LM6211 has a comparatively small input capacitance for
a high voltage CMOS design. Low input capacitance is very
beneficial in terms of driving large feedback resistors, required for higher closed loop gain. Usually, high voltage
CMOS input stages have a large input capacitance, which
when used in a typical gain configuration, interacts with the
feedback resistance to create an extra pole. The extra pole
causes gain-peaking and can compromise the stability of the
op amp. The LM6211 can, however, be used with larger
resistors due to its smaller input capacitance, and hence
provide more gain without compromising stability. This also
makes the LM6211 ideal for wideband transimpedance amplifiers, which require a wide bandwidth, low input referred
noise and low input capacitance.
RRO, Ground Sensing and Current Limiting
The LM6211 has a rail-to-rail output stage, which provides
the maximum possible output dynamic range. This is especially important for applications requiring a large output
swing, like wideband PLL synthesizers which need an active
loop filter to drive a wide frequency range VCO. The input
common mode range includes the negative supply rail which
allows direct sensing at ground in a single supply operation.
The LM6211 also has a short circuit protection circuit which
limits the output current to about 25 mA sourcing and 38 mA
sinking, and allows the LM6211 to drive short circuit loads
indefinitely. However, while driving short circuit loads care
20120337
FIGURE 1. Gain vs. Frequency for an Op Amp
An op amp, ideally, has a dominant pole close to DC, which
causes its gain to decay at the rate of 20 dB/decade with
respect to frequency. If this rate of decay, also known as the
rate of closure (ROC), remains at 20 dB/decade at the unity
gain bandwidth of the op amp, the op amp is stable. If,
however, a large capacitance is added to the output of the op
amp, it combines with the output impedance of the op amp to
create another pole in its frequency response before its unity
gain frequency (Figure 1). This increases the ROC to
40 dB/decade and causes instability.
In such a case a number of techniques can be used to
restore stability to the circuit. The idea behind all these
schemes is to modify the frequency response such that it
can be restored to a ROC of 20 dB/decade, which ensures
stability.
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Application Notes (Continued)
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LM6211
In the Loop Compensation
Figure 2 illustrates a compensation technique, known as ‘in
the loop’ compensation, that employs an RC feedback circuit
within the feedback loop to stabilize a non-inverting amplifier
configuration. A small series resistance, R
late the amplifier output from the load capacitance, C
small capacitance, C
sistor to bypass C
, is used to iso-
S
, is inserted across the feedback re-
F
at higher frequencies.
L
, and a
L
Compensation by External Resistor
In some applications it is essential to drive a capacitive load
without sacrificing bandwidth. In such a case, in the loop
compensation is not viable. A simpler scheme for compensation is shown in Figure 3. A resistor, R
, is placed in
ISO
series between the load capacitance and the output. This
introduces a zero in the circuit transfer function, which counteracts the effect of the pole formed by the load capacitance,
and ensures stability.
20120356
FIGURE 3. Compensation By Isolation Resistor
20120338
FIGURE 2. In the Loop Compensation
The values for R
zero attributed to C
attributed to C
and CFare decided by ensuring that the
S
lies at the same frequency as the pole
F
. This ensures that the effect of the second
L
pole on the transfer function is compensated for by the
presence of the zero, and that the ROC is maintained at
20 dB/decade. For the circuit shown in Figure 2 the values of
and CFare given by Equation (1). Table 1 shows different
R
S
values of R
stability with different values of C
margins to be expected. R
kΩ,R
L
and CFthat need to be used for maintaining
S
F
is taken as 2 kΩ, while R
, as well as the phase
L
and RINare assumed to be 10
is taken to be 60Ω.
OUT
(1)
TABLE 1.
CL(pF) RS(Ω)CF(pF) Phase Margin (˚)
250 60 4.5 39.8
300 60 5.4 49.5
500 60 9 53.1
Although this methodology provides circuit stability for any
load capacitance, it does so at the price of bandwidth. The
closed loop bandwidth of the circuit is now limited by R
.
C
F
and
S
The value of R
on the size of C
to be used should be decided depending
ISO
and the level of performance desired.
L
Values ranging from 5Ω to 50Ω are usually sufficient to
ensure stability. A larger value of R
will result in a system
ISO
with lesser ringing and overshoot, but will also limit the
output swing and the short circuit current of the circuit.
Stability and Input Capacitance
In certain applications, for example I-V conversion, transimpedance photodiode amplification and buffering the output of
current-output DAC, capacitive loading at the input of the op
amp can endanger stability. The capacitance of the source
driving the op amp, the op amp input capacitance and the
parasitic/wiring capacitance contribute to the loading of the
input. This capacitance, C
, interacts with the feedback
IN
network to introduce a peaking in the closed loop gain of the
circuit, and hence causes instability.
20120349
FIGURE 4. Compensating for Input Capacitance
This peaking can be eliminated by adding a feedback capacitance, C
, as shown in Figure 4. This introduces a zero
F
in the feedback network, and hence a pole in the closed loop
response, and thus maintains stability. An optimal value of
is given by Equation (2). A simpler approach is to select
C
F
=(R1/R2)CINfor a 90˚ phase margin. This approach,
C
F
however, limits the bandwidth excessively.
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Typical Applications
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ACTIVE LOOP FILTER FOR PLLs
A typical phase locked loop, or PLL, functions by creating a
negative feedback loop in terms of the phase of a signal. A
simple PLL consists of three main components: a phase
detector, a loop filter and a voltage controlled oscillator
(VCO). The phase detector compares the phase of the output of the PLL with that of a reference signal, and feeds the
error signal into the loop filter, thus performing negative
feedback. The loop filter performs the important function of
averaging (or low-pass filtering) the error and providing the
VCO with a DC voltage, which allows the VCO to modify its
frequency such that the error is minimized. The performance
of the loop filter affects a number of specifications of the PLL,
like its frequency range, locking time and phase noise.
Since a loop filter is a very noise sensitive application, it is
usually suggested that only passive components be used in
its design. Any active devices, like discrete transistors or op
amps, would add significantly to the noise of the circuit and
would hence worsen the in-band phase noise of the PLL. But
newer and faster PLLs, like National’s LMX2430, have a
power supply voltage of less than 3V, which limits the phasedetector output of the PLL. If a passive loop filter is used with
such circuits, then the DC voltage that can be provided to the
VCO is limited to couple of volts. This limits the range of
frequencies for which the VCO, and hence the PLL, is functional. In certain applications requiring a wider operating
range of frequencies for the PLL, like set-top boxes or base
stations, this level of performance is not adequate and requires active amplification, hence the need for active loop
filters.
An active loop filter typically consists of an op amp, which
provides the gain, accompanied by a three or four pole RC
filter. The non-inverting input of the op amp is biased to a
fixed value, usually the mid-supply of the PLL, while a feedback network provides the gain as well as one, or two, poles
for low pass filtering. Figure 5 illustrates a typical active loop
filter.
Certain performance characteristics are essential for an op
amp if it is to be used in a PLL loop filter. Low input referred
voltage and current noise are essential, as they directly
affect the noise of the filter and hence the phase noise of the
PLL. Low input bias current is also important, as bias current
affects the level of ‘reference spurs’, artifacts in the frequency spectrum of the PLL caused by mismatch or leakage
at the output of the phase detector. A large input and output
swing is beneficial in terms of increasing the flexibility in
biasing the op amp. The op amp can then be biased such
that the output range of the PLL is mapped efficiently onto
the input range of the VCO.
With a CMOS input, ultra low input bias currents (2 pA) and
low input referred voltage noise (5.5 nV/
is an ideal op amp for using in a PLL active loop filter. The
LM6211 has a ground sensing input stage, a rail-to-rail output stage, and an operating supply range of 5V - 24V, which
makes it a versatile choice for the design of a wide variety of
active loop filters.
Figure 7 shows the LM6211 used with the LMX2430 to
create an RF frequency synthesizer. The LMX2430 detects
the PLL output, compares it with its internal reference clock
and outputs the phase error in terms of current spikes. The
LM6211 is used to create a loop filter which averages the
error and provides a DC voltage to the VCO. The VCO
generates a sine wave at a frequency determined by the DC
voltage at its input. This circuit can provide output signal
frequencies as high as 2 GHz, much higher than a comparative passive loop filter. Compared to a similar passive loop
filter, the LM6211 doesn’t add significantly to the phase noise
of the PLL, except at the edge of the loop bandwidth, as
shown in Figure 6. A peaking of loop gain is expected, since
the loop filter is deliberately designed to have a wide bandwidth and a low phase margin so as to minimize locking time.
), the LM6211
LM6211
FIGURE 5. A Typical Active Loop Filter
20120303
20120355
FIGURE 6. Effect of LM6211 on Phase Noise of PLL
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Typical Applications (Continued)
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LM6211
FIGURE 7. LM6211 in the Active Loop Filter for LMX2430
ADC INPUT DRIVER
A typical application for a high performance op amp is as an
ADC driver, which delivers the analog signal obtained from
sensors and actuators to ADCs for conversion to the digital
domain and further processing. Important requirements in
this application are a slew rate high enough to drive the ADC
input and low input referred voltage and current noise. If an
op amp is used with an ADC, it is critical that the op amp
noise does not affect the dynamic range of the ADC. The
LM6211, with low input referred voltage and current noise,
provides a great solution for this application. For example,
the LM6211 can be used to drive an ADS121021, a 12-bit
ADC from National. If it provides a gain of 10 to a maximum
input signal amplitude of 100 mV, for a bandwidth as wide as
100 kHz, the average noise seen at the input of the ADC is
only 44.6 µVrms. Hence the dynamic range of the ADC,
20120336
measured in Effective Number of Bits or ENOB, is only
reduced by 0.3 bits, despite amplifying the input signal by a
gain of 10. Low input bias currents and high input impedance
also help as they prevent the loading of the sensor and allow
the measurement system to function over a large range.
Figure 8 shows a circuit for monitoring fluid pressure in a
hydraulic system, in which the LM6211 is used to sense the
error voltage from the pressure sensor. Two LM6211 amplifiers are used to make a difference amplifier which senses
the error signal, amplifies it by a gain of 100, and delivers it
to the ADC input. The ADC converts the error voltage into a
pressure reading to be displayed and drives the DAC, which
changes the voltage driving the resistance bridge sensor.
This is used to control the gain of the pressure measurement
circuit, such that the range of the sensor can be modified to
obtain the best resolution possible.
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Typical Applications (Continued)
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LM6211
FIGURE 8. Hydraulic Pressure Monitoring System
DAC OUTPUT AMPLIFIER
Op amps are often used to improve a DAC’s output driving
capability. High performance op amps are required as I-V
converters at the outputs of high resolution current output
DACs. Since most DACs operate with a single supply of 5V,
a rail-to-rail output swing is essential for this application. A
low offset voltage is also necessary to prevent offset errors in
the waveform generated. Also, the output impedance of
DACs is quite high, more than a few kΩ in some cases, so it
is also advisable for the op amp to have a low input bias
current. An op amp with a high input impedance also prevents the loading of the DAC, and hence, avoids gain errors.
The op amp should also have a slew rate which is fast
enough to not affect the settling time of the DAC output.
The LM6211, with a CMOS input stage, ultra low input bias
current, a wide bandwidth (20 MHz) and a rail-to-rail output
swing for a supply voltage of 24V is an ideal op amp for such
an application. Figure 9 shows a typical circuit for this application. The op amp is usually expected to add another time
constant to the system, which worsens the settling time, but
the wide bandwidth of the LM6211 (20 MHz) allows the
20120335
system performance to improve without any significant degradation of the settling time.
20120340
FIGURE 9. DAC Driver Circuit
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Typical Applications (Continued)
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LM6211
AUDIO PREAMPLIFIER
With low input referred voltage noise, low supply voltage and
low supply current, and low harmonic distortion, the LM6211
is ideal for audio applications. Its wide unity gain bandwidth
allows it to provide large gain over a wide frequency range
and it can be used to design a preamplifier to drive a load of
as low as 600Ω with less than 0.001% distortion. Two amplifier circuits are shown in Figure 10 and Figure 11. Figure
10 is an inverting amplifier, with a 10 kΩ feedback resistor,
,anda1kΩ input resistor, R1, and hence provides a gain
R
2
of −10. Figure 11 is a non-inverting amplifier, using the same
values for R
these circuits, the coupling capacitor C
frequency at which the circuit starts providing gain, while the
feedback capacitor C
gain starts dropping off. Figure 12 shows the frequency
response of the circuit in Figure 10 with different values of
.
C
F
and R2, and provides a gain of 11. In either of
1
FIGURE 10. Inverting Audio Amplifier
decides the lower
C1
decides the frequency at which the
F
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20120343
FIGURE 12. Frequency Response of the Non-Inverting
Preamplifier
TRANSIMPEDANCE AMPLIFIER
A transimpedance amplifier converts a small input current
into a voltage. This current is usually generated by a photodiode. The transimpedance gain, measured as the ratio of
the output voltage to the input current, is expected to be
large and wide-band. Since the circuit deals with currents in
the range of a few nA, low noise performance is essential.
The LM6211, being a CMOS input op amp, provides a wide
bandwidth and low noise performance while drawing very
low input bias current, and is hence ideal for transimpedance
applications.
A transimpedance amplifier is designed on the basis of the
current source driving the input. A photodiode is a very
common capacitive current source, which requires transimpedance gain for transforming its miniscule current into easily detectable voltages. The photodiode and amplifier’s gain
are selected with respect to the speed and accuracy required of the circuit. A faster circuit would require a photodiode with lesser capacitance and a faster amplifier. A more
sensitive circuit would require a sensitive photodiode and a
high gain. A typical transimpedance amplifier is shown in
Figure 13. The output voltage of the amplifier is given by the
equation V
plifier is limited, R
values of I
=−IINRF. Since the output swing of the am-
OUT
IN
should be selected such that all possible
F
can be detected.
The LM6211 has a large gain-bandwidth product (20 MHz),
which enables high gains at wide bandwidths. A rail-to-rail
output swing at 24V supply allows detection and amplification of a wide range of input currents. A CMOS input stage
with negligible input current noise and low input voltage
noise allows the LM6211 to provide high fidelity amplification
for wide bandwidths. These properties make the LM6211
ideal for systems requiring wide-band transimpedance amplification.
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FIGURE 11. Non-Inverting Audio Preamplifier
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Typical Applications (Continued)
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20120344
FIGURE 13. Photodiode Transimpedance Amplifier
The following parameters are used to design a transimped-
CM
; the
0
for the
ance amplifier: the amplifier gain-bandwidth product, A
amplifier input capacitance, C
tance, C
; the transimpedance gain required, RF; and the
D
amplifier output swing. Once a feasible R
; the photodiode capaci-
CM
is selected using
F
the amplifier output swing, these numbers can be used to
design an amplifier with the desired transimpedance gain
and a maximally flat frequency response. The input
common-mode capacitance with respect to V
LM6211 is give in Figure 14.
An essential component for obtaining a maximally flat response is the feedback capacitor, C
at the input of the amplifier, C
resistor, R
, generates a phase lag which causes gain-
F
peaking and can destabilize the circuit. C
sum of C
pole, f
zero in the noise gain, f
and CCM. The feedback capacitor CFcreates a
D
in the noise gain of the circuit, which neutralizes the
P
, created by the combination of R
Z
. The capacitance seen
F
, combined with the feedback
IN
is usually just the
IN
and CIN. If properly positioned, the noise gain pole created
can ensure that the slope of the gain remains at
by C
F
20 dB/decade till the unity gain frequency of the amplifier is
reached, thus ensuring stability. As shown in Figure 16,f
P
positioned such that it coincides with the point where the
noise gain intersects the op amp’s open loop gain. In this
case, f
pedance amplifier. The value of C
given by Equation (2). A larger value of C
is also the overall 3 dB frequency of the transim-
P
needed to make it so is
F
causes excessive
F
reduction of bandwidth, while a smaller value fails to prevent
gain peaking and maintain stability.
(2)
Calculating C
reasonably small values (
from Equation (2) can sometimes return un-
F
<
1 pF), especially for high speed
applications. In these cases, it is often more practical to use
the circuit shown in Figure 15 in order to allow more reasonable values. In this circuit, the capacitance C
times the effective feedback capacitance, C
’ is (1+ RB/RA)
F
. A larger ca-
F
pacitor can now be used in this circuit to obtain a smaller
effective capacitance.
LM6211
F
is
20120354
FIGURE 14. Input Common-Mode Capacitance vs. V
CM
FIGURE 15. Modifying C
For example, if a C
capacitor is available, R
= 9. This would convert a CF’ of 5 pF into a
R
B/RA
of 0.5 pF. This relationship holds as long as R
C
F
of 0.5 pF is needed, while onlya5pF
F
and RAcan be selected such that
B
20120347
F
<<
R
A
F
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Typical Applications (Continued)
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LM6211
SENSOR INTERFACES
The low input bias current and low input referred noise of the
LM6211 make it ideal for sensor interfaces. These circuits
are required to sense voltages of the order of a few µV, and
currents amounting to less than a nA, and hence the op amp
needs to have low voltage noise and low input bias current.
Typical applications include infra-red (IR) thermometry, thermocouple amplifiers and pH electrode buffers. Figure 17 is
an example of a typical circuit used for measuring IR radiation intensity, often used for estimating the temperature of an
object from a distance. The IR sensor generates a voltage
proportional to I, which is the intensity of the IR radiation
falling on it. As shown in Figure 17, K is the constant of
proportionality relating the voltage across the IR sensor (V
to the radiation intensity, I. The resistances R
selected to provide a high gain to amplify this voltage, while
is added to filter out the high frequency noise.
C
F
FIGURE 16. Method for CFselection
)
and RBare
A
IN
FIGURE 17. IR Radiation Sensor
20120346
20120348
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Physical Dimensions inches (millimeters) unless otherwise noted
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5-Pin SOT23
NS Package Number MF05A
LM6211 Low Noise, RRO Op-Amp with CMOS Input and 24V Operation
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves
the right at any time without notice to change said circuitry and specifications.
For the most current product information visit us at www.national.com.
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WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body, or
(b) support or sustain life, and whose failure to perform when
properly used in accordance with instructions for use
2. A critical component is any component of a life support
device or system whose failure to perform can be reasonably
expected to cause the failure of the life support device or
system, or to affect its safety or effectiveness.
provided in the labeling, can be reasonably expected to result
in a significant injury to the user.
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