Datasheet MC34067DWR2, MC33067DW, MC33067DWR2 Datasheet (MOTOROLA)

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The MC34067/MC33067 are high performance zero voltage switch
resonant mode controllers designed for off–line and dc–to–dc converter
applications that utilize frequency modulated constant off–time or constant
deadtime control. These integrated circuits feature a variable frequency
oscillator, a precise retriggerable one–shot timer, temperature compensated
reference, high gain wide bandwidth error amplifier, steering flip–flop, and
dual high current totem pole outputs ideally suited for driving power
MOSFETs.

Also included are protective features consisting of a high speed fault
comparator, programmable soft–start circuitry, input undervoltage lockout
with selectable thresholds, and reference undervoltage lockout.

These devices are available in dual–in–line and surface mount packages.

• Zero Voltage Switch Resonant Mode Operation

• Variable Frequency Oscillator with a Control Range Exceeding 1000:1

• Precision One–Shot Timer for Controlled Of f–Time

• Internally Trimmed Bandgap Reference

• 4.0 MHz Error Amplifier

• Dual High Current Totem Pole Outputs

• Selectable Undervoltage Lockout Thresholds with Hysteresis

• Enable Input

• Programmable Soft–Start Circuitry

• Low Startup Current for Off–Line Operation

HIGH PERFORMANCE

ZERO VOLTAGE SWITCH

RESONANT MODE

CONTROLLERS

SEMICONDUCTOR

TECHNICAL DATA

P SUFFIX

PLASTIC PACKAGE

CASE 648

DW SUFFIX

PLASTIC PACKAGE

CASE 751G

(SO–16L)

16

1

16

1

Simplified Block Diagram

15

V

CC

Enable /

UVLO Adjust

Osc Charge

Osc RC

Oscillator

Control Current

One–Shot
Error Amp

Output

Noninverting

Input

Inverting Input

Soft–Start

9
1
2

3

16

6
8

7

11

Error
Amp

V

UVLO /

CC

Enable

Variable

Frequency

Oscillator

One–Shot

2.5 V

Clamp

5.0 V

Reference

Soft–Start

MOTOROLA ANALOG IC DEVICE DATA

V

Steering

Flip–Flop

Fault Detector

Ground4

UVLO

ref

5

V

ref

14

Output A

12

Output B

13

Pwr Gnd

10

Fault Input

PIN CONNECTIONS

Osc Charge

Osc RC

Osc Control Current

Gnd

V

ref

Error Amp Out

Inverting Input

Noninverting Input

1
2
3
4
5
6
7
8

(Top View)

One–Shot RC

16

V

15

CC

Drive Output A

14

Power Gnd

13

Drive Output B

12

C

11

Soft–Start

Fault Input

10

Enable/UVLO

9

Adjust

ORDERING INFORMATION

Operating

Device

MC34067DW SO–16L
MC34067P Plastic DIP

MC33067DW SO–16L
MC33067P Plastic DIP

 Motorola, Inc. 1999 Rev 1, 05/99

Temperature Range

TA = 0 to + 70°C

TA = – 40° to + 85°C

Package

1

MC34067 MC33067

MAXIMUM RATINGS

Rating Symbol Value Unit

Power Supply Voltage V
Drive Output Current, Source or Sink (Note 1)

Continuous
Pulsed (0.5 µs, 25% Duty Cycle

Error Amplifier, Fault, One–Shot, Oscillator and

Soft–Start Inputs
UVLO Adjust Input V
Power Dissipation and Thermal Characteristics

DW Suffix, Plastic Package, Case 751G

TA = 25°C
Thermal Resistance, Junction–to–Air

P Suffix, Plastic Package, Case 648

TA = 25°C

Thermal Resistance, Junction–to–Air
Operating Junction Temperature T
Operating Ambient Temperature

MC34067
MC33067

Storage Temperature T

CC

I

O

V

in

in(UVLO)

P

D

R

θJA

P

D

R

θJA

J

T

A

stg

20 V

0.3

1.5

– 1.0 to + 6.0 V

– 1.0 to V

862
145

1.25
100

CC

mW

°C/W

W

°C/W

+ 150 °C

°C

0 to + 70

– 40 to + 85

– 55 to + 150 °C

A

V

ELECTRICAL CHARACTERISTICS (V

= 12 V [Note 2], R

CC

OSC

= 18.2 k, R

VFO

= 2940, C

= 300 pF, RT = 2370 k, CT = 300 pF,

OSC

CL = 1.0 nF. For typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless
otherwise noted.)

Characteristic

Symbol Min Typ Max Unit

REFERENCE SECTION

Reference Output Voltage (IO = 0 mA, TJ = 25°C) V
Line Regulation (VCC = 10 TO 18 V) Reg
Load Regulation (IO = 0 mA to 10 mA) Reg
Total Output Variation Over Line, Load, and Temperature V
Output Short Circuit Current I
Reference Undervoltage Lockout Threshold V

ref

line

load
ref
O

th

5.0 5.1 5.2 V
– 1.0 20 mV
– 1.0 20 mV

4.9 – 5.3 V

25 100 190 mA

3.8 4.3 4.8 V

ERROR AMPLIFIER

Input Offset Voltage (VCM = 1.5 V) V
Input Bias Current (VCM = 1.5 V) I
Input Offset Current (VCM = 1.5 V) I
Open Loop Voltage Gain (VCM = 1.5 V, VO = 2.0 V) A

IO
IB

IO

VOL

– 1.0 10 mV
– 0.2 1.0 µA
– 0 0.5 µA

70 100 – dB
Gain Bandwidth Product (f = 100 kHz) GBW 3.0 5.0 – MHz
Input Common Mode Rejection Ratio (VCM = 1.5 to 5.0 V) CMR 70 95 – dB
Power Supply Rejection Ratio (VCC = 10 to 18 V, f = 120 Hz) PSR 80 100 – dB
Output Voltage Swing

High State
Low State

NOTES: 1. Maximum package power dissipation limits must be observed.

2.Adjust VCC above the Startup threshold before setting to 12 V.

3.Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
T

=0°C for the MC34067 T

low

=–40°C for the MC33067 T

= + 70°C for MC34067

high

= + 85°C for MC33067

high

V

OH

V

OL

2.8
–

3.2

0.6

–

0.8

V

2

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

ELECTRICAL CHARACTERISTICS (V

= 12 V [Note 2], R

CC

OSC

= 18.2 k, R

VFO

= 2940, C

= 300 pF, RT = 2370 k, CT = 300 pF,

OSC

CL = 1.0 nF. For typical values TA = 25°C, for min/max values TA is the operating ambient temperature range that applies [Note 3], unless
otherwise noted.)

Characteristic

Symbol Min Typ Max Unit

OSCILLAT OR

Frequency (Error Amp Output Low)

TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = T

Low

to T

High

f

OSC(low)

Frequency (Error Amp Output High)

TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = T

Low

to T

High

f

OSC(high)

Oscillator Control Input Voltage, Pin 3 @ 25°C V

500
490

1900
1850

in

– 2.5 – V

525

–

540
550

2050–2150

2200

kHz

kHz

ONE–SHOT

Drive Output Off–Time

TA = 25°C
Total Variation (VCC = 10 to 18 V, TA = T

Low

to T

High

t

Blank

235
225

250

–

270
280

ns

DRIVE OUTPUTS

Output Voltage

Low State (I

Low State (I

High State (I

High State (I

Output Voltage with UVLO Activated (VCC = 6.0 V, I

= 20 mA)

Sink

= 200 mA)

Sink
Source
Source

= 20 mA)
= 200 mA)

= 1.0 mA) V

Sink

V

V

OL(UVLO)

Output Voltage Rise T ime (CL = 1.0 nF) t
Output Voltage Fall T ime (CL = 1.0 nF) t

OL

OH

r
f

–
–

9.5

9.0

0.8

1.5

10.3

9.7

1.2

2.0
–
–

– 0.8 1.2 V
– 20 50 ns
– 15 50 ns

V

FAULT COMP ARATOR

Input Threshold V
Input Bias Current (V
Propagation Delay to Drive Outputs (100 mV Overdrive) t

= 0 V) I

Pin 10

PLH(In/Out)

th

IB

0.93 1.0 1.07 V
– –2.0 –10 µA
– 60 100 ns

SOFT–START

Capacitor Charge Current (V
Capacitor Discharge Current (V

= 2.5 V) I

Pin 11

= 2.5 V) I

Pin 11

chg

dischg

4.5 9.0 14 µA

3.0 8.0 – mA

UNDERVOLTAGE LOCKOUT

Startup Threshold, VCC Increasing

Enable/UVLO Adjust Pin Open
Enable/UVLO Adjust Pin Connected to V

CC

Minimum Operating Voltage After Turn–On

Enable/UVLO Adjust Pin Open
Enable/UVLO Adjust Pin Connected to V

CC

Enable/UVLO Adjust Shutdown Threshold Voltage V
Enable/UVLO Adjust Input Current (Pin 9 = 0 V) I

V

th(UVLO)

V

CC(min)

th(Enable)

in(Enable)

14.8

8.0

8.0

7.6

16

9.0

9.0

8.6

17.2
10

10

9.6

6.0 7.0 – V
– –0.2 –1.0 mA

V

V

TOTAL DEVICE

Power Supply Current (Enable/UVLO Adjust Pin Open)

Startup (VCC = 13.5 V)
Operating (f

NOTES: 1. Maximum package power dissipation limits must be observed.

2.Adjust VCC above the Startup threshold before setting to 12 V.

3.Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
T

low

= 500 kHz) (Note 2)

OSC

=0°C for the MC34067 T
=–40°C for the MC33067 T

= + 70°C for MC34067

high

= + 85°C for MC33067

high

I

CC

–
–

0.5
27

0.8
35

mA

MOTOROLA ANALOG IC DEVICE DATA

3

MC34067 MC33067

Figure 1. Oscillator Timing Resistor

versus Discharge Time

500

Ω

400

300

200

100

, OSCILLAT OR TIMING RESIST OR (k )

OSC

R

C

= 300 pF

OSC

C

= 200 pF

OSC

C

= 500 pF

OSC

VCC = 12 V

∞

R

=

VFO

RT =

∞

CT = 500 pF

°

C

TA = 25

0

Oscillator Discharge Time is Measured at the Drive Outputs.

t

, OSCILLAT OR DISCHARGE TIME (µs)

dischg

3500
3000
2500
2000
1500
1000

, OSCILLAT OR FREQUENCY (kHz)

500

OSC

f

Figure 3. Error Amp Output Saturation

V oltage versus Oscillator Control Current

0.35

0.30

0.25

0.20

0.15

0.10

, OUTPUT SA TURATION VOLTAGE (V)

sat

V

0.05
0 0.5 1.0 1.5 2.0 2.5 3.0 0.1 0.3 0.6 1.0 3.0 6.0 10

I

, OSCILLAT OR CONTROL CURRENT (mA) tOS, ONE–SHOT PERIOD (

OSC

60

30

Ω

20

10

, TIMING RESISTOR (k )

400

T

R

3.0

Figure 2. Oscillator Frequency versus

Oscillator Control Current

VCC = 12 V

°

C

TA = 25
R

= 18.2 k

OSC

C

= 300 pF

OSC

0

0 400 800 1200 1600 20000 20406080100

I

, OSCILLAT OR CONTROL CURRENT (mA)

OSC

Figure 4. One–Shot Timing Resistor

versus Period

VCC = 12 V
C

= 500 pF

OSC

R

= 100 k

OSC

°

C

TA = 25

CT = 300 pF CT = 500 pF

CT = 200 pF

One–Shot Period is Measured
at the Drive Outputs.

µ

s)

Figure 5. Open Loop V oltage Gain and Phase

versus Frequency

50

Phase
Margin

°

= 64

VCC = 12 V
VO = 2.0 V
RL = 100 k
TA = 25

40
30

20
10

0

, OPEN LOOP VOL TAGE GAIN (dB)

–10

VOL

A

–20

10 k 100 k 1.0M 10M – 55 – 25 0 25 50 75 125100

Gain

Phase

f, FREQUENCY (Hz)

50
60
70

°

C

80
90
100

0, EXCESS PHASE (DEGREES)

110
120

, REFERENCE OUTPUT VOLTAGE CHANGE (mV)

∇

V

4

Figure 6. Reference Output V oltage Change

versus T emperature

*V

= 5.0 V

ref

0

–10

–20

*V

= 5.0 V

–30

–40

–50

ref

ref

*V

= 5.0 V

ref

TA, AMBIENT TEMPERATURE (°C)

VCC = 12 V

∞

RL =
*V

ref at TA

= 25°C

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

Figure 7. Reference V oltage Change

0

–10

–20

–30

–40

–50

, REFERENCE OUTPUT VOLTAGE CHANGE (mV)

ref

V

∇

VCC = 12 V

Figure 9. Drive Output Waveform

90%

versus Source Current

TA = – 40°C

TA = – 20°C

TA = –125°C

I

, REFERENCE SOURCE CURRENT (mA)

ref

CL = 1.0 nF

TA = 25

Figure 8. Drive Output Saturation Voltage

versus Load Current

, OUTPUT SA TURATION VOLTAGE (V)
V

sat

–1.0

– 2.0
– 3.0

3.0

2.0

1.0

0

V

CC

TA = 25°C

TA = – 40°C

0

0 0.2 0.4 0.6 0.8 1.00 20 40 60 80 100

Source Saturation

(Load to Ground)

TA = – 40°C

TA = 25°C

Source Saturation

(Load to VCC)

IO, OUTPUT LOAD CURRENT (A)

80

VCC = 12 V

µ

s Pulsed Load

120 Hz Rate

Gnd

Figure 10. Soft–Start Saturation Voltage

versus Capacitor Discharge Current

3.2

°

C

2.4

10%

2000

1600

1200

800

f, FREQUENCY (kHz)

400

20 ns/DIV

Figure 11. Operating Frequency

versus Supply Current

VCC = 12 V
CL = 1.0 nF

°

C

TA = 25

1.6

0.8

OL

0

V , SOFT–START SATURATION VOLTAGE (V)

0 2.0 4.0 6.0 8.0 10

I

, CAPACITOR DISCHARGE CURRENT (mA)

dchg

VCC = 12 V
Pin 10 = V
TA = 25 °C

Figure 12. Supply Current versus Supply V oltage

24

TA = 25 °C

20

Enable/UVLO

Adjust Pin

Open

CC

I , SUPPLY CURRENT (mA)

16

12

8.0

4.0

Enable/UVLO

Adjust Pin

to V

CC

ref

0

30 40 50 60 70 80 90 0 4.0 8.0 12 16 20

ICC, INPUT SUPPLY CURRENT (mA)

0

VCC, SUPPLY VOLTAGE (V)

MOTOROLA ANALOG IC DEVICE DATA

5

C

R

T

OSC

C

OSC

R

T

Error Amp Output

Noninverting Input

V

CC

Enable /

UVLO Adjust

OSC Charge

OSC RC

One–Shot RC

Oscillator

Control Current

I

OSO

Inverting Input

R

VFO

Soft–Start

15

16

11

9

1

2

3

6
8

7

7.0k

Error Amp

MC34067 MC33067

Figure 13. MC34067 Representative Block Diagram

50k 7.0k

V

50k

I

OSC

8.0V

Q1

One–Shot

3.1V

Error Amp

Clamp

9.0

µ

VCC UVLO

V

D1

Oscillator

4.9V/3.6V

4.9V/3.6V

A

5.1V

Reference

ref

Q2

V

ref

ref

UVLO

Steering

Flip–Flop

T

Q
Q

R

4.2/4.0V

Fault Comparator

1.0V

V

ref

5

Output A

14

Power Ground

13

Output B

12

Fault Input

10

C

OSC

One–Shot

Output A

Output B

5.1 V

3. 6 V

5.1 V

3.6 V

Timing Diagram

t

OS

Error Amp output high, minimum I
occurring at minimum input voltage, maximum load.

t

OS

OSC

current

t

OS

Ground4

t

OS

Error Amp output low, maximum I
occurring at maximum input voltage, minimum load.

t

OS

t

OS

OSC

current

6

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

OPERA TING DESCRIPTION

Introduction

As power supply designers have strived to increase power
conversion efficiency and reduce passive component size,
high frequency resonant mode power converters have
emerged as attractive alternatives to conventional
pulse–width modulated control. When compared to
pulse–width modulated converters, resonant mode control
offers several benefits including lower switching losses,
higher efficiency, lower EMI emission, and smaller size. A
new integrated circuit has been developed to support this
trend in power supply design. The MC34067 Resonant Mode
Controller is a high performance bipolar IC dedicated to
variable frequency power control at frequencies exceeding

1.0 MHz. This integrated circuit provides the features and
performance specifically for zero voltage switching resonant
mode power supply applications.

The primary purpose of the control chip is to provide a
fixed off–time to the gates of external power MOSFETs at a
repetition rate regulated by a feedback control loop.
Additional features of the IC ensure that system startup and
fault conditions are administered in a safe, controlled manner.

A simplified block diagram of the IC is shown on the front
page, which identifies the main functional blocks and the
block–to–block interconnects. Figure 13 is a detailed
functional diagram which accurately represents the internal
circuitry. The various functions can be divided into two
sections. The first section includes the primary control path
which produces precise output pulses at the desired
frequency. Included in this section are a variable frequency
Oscillator, a One–Shot, a pulse Steering Flip–Flop, a pair of
power MOSFET Drivers, and a wide bandwidth Error
Amplifier. The second section provides several peripheral
support functions including a voltage reference, undervoltage
lockout, Soft–Start circuit, and a fault detector.

Primary Control Path

The output pulse width and repetition rate are regulated
through the interaction of the variable frequency Oscillator,
One–Shot timer and Error Amplifier. The Oscillator triggers
the One–Shot which generates a pulse that is alternately
steered to a pair of totem pole output drivers by a toggle
Flip–Flop. The Error Amplifier monitors the output of the
regulator and modulates the frequency of the Oscillator. High
speed Schottky logic is used throughout the primary
control channel to minimize delays and enhance high
frequency characteristics.

Oscillator

The characteristics of the variable frequency Oscillator are
crucial for precise controller performance at high operating
frequencies. In addition to triggering the One–Shot timer and
initiating the output deadtime, the oscillator also determines
the initial voltage for the one–shot capacitor. The Oscillator is
designed to operate at frequencies exceeding 1.0 MHz. The
Error Amplifier can control the oscillator frequency over a
1000:1 frequency range, and both the minimum and
maximum frequencies are easily and accurately
programmed by the proper selection of external components.

The functional diagram of the Oscillator and One–Shot
timer is shown in Figure 14. The oscillator capacitor (C
initially charged by transistor Q1. When C

exceeds the

OSC

OSC

) is

4.9 V upper threshold of the oscillator comparator, the base
of Q1 is pulled low allowing C
external resistor, (R
(I

). When the voltage on C

OSC

and the oscillator control current,

OSC),

to discharge through the

OSC

falls below the

OSC

comparator’s 3.6 V lower threshold, Q1 turns on and again
charges C

C

OSC

high slew rate of C

.

OSC

charges from 3.6 V to 5.1 V in less than 50 ns. The

and the propagation delay of the

OSC

comparator make it difficult to control the peak voltage. This
accuracy issue is overcome by clamping the base of Q1
through a diode to a voltage reference. The peak voltage of
the oscillator waveform is thereby precisely set at 5.1 V.

Figure 14. Oscillator and One–Shot Timer

V

ref

D1

Oscillator

4.9V/3.6V

4.9V/3.6V

+

V

OE

V

+

OE

OSC Charge

1

R

OSC

C

OSC

CT

R

T

OSC RC

One–Shot RC

Oscillator

Control Current

I

OSO

R

VFO

Error Amp Output

2

10

3
6

I

OSC

Error Amp
Charge

Q1

One–Shot

3.1V

The frequency of the Oscillator is modulated by varying
the current flowing out of the Oscillator Control Current (I
pin. The I

pin is the output of a voltage regulator. The

OSC

OSC

input of the voltage regulator is tied to the variable frequency
oscillator. The discharge current of the Oscillator increases
by increasing the current out of the I

pin. Resistor R

OSC

VFO

used in conjunction with the Error Amp output to change the
I

current. Maximum frequency occurs when the Error

OSC

Amplifier output is at its low state with a saturation voltage of

0.1 V at 1.0 mA.

The minimum oscillator frequency will result when the
I

current is zero, and C

OSC

external resistor (R

OSC

is discharged through the

OSC

). This occurs when the Error
Amplifier output is at its high state of 2.5 V . The minimum and
maximum oscillator frequencies are programmed by the
proper selection of resistor R
frequency is programmed by R

1

t

–

PD

ȏ

nC

ǒǓ

R

OSC

ƒ

(min)

=

OSC

5.1

3.6

and R

OSC

using Equation 1:

OSC

t

(max)

=

0.348

. The minimum

VFO

–

70 ns

C

OSC

(1)

where tPD is the internal propagation delay .

)

is

MOTOROLA ANALOG IC DEVICE DATA

7

MC34067 MC33067

The maximum oscillator frequency is set by the current
through resistor R
C

at the maximum oscillator frequency can be calculated

OSC

. The current required to discharge

VFO

by Equation 2:

I

(max)

C

=

OSC

5.1 – 3.6
1

ƒ

(max)

The discharge current through R

1.5C

=

must also be known

OSC

OSC

ƒ

(max)

(2)

and can be calculated by Equation 3:

1

ƒ

(min)

ǒǓ

–

C

R

OSC

R

OSC

Resistor R

5.1 – 3.6

=

R

OSC

1.5

=

R

OSC

can now be calculated by Equation 4:

VFO

2.5 – V

R=

VFO

I

(max)IR

OSC

εI

ƒ

(min)

R

1

OSC

C

OSC

ǒǓ

–

ε

EAsat

–

OSC

(3)

(4)

One–Shot Timer

The One–Shot is designed to disable both outputs
simultaneously providing a deadtime before either output is
enabled. The One–Shot capacitor (CT) is charged
concurrently with the oscillator capacitor by transistor Q1, as
shown in Figure 14. The one–shot period begins when the
oscillator comparator turns off Q1, allowing CT to discharge.
The period ends when resistor RT discharges CT to the
threshold of the One–Shot comparator. The lower threshold
of the One–Shot is 3.6 V. By choosing CT, RT can by solved
by Equation 5:

t

C

ȏ

T

OS

5.1

n

ǒǓ

3.6

R

=

T

t

=

OS

0.348

C

T

(5)

Errors in the threshold voltage and propagation delays
through the output drivers will affect the One–Shot period. To
guarantee accuracy, the output pulse of the control chip is
trimmed to within 5% of 250 ns with nominal values of RT and
CT.

The outputs of the Oscillator and One–Shot comparators
are OR’d together to produce the pulse tOS, which drives the
Flip–Flop and output drivers. The output pulse (tOS) is
initiated by the Oscillator and terminated by the One–Shot
comparator. With zero–voltage resonant mode converters,
the oscillator discharge time should never be set less than
the one–shot period.

Figure 15. Error Amplifier and Clamp

Oscillator

Control Current

3

I

OSC

Error Amp Output

Noninverting Input

Inverting Input

R

VFO

6

8

7

Error

Amp

When the Error Amplifier output is coupled to the I
by R
Control Current, I

, as illustrated in Figure 15, it provides the Oscillator

VFO

. The output swing of the Error Amplifier

OSC

3.1V

Error Amp

Charge

OSC

pin

is restricted by a clamp circuit to improve its transient
recovery time.

Output Section

The pulse(tOS), generated by the Oscillator and One–Shot
timer is gated to dual totem–pole output drives by the
Steering Flip–Flop shown in Figure 16. Positive transitions of
tOS toggle the Flip–Flop, which causes the pulses to alternate
between Output A and Output B. The flip–flop is reset by the
undervoltage lockout circuit during startup to guarantee that
the first pulse appears at Output A.

Figure 16. Steering Flip–Flop and Output Drivers

V

OE

±

Steering

Flip–Flop

Q

T

Q

R

Pwr

Gnd

Pwr

Gnd

V

OE

±

Output A

14

Power Ground

13

Output B

12

Error Amplifier

A fully accessible high performance Error Amplifier is
provided for feedback control of the power supply system.
The Error Amplifier is internally compensated and features dc
open loop gain greater than 70 dB, input offset voltage of less
than 10 mV and a guaranteed minimum gain–bandwidth
product of 2.5 MHz. The input common mode range extends
from 1.5 V to 5.1 V, which includes the reference voltage.

8

The totem–pole output drivers are ideally suited for driving

power MOSFETs and are capable of sourcing and sinking

1.5 A. Rise and fall times are typically 20 ns when driving a

1.0 nF load. High source/sink capability in a totem–pole
driver normally increases the risk of high cross conduction
current during output transitions. The MC34067 utilizes a
unique design that virtually eliminates cross conduction, thus
controlling the chip power dissipation at high frequencies. A
separate power ground pin is provided to isolate the sensitive
analog circuitry from large transient currents.

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

Figure 17. Undervoltage Lockout and Reference

V

CC

15

7.0k

8.0V

VCC UVLO

Enable /

UVLO Adjust

50k

7.0k

9

50k

PERIPHERAL SUPPORT FUNCTIONS

The MC34067 Resonant Controller provides a number of
support and protection functions including a precision voltage
reference, undervoltage lockout comparators, soft–start
circuitry, and a fault detector. These peripheral circuits
ensure that the power supply can be turned on and off in a
controlled manner and that the system will be quickly
disabled when a fault condition occurs.

Undervoltage Lockout and V oltage Reference

Separate undervoltage lockout comparators sense the
input VCC voltage and the regulated reference voltage as
illustrated in Figure 17. When VCC increases to the upper
threshold voltage, the VCC UVLO comparator enables the
Reference Regulator. After the V
Regulator rises to 4.2 V , the V
the UVLO signal to a logic zero state enabling the primary
control path. Reducing VCC to the lower threshold voltage
causes the VCC UVLO comparator to disable the Reference
Regulator. The V

UVLO comparator then switches the

ref

UVLO output to a logic one state disabling the controller.

The Enable/UVLO Adjust pin allows the power supply
designer to select the VCC UVLO threshold voltages. When
this pin is open, the comparator switches the controller on at
16 V and off at 9.0 V. If this pin is connected to the V
terminal, the upper and lower thresholds are reduced to

9.0 V and 8.6 V, respectively. Forcing the Enable/UVLO
Adjust pin low will pull the VCC UVLO comparator input low
(through an internal diode) turning off the controller.

The Reference Regulator provides a precise 5.1 V
reference to internal circuitry and can deliver up to 10 mA to
external loads. The reference is trimmed to better than 2%
initial accuracy and includes active short circuit protection.

Fault Detector

The high speed Fault Comparator illustrated in Figure 18
can protect a power supply from destruction under fault
conditions. The Fault Input pin connects to the input of the
Fault Comparator. The Fault Comparator output connects to
the output drivers. This direct path reduces the propagation

output of the Reference

ref

UVLO comparator switches

ref

CC

V

V

ref

ref

UVLO

4.2/4.0V

V

ref

5

5.1V

Reference

UVLO

delay from the Fault Input to the A and B outputs to typically
70 ns. The Fault Comparator output is also OR’d with the
UVLO output from the V

UVLO comparator to produce the

ref

logic output labeled “UVLO+Fault”. This signal disables the
Oscillator and One–Shot by forcing both the C

OSC

and C

capacitors to be continually charged.

Figure 18. Fault Detector and Soft–Start

UVLO

6 Ground

Fault

Comparator

1.0V

Fault
Input

10

C

Soft–Start

11

9.0

Ea Clamp

Soft–Start

Buffer

UVLO + Fault

µ

A

Soft–Start Circuit

The Soft–Start circuit shown in Figure 18 forces the
variable frequency Oscillator to start at the maximum
frequency and ramp downward until regulated by the
feedback control loop. The external capacitor at the
C

Soft–Start

terminal is initially discharged by the UVLO+Fault
signal. The low voltage on the capacitor passes through the
Soft–Start Buffer to hold the Error Amplifier output low. After
UVLO+Fault switches to a logic zero, the soft–start
capacitor is charged by a 9.0 µA current source. The buffer
allows the Error Amplifier output to follow the soft–start
capacitor until it is regulated by the Error Amplifier inputs. The
soft–start function is generally applicable to controllers
operating below resonance and can be disabled by simply
opening the C

Soft–Start

terminal.

T

MOTOROLA ANALOG IC DEVICE DATA

9

MC34067 MC33067

APPLICATIONS INFORMATION

The MC34067 is specifically designed for zero voltage
switching (ZVS) quasi–resonant converter (QRC)
applications. The IC is optimized for double–ended push–pull
or bridge type converters operating in continuous conduction
mode. Operation of this type of ZVS with resonant properties
is similar to standard push–pull or bridge circuits in that the
energy is transferred during the transistor on–time. The
difference is that a series resonant tank is usually introduced
to shape the voltage across the power transistor prior to
turn–on. The resonant tank in this topology is not used to
deliver energy to the output as is the case with zero current
switch topologies. When the power transistor is enabled the
voltage across it should already be zero, yielding minimal
switching loss. Figure 19 shows a timing diagram for a
half–bridge ZVS QRC. An application circuit is shown in
Figure 20. The circuit built is a dc to dc half–bridge converter
delivering 75 W to the output from a 48 V source.

When building a zero voltage switch (ZVS) circuit, the
objective is to waveshape the power transistor’s voltage
waveform so that the voltage across the transistor is zero
when the device is turned on. The purpose of the control IC is
to allow a resonant tank to waveshape the voltage across the
power transistor while still maintaining regulation. This is
accomplished by maintaining a fixed deadtime and by
varying the frequency; thus the effective duty cycle is
changed.

Primary side resonance can be used with ZVS circuits. In
the application circuit, the elements that make the resonant
tank are the primary leakage inductance of the transformer
(LL) and the average output capacitance (C
MOSFET (CR). The desired resonant frequency for the
application circuit is calculated by Equation 6:

=

ƒ

r

1

LL2C

π2

R

) of a power

OSS

(6)

In the application circuit, the operating voltage is low and
the value of C
the C
of the CR is approximated as the average C
MOSFET . For the application circuit the average C
calculated by Equation 7:

achieve the desired resonant frequency .
than the leakage inductance. Figure 19 shows the primary

current ramping toward its peak value during the resonant
transition. During this time, there is circulating current
flowing through the secondary inductance, which effectively
makes the primary inductance appear shorted. Therefore,
the current through the primary will ramp to its peak value at
a rate controlled by the leakage inductance and the applied
voltage. Energy is not transferred to the secondary during
this stage, because the primary current has not overcome the
circulating current in the secondary. The larger the leakage
inductance, the longer it takes for the primary current to slew.
The practical effect of this is to lower the duty cycle, thus
reducing the operating range.

inductance, not by the MC34067. The One–Shot in the
MC34067 only assures that the power switch is turned on
under a zero voltage condition. Adjust the one–shot period so
that the output switch is activated while the primary current is
slewing but before the current changes polarity . The resonant
stage should then be designed to be as long as the time for
the primary current to go to zero amps.

of a MOSFET changes with drain voltage, the value

OSS

C

R

The MOSFET chosen fixes CR and that LL is adjusted to

However, the desired resonant frequency is less critical

The maximum duty cycle is controlled by the leakage

versus Drain Voltage is known. Because

OSS

=

2 * C

measured at

OSS

1

V

2

OSS

OSS

in

of the

can be

(7)

10

MOTOROLA ANALOG IC DEVICE DATA

5.1 V

C

OSC

3.6 V

MC34067 MC33067

Figure 19. Application Timing Diagram

One–Shot

Output A

Output B

+ I

5.1 V

3.6 V

V

in

1/2 V

in

0 V

primary

0 A

– I

primary

Vin/Turns Ratio

Output Diode

Voltage

MOTOROLA ANALOG IC DEVICE DATA

11

MC34067 MC33067

= 5.0V

out

V

230

L2 =

100ns

FB

V

µ

L1 =

1.8

= 36 – 56V

in

V

51, 0.5W

500pF

1.0

100

5

CTL

MBR2535

1.0

MTP33N10E

T2

1N5819

T3

1.0k

1N5819 x 4

100

1.0k

14

13

3.9k

470

Secondary: 6 turns center tapped #48 A WG (1300 strands litz wire)

Core: Philips 3F3 4312 020 4124

Bobbin: Philips 4322 021 3525

Primary Leakage Inductance = 1.0 Hµ

Core: Philips 3F3 EP7–3F3

Bobbin: Philips EP7PCB1–6

12

470pF

10

T1 = Primary: 12 turns #48 AWG (1300 strands litz wire)

T2 = All windings: 8 turns #36 AWG

Core: Philips 3F3 EP10–3F3

L1 = 2 turns #48 AWG (1300 strands litz wire)

T3 = Coilcraft D1870 (100 turns)

″ diameter air code

Bobbin: Philips EP10PCB1–8

Core: 0.5

Inductance = 100 nH

Inductance = 1.8 Hµ

MC34067–5803

L2 = 5 turns #48 AWG (1300 strands litz wire)

Insulators = Berquist Sil–Pad 1500

Heatsinks = AAVID Engineering Inc. 533402B02552 with clip

±0.198%

p–p

20 mV =

4.0 mV = ±0.039%

25 mV

83.5%

4

84.2%

Figure 20. Application Circuit

CC

V

Reference

15

10

Regulator

9

= 1.0 MHz

= 1.7 MHz

= 1.0 MHz

switch

switch

=15 A

switch

O

= 10 A to 15 A

O

= 40 V to 56 V, I

= 48 V, I

= 48 V, I = 15 A, f

= 48 V, I = 10 A, f

in

in

V

1

18k

330pF

2

3

6

8

10

1.1k
10k

2.7k
330pF

1.6k

FB

100pF

V

220pF

1500pF

16k

7

11

0.01

Test Conditions Results

V

Line Regulation

Load Regulation

= 48 V, I = 15 A, f

in O

in O

in O

V

V

V

Output Ripple

Efficiency

12

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

Figure 21. Printed Circuit Board and Component Layout

3.875

″

5.0

″

(Bottom View)

MOTOROLA ANALOG IC DEVICE DATA

(Top View)

13

MC34067 MC33067

OUTLINE DIMENSIONS

PLASTIC PACKAGE

–A–

916

B

18

H

G

16 9

M

B

H8X

M

0.25

0.25 B

14X

F

D

16 PL

0.25 (0.010) T

D

B16X

M

S

A

T

e

C

S

SEATING

–T–

PLANE

K

M

M

A

PLASTIC PACKAGE

CASE 751G–03

A

E

81

B

S

A

SEATING
PLANE

A1

T

P SUFFIX

CASE 648–08

ISSUE R

J

DW SUFFIX

(SO–16L)

ISSUE B

_

h X 45

C

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION L TO CENTER OF LEADS WHEN
FORMED PARALLEL.

4. DIMENSION B DOES NOT INCLUDE MOLD FLASH.

5. ROUNDED CORNERS OPTIONAL.

DIM MIN MAX MIN MAX

L

M

q

L

A 0.740 0.770 18.80 19.55
B 0.250 0.270 6.35 6.85
C 0.145 0.175 3.69 4.44
D 0.015 0.021 0.39 0.53
F 0.040 0.70 1.02 1.77

G 0.100 BSC 2.54 BSC

H 0.050 BSC 1.27 BSC
J 0.008 0.015 0.21 0.38
K 0.110 0.130 2.80 3.30
L 0.295 0.305 7.50 7.74

M 0 10 0 10

S 0.020 0.040 0.51 1.01

NOTES:

1. DIMENSIONS ARE IN MILLIMETERS.

2. INTERPRET DIMENSIONS AND TOLERANCES
PER ASME Y14.5M, 1994.

3. DIMENSIONS D AND E DO NOT INLCUDE MOLD
PROTRUSION.

4. MAXIMUM MOLD PROTRUSION 0.15 PER SIDE.

5. DIMENSION B DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 TOTAL IN EXCESS
OF THE B DIMENSION AT MAXIMUM MATERIAL
CONDITION.

MILLIMETERS

DIM MIN MAX

A 2.35 2.65

A1 0.10 0.25

B 0.35 0.49
C 0.23 0.32
D 10.15 10.45
E 7.40 7.60
e 1.27 BSC
H 10.05 10.55
h 0.25 0.75
L 0.50 0.90

q

0 7

MILLIMETERSINCHES

____

__

14

MOTOROLA ANALOG IC DEVICE DATA

MC34067 MC33067

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty , representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “T ypical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that
Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.

MOTOROLA ANALOG IC DEVICE DATA

15

MC34067 MC33067

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HOME PAGE: http://motorola.com/sps/

16

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MOTOROLA ANALOG IC DEVICE DATA

Mfax is a trademark of Motorola, Inc.

MC34067/D