MIC2169 Micrel
50
55
60
65
70
75
80
85
90
95
100
0 2 4 6 8 10 12 14 16
EFFICIENCY (%)
I
LOAD
(A)
MIC2169 Efficiency
V
IN
= 5V
V
OUT
= 3.3V
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MIC2169
500kHz PWM Synchronous Buck Control IC
General Description
The MIC2169 is a high-efficiency, simple to use 500kHz
PWM synchronous buck control IC housed in a small MSOP10 package. The MIC2169 allows compact DC/DC solutions
with a minimal external component count and cost.
The MIC2169 operates from a 3V to 14.5V input, without the
need of any additional bias voltage. The output voltage can
be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range.
The MIC2169 senses current across the high-side N-Channel MOSFET, eliminating the need for an expensive and
lossy current-sense resistor. Current limit accuracy is maintained by a positive temperature coefficient that tracks the
increasing R
and space are saved by the internal in-rush-current limiting
digital soft-start.
The MIC2169 is available in a 10-pin MSOP package, with a
wide junction operating range of –40°C to +125°C.
All support documentation can be found on Micrel’s web
site at www.micrel.com.
of the external MOSFET. Further cost
DS(ON)
Features
• 3V to 14.5V input voltage range
• Adjustable output voltage down to 0.8V
• Up to 95% efficiency
• 500kHz PWM operation
• Adjustable current limit senses high-side N-Channel
MOSFET current
• No external current-sense resistor
• Adaptive gate drive increases efficiency
• Ultra-fast response with hysteretic transient recovery
mode
• Overvoltage protection protects the load in fault
conditions
• Dual mode current limit speeds up recovery time
• Hiccup mode short-circuit protection
• Internal soft-start
• Dual function COMP and EN pin allows low-power
shutdown
• Small size MSOP 10-lead package
Applications
• Point-of-load DC/DC conversion
• Set-top boxes
• Graphic cards
• LCD power supplies
• Telecom power supplies
• Networking power supplies
• Cable modems and routers
T ypical Application
V
= 5V
IN
100µF
150pF
Micrel, Inc. • 1849 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 944-0970 • http://www.micrel.com
November 2003 1 M9999-111803
SD103BWS
4.7µF
0.1µF
4kΩ
100nF
VDD
VIN
COMP/EN
MIC2169
GND
BST
HSD
VSW
LSD
FB
CS
1kΩ
IRF7821
IRF7821
MIC2169 Adjustable Output 500kHz Converter
2.5µH
3.3V
10kΩ
150µF x 2
3.24kΩ
MIC2169 Micrel
Ordering Information
Part Number Frequency Junction Temp. Range Package
MIC2169BMM 500kHz –40°C to +125°C 10-lead MSOP
Pin Configuration
VDD
CS
COMP/EN
FB GND65
Pin Description
Pin Number Pin Name Pin Function
1 VIN Supply Voltage (Input): 3V to 14.5V.
2 VDD 5V Internal Linear Regulator (Output): V
3 CS Current Sense / Enable (Input): Current-limit comparator noninverting input.
4 COMP/EN Compensation (Input): Dual function pin. Pin for external compensation. If
5 FB Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
6 GND Ground (Return).
7 LSD Low-Side Drive (Output): High-current driver output for external synchro-
8 VSW Switch (Return): High-side MOSFET driver return.
9 HSD High-Side Drive (Output): High-current output-driver for the high-side
10 BST Boost (Input): Provides the drive voltage for the high-side MOSFET driver.
1VIN
2
3
4
10 BST
HSD
9
VSW
8
LSD
7
10-Pin MSOP (MM)
is the external MOSFET gate
drive supply voltage and an internal supply bus for the IC. When V
DD
this regulator operates in dropout mode.
The current limit is sensed across the MOSFET during the ON time. The
current can be set by the resistor in series with the CS pin.
this pin is pulled below 0.2V, with the reference fully up the device shuts
down (50µA typical current draw).
nous MOSFET.
MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs
should be used. At VIN > 5V, 5V threshold MOSFETs should be used.
The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.
is <5V,
IN
M9999-111803 2 November 2003
MIC2169 Micrel
Absolute Maximum Ratings
(1)
Supply Voltage (VIN) ..................................................15.5V
Booststrapped Voltage (V
Junction Temperature (T
) ............................... VIN +5V
BST
) ................ –40°C ≤ TJ ≤ +125°C
J
Storage Temperature (TS) ....................... –65°C to +150°C
Electrical Characteristics
(3)
Operating Ratings
Supply Voltage (VIN) .................................... +3V to +14.5V
Output Voltage Range...........................0.8V to V
Package Thermal Resistance
θJA 10-lead MSOP ............................................180°C/W
(2)
IN
× D
MAX
TJ = 25°C, VIN = 5V; bold values indicate –40°C < TJ < +125°C; unless otherwise specified.
Parameter Condition Min Typ Max Units
Feedback Voltage Reference (± 1%) 0.792 0.8 0.808 V
Feedback Voltage Reference (± 2% over temp) 0.784 0.8 0.816 V
Feedback Bias Current 30 100 nA
Output Voltage Line Regulation 0.03 % / V
Output Voltage Load Regulation 0.5 %
Output Voltage Total Regulation 3V ≤ VIN ≤ 14.5V; 1A ≤ I
≤ 10A; (V
OUT
OUT
= 2.5V)
(4)
0.6 %
Oscillator Section
Oscillator Frequency 450 500 550 kHz
Maximum Duty Cycle 92 %
Minimum On-Time
(4)
30 60 ns
Input and VDD Supply
PWM Mode Supply Current V
= VIN –0.25V; VFB = 0.7V (output switching but excluding 1.5 3 mA
CS
external MOSFET gate current.)
Shutdown Quiescent Current V
V
Shutdown Threshold 0.1 0.25 0.4 V
COMP
Shutdown Blanking C
V
COMP
Period
Digital Supply Voltage (VDD)V
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max),
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature, and the regulator will go into thermal shutdown.
2. Devices are ESD sensitive, handling precautions required.
3. Specification for packaged product only.
4. Guaranteed by design.
COMP/EN
COMP
IN
= 0V 50 150 µA
= 100nF 4 ms
≥ 6V 4.7 5 5.3 V
November 2003 3 M9999-111803
MIC2169 Micrel
Electrical Characteristics
(5)
Parameter Condition Min Typ Max Units
Error Amplifier
DC Gain 70 dB
Transconductance 1ms
Soft-Start
Soft-Start Current After timeout of internal timer. See
“Soft-Start”
section. 8.5 µA
Current Sense
CS Over Current Trip Point VCS = VIN –0.25V 160 200 240 µA
Temperature Coefficient +1800 ppm/°C
Output Fault Correction Thresholds
Upper Threshold, V
Lower Threshold, V
FB_OVT
FB_UVT
(relative to VFB)+3%
(relative to VFB) –3%
Gate Drivers
Rise/Fall Time Into 3000pF at VIN > 5V 30 ns
Output Driver Impedance Source, V
= 5V 6 Ω
IN
Sink, VIN = 5V 6 Ω
Source, V
= 3V 10 Ω
IN
Sink, VIN = 3V 10 Ω
Driver Non-Overlap Time Note 6 10 20 ns
Notes:
5. Specification for packaged product only.
6. Guaranteed by design.
M9999-111803 4 November 2003
MIC2169 Micrel
0.7980
0.7985
0.7990
0.7995
0.8000
0.8005
0.8010
0 5 10 15
V
FB
(V)
VIN (V)
VFB Line Regulation
4.90
4.92
4.94
4.96
4.98
5.00
5.02
0 5 10 15 20 25 30
V
DD
REGULATOR VOLTAGE (V)
LOAD CURRENT (mA)
VDD Load Regulation
450
460
470
480
490
500
510
520
530
540
550
-60 -30 0 30 60 90 120 150
FREQUENCY (kHz)
TEMPERATURE (°C)
-1.5
-1.0
-0.5
0
0.5
1.0
1.5
051015
FREQUENCY VARIATION (%)
VIN (V)
Oscillator Frequency
vs. Supply Voltage
100
120
140
160
180
200
220
240
260
-60 -30 0 30 60 90 120 150
I
CS
(µA)
TEMPERATURE (°C)
Typical Characteristics
VIN = 5V
PWM Mode Supply Current
2.9
2.7
2.5
2.3
2.1
1.9
1.7
(mA)
1.5
DD
I
1.3
1.1
0.9
0.7
0.5
vs. Temperature
-40 -20 0 20 40 60 80 100120140
TEMPERATURE (°C)
VFB vs. Temperature
0.806
0.804
0.802
0.800
(V)
FB
0.798
V
0.796
0.794
0.792
-60 -30 0 30 60 90 120 150
TEMPERATURE (°C)
PWM Mode Supply Current
vs. Supply Voltage
2.0
1.5
1.0
QUIESCENT CURRENT (mA)
0.5
0 5 10 15
SUPPLY VOLTAGE (V)
VDD Line Regulation
6
5
4
(V)
3
DD
V
2
1
0
0 5 10 15
VIN (V)
VDD Line Regulation
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
LINE REGULATION (%)
1.0
DD
0.5
V
0.0
-60 -30 0 30 60 90 120 150
Current Limit Foldback
4
3
(V)
2
OUT
November 2003 5 M9999-111803
V
1
Top MOSFET = Si4800
RCS= 1kΩ
0
0246810
vs. Temperature
TEMPERATURE (°C)
I
(A)
LOAD
Oscillator Frequency
vs. Temperature
Overcurrent Trip Point
vs. Temperature
MIC2169 Micrel
Functional Diagram
C
IN
5V
Bandgap
Reference
0.8V
V
IN
5V LDO
5V
Soft-Start &
Digital Delay
Counter
Enable
Error
Loop
Error
Amp
V
DD
5V
0.8V
BG Valid
Clamp &
Startup
Current
COMP
RCS
CS
Current Limit
Reference
Ramp
Clock
V
REF
V
REF
PWM
Comparator
+3%
3%
Current Limit
Comparator
Hys
Comparator
Driver
Logic
GND
High-Side
Driver
5V
Low-Side
Driver
V
DD
MIC2169
HSD
BOOST
SW
LSD
FB
4Ω
RSW
R2
D1
Q1
C
BST
L1
Q2
R3
V
OUT
C
OUT
C1
C2
R1
MIC2169 Block Diagram
Functional Description
The MIC2169 is a voltage mode, synchronous step-down
switching regulator controller designed for high power without the use of an external sense resistor. It includes an
internal soft-start function which reduces the power supply
input surge current at start-up by controlling the output
voltage rise time, a PWM generator, a reference voltage, two
MOSFET drivers, and short-circuit current limiting circuitry to
form a complete 500kHz switching regulator.
Theory of Operation
The MIC2169 is a voltage mode step-down regulator. The
figure above illustrates the block diagram for the voltage
control loop. The output voltage variation due to load or line
changes will be sensed by the inverting input of the
transconductance error amplifier via the feedback resistors
R3, and R2 and compared to a reference voltage at the noninverting input. This will cause a small change in the DC
voltage level at the output of the error amplifier which is the
input to the PWM comparator. The other input to the comparator is a 0 to 1V triangular waveform. The comparator
generates a rectangular waveform whose width tON is equal
to the time from the start of the clock cycle t0 until t1, the time
the triangle crosses the output waveform of the error amplifier. To illustrate the control loop, let us assume the output
voltage drops due to sudden load turn-on, this would cause
the inverting input of the error amplifier which is divided down
version of V
to be slightly less than the reference voltage
OUT
causing the output voltage of the error amplifier to go high.
This will cause the PWM comparator to increase tON time of
the top side MOSFET, causing the output voltage to go up
and bringing V
back in regulation.
OUT
Soft-Start
The COMP/EN pin on the MIC2169 is used for the following
three functions:
1. Disables the part by grounding this pin
2. External compensation to stabilize the voltage
control loop
3. Soft-start
For better understanding of the soft-start feature, let’s assume V
= 12V, and the MIC2169 is allowed to power-up by
IN
un-grounding the COMP/EN pin. The COMP pin has an
internal 6.5µA current source that charges the external compensation capacitor. As soon as this voltage rises to 180mV
(t = Cap_COMP × 0.18V/8.5µA), the MIC2169 allows the
internal VDD linear regulator to power up and as soon as it
crosses the undervoltage lockout of 2.6V, the chip’s internal
oscillator starts switching. At this point in time, the COMP pin
current source increases to 40µA and an internal 11-bit
counter starts counting which takes approximately 2ms to
complete. During counting, the COMP voltage is clamped at
M9999-111803 6 November 2003
MIC2169 Micrel
V
V – V
VF L
OUT
IN
OUT
IN
SWITCHING
×
()
××
0.65V. After this counting cycle the COMP current source is
reduced to 8.5µA and the COMP pin voltage rises from 0.65V
to 0.95V, the bottom edge of the saw-tooth oscillator. This is
the beginning of 0% duty cycle and it increases slowly
causing the output voltage to rise slowly. The MIC2169 has
two hysteretic comparators that are enabled when V
OUT
is
within ±3% of steady state. When the output voltage reaches
97% of programmed output voltage then the gm error amplifier is enabled along with the hysteretic comparator. This
point onwards, the voltage control loop (g
error amplifier) is
m
fully in control and will regulate the output voltage.
Soft-start time can be calculated approximately by adding the
following four time frames:
t1 = Cap_COMP × 0.18V/8.5µA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5µA
t4
=
V
OUT
V
IN
××
0.5
Cap_COMP
µ
8.5 A
Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 +
t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
Current Limit
The MIC2169 uses the R
of the top power MOSFET to
DS(ON)
measure output current. Since it uses the drain to source
resistance of the power MOSFET, it is not very accurate. This
scheme is adequate to protect the power supply and external
components during a fault condition by cutting back the time
the top MOSFET is on if the feedback voltage is greater than
0.67V. In case of a hard short when feedback voltage is less
than 0.67V, the MIC2169 discharges the COMP capacitor to
0.65V, resets the digital counter and automatically shuts off
the top gate drive, and the gm error amplifier and the –3%
hysteretic comparators are completely disabled and the softstart cycles restarts. This mode of operation is called the
“hiccup mode” and its purpose is to protect the down stream
load in case of a hard short. The circuit in Figure 1 illustrates
the MIC2169 current limiting circuit.
V
IN
C2
C
IN
0
RCS
CS
200µA
HSD
LSD
Q1
MOSFET N
L1 Inductor
Q2
MOSFET N
C1
C
OUT
V
OUT
Figure 1. The MIC2169 Current Limiting Circuit
The current limiting resistor R
is calculated by the following
CS
equation:
RI
R
II
L
DS(ON) Q1 L
=
CS
=+
LOAD
×
200 A
µ
2 Inductor Ripple Current
()
Equation (1)
1
where:
Inductor Ripple Current =
F
SWITCHING
= 500kHz
200µA is the internal sink current to program the MIC2169
current limit.
The MOSFET R
varies 30% to 40% with temperature;
DS(ON)
therefore, it is recommended to add a 50% margin to the load
current (I
) in the above equation to avoid false current
LOAD
limiting due to increased MOSFET junction temperature rise.
It is also recommended to connect R
resistor directly to the
CS
drain of the top MOSFET Q1, and the RSW resistor to the
source of Q1 to accurately sense the MOSFETs R
DS(ON)
. A
0.1µF capacitor in parallel with RCS should be connected to
filter some of the switching noise.
Internal VDD Supply
The MIC2169 controller internally generates VDD for self
biasing and to provide power to the gate drives. This V
DD
supply is generated through a low-dropout regulator and
generates 5V from VIN supply greater than 5V. For supply
voltage less than 5V, the VDD linear regulator is approximately 200mV in dropout. Therefore, it is recommended to
short the VDD supply to the input supply through a 10Ω
resistor for input supplies between 2.9V to 5V.
MOSFET Gate Drive
The MIC2169 high-side drive circuit is designed to switch an
N-Channel MOSFET. The block diagram in Figure 2 shows
a bootstrap circuit, consisting of D2 and CBST, supplies
energy to the high-side drive circuit. Capacitor CBST is
charged while the low-side MOSFET is on and the voltage on
the VSW pin is approximately 0V. When the high-side
MOSFET driver is turned on, energy from CBST is used to
turn the MOSFET on. As the MOSFET turns on, the voltage
on the VSW pin increases to approximately VIN. Diode D2 is
reversed biased and CBST floats high while continuing to
keep the high-side MOSFET on. When the low-side switch is
turned back on, CBST is recharged through D2. The drive
voltage is derived from the internal 5V VDD bias supply. The
nominal low-side gate drive voltage is 5V and the nominal
high-side gate drive voltage is approximately 4.5V due the
voltage drop across D2. An approximate 20ns delay between
the high- and low-side driver transitions is used to prevent
current from simultaneously flowing unimpeded through both
MOSFETs.
MOSFET Selection
The MIC2169 controller works from input voltages of 3V to
13.2V and has an internal 5V regulator to provide power to
turn the external N-Channel power MOSFETs for high- and
November 2003 7 M9999-111803
MIC2169 Micrel
PP P
AC AC(off) AC(on)
=+
low-side switches. For applications where V
< 5V, the
IN
internal VDD regulator operates in dropout mode, and it is
necessary that the power MOSFETs used are sub-logic level
and are in full conduction mode for VGS of 2.5V. For applications when V
> 5V; logic-level MOSFETs, whose operation
IN
is specified at VGS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in junction temperature will increase the channel resistance of the
MOSFET by 50% to 75% of the resistance specified at 25°C.
This change in resistance must be accounted for when
calculating MOSFET power dissipation and in calculating the
value of current-sense (CS) resistor. Total gate charge is the
charge required to turn the MOSFET on and off under
specified operating conditions (V
and VGS). The gate
DS
charge is supplied by the MIC2169 gate-drive circuit. At
500kHz switching frequency and above, the gate charge can
be a significant source of power dissipation in the MIC2169.
At low output load, this power dissipation is noticeable as a
reduction in efficiency. The average current required to drive
the high-side MOSFET is:
IQf
G[high-side](avg) G S
=×
where:
I
G[high-side](avg)
= average high-side MOSFET gate
current.
QG = total gate charge for the high-side MOSFET taken from
manufacturer’s data sheet for VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching loss for the low-side MOSFET is usually
negligible. Also, the gate-drive current for the low-side
MOSFET is more accurately calculated using CISS at VDS =
0 instead of gate charge.
For the low-side MOSFET:
ICVf
G[low-side](avg) ISS GS S
=××
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2169 due to gate
drive is:
PVI I
GATEDRIVE IN G[high-side](avg) G[low-side](avg)
=+
()
A convenient figure of merit for switching MOSFETs is the on
resistance times the total gate charge R
DS(ON)
× QG. Lower
numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2169.
Parameters that are important to MOSFET switch selection
are:
• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
should be added to the VDS(max) of the MOSFETs to account
for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the sum of
the conduction losses during the on-time (P
CONDUCTION
) and
the switching losses that occur during the period of time when
the MOSFETs turn on and off (PAC).
PP P
=+
SW CONDUCTION AC
where:
PIR
CONDUCTION
=×
SW(rms)
2
SW
RSW = on-resistance of the MOSFET switch
V
D duty cycle
=
O
V
IN
Making the assumption the turn-on and turn-off transition
times are equal; the transition times can be approximated by:
CVC V
×+ ×
ISS GS OSS IN
t
=
T
I
G
where:
C
ISS
and C
are measured at VDS = 0
OSS
IG = gate-drive current (1A for the MIC2169)
The total high-side MOSFET switching loss is:
P(VV)Itf
=+×××
AC IN D PK T S
where:
tT = switching transition time (typically 20ns to 50ns)
VD = freewheeling diode drop, typically 0.5V
fS it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple
current. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.
V (V max V )
×−
L
=
OUT IN OUT
V max f 0.2 I max
() ()
IN S OUT
()
×× ×
where:
fS = switching frequency, 500kHz
0.2 = ratio of AC ripple current to DC output current
VIN(max) = maximum input voltage
M9999-111803 8 November 2003
MIC2169 Micrel
The peak-to-peak inductor current (AC ripple current) is:
V (V max V )
×−
I
OUT IN OUT
=
PP
()
V max f L
IN S
××
()
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor ripple
current.
I I max 0.5 I
=+×()
PK OUT PP
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
2
P
I I max 1
INDUCTOR(rms) OUT
=×+
()
1
3II max
OUT
()
Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2169 requires the use of ferrite
materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply. This
is especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the core
and copper losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
currents, the core losses can be a significant contributor.
Core loss information is usually available from the magnetics
vendor. Copper loss in the inductor is calculated by the
equation below:
PI R
INDUCTORCu
The resistance of the copper wire, R
=×
INDUCTOR(rms)
2
WINDING
WINDING
, increases with
temperature. The value of the winding resistance used should
be at the operating temperature.
R R 1 0.0042 (T T )
WINDING(hot) WINDING(20 C) HOT 20 C
=×+×−
()
°°
where:
T
= temperature of the wire under operating load
HOT
T
= ambient temperature
20°C
R
WINDING(20°C)
is room temperature winding resistance (usu-
ally specified by the manufacturer)
Output Capacitor Selection
The output capacitor values are usually determined capacitors ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors selecting
the output capacitor. Recommended capacitors tantalum,
low-ESR aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output ripple.
The output capacitor ESR also affects the overall voltage
feedback loop from stability point of view. See
Loop Compensation”
section for more information. The
“Feedback
maximum value of ESR is calculated:
V
∆
R
ESR
OUT
≤
I
PP
where:
V
= peak-to-peak output voltage ripple
OUT
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ESR output
capacitance. The total ripple is calculated below:
∆V
OUT
I(1D)
×−
PP
=
Cf
OUT S
×
2
IR
+×
()
PP ESR
2
where:
D = duty cycle
C
= output capacitance value
OUT
fS = switching frequency
The voltage rating of capacitor should be twice the voltage for
a tantalum and 20% greater for an aluminum electrolytic.
The output capacitor RMS current is calculated below:
I
I
C
OUT(rms)
PP
=
12
The power dissipated in the output capacitor is:
PIR
DISS(C C ESR(C )
=×
OUT
)
OUT(rms)
2
OUT
Input Capacitor Selection
The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may fail
when subjected to high inrush currents, caused by turning the
input supply on. Tantalum input capacitor voltage rating
should be at least 2 times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage derating. The input voltage
ripple will primarily depend on the input capacitor’s ESR. The
peak input current is equal to the peak inductor current, so:
∆VI R
=×
IN INDUCTOR(peak) ESR(C )
IN
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at the
maximum output current. Assuming the peak-to-peak inductor ripple current is low:
I I max D (1 D)
≈××−()
C (rms) OUT
IN
The power dissipated in the input capacitor is:
PIR
DISS(C )
=×
C (rms)
IN
2
IN
ESR(C )
IN
November 2003 9 M9999-111803
MIC2169 Micrel
Voltage Setting Components
The MIC2169 requires two resistors to set the output voltage
as shown in Figure 2.
Error
Amp
MIC2169 [adj.]
V
REF
0.8V
FB
7
R1
R2
Figure 2. Voltage-Divider Configuration
Where:
V
for the MIC2169 is typically 0.8V
REF
The output voltage is determined by the equation:
VV 1
=×+
O REF
R1
R2
A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is
too large, it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small, in value, it will decrease the
efficiency of the power supply, especially at light loads. Once
R1 is selected, R2 can be calculated using:
VR1
×
R2
REF
=
VV
−
OREF
External Schottky Diode
An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 15ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must be
able to handle the peak current.
I I 2 80ns f
=×××
D(avg) OUT S
The reverse voltage requirement of the diode is:
VV
DIODE(rrm)
=
IN
The power dissipated by the Schottky diode is:
PIV
=×
DIODE D(avg) F
where:
VF = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode is
used, it must be rated to handle the peak and average current.
The body diode has a relatively slow reverse recovery time
and a relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of the
diode. As the high-side MOSFET starts to turn on, the body
diode becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts at a
lower forward voltage preventing the body diode in the
MOSFET from turning on. The lower forward voltage drop
dissipates less power than the body diode. The lack of a
reverse recovery mechanism in a Schottky diode causes less
ringing and less power loss. Depending on the circuit components and operating conditions, an external Schottky diode
will give a 1/2% to 1% improvement in efficiency.
Feedback Loop Compensation
The MIC2169 controller comes with an internal
transconductance error amplifier used for compensating the
voltage feedback loop by placing a capacitor (C1) in series
with a resistor (R1) and another capacitor C2 in parallel from
the COMP pin to ground. See
“Functional Block Diagram.”
Power Stage
The power stage of a voltage mode controller has an inductor,
L1, with its winding resistance (DCR) connected to the output
capacitor, C
, with its electrical series resistance (ESR) as
OUT
shown in Figure 3. The transfer function G(s), for such a
system is:
DCRL
ESR
C
OUT
V
O
Figure 3. The Output LC Filter in a Voltage Mode
Buck Converter
G(s)
=
××+ ××++ ××
DCR s C s L C 1 ESR s C
1 ESR s C
+××
()
2
Plotting this transfer function with the following assumed
values (L=2 µH, DCR=0.009Ω, C
=1000µF, ESR=0.050Ω)
OUT
gives lot of insight as to why one needs to compensate the
loop by adding resistor and capacitors on the COMP pin.
Figures 4 and 5 show the gain curve and phase curve for the
above transfer function.
M9999-111803 10 November 2003
MIC2169 Micrel
Error Amplifier(z) g
1R1S C1
s C1 1 R1 C2 S
m
=×
+××
×
()
+××
()
30
30
0
0
7.5
15
GAIN
37.5
60
60
100 1
3
.
1
0
4
.
1
1
0
f
5
.
1
1
0
6
.
1
1
0
1000000100
Figure 4. The Gain Curve for G(s)
0
0
50
100
PHASE
150
180
100 1
3
.
10
4
.
10
1
1.10
5
6
.
10
1
1000000100 f
Figure 5. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the gain
curve that the output inductor and capacitor create a two pole
system with a break frequency at:
f
=
LC
2LC
××π
1
OUT
Therefore, fLC = 3.6kHz
By looking at the phase curve, it can be seen that the output
capacitor ESR (0.050Ω) cancels one of the two poles (LC
OUT
system by introducing a zero at:
50
100
PHASE
150
180
100 1.10
3
1.10
4
1.10
5
Figure 6. The Phase Curve with ESR = 0.002
It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number is
–150°. This means that the transconductance error amplifier
has to provide a phase boost of about 45° to achieve a closed
loop phase margin of 45° at a crossover frequency of 50kHz
for Figure 4, versus 105° for Figure 6. The simple RC and C2
compensation scheme allows a maximum error amplifier
phase boost of about 90°. Therefore, it is easier to stabilize
the MIC2169 voltage control loop by using high ESR value
output capacitors.
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would be
picked up and transmitted at large amplitude to the output,
thus, gain should be permitted to fall off at high frequencies.
At low frequency, it is desired to have high open-loop gain to
attenuate the power line ripple. Thus, the error amplifier gain
should be allowed to increase rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal g
error amplifier can be approximated by the following equation:
Error Amplifier(z) g
)
=×
m
sC1C21R1
×+
1R1S C1
+××
()
C1 C2 S
+×
C1 C2
××
6
1.10
1000000100 f
ΩΩ
Ω
ΩΩ
m
+
f
ZERO
Therefore, F
From the point of view of compensating the voltage loop, it is
recommended to use higher ESR output capacitors since
they provide a 90° phase gain in the power path. For comparison purposes, Figure 6, shows the same phase curve with an
ESR value of 0.002Ω.
November 2003 11 M9999-111803
=
2 ESR C
×× ×π
ZERO
1
= 6.36kHz.
The above equation can be simplified by assuming C2<<C1,
OUT
From the above transfer function, one can see that R1 and C1
introduce a zero and R1 and C2 a pole at the following
frequencies:
Fzero= 1/2 π × R1 × C1
Fpole = 1/2 π × C2 × R1
Fpole@origin = 1/2 π × C1
MIC2169 Micrel
Figures 7 and 8 show the gain and phase curves for the above
transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF,
and gm = .005Ω–1. It can be seen that at 50kHz, the error
amplifier exhibits approximately 45° of phase margin.
60
60
40
20
ERROR AMPLIFIER GAIN
.001
1.10
3
4
.
10
1
5
.
10
1
1.10
6
100000001000 f
7
.
10
1
Figure 7. Error Amplifier Gain Curve
200
215.856
100
71.607
50
0
OPEN LOOP GAIN MARGIN
42.933
50
250
269.097
300
100 1.10
3
1.10
4
1.10
Figure 9. Open-Loop Gain Margin
5
1.10
6
1000000100 f
220
240
ERROR AMPLIFIER PHASE
260
270
10 100 1
.
1
031.1041.10
5
6
.
1
1
0
100000010 f
Figure 8. Error Amplifier Phase Curve
Total Open-Loop Response
The open-loop response for the MIC2169 controller is easily
obtained by adding the power path and the error amplifier
gains together, since they already are in Log scale. It is
desirable to have the gain curve intersect zero dB at tens of
kilohertz, this is commonly called crossover frequency; the
phase margin at crossover frequency should be at least 45°.
Phase margins of 30° or less cause the power supply to have
substantial ringing when subjected to transients, and have
little tolerance for component or environmental variations.
Figures 9 and 10 show the open-loop gain and phase margin.
It can be seen from Figure 9 that the gain curve intersects the
0dB at approximately 50kHz, and from Figure 10 that at
50kHz, the phase shows approximately 50° of margin.
OPEN LOOP PHASE MARGIN
350
360
10 100 1
3
.
1
0
4
.
1
1
0
Figure 10. Open-Loop Phase Margin
1.10
5
6
.
1
1
0
100000010 f
M9999-111803 12 November 2003
MIC2169 Micrel
Design Example
Layout and Checklist:
1. Connect the current limiting (CS) resistor directly
to the drain of top MOSFET Q1.
2. Connect the VSW pin directly to the source of top
MOSFET Q1 thru a 4Ω to 10Ω resistor. The purpose of the resistor is to filter the switch node.
3. The feedback resistors R1 and R2 should be
placed close to the FB pin. The top side of R1
should connect directly to the output node. Run
this trace away from the switch node (junction of
Q1, Q2, and L1). The bottom side of R1 should
connect to the GND pin on the MIC2169.
4. The compensation resistor and capacitors should
be placed right next to the COMP/EN pin and the
other side should connect directly to the GND pin
on the MIC2169 rather than going to the plane.
5. The input bulk capacitors should be placed close to
the drain of the top MOSFET.
6. The 1µF ceramic capacitor should be placed right
on the VIN pin of the MIC2169.
7. The 4.7µF to 10µF ceramic capacitor should be
placed right on the VDD pin.
8. The source of the bottom MOSFET should connect
directly to the input capacitor GND with a thick
trace. The output capacitor and the input capacitor
should connect directly to the GND plane.
9. Place a 0.1µF ceramic capacitor in parallel with the
CS resistor to filter any switching noise.
November 2003 13 M9999-111803
MIC2169 Micrel
Package Information
10-Pin MSOP (MM)
Rev. 00
MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131 USA
The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
M9999-111803 14 November 2003
TEL + 1 (408) 944-0800 FAX + 1 (408) 944-0970 WEB http://www.micrel.com
Micrel for any damages resulting from such use or sale.
© 2003 Micrel, Incorporated.
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