Maxim MAX767TCAP, MAX767RCAP, MAX767REAP, MAX767SCAP, MAX767SEAP Datasheet

19-0224; Rev 2; 8/94

Evaluation Kit Manual

Follows Data Sheet

5V-to-3.3V, Synchronous, Step-Down

_______________General Description

The MAX767 is a high-efficiency, synchronous buck
controller IC dedicated to converting a fixed 5V supply
into a tightly regulated 3.3V output. Two key features set
this device apart from similar, low-voltage step-down
switching regulators: high operating frequency and all
N-channel construction in the application circuit. The
300kHz operating frequency results in very small, low­cost external surface-mount components.

The inductor, at 3.3µH for 5A, is physically at least five
times smaller than inductors found in competing solu­tions. All N-channel construction and synchronous rectifi­cation result in reduced cost and highest efficiency.
Efficiency exceeds 90% over a wide range of loading,
eliminating the need for heatsinking. Output capacitance
requirements are low, reducing board space and cost.

The MAX767 is a monolithic BiCMOS IC available in
20-pin SSOP packages. For other fixed output voltages
and package options, please consult the factory.

________________________Applications

Local 5V-to-3.3V DC-DC Conversion
Microprocessor Daughterboards
Power Supplies up to 10A or More

________Typical Application Circuit

Power-Supply Controller

____________________________Features

♦ >90% Efficiency
♦ 700µA Quiescent Supply Current
♦ 120µA Standby Supply Current
♦ 4.5V-to-5.5V Input Range
♦ Low-Cost Application Circuit
♦ All N-Channel Switches
♦ Small External Components
♦ Tiny Shrink-Small-Outline Package (SSOP)
♦ Predesigned Applications:

Standard 5V to 3.3V DC-DC Converters up to 10A
High-Accuracy Pentium P54C VR-Spec Supply

♦ Fixed Output Voltages Available:

3.3V (Standard)

3.45V (High-Speed Pentium™)

3.6V (PowerPC™)

______________Ordering Information

PART TEMP. RANGE

MAX767CAP 0°C to +70°C 20 SSOP
MAX767RCAP 0°C to +70°C 20 SSOP
MAX767SCAP 0°C to +70°C 20 SSOP
MAX767TCAP 0°C to +70°C 20 SSOP ±1.2% 3.3V
MAX767C/D 0°C to +70°C Dice*

Ordering Information continued at end of data sheet.

*

Contact factory for dice specifications.

PIN-

PACKAGE

REF.
TOL.

±1.8%
±1.8%
±1.8%

–

V

OUT

3.3V

3.45V

3.6V

–

MAX767

INPUT

4.5V TO 5.5V

V

ON

CC

MAX767

REF

™ Pentium is a trademark of Intel. PowerPC is a trademark of IBM.

BST

DH

LX

DL

PGND

CS

FB

GND

________________________________________________________________

3.3µH

OUTPUT

3.3V

AT 5A

__________________Pin Configuration

TOP VIEW

CS

1

SS

2

ON

3

GND
GND
GND

GND

REF

SYNC

V

CC

MAX767

4
5
6
7
8
9

10

SSOP

Maxim Integrated Products

Call toll free 1-800-998-8800 for free samples or literature.

FB

20

DH

19

LX

18

BST

17

DL

16

V

15

CC

V

14

CC

PGND

13

N.C.

12

GND

11

1

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

ABSOLUTE MAXIMUM RATINGS

VCCto GND.................................................................-0.3V, +7V

PGND to GND........................................................................±2V

BST to GND...............................................................-0.3V, +15V

LX to BST.....................................................................-7V, +0.3V

Inputs/Outputs to GND

(ON, REF, SYNC, CS, FB, SS) .....................-0.3V, V

DL to PGND .....................................................-0.3V, V

MAX767

DH to LX...........................................................-0.3V, BST + 0.3V

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.

CC
CC

+ 0.3V
+ 0.3V

ELECTRICAL CHARACTERISTICS

(VCC= ON = 5V, GND = PGND = SYNC = 0V, I

PARAMETER

VCCInput Supply Range

0mV < (CS - FB) < 80mV,

Output Voltage (FB)

Load Regulation 2.5 %(CS - FB) = 0mV to 80mV
Line Regulation
VCCFault Lockout Voltage
Current-Limit Voltage
SS Source Current
SS Fault Sink Current

Reference Voltage (REF)
VCCStandby Current

VCCQuiescent Current
Oscillator Frequency
Oscillator SYNC Range

SYNC High Pulse Width
SYNC Low Pulse Width 200 ns
SYNC Rise/Fall Time

Oscillator Maximum Duty Cycle
Input Low Voltage
Input High Voltage
Input Current ±1 µA

DL Sink/Source Current 1 A
DH Sink/Source Current
DL On Resistance
DH On Resistance

4.5V < V
(includes load and
line regulation)

VCC= 4.5V to 5.5V
Falling edge, hysteresis = 1%
CS - FB

MAX767, MAX767R, MAX767S
MAX767T
ON = 0V, VCC= 5.5V
FB = CS = 3.5V
SYNC = 3.3V
SYNC = 0V or 5V

Not tested
SYNC = 3.3V
SYNC = 0V
SYNC, ON
ON
SYNC
SYNC, ON = 0V or 5V
DL = 2V
(BST - LX) = 4.5V, DH = 2V
High or low
High or low, (BST - LX) = 4.5V

CC

REF

< 5.5V

= 0mA, TA= T

CONDITIONS

REF Short to GND.......................................................Momentary

REF Current.........................................................................20mA

Continuous Power Dissipation (TA= +70°C)

20-Pin SSOP (derate 8.00mW/°C above +70°C) ..........640mW

Operating Temperature Ranges:

MAX767CAP/MAX767_CAP.................................0°C to +70°C

MAX767EAP/MAX767_EAP ..............................-40°C to +85°C

Lead Temperature (soldering, 10sec).............................+300°C

to T

MIN

MAX767, MAX767T
MAX767R
MAX767S

, unless otherwise noted. Typical values are at TA= +25°C.)

MAX

MIN TYP MAX

4.5 5.5 V

3.17 3.35 3.46

3.32 3.50 3.60

3.46 3.65 3.75

0.1 %

3.80 4.20
80 100 120 mV

2.50 4 6.5

2 mA

3.24 3.30 3.36

3.26 3.30 3.34

120 200

0.7 1.0 mA

260 300 340

200
240 350 kHz
200

89 92

95

2.40

V

- 0.5

CC

1

UNITS

200 ns

0.8 V

7
7

V

V

µA

V

µA

kHz

ns

%

V

A

Ω
Ω

2 _______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

__________________________________________Typical Operating Characteristics

(Circuit of Figure 1 (5A configuration), VIN= 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

EFFICIENCY vs. OUTPUT CURRENT

(1.5A CIRCUIT)

100

90

80

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

EFFICIENCY vs. OUTPUT CURRENT

(7A CIRCUIT)

100

90

MAX767-01

EFFICIENCY (%)

MAX767-04

EFFICIENCY vs. OUTPUT CURRENT

(3A CIRCUIT)

100

90

80

70

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

EFFICIENCY vs. OUTPUT CURRENT

(10A CIRCUIT)

100

90

100

MAX767-02

EFFICIENCY (%)

1000

MAX767-05

100

EFFICIENCY vs. OUTPUT CURRENT

(5A CIRCUIT)

90

80

70

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

SWITCHING FREQUENCY vs.

PERCENT OF FULL LOAD

SYNC = REF (300kHz)

MAX767

MAX767-03

MAX767-06

80

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

IDLE-MODE WAVEFORMS



= 300mA

I

LOAD

5µs/div

80

70

EFFICIENCY (%)

60

50

0.001 0.1 10

0.01 1
OUTPUT CURRENT (A)

3.3V OUTPUT
50mV/div, AC COUPLED

LX
5V/div

I

LOAD

10

1

0.1

SWITCHING FREQUENCY (kHz)

0.01

0.001 1 100

PWM-MODE WAVEFORMS

= 5A

1µs/div

0.01 0.1 10

LOAD CURRENT (% FULL LOAD)

3.3V OUTPUT
50mV/div, AC COUPLED

LX
5V/div



_______________________________________________________________________________________

3

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

____________________________Typical Operating Characteristics (continued)

(Circuit of Figure 1 (5A configuration), VIN= 5V, oscillator frequency = 300kHz, TA= +25°C, unless otherwise noted.)

1.5A CIRCUIT LOAD-TRANSIENT RESPONSE

MAX767

5A CIRCUIT LOAD-TRANSIENT RESPONSE

200µs/div

1.5A
LOAD CURRENT

0A

3.3V OUTPUT

50mV/div
AC-COUPLED

5A

LOAD CURRENT

0A

3.3V OUTPUT

50mV/div
AC-COUPLED

3A CIRCUIT LOAD-TRANSIENT RESPONSE

3A

LOAD CURRENT

0A

3.3V OUTPUT
50mV/div
AC-COUPLED

200µs/div

7A CIRCUIT LOAD-TRANSIENT RESPONSE

7A

LOAD CURRENT

0A

3.3V OUTPUT
50mV/div
AC-COUPLED

200µs/div

200µs/div

10A CIRCUIT LOAD-TRANSIENT RESPONSE

10A

LOAD CURRENT

0A

3.3V OUTPUT
50mV/div
AC-COUPLED

200µs/div

4 _______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

______________________________________________________________Pin Description

PIN

1

2 SS Soft-start input. Ramp time to full current limit is 1ms/nF of capacitance to GND.
3 ON

4–7, 11 GND Low-current analog ground. Feedback reference point for the output.

8 REF 3.3V internal reference output. Bypass to GND with 0.22µF minimum capacitor.

9 SYNC

10, 14, 15 V

12 N.C. No internal connection
13 PGND Power ground
16
17 BST Boost capacitor connection (0.1µF)
18 LX Inductor connection. Can swing 2V below GND without latchup.
19 DH Gate-drive output for the high-side MOSFET
20 FB Feedback and current-sense input for the PWM

NAME FUNCTION

CS Current-sense input: +100mV = nominal current-limit level referred to FB.

ON/O—F—F–control input to disable the PWM. Tie directly to VCCfor automatic start-up.

Oscillator control/synchronization input. Connect to VCCor GND for 200kHz; connect to REF for
300kHz. For external clock synchronization in the 240kHz to 350kHz range, a high-to-low transition
causes a new cycle to start.

CC

Supply voltage input: 4.5V to 5.5V

DL Gate-drive output for the low-side synchronous rectifier MOSFET

MAX767

SHUTDOWN

ON/OFF

C5

(OPTIONAL)

C6

0.22µF

Figure 1. Standard Application Circuit

0.01µF

4.7µF

C4

ON

SS

SYNC

REF

V

CC

MAX767

GND

R2

10Ω

BST

PGND

INPUT

4.5V TO 5.5V

D1
SMALL-
SIGNAL 
SCHOTTKY

DH

LX

DL

CS

FB

C3

N1

0.1µF

D2

N2

C1

OUTPUT

L1

R1

3.3V

C2

_______________________________________________________________________________________ 5

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

_____Standard Application Circuits

This data sheet shows five predesigned circuits with
output current capabilities from 1.5A to 10A. Many
users will find one of these standard circuits appropri­ate for their needs. If a standard circuit is used, the
remainder of this data sheet (

Applications Information and Design Procedure

MAX767

be bypassed.
Figure 1 shows the Standard Application Circuit. Table 1

gives component values and part numbers for five dif­ferent implementations of this circuit: 1.5A, 3A, 5A, 7A,
and 10A output currents.

Each of these circuits is designed to deliver the full
rated output load current over the temperature range
listed. In addition, each will withstand a short circuit of
several seconds duration from the output to ground. If
the circuit must withstand a continuous short circuit,
refer to the
required changes.

Good layout is necessary to achieve the designed out­put power, high efficiency, and low noise. Good layout
includes the use of a ground plane, appropriate com­ponent placement, and correct routing of traces using
appropriate trace widths. The following points are in
order of decreasing importance.

1. A ground plane is essential for optimum perfor­mance. In most applications, the circuit will be
located on a multilayer board and full use of the four
or more copper layers is recommended. Use the
top and bottom layers for interconnections and the
inner layers for an uninterrupted ground plane.

2. Because the sense resistance values are similar to
a few centimeters of narrow traces on a printed cir­cuit board, trace resistance can contribute signifi­cant errors. To prevent this, Kelvin connect CS and
FB to the sense resistor; i.e., use separate traces
not carrying any of the inductor or load current, as
shown in Figure 2. These signals must be carefully
shielded from DH, DL, BST, and the LX node.

Important: place the sense resistor as close as pos­sible to and no further than 10mm from the MAX767.

3. Place the LX node components N1, N2, L1, and D2
as close together as possible. This reduces resis­tive and switching losses and confines noise due to
ground inductance.

4. The input filter capacitor C1 should be less than
10mm away from N1’s drain. The connecting cop­per trace carries large currents and must be at least
2mm wide, preferably 5mm.

Short-Circuit Duration

Detailed Description

section for the

Layout and Grounding

and

) can

5. Keep the gate connections to the MOSFETs short
for low inductance (less than 20mm long and more
than 0.5mm wide) to ensure clean switching.

6. To achieve good shielding, it is best to keep all
switching signals (MOSFET gate drives DH and DL,
BST, and the LX node) on one side of the board
and all sensitive nodes (CS, FB, and REF) on the
other side.

7. Connect the GND and PGND pins directly to the
ground plane, which should ideally be an inner
layer of a multilayer board.

_______________Detailed Description

Note:

The remainder of this document contains the
detailed information necessary to design a circuit that
differs substantially from the five standard application
circuits. If you are using one of the predesigned stan­dard circuits, the following sections are provided only
for your reading pleasure.

The MAX767 converts a 4.5V to 5.5V input to a 3.3V
output. Its load capability depends on external compo­nents and can exceed 10A. The 3.3V output is generat­ed by a current-mode, pulse-width-modulation (PWM)
step-down regulator. The PWM regulator operates at
either 200kHz or 300kHz, with a corresponding trade­off between somewhat higher efficiency (200kHz) and
smaller external component size (300kHz). The
MAX767 also has a 3.3V, 5mA reference voltage. Fault­protection circuitry shuts off the output should the refer­ence lose regulation or the input voltage go below 4V
(nominally).

External components for the MAX767 include two N­channel MOSFETs, a rectifier, and an LC output filter.
The gate-drive signal for the high-side MOSFET, which
must exceed the input voltage, is provided by a boost
circuit that uses a 0.1µF capacitor. The synchronous
rectifier keeps efficiency high by clamping the voltage
across the rectifier diode. An external low-value cur­rent-sense resistor sets the maximum current limit, pre­venting excessive inductor current during start-up or
under short-circuit conditions. An optional external
capacitor sets the programmable soft-start, reducing
in-rush surge currents upon start-up and providing
adjustable power-up time.

The PWM regulator is a direct-summing type, lacking a
traditional integrator-type error amplifier and the phase
shift associated with it. It therefore does not require
external feedback-compensation components, as long
as you follow the ESR guidelines in the

Information and Design Procedure

sections.

Applications

6 _______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

Table 1. Component Values

Part 1.5A Circuit 3A Circuit 5A Circuit 7A Circuit 10A Circuit

L1

R1

N1,

N2

C1

C2

D2

Temp.
Range

10µH
Sumida CDR74B-100

0.04Ω
IRC LR2010-01-R040
or DD WSL-2512-R040

International
Rectifier IRF7101,
Siliconix Si9936DY
or Motorola
MMDF3N03HD
(dual N-channel)

47µF, 20V
AVX TPSD476K020R

220µF, 6.3V
Sprague
595D227X06R3D2B

1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3

to +85°C to +85°C to +85°C to +85°C to +85°C

5µH
Sumida CDR125
DRG# 4722-JPS-001

0.02Ω
IRC LR2010-01-R020
or DD WSL-2512-R020

Siliconix Si9410DY,
International
Rectifier IRF7101
or Motorola
MMDF3N03HD
(both FETs in parallel)

2 x 47µF, 20V
AVX TPSD476K020R

2 x 150µF, 10V
Sprague
595D157X0010D7T

1N5817
Nihon EC10QS02,
or Motorola
MBRS120T3

3.3µH
CoilCraft
DO3316-332

0.012Ω
DD WSL-2512-R012
or 2 x 0.025Ω
IRC LR2010-01-R025
(in parallel)

Motorola
MTD20N03HDL

220µF, 10V
Sanyo
OS-CON 10SA220M

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

2.1µH, 5mΩ
Coiltronics
CTX03-12338-1

3 x 0.025Ω
IRC LR2010-01-R025
or DD WSL-2512-R025
(in parallel)

Motorola
MTD75N03HDL
(N1)
MTD20N03HDL
(N2)

2 x 100µF, 10V
Sanyo
OS-CON 10SA100M

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

1.5µH, 3.5mΩ
Coiltronics
CTX03-12357-1

3 x 0.020Ω
IRC LR2010-01-R020
or 2 x 0.012Ω
DD WSL-2512-R012
(in parallel)

Motorola
MTD75N03HDL

2 x 220µF, 10V
Sanyo
OS-CON 10SA220M

4 x 220µF, 10V
Sanyo
OS-CON 10SA220M

1N5820
Nihon NSQ03A02,
or Motorola
MBRS340T3

MAX767

Table 2. Component Suppliers

Company Factory Fax [Country Code] USA Telephone

AVX [1] [phone] [phone]
CoilCraft [1] [phone] [phone]
Coiltronics [1] [phone] [phone]
DD [1] [phone] [phone]
IRC [1] [phone] [phone]
International

Rectifier
Motorola [1] [phone] (602) 244- 3576
Nihon [81] 3-3494-7414 [phone]
Sanyo [81] 7-2070-1174 [phone]
Siliconix [1] [phone] [phone]
Sprague [1] [phone] [phone]
Sumida [81] 3-3607-5144 [phone]

[1] [phone] [phone]

_______________________________________________________________________________________ 7

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

because the minimum-current comparator immediately
resets the high-side latch at the beginning of each

FAT, HIGH-CURRENT TRACES

MAIN CURRENT PATH

MAX767

MAX767

Figure 2. Kelvin Connections for the Current-Sense Resistor

The main gain block is an open-loop comparator that
sums four signals: output voltage error signal, current­sense signal, slope-compensation ramp, and the 3.3V
reference. This direct-summing method approaches
the ideal of cycle-by-cycle control of the output voltage.
Under heavy loads, the controller operates in full PWM
mode. Every pulse from the oscillator sets the output
latch and turns on the high-side switch for a period
determined by the duty factor (approximately
V

/ VIN).

OUT

As the high-side switch turns off, the synchronous recti­fier latch is set; 60ns later, the low-side switch turns on.
The low-side switch stays on until the beginning of the
next clock cycle (in continuous-conduction mode) or
until the inductor current reaches zero (in discontinu­ous-conduction mode). Under fault conditions where
the inductor current exceeds the 100mV current-limit
threshold, the high-side latch resets and the high-side
switch turns off.

At light loads, the inductor current fails to exceed the
25mV threshold set by the minimum-current compara­tor. When this occurs, the PWM goes into Idle-Mode™,
skipping most of the oscillator pulses to reduce the
switching frequency and cut back switching losses.
The oscillator is effectively gated off at light loads

SENSE RESISTOR

cycle, unless the FB signal falls below the reference
voltage level.

Connecting a capacitor from the soft-start pin (SS) to
ground allows a gradual build-up of the 3.3V output
after power is applied or ON is driven high. When ON is
low, the soft-start capacitor is discharged to GND.
When ON is driven high, a 4µA constant current source
charges the capacitor up to 4V. The resulting ramp volt­age on SS linearly increases the current-limit compara­tor set-point, increasing the duty cycle to the external
power MOSFETs. With no soft-start capacitor, the full
output current is available within 10µs (see

Information and Design Procedure

Synchronous rectification allows for high efficiency by
reducing the losses associated with the Schottky rectifi­er. Also, the synchronous-rectifier MOSFET is neces­sary for correct operation of the MAX767’s boost gate­drive supply.

When the external power MOSFET (N1) turns off, ener­gy stored in the inductor causes its terminal voltage to
reverse instantly. Current flows in the loop formed by
the inductor (L1), Schottky diode (D2), and the load—
an action that charges up the output filter capacitor
(C2). The Schottky diode has a forward voltage of
about 0.5V which, although small, represents a signifi­cant power loss and degrades efficiency. The synchro­nous-rectifier MOSFET parallels the diode and is turned
on by DL shortly after the diode conducts. Since the
synchronous rectifier’s on resistance (r
low, the losses are reduced. The synchronous-rectifier
MOSFET is turned off when the inductor current falls to
zero.

The MAX767’s internal break-before-make timing
ensures that shoot-through (both external switches
turned on at the same time) does not occur. The
Schottky rectifier conducts during the time that neither
MOSFET is on, which improves efficiency by preventing
the synchronous-rectifier MOSFET’s lossy body diode
from conducting.

The synchronous rectifier works under all operating
conditions, including discontinuous-conduction mode
and idle-mode.

Soft-Start

Applications

section).

Synchronous Rectifier

) is very

DS(ON)

™ Idle-Mode is a trademark of Maxim Integrated Products.

8 _______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

CS

1X

MAX767

2.8V

MAIN PWM
COMPARATOR

MINIMUM
CURRENT

(IDLE-MODE)

3.3V

N

60kHz

LPF

30R

R

Q

S

V

CC



CURRENT

LIMIT

0mV TO 100mV

200kHz/300kHz

OSCILLATOR



1R

ON

LEVEL
SHIFT

SHOOT-
THROUGH
CONTROL

REFV

CC

+3.3V

REFERENCE

SLOPE COMP

4V

25mV

4µA

FAULT

SS

ON

FB

BST

DH

LX

MAX767

Figure 3. MAX767 Block Diagram

_______________________________________________________________________________________ 9

SYNCHRONOUS

RECTIFIER CONTROL

V

R

Q

S

SYNC

LEVEL
SHIFT

CC

DL

PGND

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

Under heavy loads—over approximately 25% of full
load—the supply operates as a continuous-current
PWM supply (see
The duty cycle, %ON, is approximately:

V

%ON = ________

OUT

V

IN

Current flows continuously in the inductor: first, it ramps
up when the power MOSFET conducts; second, it
ramps down during the flyback portion of each cycle as
energy is put into the inductor and then discharged into
the load. Note that the current flowing into the inductor
when it is being charged is also flowing into the load,
so the load is continuously receiving current from the
inductor. This minimizes output ripple and maximizes
inductor use, allowing very small physical and electrical
sizes. Output ripple is primarily a function of the filter
capacitor’s effective series resistance (ESR), and is
typically under 50mV (see

Under light loads (<25% of full load), the MAX767
enhances efficiency by turning the drive voltage on and
off for only a single clock period, skipping most of the
clock pulses entirely. Asynchronous switching, seen as
“ghosting” on an oscilloscope, is thus a normal operat­ing condition whenever the load current is less than
approximately 25% of full load.

At certain input voltage and load conditions, a transition
region exists where the controller can pass back and
forth from idle-mode to PWM mode. In this situation,
short pulse bursts occur, which make the current wave­form look erratic but do not materially affect the output
ripple. Efficiency remains high.

The voltage between CS and FB is continuously moni­tored. An external, low-value shunt resistor is connect­ed between these pins, in series with the inductor,
allowing the inductor current to be continuously mea­sured throughout the switching cycle. Whenever this
voltage exceeds 100mV, the drive voltage to the exter­nal high-side MOSFET is cut off. This protects the MOS­FET, the load, and the input supply in case of short cir­cuits or temporary load surges. The current-limiting
resistance is typically 20mΩ for 3A.

MAX767

V

MAX767

Figure 4. Boost Supply for High-Side Gate Driver

CC

LEVEL

TRANSLATOR

PWM

BST

DH

LX

V

CC

DL

D1

C3

N1

L1

N2

Gate-Driver Boost Supply

Gate-drive voltage for the high-side N-channel switch is
generated with the flying-capacitor boost circuit shown
in Figure 4. The capacitor (C3) is alternately charged
from the 5V input via the diode (D1) and placed in par­allel with the high-side MOSFET’s gate-source termi­nals. On start-up, the synchronous rectifier (low-side)
MOSFET (N2) forces LX to 0V and charges the BST
capacitor to 5V. On the second half-cycle, the PWM
turns on the high-side MOSFET (N1); it does this by
closing an internal switch between BST and DH, which
connects the capacitor to the MOSFET gate. This pro­vides the necessary enhancement voltage to turn on
the high-side switch, an action that “boosts” the 5V
gate-drive signal above the input voltage.

Ringing seen at the high-side MOSFET gates (DH) in
discontinuous-conduction mode (light loads) is a natur­al operating condition. It is caused by the residual
energy in the tank circuit, formed by the inductor and
stray capacitance at the LX node. The gate-driver neg­ative rail is referred to LX, so any ringing there is direct­ly coupled to the gate-drive supply.

V

IN

C1

Modes of Operation

PWM Mode

Typical Operating Characteristics

Design Procedure

section).

Idle-Mode

Current Limiting

).

10 ______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

Oscillator Frequency

The SYNC input controls the oscillator frequency.
Connecting SYNC to GND or to VCCselects 200kHz
operation; connecting it to REF selects 300kHz opera­tion. SYNC can also be driven with an external 240kHz
to 350kHz CMOS/TTL source to synchronize the inter­nal oscillator. Normally, 300kHz operation is chosen to
minimize the inductor and output filter capacitor sizes,
but 200kHz operation may be chosen for a small (about
1%) increase in efficiency at heavy loads.

Internal Reference

The internal 3.3V bandgap reference (REF) remains
active, even when the switching regulator is turned off.
It can furnish up to 5mA, and can be used to supply
memory keep-alive power or for other purposes.
Bypass REF to GND with 0.22µF, plus 1µF/mA of load
current.

Applications Information and

__________________Design Procedure

Most users will be able to work with one of the standard
application circuits; others may want to implement a
circuit with an output current rating that lies between or
beyond the standard values.

If you want an output current level that lies between two
of the standard application circuits, you can interpolate
many of the component values from the values given
for the two circuits. These components include the
input and output filter capacitors, the inductor, and the
sense resistor. The capacitors must meet ESR and rip­ple current requirements (see

Output Filter Capacitor

meet the required current rating (see
You may use the rectifier and MOSFETs specified for

the circuit with the greater output current capability, or
choose a new rectifier and MOSFETs according to the
requirements detailed in the

Switches

for output currents in excess of 10A, refer to the design
information in the following sections.

Three inductor parameters are required: the inductance
value (L), the peak inductor current (I
coil resistance (RL). The inductance is:

sections. For more complete information, or

f x I

1.32

OUT

x LIR

L1 = ______________

Input Filter Capacitor

sections). The inductor must

Inductor

Rectifier

and

and

section).

MOSFET

Inductor, L1

), and the

LPEAK

where:

f = switching frequency, normally 300kHz

= maximum 3.3V DC load current (A)

I

OUT

LIR = ratio of inductor peak-to-peak AC

current to average DC load current,
typically 0.3.

A higher LIR value allows smaller inductance, but
results in higher losses and ripple.

The highest peak inductor current (I
DC load current (I
inductor current (I
current is typically chosen as 30% of the maximum DC
load current, so the peak inductor current is 1.15 x I

The peak inductor current at any load is given by:

I

The coil resistance should be as low as possible,
preferably in the low milliohms. The coil is effectively in
series with the load at all times, so the wire losses alone
are approximately:

Power Loss = I

In general, select a standard inductor that meets the L,
I

LPEAK

unavailable, choose a core with an LI2parameter
greater than L x I
will fit the core.

= I

LPEAK

, and RLrequirements. If a standard inductor is

) plus half the peak-to-peak AC

OUT

). The peak-to-peak AC inductor

LPP

1.32

+ __________

OUT

OUT

LPEAK

2 x f x L1

2

x R

L

2

, and use the largest wire that

LPEAK

) equals the

OUT

Current-Sense Resistor, R1

The current-sense resistor must carry the peak current
in the inductor, which exceeds the full DC load current.
The internal current limiting starts when the voltage
across the sense resistors exceeds 100mV nominally,
80mV minimum. Use the minimum value to ensure ade­quate output current capability: R1 = 80mV / I

The low VIN/V
start-up under full load or with load transients from no­load to full load. If the supply is subjected to these con­ditions, reduce the sense resistor:

R1 = ———

Since the sense-resistance values are similar to a few
centimeters of narrow traces on a printed circuit board,
trace resistance can contribute significant errors. To
prevent this, Kelvin connect the CS and FB pins to the
sense resistors; i.e., use separate traces not carrying
any of the inductor or load current, as shown in Figure 2.

ratio creates a potential problem with

OUT

70mV
I

LPEAK

LPEAK

.

MAX767

.

______________________________________________________________________________________ 11

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

Place R1 as close as possible to the MAX767, prefer­ably less than 10mm. Run the traces at minimum spac­ing from one another. If they are longer than 20mm,
bypass CS to FB with a 1nF capacitor placed as close
as possible to these pins. The wiring layout for these
traces is critical for stable, low-ripple outputs (see

Layout and Grounding

MAX767

Use at least 6µF per watt of output power for C1. If the
5V input is some distance away or comes through a PC
bus, greater capacitance may be desirable to improve
the load-transient response. Use a low-ESR capacitor
located no further than 10mm from the MOSFET switch
(N1) to prevent ringing. The ripple current rating must
be at least I
tions, two or more capacitors in parallel may be needed
to meet these requirements.

The ESR of C1 is effectively in series with the input. The
resistive dissipation of C1, I
cantly impact the circuit’s efficiency.

RMS

section).

Input Filter Capacitor, C1

= 0.5 x I

. For high-current applica-

OUT

2

x ESRC1, can signifi-

RMS

Output Filter Capacitor, C2

The output filter capacitor determines the loop stability,
output voltage ripple, and output load-transient
response.

To ensure stability, stay above the minimum capaci­tance value and below the maximum ESR value. These
values are:

C2 > —— µF

and

ESRC2< R1

Be sure to satisfy both these requirements. To achieve
the low ESR required, it may be appropriate to parallel
two or more capacitors and/or use a total capacitance
2 or 3 times larger than the calculated minimum.

3Ω
R1

Output Ripple

The output ripple in continuous-conduction mode is:

V

OUT(RPL)

where f is the switching frequency (200kHz or 300kHz).

= I

OUT

(ESR

(max) x LIR x

+ ———————)

C2

1

2 x π x f x C2

Stability

In idle-mode, the ripple has a capacitive and a resistive
component:

.

V

OUT(RPL)

V

OUT(RPL)

The total ripple, V
follows:

if

V

OUT(RPL)

then

V

OUT(RPL)

otherwise

V

OUT(RPL)

(C) = _____________ x 0.89 Volts

(R) = _____________

(R) < 0.5 V

= V

= 0.5 V

0.0004 x L
R12x C2

0.02 x ESR

OUT(RPL)

OUT(RPL)

OUT(RPL)

V

OUT(RPL)

C2

R1

, can be approximated as

OUT(RPL)

(C)

(R)

(C)

(C) +

Load-Transient Performance

In response to a large step increase in load current, the
output voltage will sag for several microseconds unless
C2 is increased beyond the values that satisfy the
above requirements. Note that an increase in capaci­tance is all that’s required to improve the transient
response, and that the ESR requirements don’t change.
Therefore, the added capacitance can be supplied by
an additional low-cost bulk capacitor in parallel with the
normal low-ESR switching-regulator capacitor. The
equation for voltage sag under a step load change is:

2

I

x L

V

= ________________________________

SAG

2 x C2 x (VIN(min) x DMAX - 3.3V)

where DMAX is the maximum duty cycle. Higher duty
cycles are possible when the oscillator frequency is
reduced to 200kHz, since fixed propagation delays
through the PWM comparator become a lesser part of
the whole period. The tested worst-case limit for DMAX
is 92% at 200kHz or 89% at 300kHz. Lower inductance
values can reduce the filter capacitance requirement,
but only at the expense of increased output ripple (due
to higher peak currents).

STEP

12 ______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

RC Filter for V

R2 and C4 form a lowpass filter to remove switching
noise from the VCCinput to the MAX767. C4 must have
fairly low ESR (<5Ω). Switching noise can interfere with
proper output voltage regulation, resulting in an exces­sive output voltage decrease (>100mV) at full load.

Overheating during soldering can damage the surface­mount capacitors specified for C4, causing the regula­tion problems described above. Take care to heat the
capacitor for as short a time as possible, especially if it
is soldered by hand.

CC

Rectifier, D2

Use a 1N5817 or similar Schottky diode for applications
up to 3A, or a 1N5820 for up to 10A. Surface-mount
equivalents are available from N.I.E.C. with part num­bers EC10QS02 and NSQ03A02, or from Motorola with
part numbers MBRS120T3 and MBRS320T3. D2 must
be a Schottky diode to prevent the lossy MOSFET body
diode from turning on.

Soft-Start

A capacitor connected from GND to SS causes the
supply’s current-limit level to ramp up slowly. The ramp
time to full current limit is approximately 1ms for every
nF of capacitance on SS, with a minimum value of
10µs. Typical values for the soft-start capacitor are in
the 10nF to 100nF range; a 5V rating is sufficient.

The time required for the output voltage to ramp up to
its rated value depends upon the output load, and is
not necessarily the same as the time it takes for the cur­rent limit to reach full capacity.

Duty Cycle

The duty cycle for the high-side MOSFET (N1) in con­tinuous-conduction mode is:

100% x ( V

___________________

VIN- V

where:

V

= 3.3V

OUT

VIN= 5V
VN1and VN2= I

It is apparent that, in continuous-conduction mode, N1
will conduct for about twice the time as N2. Under short­circuit conditions, however, N2 can conduct as much
90% of the time. If there is a significant chance of short
circuiting the output, select N2 to handle the resulting
duty cycle (see

+ VN2)

OUT

N1

x r

LOAD

DS(ON)

Short-Circuit Duration

for each MOSFET.

section).

MOSFET Switches, N1 and N2

The two N-channel MOSFETs must be “logic-level”
FETs; that is, they must be fully on (have low
r

DS(ON)

high-current applications, FETs with low gate­threshold voltage specifications (i.e., maximum
V
tion, they should have low total gate charge (<70nC)
to minimize switching losses.

For output currents in excess of the five standard appli­cation circuits, placing MOSFETs with very low gate
charge in parallel increases output current and lowers
resistive losses. N2 does not normally require the same
current capacity as N1 because it conducts only about
33% of the time, while N1 conducts about 66% of the
time.

) with only 4V gate-source drive voltage. For

GS(TH)

= 2V rather than 3V) are preferred. In addi-

Short-Circuit Duration

At their highest rated temperatures (+70°C or +85°C),
each of the five standard application circuits will with­stand a short circuit of several seconds duration. In
most cases, the MAX767 will be used in applications
where long-term short circuiting of the output is unlikely.

If it is desirable for the circuit to withstand a continuous
short circuit, the MOSFETs must be able to dissipate
the required power. This depends on physical factors
such as the mounting of the transistor, any heat­sinking used, and ventilation provided, as well as the
actual current the transistor must deliver. The short­circuit current is approximately 100mV / R1, but may
vary by ±20%.

Cautious design requires that the transistors
withstand the maximum possible current, which is
ISC= 120mV / R1. N1 and N2 must withstand this
current scaled by their maximum duty factors. The
maximum duty factor for N1 occurs under high­load (but not short-circuit) conditions, and is approxi­mately V
imum duty factor for N2 occurs during short-circuit
conditions and is:

1 - —————————————

which can exceed 0.9. The total power dissipated in
both MOSFETs together is I

/ VIN(min) or about 0.7. The max-

OUT

ISCx r

DS(ON)N2

VIN(max) - ISCx r

DS(ON)N1

2

x r

SC

DS(ON)

.

MAX767

______________________________________________________________________________________ 13

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

Proper circuit operation requires that the short-circuit
current be at least I

x (1 + LIR / 2). However, the

LOAD

standard application circuits are designed for a short­circuit current slightly in excess of this amount. This
excess design current guarantees proper start-up
under constant full-load conditions and proper full-load
transient response, and is particularly necessary with
low input voltages. If the circuit will not be subjected to

MAX767

full-load transients or to loads approaching the full-load
at start-up, you can decrease the short-circuit current
by increasing R1, as described in the

Resistor

section. This may allow use of MOSFETs with a

Current-Sense

lower current-handling capability.

Heavy-Load Efficiency

Losses due to parasitic resistances in the switches,
coil, and sense resistor dominate at high load-current
levels. Under heavy loads, the MAX767 operates deep
in the continuous-conduction mode, where there is a
large DC offset to the inductor current (plus a small
sawtooth AC component) (see
DC current is exactly equal to the load current, a fact
which makes it easy to estimate resistive losses via the
simplifying assumption that the total inductor current is
equal to this DC offset current. The major loss mecha­nisms under heavy loads, in usual order of importance,
are:

♦ I2R losses
♦ gate-charge losses
♦ diode-conduction losses
♦ transition losses
♦ capacitor-ESR losses
♦ losses due to the operating supply current of the IC.

Inductor-core losses, which are fairly low at heavy
loads because the AC component of the inductor cur­rent is small, are not accounted for in this analysis.

P

Efficiency = ______ x 100% =

PD

TOTAL

OUT

P

IN

POUT

_______________ x 100%
P

OUT

= PD

PD

(I2R)
TRAN

+ PD

+ PD

+ PD

Inductor

TOTAL

GATE

CAP

section). This

+ PD

DIODE

+ PD

IC

+

I2R Losses

PD

= resistive loss = (I

(I2R)

(R

where R
r

DS(ON)

is the DC resistance of the coil and

COIL

is the drain-source on resistance of the MOS-

FET. Note that the r

COIL

DS(ON)

+ r

DS(ON)

term assumes that identical

LOAD

+ R1)

2

) x

MOSFETs are employed for both the synchronous recti­fier and high-side switch, because they time-share the
inductor current. If the MOSFETs are not identical, esti­mate losses by averaging the two individual r

DS(ON)

terms according to their duty factors: 0.66 for N1 and

0.34 for N2.

Gate-Charge Losses

PD

= gate driver loss = qGx f x 5V

GATE

where qGis the sum of the gate charge for low- and
high-side switches. Note that gate-charge losses are
dissipated in the IC, not the MOSFETs, and therefore
contribute to package temperature rise. For a pair of
matched MOSFETs, qGis simply twice the gate capaci­tance of a single MOSFET (a data sheet specification).

Diode Conduction Losses

PD

= diode conduction losses =

DIODE

I

x VDx tDx f

LOAD

where VDis the forward voltage of the Schottky diode
at the output current, tDis the diode’s conduction time
(typically 110ns), and f is the switching frequency.

Transition Losses

PD

where C

= transition loss =

TRAN

2

V

x C

IN

______________________

is the reverse transfer capacitance of the

RSS

RSS

I

DRIVE

x I

LOAD

x f

high-side MOSFET (a data sheet parameter), f is the
switching frequency, and I

is the peak current

DRIVE

available from the high-side gate driver output (approx­imately 1A).

Additional switching losses are introduced by other
sources of stray capacitance at the switching node,
including the catch-diode capacitance, coil interwind­ing capacitance, and low-side switch drain capaci­tance, and are given as PDSW= V
these are usually negligible compared to C

IN

2

x C

STRAY

RSS

x f, but

losses.
The low-side switch introduces only tiny switching loss­es, since its drain-source voltage is already low when it
turns on.

14 ______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

Capacitor ESR Losses

PD

where I
I

/ 2.

LOAD

Note that losses in the output filter capacitors are small
when the circuit is heavily loaded, because the current
into the capacitor is not chopped. The output capacitor
sees only the small AC sawtooth ripple current. Ensure
that the input bypass capacitor has a ripple current rat­ing that exceeds the value of I

PDICis the quiescent power dissipation of the IC and is
5V times the quiescent supply current (a data sheet
parameter), or about 5mW.

= capacitor ESR loss = I

CAP

= RMS AC input current, approximately

RMS

.

RMS

IC Supply-Current Losses

RMS

2

x ESR

Light-Load Efficiency

Under light loads, the PWM will operate in discontinu­ous-conduction mode, where the inductor current dis­charges to zero at some point during each switching
cycle. New loss mechanisms, insignificant at heavy
loads, begin to become important. The basic difference
is that in discontinuous mode, the AC component of the
inductor current is large compared to the load current.
This increases losses in the core and in the output filter
capacitors. Ferrite cores are recommended over pow­dered-material types for best light-load efficiency.

At light loads, the inductor delivers triangular current
pulses rather than the nearly square waves found in
continuous-conduction mode. These pulses ramp up to
a point set by the idle-mode current comparator, which
is internally fixed at approximately 25% of the full-scale
current-limit level. This 25% threshold provides an opti­mum balance between low-current efficiency and out­put voltage noise (the efficiency curve would actually
look better with this threshold set at about 45%, but the
output noise would be too high).

____Additional Application Circuits

High-Accuracy Power Supplies

The standard application circuit’s accuracy is dominat­ed by reference voltage error (±1.8%) and load regula­tion error (-2.5%). Both of these parameters can be
improved as shown in Figures 5 and 6. Both circuits
rely on an external integrator amplifier to increase the
DC loop gain in order to reduce the load regulation
error to 0.1%. Reference error is improved in the first
circuit by employing a version of the MAX767 (“T”
grade) which has a ±1.2% reference voltage tolerance.

Reference error of the second circuit is further
improved by substituting a highly accurate external ref­erence chip (MAX872), which contributes ±0.38% total
error over temperature.

These two circuits were designed with the latest gener­ation of dynamic-clock µPs in mind, which place great
demands on the transient-response performance of the
power supply. As the µP clock starts and stops, the
load current can change by several amps in less than
100ns. This tremendous ∆i/∆t can cause output voltage
overshoot or sag that results in the CPU VCC going out
of tolerance unless the power supply is carefully
designed and located close to the CPU. These circuits
have excellent dynamic response and low ripple, with
transient excursions of less than 40mV under zero to
full-load step change. In particular, these two circuits
support the “VR” (voltage regulator) version of the Intel
P54C Pentium™ CPU, which requires that its supply
voltage, including noise and transient errors, be within
the 3.30V to 3.45V range.

To configure these circuits for a given load current
requirement, substitute standard components from
Table 1 for the power switching elements (N1, N2, L1,
C1, C2) or use the
taken from Table 1, but must be adjusted approximate­ly 10% higher in order to maintain the correct current­limit threshold. This increased value is due to the 0.9
gain factor introduced by the H-bridge resistor divider
(R3–R6).

If the remote sense line must sense the output voltage
on the far side of a connector or jumper that has the
possibility of becoming disconnected while the power
supply is operating, an additional 10kΩ resistor should
connect the sense line to the output voltage in the con­nector’s power-supply side in order to prevent acciden­tal overvoltage at the CPU.

For applications that are powered from a fixed +12V or
battery input rather than from +5V, use a MAX797 IC
instead of the MAX767. The MAX797 is capable of
accepting inputs up to 30V. See the MAX796–MAX799
data sheet for a high-accuracy circuit schematic.

Design Procedure

. R1 can also be

MAX767

______________________________________________________________________________________ 15

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

INPUT

4.75V TO 5.5V

R2

10Ω

MAX767

C4

4.7µF

SHUTDOWN

ON/OFF

V

CC

MAX767T

ON

C10

0.01µF

(OPTIONAL)

V

0.01µF

V

CC

OUT

C5

C8
620pF

MAX495

D1

DH

BST

LX

DL

PGND



CS

FB

REF

SYNC

SS

GND

N1

C3

0.1µF

N2

R8

10k

C9

0.22µF

R9

332k, 1%

C1

L1

D2

TO MAX767

= V

REF

(

R10

+ 1

R9

R3
1k,
1%

R7

330k

R5
10k,
1%

R10

8.06k, 1%

)

R4
1k,
1%

R6
10k,
1%

C2R1

C6

0.01µF

REMOTE SENSE LINE

R11

5.1k
MIN

LOAD

3.38V OUTPUT

3.427V MAX

3.330V MIN

C7

10µF CERAMIC

(LOCATE AT 

µP PINS)



Figure 5. High-Accuracy CPU Power Supply with Internal Reference

16 ______________________________________________________________________________________

INPUT

4.75V TO 5.5V

R2

20Ω

C4

22µF

SHUTDOWN

ON/OFF

0.22µF

V

ON

REF
SYNC

C10

0.01µF

(OPTIONAL)

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

MAX767

R10

= V

REF

(

R9

R3
1k,
1%

330k

R5
10k,
1%

118k, 0.1%

R7

+ 1

R10

)

3.38V OUTPUT

R11

5.1k
MIN

LOAD

3.408V MAX

3.369V MIN

C7

10µF CERAMIC

(LOCATE AT 

µP PINS)



C2R1

R4
1k,
1%

C6

0.01µF

R6
10k,
1%

REMOTE SENSE LINE

V

D1

C3

0.1µF

TO MAX767

V

CCC9

R8

10k

R9

332k, 0.1%

N1

N2

DH

GND

VIN

MAX872

GND

BST

DL

PGND

CS

VOUT

LX



FB

CC

MAX767

SS

C1

L1

D2

OUT

C5

0.01µF

C8
1000pF

MAX495

Figure 6. High-Accuracy CPU Power Supply with External Reference

______________________________________________________________________________________ 17

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

___________________Chip Topography_Ordering Information (continued)

TEMP. RANGEPART

MAX767

PIN-

PACKAGE

20 SSOP0°C to +70°CMAX767EAP
20 SSOP0°C to +70°CMAX767REAP
20 SSOP0°C to +70°CMAX767SEAP

REF.
TOL.

±1.8%
±1.8%
±1.8%

V

OUT

3.3V

3.45V

3.6V

3.3V±1.2%20 SSOP0°C to +70°CMAX767TEAP

GND
GND

GND

GND
GND

REF

SYNC

ON

SS CS FB DH

LX

BST

DL

0.181"

V

CC

(4.597mm)

V

CC

V

CC

PGND

CC

GND

0.109"

(2.769mm)

V

TRANSISTOR COUNT: 1294
SUBSTRATE CONNECTED TO GND

18 ______________________________________________________________________________________

5V-to-3.3V, Synchronous, Step-Down

Power-Supply Controller

________________________________________________________Package Information

DIM

e

HE

A

A1

B
C
D
E
e
H
L

α

D

α

A

0.127mm

0.004in.

A1

B

C

L

INCHES MILLIMETERS



MIN

0.068

0.002

0.010

0.005

0.278

0.205

0.301

0.022

MAX

0.078

0.008

0.015

0.009

0.289

0.212



0˚



0.311

0.037
8˚

20-PIN PLASTIC

SHRINK

SMALL-OUTLINE

PACKAGE

MIN

1.73

0.05

0.25

0.13

7.07

5.20

0.65 BSC0.0256 BSC



7.65

0.55
0˚

MAX

1.99

0.21

0.38

0.22

7.33

5.38


7.90

0.95

8˚

21-0003A

MAX767

______________________________________________________________________________________ 19

5V-to-3.3V, Synchronous, Step-Down
Power-Supply Controller

MAX767

20 ______________________________________________________________________________________