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General Description
The MAX1714 pulse-width modulation (PWM) controller
provides the high efficiency, excellent transient
response, and high DC output accuracy needed for
stepping down high-voltage batteries to generate lowvoltage CPU core or chip-set/RAM supplies in notebook
computers.
Maxim’s proprietary Quick-PWM™ quick-response,
constant-on-time PWM control scheme handles wide
input/output voltage ratios with ease and provides
100ns “instant-on” response to load transients while
maintaining a relatively constant switching frequency.
The MAX1714 achieves high efficiency at a reduced
cost by eliminating the current-sense resistor found in
traditional current-mode PWMs. Efficiency is further
enhanced by an ability to drive very large synchronousrectifier MOSFETs.
Single-stage buck conversion allows these devices to
directly step down high-voltage batteries for the highest
possible efficiency. Alternatively, two-stage conversion
(stepping down the +5V system supply instead of the
battery) at a higher switching frequency allows the minimum possible physical size.
The MAX1714 is intended for CPU core, chipset,
DRAM, or other low-voltage supplies as low as 1V. The
MAX1714A is available in a 20-pin QSOP package and
includes overvoltage protection. The MAX1714B is
available in a 16-pin QSOP package with no overvoltage protection. For applications requiring VID compliance or DAC control of output voltage, refer to the
MAX1710/MAX1711 data sheet. For a dual output version, refer to the MAX1715†data sheet.
Applications
Notebook Computers
CPU Core Supply
Chipset/RAM Supply as Low as 1V
1.8V and 2.5V I/O Supply
Features
♦ Ultra-High Efficiency
♦ No Current-Sense Resistor (Lossless I
LIMIT
)
♦ Quick-PWM with 100ns Load-Step Response
♦ 1% V
OUT
Accuracy Over Line and Load
♦ 2.5V/3.3V Fixed or 1V to 5.5V Adjustable Output
Range
♦ 2V to 28V Battery Input Range
♦ 200/300/450/600kHz Switching Frequency
♦ Overvoltage Protection (MAX1714A)
♦ Undervoltage Protection
♦ 1.7ms Digital Soft-Start
♦ Drives Large Synchronous-Rectifier FETs
♦ 2V ±1% Reference Output
♦ Power-Good Indicator
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
________________________________________________________________ Maxim Integrated Products 1
19-1536; Rev 1; 12/99
Pin Configurations appear at end of data sheet.
Quick-PWM is a trademark of Maxim Integrated Products.
†
Future product—contact factory for availability.
-40°C to +85°C
PART
MAX1714AEEP
TEMP. RANGE PIN-PACKAGE
20 QSOP
Ordering Information
EVALUATION KIT MANUALS
FOLLOW DATA SHEET
V
CC
+5V INPUT
BATTERY
4.5V TO 28V
OUTPUT
1.25V TO 2V
SHDN
ILIM
DL
LX
V+
BST
DH
PGND
(GND)
OUT
SKIP
V
DD
MAX1714
REF
PGOOD
FB
AGND
(GND)
( ) ARE FOR THE MAX1714B ONLY.
Minimal Operating Circuit
MAX1714BEEE -40°C to +85°C 16 QSOP
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
V+ to AGND (Note 1)..............................................-0.3V to +30V
V
DD
, VCCto AGND (Note 1).....................................-0.3V to +6V
PGND to AGND (Note 1) ................................................... ±0.3V
SHDN, PGOOD, OUT to AGND (Note 1)..................-0.3V to +6V
ILIM, FB, REF, SKIP,
TON to AGND (Notes 1, 2)....................-0.3V to (V
CC
+ 0.3V)
DL to PGND (Note 1)..................................-0.3V to (V
DD
+ 0.3V)
BST to AGND (Note 1) ...........................................-0.3V to +36V
DH to LX.....................................................-0.3V to (BST + 0.3V)
LX to BST..................................................................-6V to +0.3V
REF Short Circuit to AGND.........................................Continuous
Continuous Power Dissipation (T
A
= +70°C)
16-Pin QSOP (derate 8.3mW/°C above +70°C)..........667mW
20-Pin QSOP (derate 9.1mW/°C above +70°C)..........727mW
Operating Temperature Range ..........................-40°C to +85°C
Junction Temperature......................................................+150°C
Storage Temperature Range ............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
V+ = 4.5V to 28V,
SKIP = V
CC
V
SHDN
= 0 , V+ = 28V, VCC= VDD= 0 or 5V
V
SHDN
= 0
V
CC,VDD
V
SHDN
= 0
Battery voltage, V+
FB forced above the regulation point
FB forced above the regulation point
Rising edge of SHDN to full I
LIM
(Note 4)
V+ = 24V,
V
OUT
= 2V
(Note 4)
I
LOAD
= 0 to 3A, SKIP = V
CC
VCC= 4.5V to 5.5V, V+ = 4.5V to 28V
CONDITIONS
µA<1 5Shutdown Supply Current (V+)
µA<1 5Shutdown Supply Current (VDD)
µA<1 5Shutdown Supply Current (VCC)
µA25 40Quiescent Supply Current (V+)
µA<1 5Quiescent Supply Current (VDD)
µA550 750Quiescent Supply Current (VCC)
ns400 500Minimum Off-Time
380 425 470
260 290 320
175 200 225
V
0.99 1.0 1.01
Error Comparator Threshold
(DC Output Voltage Accuracy)
(Note 3)
TON = REF (450kHz)
4.5 5.5
V
228
Input Voltage Range
TON = AGND (600kHz)
ns
140 160 180
On-Time
ms1.7Soft-Start Ramp Time
µA-0.1 0.1FB Input Bias Current
TON = unconnected (300kHz)
mV9Load Regulation Error
mV5Line Regulation Error
UNITSMIN TYP MAXPARAMETER
Falling edge, hysteresis = 40mV
REF in regulation
I
REF
= 0 to 50µA
VCC= 4.5V to 5.5V, no external REF load
V1.6REF Fault Lockout Voltage
µA10REF Sink Current
V0.01Reference Load Regulation
V1.98 2 2.02Reference Voltage
TON = VCC(200kHz)
Note 1: For the MAX1714B, AGND and PGND refer to a single pin designated GND.
Note 2: SKIP may be forced below -0.3V, temporarily exceeding the absolute maximum rating, disabling over/undervoltage fault
detection for the purpose of debugging prototypes (Figure 6). Limit the current drawn to 5mA maximum.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, 4A components from Table 1, V+ = +15V, VCC= VDD= +5V, SKIP = AGND, TA= 0°C to +85°C, unless otherwise
noted.) (Note 1)
FB = AGND kΩ100 190 300OUT Input Resistance
FB = OUT
2.475 2.5 2.525FB = AGND
FB = V
CC
3.267 3.3 3.333
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, 4A components from Table 1, V+ = +15V, VCC= VDD= +5V, SKIP = AGND, TA= 0°C to +85°C, unless otherwise
noted.) (Note 1)
CONDITIONS UNITSMIN TYP MAXPARAMETER
PGND - LX
PGND - LX, I
LIM
= V
CC
From SHDN signal going high
mV
40 50 60
Current-Limit Threshold
(Positive Direction, Adjustable)
mV90 100 110
Current-Limit Threshold
(Positive Direction, Fixed)
ms10 30
Output Undervoltage Protection
Blanking Time
%65 70 75
Output Undervoltage Protection
Threshold
PGND - LX, SKIP = VCC, TA= +25°C,
with respect to positive current-limit threshold
%-90 -120 -140
Current-Limit Threshold
(Negative Direction)
V
ILIM
= 0.5V
V
ILIM
= 2.0V 170 200 230
Rising edge, hysteresis = 20mV,
PWM disabled below this level
V4.1 4.4
VCCUndervoltage Lockout
Threshold
BST - LX forced to 5V Ω1.5 5DH Gate-Driver On-Resistance
DL, high state Ω1.5 5
DL Gate-Driver On-Resistance
(Pull-Up)
DL, low state Ω0.5 1.7
DL Gate-Driver On-Resistance
(Pull-Down)
DH forced to 2.5V, BST - LX forced to 5V A1
DH Gate-Driver Source/Sink
Current
DL forced to 2.5V A1DL Gate-Driver Source Current
DL forced to 2.5V A3DL Gate-Driver Sink Current
FB forced 2% above trip threshold (MAX1714A only) µs1.5
Overvoltage Fault Propagation
Delay
%10.5 12.5 14.5Overvoltage Trip Threshold
FB forced 2% below PGOOD trip threshold, falling edge µs1.5PGOOD Propagation Delay
PGND - LX, SKIP = AGND
mV3
Current-Limit Threshold
(Zero Crossing)
I
SINK
= 1mA V0.4PGOOD Output Low Voltage
High state, forced to 5.5V µA1PGOOD Leakage Current
Hysteresis = 10°C °C150Thermal Shutdown Threshold
With respect to error comparator threshold (MAX1714A only)
With respect to error comparator threshold
DL rising
ns
35
Dead Time
DH rising 26
mA
SKIP Input Current Logic
Threshold
To disable overvoltage and undervoltage fault detection,
T
A
= +25°C
-1.5 -0.1
%PGOOD Trip Threshold
Measured at FB with respect to error comparator
threshold, falling edge
-8 -6 -4
VLogic Input High Voltage
SHDN, SKIP
2.4
VLogic Input Low Voltage
SHDN, SKIP
0.8
µALogic Input Current
SHDN, SKIP
-1 1
nAILIM Input Current
SHDN, SKIP
±
10
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
4 _______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, 4A components from Table 1, V+ = 15V, V
CC
= VDD= +5V, SKIP = AGND, TA= -40°C to +85°C, unless other-
wise noted.) (Notes 1, 5)
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, 4A components from Table 1, V+ = +15V, VCC= VDD= +5V, SKIP = AGND, TA= 0°C to +85°C, unless otherwise
noted.) (Note 1)
CONDITIONS
VVCC- 0.4TON VCCLevel
V3.15 3.85TON Float Voltage
V1.65 2.35TON Reference Level
V0.5TON AGND Level
Forced to AGND or V
CC
µA-3 3TON Input Current
UNITSMIN TYP MAXPARAMETER
TON = VCC(200kHz)
Reference Voltage 1.98 2.02 VVCC= 4.5V to 5.5V, no external REF load
PARAMETER MIN TYP MAX UNITS
TON = unconnected (300kHz)
On-Time
140 180
ns
TON = AGND (600kHz)
Input Voltage Range
228
V
4.5 5.5
TON = REF (450kHz)
Error Comparator Threshold
(DC Output Voltage Accuracy)
(Note 3)
0.985 1.015
V
175 225
With respect to error comparator threshold
(MAX1714A only)
260 320
380 470
Minimum Off-Time
FB = AGND
500 ns
Quiescent Supply Current (VCC)
FB = OUT
750 µA
3.25 3.35
2.462 2.538
Logic Input Current -1 1 µA
SHDN, SKIP
Logic Input Low Voltage 0.8 V
SHDN, SKIP
Logic Input High Voltage
FB = V
DD
V
2.4 V
SHDN, SKIP
VCCUndervoltage Lockout
Threshold
4.1 4.4
Rising edge, hysteresis = 20mV, PWM disabled below
this level
Output Undervoltage
Protection Threshold
65 75
CONDITIONS
%
160 240
With respect to error comparator threshold
V
ILIM
= 2.0V
V
ILIM
= 0.5V
Current-Limit Threshold
(Positive Direction, Adjustable)
35 65
V+ = 24V,
V
OUT
= 2V
(Note 4)
(Note 4)
mV
Overvoltage Trip Threshold
FB forced above the regulation point
PGND - LX
Current-Limit Threshold (Positive
Direction, Fixed)
Battery voltage, V+
85 115
V
CC,VDD
mVPGND - LX, ILIM = V
CC
V+ = 4.5V to 28V,
SKIP = V
CC
10 15 %
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
_______________________________________________________________________________________ 5
Note 1: For the MAX1714B, AGND and PGND refer to a single pin designated GND.
Note 2: SKIP may be forced below -0.3V, temporarily exceeding the absolute maximum rating, disabling over/undervoltage fault
detection for the purpose of debugging prototypes (Figure 6). Limit the current drawn to 5mA maximum.
Note 3: When the inductor is in continuous conduction, the output voltage will have a DC regulation level higher than the error-
comparator threshold by 50% of the ripple. In discontinuous conduction (SKIP = AGND, light-loaded), the output voltage
will have a DC regulation level higher than the trip level by approximately 1.5% due to slope compensation.
Note 4: On-time and off-time specifications are measured from 50% point to 50% point at the DH pin with LX = PGND, V
BST
= 5V,
and a 250pF capacitor connected from DH to LX. Actual in-circuit times may differ due to MOSFET switching speeds.
Note 5: Specifications to -40°C are guaranteed by design, not production tested.
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1, components from Table 1, VIN= +15V, SKIP = AGND, TON = unconnected, TA= +25°C, unless otherwise noted.)
CONDITIONS
Measured at FB with respect to error comparator
threshold, falling edge
%-8 -4PGOOD Trip Threshold
I
SINK
= 1mA V0.4PGOOD Output Low Voltage
High state, forced to 5.5V µA1PGOOD Leakage Current
UNITMIN TYP MAXPARAMETER
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, 4A components from Table 1, V+ = +15V, V
CC
= VDD= +5V, SKIP = AGND, TA= -40°C to +85°C, unless other-
wise noted.) (Notes 1, 5)
EFFICIENCY vs. LOAD CURRENT
(4A COMPONENTS, V
100
VIN = 7V
90
= 2.5V, 300kHz)
OUT
MAX1714B-01
EFFICIENCY vs. LOAD CURRENT
(8A COMPONENTS, V
100
90
= 1.6V, 300kHz)
OUT
VIN = 7V
MAX1714B-02
EFFICIENCY vs. LOAD CURRENT
(1.5A COMPONENTS, V
TON = GND, 600kHz)
100
VIN = 5V
90
OUT
= 2.5V,
MAX1714B-03
80
EFFICIENCY (%)
70
60
0.01 1010.1
VIN = 12V
LOAD CURRENT (A)
VIN = 20V
80
EFFICIENCY (%)
70
60
0.01 1010.1
VIN = 12V
LOAD CURRENT (A)
VIN = 20V
80
EFFICIENCY (%)
70
60
0.01 1010.1
LOAD CURRENT (A)
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
6 _______________________________________________________________________________________
_____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1, components from Table 1, VIN= +15V, SKIP = AGND, TON = unconnected, TA= +25°C, unless otherwise noted.)
0
2
1
3
4
5
6
7
8
9
0105 15202530
INDUCTOR CURRENT PEAKS AND VALLEYS
vs. INPUT VOLTAGE (4A COMPONENTS,
AT CURRENT-LIMIT TRIP POINT)
MAX1714B-09
INPUT VOLTAGE (V)
INDUCTOR CURRENT (A)
I
PEAK
I
VALLEY
0
200
400
600
800
0 10203051525
NO-LOAD SUPPLY CURRENT
(4A COMPONENTS, SKIP MODE, 300kHz)
MAX1714B-10
INPUT VOLTAGE (V)
SUPPLY CURRENT (µA)
I
CC
I
IN
I
DD
10
8
0
0 5 20 30
NO-LOAD SUPPLY CURRENT vs.
INPUT VOLTAGE
(4A COMPONENTS, PWM MODE, 300kHz)
4
6
2
MAX1714B-11
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
10 15 25
I
CC
I
DD
I
IN
0
200
100
400
300
500
600
4.0 5.04.5 5.5 6.0
NO-LOAD SUPPLY CURRENT vs. INPUT
VOLTAGE (1.5A COMPONENTS,
SKIP MODE, V
OUT
= 2.5V, 600kHz)
MAX1714B-12
VCC, VDD, VIN INPUT VOLTAGE (V)
SUPPLY CURRENT (µA)
I
CC
I
DD
I
IN
0
12
10
2
4
8
6
I
OUT
AT CURRENT LIMIT vs.
TEMPERATURE
(4A COMPONENTS, V
OUT =
2.5V)
MAX1714B-07
TEMPERATURE (°C)
OUTPUT CURRENT (A)
-40
20 40-20 0 60 80
V
ILIM
= 1V
V
ILIM
= 0.5V
0
200
100
300
400
500
600
700
0105 15202530
CONTINUOUS TO DISCONTINUOUS INDUCTOR
CURRENT POINT vs. INPUT VOLTAGE
(4A COMPONENTS, V
OUT
= 2.5V)
MAX1714B-08
INPUT VOLTAGE (V)
LOAD CURRENT (mA)
300
0
0.01 1010.1
FREQUENCY vs. LOAD CURRENT
(4A COMPONENTS, V
OUT
= 2.5V)
100
50
250
200
350
150
MAX1714B-04
LOAD CURRENT (A)
FREQUENCY (kHz)
VIN = 7V, 15V, PWM MODE
VIN = 15V
SKIP MODE
VIN = 7V
SKIP MODE
280
290
300
310
320
FREQUENCY vs. INPUT VOLTAGE
(4A COMPONENTS, V
OUT
= 2.5V, I
OUT
= 1A)
MAX1714B-05
INPUT VOLTAGE (V)
FREQUENCY (kHz)
0
10 15520
3025
TEMPERATURE (°C)
FREQUENCY (kHz)
300
310
305
320
315
330
325
FREQUENCY vs. TEMPERATURE
(4A COMPONENTS, V
OUT
= 2.5V)
MAX1714B-06
-40
20 40-20 0 60 80
I
OUT
= 4A
I
OUT
= 1A
C
B
A
A = V
OUT
, AC-COUPLED, 100mV/div
B = INDUCTOR CURRENT, 2A/div
C = DL, 10V/div
10µs/div
LOAD-TRANSIENT RESPONSE
(4A COMPONENTS, V
OUT
= 2.5V, 300kHz)
MAX1714B-13
C
B
A
A = V
OUT
, AC-COUPLED, 100mV/div
B = INDUCTOR CURRENT, 5A/div
C = DL, 5V/div
10µs/div
LOAD-TRANSIENT RESPONSE
(8A COMPONENTS, V
OUT
= 1.6V, 300kHz)
MAX1714B-14
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
_______________________________________________________________________________________ 7
_____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1, components from Table 1, VIN= +15V, SKIP = AGND, TON = unconnected, TA= +25°C, unless otherwise noted.)
C
B
A
A = V
OUT
, 1V/div
B = INDUCTOR CURRENT, 5A/div
C = DL, 5V/div
500µs/div
START-UP WAVEFORM
(4A COMPONENTS, I
OUT
= 4A, ACTIVE LOAD,
V
OUT
= 2.5V, 300kHz)
MAX1714B-16
C
B
A
A = V
OUT
, 1V/div
B = INDUCTOR CURRENT, 5A/div
C = DL, 5V/div
200µs/div
OUTPUT OVERLOAD WAVEFORM
(4A COMPONENTS, V
OUT
= 2.5V, 300kHz)
MAX1714B-17
OUTPUT
UNDERVOLTAGE
PROTECTION
THRESHOLD
C
B
A
A = V
OUT
, 1V/div
B = INDUCTOR CURRENT, 5A/div
C = DL, 5V/div
50µs/div
SHUTDOWN WAVEFORM
(4A COMPONENTS, V
OUT
= 2.5V, 300kHz)
MAX1714B-18
C
B
A
A = V
OUT
, AC-COUPLED, 100mV/div
B = INDUCTOR CURRENT, 1A/div
C = DL, 5V/div
5µs/div
LOAD-TRANSIENT RESPONSE
(1.5A COMPONENTS, V
IN
= 5V,
V
OUT
= 2.5V, 600kHz)
MAX1714B-15
PIN
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
8 _______________________________________________________________________________________
Pin Description
Analog and Power Ground. AGND and PGND connect together internally.GND–
No Connection. These pins are not connected to any internal circuitry. Connect N.C. pins
to the ground plane to enhance thermal conductivity.
N.C.
2, 9,
11
Power-Good Open-Drain Output. PGOOD is low when the output voltage is more than 6%
below the normal regulation point or during soft-start. PGOOD is high impedance when
the output is in regulation and the soft-start circuit has terminated.
PGOOD10
High-Side Gate Driver Output. Swings from LX to BST.DH1
Current-Limit Threshold Adjustment. Connect ILIM to VCCfor 100mV current-limit threshold.
For an adjustable threshold, connect an external voltage source to ILIM, or use a two-resistor divider from REF to AGND. The external adjustment range of 0.5V to 2.0V corresponds to
a current-limit threshold of 50mV to 200mV.
ILIM6
+2.0V Reference Voltage Output. Bypass to AGND with 0.22µF (minimum) capacitor. Can
supply 50µA for external loads.
REF7
Analog Ground.AGND8
Shutdown Control Input. Drive SHDN to AGND to force the MAX1714 into shutdown. Drive
or connect to V
CC
for normal operation. A rising edge on SHDN clears the fault latch.
SHDN
3
Feedback Input. Connect to AGND for a +2.5V fixed output or to VCCfor a +3.3V fixed
output, or connect FB to a resistor divider from OUT for an adjustable output.
FB4
PIN
Output Voltage Connection. Connect directly to the junction of the external and output filter capacitors. OUT senses the output voltage to determine the on-time and also serves
as the feedback input in fixed-output modes.
OUT5
FUNCTIONNAME
Low-Side Gate-Driver Output. Swings from PGND to VDD. DL13
Supply Input for the DL Gate Drive. Connect to the system supply voltage, +4.5V to +5.5V.
Bypass to PGND with a 1µF (min) ceramic capacitor.
V
DD
14
Analog-Supply Input. Connect to the system supply voltage, +4.5V to +5.5V, with a series
20Ω resistor. Bypass to AGND with a 1µF (min) ceramic capacitor.
V
CC
15
On-Time Selection-Control Input. This is a four-level input used to determine DH on-time.
Connect to AGND, REF, or VCC, or leave TON unconnected to set the following switching
frequencies: AGND = 600kHz, REF = 450kHz, floating = 300kHz, and VCC= 200kHz.
TON16
Battery Voltage Sense Connection. Connect to input power source. V+ is used only to set
the PWM one-shot timing.
V+17
Pulse-Skipping Control Input. Connect to VCCfor low-noise forced-PWM mode. Connect
to AGND to enable pulse-skipping operation.
SKIP
18
Boost Flying-Capacitor Connection. Connect to an external capacitor and diode according
to the Standard Application Circuit (Figure 1). See MOSFET Gate Drivers (DH, DL) section.
BST19
External Inductor Connection. Connect LX to the switched side of the inductor. LX serves
as the lower supply rail for the DH high-side gate driver. LX is also the positive input to the
current-limit comparator.
LX20
8
–
7
1
5
6
–
2
3
4
9
10
11
12
13
14
15
16
Power Ground. Connect directly to the low-side MOSFET’s source. Serves as the negative
input of the current-limit comparator.
PGND12 –
MAX1714A MAX1714B
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
_______________________________________________________________________________________ 9
Figure 1. Standard Application Circuit
Standard Application Circuit
The standard application circuit (Figure 1) generates a
low-voltage rail for general-purpose use in a notebook
computer (I/O supply, fixed CPU core supply, DRAM
supply). This DC-DC converter steps down a battery or
AC adapter voltage to voltages from 1.0V to 5.5V with
high efficiency and accuracy.
See Table 1 for a list of component selections for common applications. Table 2 lists component manufacturers.
Detailed Description
The MAX1714 buck controller is targeted for low-voltage
power supplies for notebook computers. Maxim‘s proprietary Quick-PWM pulse-width modulator in the MAX1714
is specifically designed for handling fast load steps
while maintaining a relatively constant operating frequency and inductor operating point over a wide range
of input voltages. The Quick-PWM architecture circumvents the poor load-transient timing problems of fixedfrequency current-mode PWMs while also avoiding the
problems caused by widely varying switching frequencies in conventional constant-on-time and constant-offtime PWM schemes.
V
IN
4.5V TO 28V
+5V
BIAS SUPPLY
ON/OFF
CONTROL
LOW-NOISE
CONTROL
C5
4.7µF
V
CC
V+
SHDN
SKIP
R1
20Ω
MAX1714
V
DD
BST
DH
LX
C6
3.3µF
C1
D2
CMPSH-3
Q1
C7
0.1µF
L1
V
OUT
C2
PGND
(GND)
TON
REF
C4
0.22µF
AGND
(GND)
ILIM
+5V
( ) = ARE FOR THE MAX1714B ONLY.
NOTE: IN THE MAX 1714B, AGND AND PGND ARE INTERNALLY CONNECTED TO THE GND PIN.
PGOOD
OUT
DL
FB
+5V
R2
100k
POWER-GOOD
INDICATOR
D1
Q2
MAX1714
+5V Bias Supply (VCCand VDD)
The MAX1714 requires an external +5V bias supply in
addition to the battery. Typically, this +5V bias supply is
the notebook’s 95% efficient +5V system supply.
Keeping the bias supply external to the IC improves efficiency and eliminates the cost associated with the +5V
linear regulator that would otherwise be needed to sup-
ply the PWM circuit and gate drivers. If stand-alone
capability is needed, the +5V supply can be generated
with an external linear regulator such as the MAX1615.
The battery and +5V bias inputs can be tied together if
the input source is a fixed +4.5V to +5.5V supply. If the
+5V bias supply is powered up prior to the battery supply, the enable signal (SHDN) must be delayed until the
battery voltage is present in order to ensure startup. The
+5V bias supply must provide VCCand gate-drive
power, so the maximum current drawn is:
I
BIAS
= ICC+ f (QG1+ QG2) = 5mA to 30mA (typ)
where ICCis 600µA typical, f is the switching frequency,
and QG1and QG2are the MOSFET data sheet total
gate-charge specification limits at VGS= 5V.
Free-Running, Constant-On-Time PWM
Controller with Input Feed-Forward
The Quick-PWM control architecture is a pseudo-fixed-frequency, constant-on-time current-mode type with voltage
feed-forward (Figure 2). This architecture relies on the output filter capacitor’s ESR to act as the current-sense resistor, so the output ripple voltage provides the PWM ramp
signal. The control algorithm is simple: the high-side
switch on-time is determined solely by a one-shot whose
period is inversely proportional to input voltage and directly proportional to output voltage. Another one-shot sets a
minimum off-time (400ns typical). The on-time one-shot is
triggered if the error comparator is low, the low-side switch
current is below the current-limit threshold, and the minimum off-time one-shot has timed out.
High-Speed Step-Down Controller
for Notebook Computers
10 ______________________________________________________________________________________
Table 1. Component Selection for Standard Applications
Table 2. Component Suppliers
4.5V to 5.5V7V to 20V7V to 20V
Input Range
100µF, 10V
Sanyo POSCAP 10TPA100M
(2) 470µF 6V
Kemet T510X477108M006AS
470µF, 6V
Kemet T510X477108M006AS
C2 Output Capacitor
100µF, 10V
Sanyo POSCAP 10TPA100M
(2) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
10µF, 25V
Taiyo Yuden TMK432BJ106KM
C1 Input Capacitor
3.3µH
Coiltronics UP1B-3R3
1.5µH
Sumida CEP1251R5MC
6.8µH
Coilcraft DO3316P-682
L1 Inductor
Motorola MBR0520LT1Motorola MBRS340T3Nihon EP10QY03D2 Rectifier
International Rectifier
1/2 IRF7301
Fairchild Semiconductor
FDS6670A
Fairchild Semiconductor
1/2 FDS6982A
Q2 Low-Side
MOSFET
International Rectifier
1/2 IRF7301
International Rectifier
IRF7811
Fairchild Semiconductor
1/2 FDS6982A
Q1 High-Side
MOSFET
600kHz300kHz300kHz
Frequency
2.5V AT 1.5A1.6V AT 8A2.5V AT 4ACOMPONENT
[1] 602-994-6430602-303-5454Motorola
[1] 408-986-1442408-986-0424Kemet
[1] 408-721-1635408-822-2181Fairchild
[1] 561-241-9339561-241-7876Coiltronics
[1] 847-639-1469847-639-6400Coilcraft
[1] 516-435-1824516-435-1110Central Semiconductor
[1] 803-626-3123803-946-0690AVX
FACTORY FAX
[Country Code]
USA PHONEMANUFACTURER
[1] 714-960-6492714-969-2491Matsuo
[1] 310-322-3332310-322-3331International Rectifier
[1] 408-573-4159408-573-4150Taiyo Yuden
[1] 603-224-1430603-224-1961Sprague
[1] 408-970-3950
408-988-8000
800-554-5565
Siliconix
[81] 7-2070-1174619-661-6835Sanyo
[81] 3-3494-7414805-867-2555*NIEC (Nihon)
[1] 814-238-0490
814-237-1431
800-831-9172
Murata
[81] 3-3607-5144847-956-0666Sumida
[1] 847-390-4405847-390-4461TDK
*Distributor
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 11
On-Time One-Shot (TON)
The heart of the PWM core is the one-shot that sets the
high-side switch on-time. This fast, low-jitter, adjustable
one-shot includes circuitry that varies the on-time in
response to battery and output voltage. The high-side
switch on-time is inversely proportional to the battery
voltage as measured by the V+ input, and proportional
to the output voltage. This algorithm results in a nearly
constant switching frequency despite the lack of a fixed-
frequency clock generator. The benefits of a constant
switching frequency are twofold: first, the frequency can
be selected to avoid noise-sensitive regions such as the
455kHz IF band; second, the inductor ripple-current
operating point remains relatively constant, resulting in
easy design methodology and predictable output voltage ripple.
On-Time = K (V
OUT
+ 0.075V) / V
IN
Figure 2. MAX1714 Functional Diagram
IN
2V TO 28V
TON
SKIP
SHDN
-6%
TRIG
REF
V+
ON-TIME
COMPUTE
TON
TON
Q
1-SHOT
MAX1714A ONLY
REF
+12%
FROM
OUT
REF
ERROR
AMP
REF
-30%
Q
TOFF
1-SHOT
S
R
Q
TRIG
S
R
CURRENT
LIMIT
Q
x2
Σ
I
LIM
MAX1714
9R
R
ZERO CROSSING
(MAX1714B ONLY)
CHIP
SUPPLY
BST
DH
V
PGND
(GND)
OUT
V
+5V
LX
OUTPUT
+5V
DD
DL
+5V
CC
PGOOD
S1
S2 TIMER
Q
OVP/UVLO
LATCH
FB
NOTE: IN THE MAX1714B, AGND AND PGND ARE INTERNALLY CONNECTED TO THE GND PIN.
FEEDBACK
MUX
(SEE FIGURE 9)
2V
REF
REF
(GND)
AGND
( ) ARE FOR THE MAX1714B ONLY.
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
12 ______________________________________________________________________________________
where K is set by the TON pin-strap connection and
0.075V is an approximation to accommodate for the
expected drop across the low-side MOSFET switch.
One-shot timing error increases for the shorter on-time
settings due to fixed propagation delays; it is approximately ±12.5% at 600kHz and 450kHz, and ±10% at the
two slower settings. This translates to reduced switchingfrequency accuracy at higher frequencies (Table 5).
Switching frequency increases as a function of load current due to the increasing drop across the low-side
MOSFET, which causes a faster inductor-current discharge ramp. The on-times guaranteed in the Electrical
Characteristics are influenced by switching delays in the
external high-side power MOSFET.
Two external factors that influence switching-frequency
accuracy are resistive drops in the two conduction loops
(including inductor and PC board resistance) and the
dead-time effect. These effects are the largest contributors to the change of frequency with changing load current. The dead-time effect increases the effective
on-time, reducing the switching frequency as one or
both dead times are added to the effective on-time. It
occurs only in PWM mode (SKIP = high) when the inductor current reverses at light or negative load currents.
With reversed inductor current, the inductor’s EMF causes LX to go high earlier than normal, extending the ontime by a period equal to the low-to-high dead time.
For loads above the critical conduction point, the actual
switching frequency is:
where V
DROP1
is the sum of the parasitic voltage drops
in the inductor discharge path, including synchronous
rectifier, inductor, and PC board resistances; V
DROP2
is
the sum of the resistances in the charging path, and t
ON
is the on-time calculated by the MAX1714.
Automatic Pulse-Skipping Switchover
In skip mode (SKIP low), an inherent automatic
switchover to PFM takes place at light loads. This
switchover is effected by a comparator that truncates the
low-side switch on-time at the inductor current’s zero
crossing. This mechanism causes the threshold between
pulse-skipping PFM and nonskipping PWM operation to
coincide with the boundary between continuous and discontinuous inductor-current operation (also known as the
“critical conduction” point; see the Continuous to
Discontinuous Inductor Current Point vs. Input Voltage
graph in the Typical Operating Characteristics). In low-
duty-cycle applications, this threshold is relatively constant, with only a minor dependence on battery voltage.
where K is the on-time scale factor (Table 5). The loadcurrent level at which PFM/PWM crossover occurs,
I
LOAD(SKIP)
, is equal to 1/2 the peak-to-peak ripple current, which is a function of the inductor value (Figure 3).
For example, in the standard application circuit with
K = 3.3µs (Table 5), V
OUT
= 2.5V, VIN= 15V, and L =
6.8µH, switchover to pulse-skipping operation occurs at
I
LOAD
= 0.51A or about 1/8 full load. The crossover point
occurs at an even lower value if a swinging (soft-saturation) inductor is used.
The switching waveforms may appear noisy and asynchronous when light loading causes pulse-skipping
operation, but this is a normal operating condition that
results in high light-load efficiency. Trade-offs in PFM
noise vs. light-load efficiency are made by varying the
inductor value. Generally, low inductor values produce a
broader efficiency vs. load curve, while higher values
result in higher full-load efficiency (assuming that the coil
resistance remains fixed) and less output voltage ripple.
Penalties for using higher inductor values include larger
physical size and degraded load-transient response
(especially at low input voltage levels).
DC output accuracy specifications refer to the error-comparator threshold of the error comparator. When the
inductor is in continuous conduction, the output voltage
will have a DC regulation level higher than the trip level
by 50% of the ripple. In discontinuous conduction (SKIP
= AGND, light-loaded), the output voltage will have a DC
regulation level higher than the error-comparator threshold by approximately 1.5% due to slope compensation.
Forced-PWM Mode (
SKIP
= High)
The low-noise forced-PWM mode (SKIP = high) disables
the zero-crossing comparator, which controls the lowside switch on-time. This causes the low-side gate-drive
waveform to become the complement of the high-side
gate-drive waveform. This in turn causes the inductor
current to reverse at light loads while DH maintains a
duty factor of V
OUT/VIN
. The benefit of forced-PWM
mode is to keep the switching frequency fairly constant,
but it comes at a cost: the no-load battery current can be
10mA to 40mA, depending on the external MOSFETs.
Forced-PWM mode is most useful for reducing audiofrequency noise, improving load-transient response, providing sink-current capability for dynamic output voltage
adjustment, and improving the cross-regulation of
I
KV
2L
V-V
V
LOAD(SKIP)
OUT IN OUT
IN
≈ ⋅
f
VV
tV V
OUT DROP
ON IN DROP
=
+
+
()
1
2
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 13
multiple-output applications that use a flyback transformer or coupled inductor.
Current-Limit Circuit (ILIM)
The current-limit circuit employs a unique “valley” current-sensing algorithm that uses the on-state resistance
of the low-side MOSFET as a current-sensing element
(Figure 4). If the current-sense voltage (PGND - LX) is
above the current-limit threshold, the PWM is not allowed
to initiate a new cycle. The actual peak current is greater
than the current-limit threshold by an amount equal to
the inductor ripple current. Therefore, the exact currentlimit characteristic and maximum load capability are a
function of the MOSFET on-resistance, inductor value,
and battery voltage. The reward for this uncertainty is
robust, lossless overcurrent sensing. When combined
with the UVP protection circuit, this current-limit method
is effective in almost every circumstance.
There is also a negative current limit that prevents
excessive reverse inductor currents when V
OUT
is sinking current. The negative current-limit threshold is set to
approximately 120% of the positive current limit, and
therefore tracks the positive current limit when ILIM is
adjusted.
The current-limit threshold is adjusted with an external
resistor-divider at ILIM. A 1µA min divider current is recommended. The current-limit threshold adjustment
range is from 50mV to 200mV. In the adjustable mode,
the current-limit threshold voltage is precisely 1/10 the
voltage seen at ILIM. The threshold defaults to 100mV
when ILIM is connected to VCC. The logic threshold for
switchover to the 100mV default value is approximately
VCC- 1V.
The adjustable current limit accommodates MOSFETs
with a wide range of on-resistance characteristics (see
Design Procedure).
Carefully observe the PC board layout guidelines to
ensure that noise and DC errors don’t corrupt the current-sense signals seen by LX and PGND. Mount or
place the IC close to the low-side MOSFET with short,
direct traces, making a Kelvin sense connection to the
source and drain terminals.
MOSFET Gate Drivers (DH, DL)
The DH and DL drivers are optimized for driving moderate-sized high-side, and larger low-side power
MOSFETs. This is consistent with the low duty factor
seen in the notebook environment, where a large V
BATT
-
V
OUT
differential exists. An adaptive dead-time circuit
monitors the DL output and prevents the high-side FET
from turning on until DL is fully off. There must be a lowresistance, low-inductance path from the DL driver to the
MOSFET gate for the adaptive dead-time circuit to work
properly; otherwise, the sense circuitry in the MAX1714
will interpret the MOSFET gate as “off” while there is
actually still charge left on the gate. Use very short, wide
traces measuring no more than 20 squares (50 to 100
mils wide if the MOSFET is 1 inch from the MAX1714).
The dead time at the other edge (DH turning off) is
determined by a fixed 35ns (typical) internal delay.
The internal pull-down transistor that drives DL low is
robust, with a 0.5Ω typical on-resistance. This helps prevent DL from being pulled up during the fast rise-time of
the inductor node, due to capacitive coupling from the
drain to the gate of the low-side synchronous-rectifier
MOSFET. However, for high-current applications, you
might still encounter some combinations of high- and
low-side FETs that will cause excessive gate-drain coupling, which can lead to efficiency-killing, EMI-producing
shoot-through currents. This is often remedied by adding
a resistor in series with BST, which increases the turn-on
time of the high-side FET without degrading the turn-off
time (Figure 5).
Figure 3. Pulse-Skipping/Discontinuous Crossover Point
Figure 4. ‘‘Valley’’ Current-Limit Threshold Point
∆i
V
BATT -VOUT
=
∆t
INDUCTOR CURRENT
ON-TIME0 TIME
L
-I
PEAK
I
= I
/2
LOAD
PEAK
-I
PEAK
I
LOAD
I
LIMIT
INDUCTOR CURRENT
0 TIME
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
14 ______________________________________________________________________________________
POR, UVLO, and Soft-Start
Power-on reset (POR) occurs when VCCrises above
approximately 2V, resetting the fault latch and soft-start
counter, and preparing the PWM for operation. V
CC
undervoltage lockout (UVLO) circuitry inhibits switching
and forces the DL gate driver high (to enforce output
overvoltage protection) until VCCrises above 4.2V,
whereupon an internal digital soft-start timer begins to
ramp up the maximum allowed current limit. The ramp
occurs in five steps: 20%, 40%, 60%, 80%, and 100%;
100% current is available after 1.7ms ±50%.
A continuously adjustable analog soft-start function can
be realized by adding a capacitor in parallel with the
ILIM resistor. This soft-start method requires a minimum
interval between power-down and power-up to discharge the capacitor.
Power-Good Output (PGOOD)
The output voltage is continuously monitored for undervoltage by the PGOOD comparator. In shutdown,
standby, and soft-start, PGOOD is actively held low.
After digital soft-start has terminated, PGOOD is
released if the digital output is within 6% of the errorcomparator threshold. The PGOOD output is a true
open-drain type with no parasitic ESD diodes. Note that
the PGOOD undervoltage detector is completely independent of the output UVP fault detector.
Output Overvoltage Protection
The overvoltage protection (OVP) circuit is available in
the MAX1714A only, and is designed to protect against
a shorted high-side MOSFET by drawing high current
and blowing the battery fuse. The output voltage is continuously monitored for overvoltage. If the output is more
than 12.5% above the trip level of the error amplifier,
overvoltage protection (OVP) is triggered and the circuit
shuts down. The DL low-side gate-driver output is
then latched high until SHDN is toggled or VCCpower is
cycled below 1V. This action turns on the synchronousrectifier MOSFET with 100% duty and, in turn, rapidly
discharges the output filter capacitor and forces the output to ground. If the condition that caused the overvoltage (such as a shorted high-side MOSFET) persists, the
battery fuse will blow. DL is also kept high continuously
when VCCUVLO is active, as well as in shutdown mode
(Table 3).
Note that DL latching high causes the output voltage to
go slightly negative, due to energy stored in the output
LC tank circuit when OVP activates. If the load can’t tolerate being forced to a negative voltage, it may be desirable to place a power Schottky diode across the output
to act as a reverse-polarity clamp.
Overvoltage protection can be defeated using the nofault test mode (see No-Fault Test Mode section).
Output Undervoltage Protection
The output undervoltage protection (OVP) function is
similar to foldback current limiting, but employs a timer
rather than a variable current limit. If the MAX1714 output voltage is under 70% of the nominal value 20ms after
coming out of shutdown, the PWM is latched off and
won’t restart until VCCpower is cycled or SHDN is toggled. Under- voltage protection can be defeated using
the no-fault test mode.
No-Fault Test Mode
The over/undervoltage protection features can complicate the process of debugging prototype breadboards,
since there are (at most) a few milliseconds in which to
determine what went wrong. Therefore, a test mode is
provided to totally disable the OVP, UVP, and thermal
shutdown features, and clear the fault latch if it has been
set. The PWM operates as if SKIP were grounded
(PFM/PWM mode).
The no-fault test mode is entered by sinking 1.5mA from
SKIP through an external negative voltage source in
series with a resistor (Figure 6). SKIP is clamped to
AGND with a silicon diode, so choose a resistor value of
approximately (V
FORCE
- 0.65V) / 1.5mA.
Design Procedure
Component selection for the MAX1714 is primarily dictated by the following four criteria:
1) Input voltage range. The maximum value (V
IN(MAX)
)
must accommodate the worst-case high AC adapter
voltage. The minimum value (V
IN(MIN)
) must account
for the lowest battery voltage after drops due to connectors, fuses, and battery selector switches. If
there is a choice at all, lower input voltages result in
better efficiency.
Figure 5. Reducing the Switching-Node Rise Time
+5V
5Ω
BST
DH
LX
MAX1714
V
IN
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 15
2) Maximum load current. There are two values to con-
sider. The peak load current (I
LOAD(MAX)
) determines
the instantaneous component stresses and filtering
requirements, and thus drives output capacitor selection, inductor saturation rating, and the design of the
current-limit circuit. The continuous load current
(I
LOAD
) determines the thermal stresses and thus drives the selection of input capacitors, MOSFETs, and
other critical heat-contributing components. Modern
notebook CPUs generally exhibit:
I
LOAD
= I
LOAD(MAX)
· 80%
3) Switching frequency. This choice determines the
basic trade-off between size and efficiency. The optimal frequency is largely a function of maximum input
voltage, due to MOSFET switching losses that are
proportional to frequency and V
IN
2
. The optimum frequency is also a moving target, due to rapid improvements in MOSFET technology that are making higher
frequencies more practical (Table 4).
4) Inductor operating point. This choice provides
trade-offs between size vs. efficiency. Low inductor
values cause large ripple currents, resulting in the
smallest size, but poor efficiency and high output ripple. The minimum practical inductor value is one that
causes the circuit to operate at the edge of critical
conduction (where the inductor current just touches
zero with every cycle at maximum load). Inductor values lower than this grant no further size-reduction
benefit.
The MAX1714’s pulse-skipping algorithm initiates skip
mode at the critical conduction point. So, the inductor
operating point also determines the load-current value
at which PFM/PWM switchover occurs.
These four factors impact the component selection
process. Selecting components and calculating their
effect on the MAX1714’s operation is best done with a
spreadsheet. Using the formulas provided, calculate the
LIR (the ratio of the inductor ripple current to the
designed maximum load current) for both the minimum
and maximum input voltages. Maintaining an LIR within a
20% to 50% range is prudent. The use of a spreadsheet
allows quick evaluation of component selection.
Table 3. Operating Mode Truth Table
SHDN SKIP
DL MODE COMMENTS
0 X High Shutdown Low-power shutdown state. DL is forced to VDD, enforcing OVP. ICC< 1µA typ.
1
Below
AGND
Switching No Fault
Test mode with OVP, UVP, and thermal faults disabled and latches cleared. Otherwise
normal operation, with automatic PWM/PFM switchover for pulse skipping at light loads
(Figure 6).
1 V
CC
Switching
Run (PWM),
Low Noise
Low-noise operation with no automatic switchover. Fixed-frequency PWM action is
forced regardless of load. Inductor current reverses at light load levels. Low noise,
high I
Q
.
1 AGND Switching
Run
(PFM/PWM)
Normal operation with automatic PWM/PFM switchover for pulse skipping at light loads.
Best light-load efficiency.
1 X High Fault
Fault latch has been set by OVP, output UVLO, or thermal shutdown. Device will remain
in FAULT mode until V
CC
power is cycled, SKIP is forced below ground (Figure 6), or
SHDN is toggled.
Figure 6. Disabling Over/Undervoltage Protection
(No-Fault Test Mode)
( ) ARE FOR THE MAX1714B ONLY.
MAX1714
APPROXIMATELY
-0.65V
SKIP
1.5mA
(GND)
AGND
V
FORCE
Inductor Selection
The switching frequency and inductor operating point
determine the inductor value as follows:
Example: I
LOAD(MAX)
= 8A, V
IN =
7V, V
OUT
= 1.5V,
f = 300kHz, 33% ripple current or LIR = 0.33.
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered iron
is inexpensive and can work well at 200kHz. The core
must be large enough not to saturate at the peak inductor current (I
PEAK
).
I
PEAK
= I
LOAD(MAX)
+ [(LIR / 2) · I
LOAD(MAX)
]
Most inductor manufacturers provide inductors in standard values, such as 1.0µH, 1.5µH, 2.2µH, 3.3µH, etc.
Also look for nonstandard values, which can provide a
better compromise in LIR across the input voltage range.
For example, Sumida offers 3.1µH and 4.4µH in their
CDRH125 series. If using a swinging inductor (where the
no-load inductance decreases linearly with increasing
current), evaluate the LIR with properly scaled inductance values.
Transient Response
The inductor ripple current also impacts transientresponse performance, especially at low V
IN
- V
OUT
differentials. Low inductor values allow the inductor
current to slew faster, replenishing charge removed
from the output filter capacitors by a sudden load step.
The amount of output sag is also a function of the maximum duty factor, which can be calculated from the ontime and minimum off-time:
where
and minimum off-time = 400ns typ (see Table 5 for K values).
The amount of overshoot during a full-load to no-load
transient due to stored inductor energy can be calculated
as:
where I
PEAK
is the peak inductor current.
Setting the Current Limit
The minimum current-limit threshold must be high
enough to support the maximum load current. The valley
of the inductor current occurs at I
LOAD(MAX)
minus half
of the ripple current (Figure 4); therefore:
I
LIMIT(LOW)
> I
LOAD(MAX)
- (LIR / 2) I
LOAD(MAX)
where I
LIMIT(LOW)
equals minimum current-limit thresh-
old voltage divided by the R
DS(ON)
of Q2. For the
MAX1714, the minimum current-limit threshold using the
100mV default setting is 90mV. Use the worst-case maximum value for R
DS(ON)
from the MOSFET Q2 data sheet,
and add some margin for the rise in R
DS(ON)
with temperature. A good general rule is to allow 0.5% additional
resistance for each °C of temperature rise.
Examining the 8A circuit example with a maximum
R
DS(ON)
= 12mΩ at high temperature reveals the follow-
ing:
I
LIMIT(LOW)
= 90mV / 12mΩ = 7.5A
This 7.5A is greater than the valley current of 6.7A, so the
circuit can easily deliver the full rated 8A using the default
100mV nominal ILIM threshold.
For an adjustable threshold, connect a two-resistor
divider from REF to AGND, with ILIM connected at the
center tap. The external adjustment range of 0.5V to 2.0V
corresponds to a current-limit threshold of 50mV to
200mV. When adjusting the current limit, use 1% tolerance resistors to prevent a significant increase of errors in
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
16 ______________________________________________________________________________________
Good operating point for
compound buck designs
or desktop circuits.
+5V input
600
TON = AGND
450
TON = REF
3-cell Li+ notebook
Useful in 3-cell systems
for lighter loads than the
CPU core or where size is
key.
Considered mainstream
by current standards.
4-cell Li+ notebook
300
TON = Float
200
TON = V
CC
4-cell Li+ notebook
Use for absolute best
efficiency.
COMMENTS
TYPICAL
APPLICATION
FREQUENCY
(kHz)
Table 4. Frequency Selection Guidelines
DUTY
K (V + 0.075V) V
K (V + 0.075V) V + min off - time
OUT IN
OUT OUT
=
L =
L
V(V- V)
OUT IN OUT
⋅⋅ ⋅
V f LIR I
IN LOAD(MAX)
1.5V (7V -1.5V)
⋅⋅⋅
7V 300kHz 0.33 8A
1.49 H==
µ
V
=
SAG
(I ) L
∆
LOAD(MAX)
2 C DUTY (V - V )
⋅⋅
F IN(MIN) OUT
2
⋅
V
SOAR
LI
≈
2C V
2
⋅
PEAK
OUT OUT
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 17
the current-limit tolerance. A 1µA minimum divider current
is recommended.
Output Capacitor Selection
The output filter capacitor must have low enough effective
series resistance (ESR) to meet output ripple and loadtransient requirements, yet have high enough ESR to satisfy stability requirements. Also, the capacitance value
must be high enough to absorb the inductor energy
going from a full-load to no-load condition without tripping
the overvoltage protection circuit.
In CPU V
CORE
converters and other applications where
the output is subject to violent load transients, the output
capacitor’s size depends on how much ESR is needed to
prevent the output from dipping too low under a load
transient. Ignoring the sag due to finite capacitance:
In non-CPU applications, the output capacitor’s size
depends on how much ESR is needed to maintain an
acceptable level of output voltage ripple:
The actual microfarad capacitance value required relates
to the physical size needed to achieve low ESR, as well
as to the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value (this is true of tantalums,
OS-CONs, and other electrolytics).
When using low-capacity filter capacitors such as ceramic or polymer types, capacitor size is usually determined
by the capacity needed to prevent V
SAG
and V
SOAR
from
causing problems during load transients. Generally, once
enough capacitance is added to meet the overshoot
requirement, undershoot at the rising load edge is no
longer a problem (also, see the V
SAG
and V
SOAR
equa-
tion in the Transient Response section).
Output Capacitor Stability Considerations
Stability is determined by the value of the ESR zero relative to the switching frequency. The point of instability is
given by the following equation:
For a typical 300kHz application, the ESR zero frequency
must be well below 95kHz, preferably below 50kHz.
Tantalum and OS-CON capacitors in widespread use at
the time of publication have typical ESR zero frequencies
of 25kHz. In the design example used for inductor selection, the ESR needed to support 50mVp-p ripple is
60mV/2.7A = 22mΩ. Two 470µF/4V Kemet T510 low-ESR
tantalum capacitors in parallel provide 22mΩ max ESR.
Their typical combined ESR results in a zero at 27kHz,
well within the bounds of stability.
Don’t put high-value ceramic capacitors directly across
the feedback sense point without taking precautions to
ensure stability. Large ceramic capacitors can have a
high ESR zero frequency and cause erratic, unstable
operation. However, it’s easy to add enough series resistance by placing the capacitors a couple of inches
downstream from the feedback sense point, which
should be as close as possible to the inductor (see the
All-Ceramic-Capacitor Application section).
Unstable operation manifests itself in two related but distinctly different ways: double-pulsing and fast-feedback
loop instability.
Double-pulsing occurs due to noise on the output or
because the ESR is so low that there isn’t enough voltage ramp in the output voltage signal. This “fools” the
error comparator into triggering a new cycle immediately
after the 400ns minimum off-time period has expired.
Double-pulsing is more annoying than harmful, resulting
in nothing worse than increased output ripple. However,
it can indicate the possible presence of loop instability,
which is caused by insufficient ESR.
Loop instability can result in oscillations at the output
after line or load perturbations that can trip the overvoltage protection latch or cause the output voltage to fall
below the tolerance limit.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient and carefully
observe the output voltage ripple envelope for overshoot and ringing. It can help to simultaneously monitor
the inductor current with an AC current probe. Don’t
allow more than one cycle of ringing after the initial
step-response under- or overshoot.
Input Capacitor Selection
The input capacitor must meet the ripple current
requirement (I
RMS
) imposed by the switching currents.
Nontantalum chemistries (ceramic, aluminum, or OSCON) are preferred due to their resistance to power-up
surge currents.
I I
VV-V
V
RMS LOAD
OUT IN OUT
IN
=
()
f
f
where f
1
2R C
ESR
ESR
ESR F
=
=
⋅⋅ ⋅
π
π
R
Vp p
LIR I
ESR
LOAD MAX
≤
⋅
-
()
R
V
I
ESR
DIP
LOAD MAX
≤
()
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
18 ______________________________________________________________________________________
For optimal circuit reliability, choose a capacitor that
has less than 10°C temperature rise at the peak ripple
current.
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability (>5A)
when using high-voltage (>20V) AC adapters. Low-current applications usually require less attention.
For maximum efficiency, choose a high-side MOSFET
(Q1) that has conduction losses equal to the switching
losses at the optimum battery voltage (15V). Check to
ensure that the conduction losses at minimum input
voltage don’t exceed the package thermal limits or
violate the overall thermal budget. Check to ensure that
conduction losses plus switching losses at the maxi-
mum input voltage don’t exceed the package ratings or
violate the overall thermal budget.
Choose a low-side MOSFET (Q2) that has the lowest
possible R
DS(ON)
, comes in a moderate to small package (i.e., SO-8), and is reasonably priced. Ensure that
the MAX1714 DL gate driver can drive Q2; in other
words, check that the gate isn’t pulled up by the highside switch turn on, due to parasitic drain-to-gate capacitance, causing cross-conduction problems. Switching
losses aren’t an issue for the low-side MOSFET, since it’s
a zero-voltage switched device when used in the buck
topology.
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty factor
extremes. For the high-side MOSFET, the worst-case
power dissipation due to resistance occurs at minimum
battery voltage:
PD(Q1 Resistive) = (V
OUT
/ V
IN(MIN)
) · I
LOAD
2
· R
DS(ON)
Generally, a small high-side MOSFET is desired to
reduce switching losses at high input voltages. However,
the R
DS(ON)
required to stay within package power-dissipation limits often limits how small the MOSFET can be.
Again, the optimum occurs when the switching (AC)
losses equal the conduction (R
DS(ON)
) losses. High-side
switching losses don’t usually become an issue until the
input is greater than approximately 15V.
Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum AC adapter
voltages are applied, due to the squared term in the
CV2F switching loss equation. If the high-side MOSFET
you’ve chosen for adequate R
DS(ON)
at low battery voltages becomes extraordinarily hot when subjected to
V
IN(MAX)
, you must reconsider your choice of MOSFET.
Calculating the power dissipation in Q1 due to switching
losses is difficult, since it must allow for difficult-to-quanti-
fy factors that influence the turn-on and turn-off times.
These factors include the internal gate resistance, gate
charge, threshold voltage, source inductance, and PC
board layout characteristics. The following switching loss
calculation provides only a very rough estimate and is no
substitute for breadboard evaluation, preferably including
a sanity check using a thermocouple mounted on Q1.
where C
RSS
is the reverse transfer capacitance of Q1
and I
GATE
is the peak gate-drive source/sink current (1A
typical).
For the low-side MOSFET, Q2, the worst-case power dissipation always occurs at maximum battery voltage:
PD(Q2) = (1 - V
OUT
/ V
IN(MAX)
) · I
LOAD
2
· R
DS(ON)
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
I
LOAD(MAX)
but are not quite high enough to exceed the
current limit and cause the fault latch to trip. To protect
against this possibility, you must “overdesign” the circuit
to tolerate I
LOAD
= I
LIMIT(HIGH)
+ [(LIR / 2) · I
LOAD(MAX)
],
where I
LIMIT(HIGH)
is the maximum valley current allowed
by the current-limit circuit, including threshold tolerance
and on-resistance variation. This means that the
MOSFETs must be very well heatsinked. If short-circuit
protection without overload protection is enough, a normal I
LOAD
value can be used for calculating component
stresses.
Choose a Schottky diode D1 having a forward voltage
low enough to prevent the Q2 MOSFET body diode from
turning on during the dead time. As a general rule, a
diode having a DC current rating equal to 1/3 of the load
current is sufficient. This diode is optional, and if efficiency isn’t critical it can be removed.
Application Issues
Dropout Performance
The output voltage adjust range for continuous-conduction operation is restricted by the nonadjustable 500ns
(max) minimum off-time one-shot. For best dropout performance, use the slowest (200kHz) on-time setting.
When working with low input voltages, the duty-factor
limit must be calculated using worst-case values for onand off-times. Manufacturing tolerances and internal
propagation delays introduce an error to the TON K-factor. This error is greater at higher frequencies (Table 5).
Also, keep in mind that transient response performance
of buck regulators operated close to dropout is poor,
PD(Q1 switching)
CV fI
RSS IN(MAX)2LOAD
=
⋅⋅⋅
I
GATE
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 19
and bulk output capacitance must often be added (see
the V
SAG
equation in Transient Response section).
Dropout Design Example: VIN= 3V min, V
OUT
= 2V,
f = 300kHz. The required duty is (V
OUT
+ VSW) / (VINVSW) = (2V + 0.1V) / (3.0V - 0.1V) = 72.4%. The worstcase on-time is (V
OUT
+ 0.075) / VIN· K = 2.075V / 3V ·
3.35µs-V · 90% = 2.08µs. The IC duty-factor limitation is:
which meets the required duty.
Remember to include inductor resistance and MOSFET
on-state voltage drops (V
SW
) when doing worst-case
dropout duty-factor calculations.
All-Ceramic-Capacitor Application
Ceramic capacitors have advantages and disadvantages. They have ultra-low ESR, are noncombustible, are
relatively small, and are nonpolarized. On the other
hand, they’re expensive and brittle, and their ultra-low
ESR characteristic can result in excessively high ESR
zero frequencies (affecting stability). In addition, their relatively low capacitance value can cause output overshoot when going abruptly from full-load to no-load
conditions, unless there are some bulk tantalum or electrolytic capacitors in parallel to absorb the stored energy
in the inductor. In some cases, there may be no room for
electrolytics, creating a need for a DC-DC design that
uses nothing but ceramics.
The all-ceramic-capacitor application of Figure 7
replaces the standard, typical tantalum output capacitors
with ceramics in a 7A circuit. This design relies on having a minimum of 5mΩ parasitic PC board trace resistance in series with the capacitor in order to reduce the
ESR zero frequency. This small amount of resistance is
easily obtained by locating the MAX1714 circuit 2 or 3
inches away from the CPU, and placing all the ceramic
DUTY
t
tt
2.08 s
2.08 s 500ns
80.6%,
ON(MIN)
ON(MIN) OFF(MAX)
=
+
=
+
=
µ
µ
Figure 7. All-Ceramic-Capacitor Application
Table 5. Approximate K-Factor Errors
TON
SETTING
(kHz)
APPROXIMATE
K-FACTOR
ERROR (%)
MIN V
IN
AT V
OUT
= 2V
(V)
200 ±10 2.6
300 ±10 2.9
450 ±12.5 3.2
600 ±12.5 3.6
K
FACTOR
(µs)
5
3.3
2.2
1.7
V
= 7V TO 24V*
IN
* FOR HIGHER MINIMUM INPUT VOLTAGE,
* LESS OUTPUT CAPACITANCE IS ACCEPTABLE.
ON/OFF
0.22µF
1µF
V+ V
SHDN
SKIP
REF
TON
20Ω
ILIM
CC
MAX1714
( ) ARE FOR THE MAX1714B ONLY.
V
BST
PGND (GND)
AGND (GND)
OUT
DD
DH
0.1µF
LX
FB
DL
5Ω
+5V
1µF
C1 = 2 x 10µF/25V TAIYO YUDEN (1812) (TMK432BJ106AM)
C2 = 6 x 47µF/6.3V TAIYO YUDEN (1812) (JMK432BJ476MN)
R1 + R2 = 5mΩ MINIMUM OF PCB TRACE RESISTANCE (TOTAL)
C1
Q1
0.5µH
Q2
1k
R1
C2 CPU
R2
2.5V AT 7A
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
20 ______________________________________________________________________________________
capacitors close to the CPU. Resistance values higher
than 5mΩ just improve the stability (which can be
observed by examining the load-transient response
characteristic as shown in the Typical Operating
Characteristics). Avoid adding excess PC board trace
resistance, as there’s an efficiency penalty; 5mΩ is sufficient for a 7A circuit.
V
SOAR
determines the minimum output capacitance
requirement. In this example, the switching frequency
has been increased to 600kHz and the inductor value
has been reduced to 0.5µH (compared to 300kHz and
1.5µH for the standard 8A circuit) in order to minimize the
energy transferred from inductor to capacitor during
load-step recovery. The overshoot must be calculated to
avoid tripping the OVP latch. The efficiency penalty for
operating at 600kHz is about 2% to 3%, depending on
the input voltage.
An optional 1kΩ resistor is placed in series with OUT.
This resistor attenuates high-frequency noise in some
boards, which can cause double pulsing.
Fixed Output Voltages
The MAX1714’s Dual Mode™ operation allows the selection of common voltages without requiring external components (Figure 8). Connect FB to AGND for a fixed
+2.5V output or to VCCfor a +3.3V output, or connect FB
directly to OUT for a fixed +1.0V output.
Setting V
OUT
with a Resistor-Divider
The output voltage can be adjusted with a resistordivider if desired (Figure 9). The equation for adjusting
the output voltage is:
where VFBis 1.0V.
2-Stage (5V Powered) Notebook CPU
Buck Regulator
The most efficient and overall cost-effective solution for
stepping down a high-voltage battery to a very low output voltage is to use a single-stage buck regulator that’s
powered directly from the battery. However, there may
be situations where the battery bus can’t be routed near
the CPU, or where space constraints dictate the smallest
possible local DC-DC converter. In such cases, the 5V
powered circuit of Figure 10 may be appropriate. The
reduced input voltage allows a higher switching frequency and a much smaller inductor value.
PC Board Layout Guidelines
Careful PC board layout is critical to achieving low
switching losses and clean, stable operation. The switching power stage requires particular attention (Figure 11).
If possible, mount all of the power components on the
top side of the board with their ground terminals flush
against one another. Follow these guidelines for good PC
board layout:
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
• Connect AGND and PGND together close to the IC.
For the MAX1714B, these grounds are connected
internally to the GND pin. Carefully follow the grounding instructions under step 4 of the Layout Procedure.
• Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PC boards (2 oz vs. 1 oz) can enhance
full-load efficiency by 1% or more. Correctly routing
PC board traces is a difficult task that must be
approached in terms of fractions of centimeters,
V V1
R1
R2
OUT FB
=+
Figure 8. Feedback Mux
Dual Mode is a trademark of Maxim Integrated Products.
R
1
2FC
ESR
OUT
≥
OUT
TO ERROR AMP
FIXED
2.5V
MAX1714
FIXED
3.3V
FB
2V
0.2V
where a single milliohm of excess trace resistance
causes a measurable efficiency penalty.
• LX and PGND connections to Q2 for current limiting
must be made using Kelvin sense connections to
guarantee the current-limit accuracy. With SO-8
MOSFETs, this is best done by routing power to the
MOSFETs from outside using the top copper layer,
while tying in PGND and LX inside (underneath) the
SO-8 package.
• When trade-offs in trace lengths must be made, it’s
preferable to allow the inductor charging path to be
made longer than the discharge path. For example,
it’s better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
allow distance between the inductor and the lowside MOSFET or between the inductor and the output filter capacitor.
• Ensure that the OUT connection to C
OUT
is short and
direct. However, in some cases it may be desirable to
deliberately introduce some trace length between the
OUT inductor node and the output filter capacitor (see
the All-Ceramic-Capacitor Application section).
• Route high-speed switching nodes (BST, LX, DH, and
DL) away from sensitive analog areas (REF, FB).
• Make all pin-strap control input connections (SKIP,
ILIM, etc.) to AGND or V
CC
rather than PGND or VDD.
Layout Procedure
1) Place the power components first, with ground terminals adjacent (Q2 source, CIN-, COUT-, D1 anode). If
possible, make all these connections on the top layer
with wide, copper-filled areas.
2) Mount the controller IC adjacent to MOSFET Q2,
preferably on the back side opposite Q2 in order to
keep LX, PGND, and the DL gate-drive lines short and
wide. The DL gate trace must be short and wide,
measuring 10 to 20 squares (50 to 100 mils wide if the
MOSFET is 1 inch from the controller IC).
3) Group the gate-drive components (BST diode and
capacitor, VDDbypass capacitor) together near the
controller IC.
4) Make the DC-DC controller ground connections as
shown in Figure 11. This diagram can be viewed as
having three separate ground planes: output ground,
where all the high-power components go; the PGND
plane, where the PGND pin and VDDbypass capacitor go; and an analog AGND plane, where sensitive
analog components go. The analog ground plane and
PGND plane must meet only at a single point directly
beneath the IC. For the MAX1714B, this point should
be the GND pin. These two planes are then connected to the high-power output ground with a short connection from VDDcap/PGND to the source of the
low-side MOSFET, Q2 (the middle of the star ground).
This point must also be very close to the output
capacitor ground terminal.
5) Connect the output power planes (V
CORE
and system
ground planes) directly to the output filter capacitor
positive and negative terminals with multiple vias.
Place the entire DC-DC converter circuit as close to
the CPU as is practical.
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 21
Figure 9. Setting V
OUT
with a Resistor-Divider
V
BATT
DH
V
MAX1714
DL
PGND
(GND)
OUT
FB
AGND
(GND)
R1
R2
( ) ARE FOR THE MAX1714B ONLY.
OUT
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
22 ______________________________________________________________________________________
Figure 10. 5V Powered, 7A CPU Buck Regulator
Pin Configurations
TOP VIEW
20
19
18
17
16
15
14
13
1
2
3
4
5
6
7
8
LX
BST
SKIP
V+FB
SHDN
N.C.
DH
TON
V
CC
V
DD
DLAGND
REF
ILIM
OUT
12
11
9
10
PGND
N.C.PGOOD
N.C.
MAX1714A
QSOP
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
LX
BST
SKIP
V+OUT
FB
SHDN
DH
TON
V
CC
V
DD
DLGND
PGOOD
REF
ILIM
QSOP
MAX1714B
4.5V TO 5.5V
1µF
0.1µF
ON/OFF
1µF20Ω
I
LIMVCC
SHDN
0.22µF
REF
V
CC
MAX1714
V+
V
DD
BST
DH
LX
DL
PGND
(GND)
FB
V
IN
C1
4 x 10µF/25V
IRF7805
0.5µH
IRF7805
V
L1
OUT
2.5V AT 7A
C2
3 x 470µF
KEMET T510
(GND)
SKIP
OUT
AGND
( ) = ARE FOR THE MAX1714B ONLY.
NOTE: IN THE MAX1714B, AGND AND PGND ARE
INTERNALLY CONNECTED TO THE GND PIN.
100k
PGOOD
TON
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
______________________________________________________________________________________ 23
Figure 11. Power-Stage PC Board Layout Example
ALL ANALOG GROUNDS
CONNECT TO AGND ONLY
MAX1714
V
CC
REF
I
LIM
AGND
CONNECT AGND TO PGND
BENEATH IC, 1 POINT ONLY.
SPLIT ANALOG GND PLANE AS SHOWN.
NOTES: "STAR" GROUND IS USED.
D1 IS DIRECTLY ACROSS Q2.
V
DD
VIA TO PGND
NEAR Q2 SOURCE
VIA TO SOURCE
OF Q2
VIA TO LX
V
BATT
CIN
Q1
D1 Q2
L1
GND IN
GND
OUT
C
OUT
V
OUT
VIA TO OUT
+
NEAR C
OUT
INDUCTOR DISCHARGE PATH HAS LOW DC RESISTANCE.
MAX1714
High-Speed Step-Down Controller
for Notebook Computers
24 ______________________________________________________________________________________
Package Information
QSOP.EPS
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