Linear Technology LTC3558 User Manual

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Page 1

LTC3558

Linear USB Battery Charger

with Buck and

Buck-Boost Regulators

FEATURES

Battery Charger

Standalone USB Charger

Up to 950mA Charge Current Programmable via

Single Resistor

HPWR Input Selects 20% or 100% of Programmed

Charge Current

NTC Input for Temperature Qualiﬁ ed Charging

Internal Timer Termination

Bad Battery Detection

Switching Regulators (Buck and Buck-Boost)

Up to 400mA Output Current per Regulator

2.25MHz Constant-Frequency Operation

Power Saving Burst Mode® Operation

Low Proﬁ le, 20-Lead, 3mm × 3mm QFN Package

APPLICATIONS

SD/Flash-Based MP3 Players

Low Power Handheld Applications

DESCRIPTION

The LTC®3558 is a USB battery charger with dual high efﬁ ciency switching regulators. The device is ideally suited to power single-cell Li-Ion/Polymer based handheld applications needing multiple supply rails.

Battery charge current is programmed via the PROG pin and the HPWR pin with capability up to 950mA of current at the BAT pin. The CHRG pin allows battery status to be monitored continuously during the charging process. An internal timer controls charger termination.

The part includes monolithic synchronous buck and buckboost regulators that can provide up to 400mA of output current each and operate at efﬁ ciencies greater than 90% over the entire Li-Ion/Polymer battery range. The buckboost regulator can regulate its programmed output voltage at its rated deliverable current over the entire Li-Ion range without drop out, increasing battery runtime.

The LTC3558 is offered in a low proﬁ le (0.75mm), thermally enhanced, 20-lead (3mm × 3mm) QFN package.

, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology

Corporation. All other trademarks are the property of their respective owners.

TYPICAL APPLICATION

USB Charger Plus Buck Regulator and Buck-Boost Regulator

USB (4.3V TO 5.5V)

DIGITAL

CONTROL

1.74k

PROG

NTC

CHRG

SUSP

HPWR

MODE

EN1

EN2

GND

LTC3558

EXPOSED

1µF

PAD

BAT

PV PV

SW1

FB1

SWAB2

SWCD2

OUT2

FB2

3558 TA01

IN1

IN2

4.7µH

2.2µH

324k

105k

10µF

324k

649k

121k

15k

33pF

330pF

SINGLE

Li-lon CELL (2.7V TO 4.2V)

1.2V AT 400mA

10pF

3.3V AT 400mA

10pF

Demo Board

10µF

22µF

3558f

Page 2

LTC3558

ABSOLUTE MAXIMUM RATINGS

(Note 1)

VCC (Transient);

t < 1ms and Duty Cycle < 1% ....................... –0.3V to 7V

(Static) .................................................. –0.3V to 6V

BAT, CHRG ................................................... –0.3V to 6V

PROG, SUSP .................................–0.3V to (V

HPWR, NTC ...................–0.3V to Max (V

PROG Pin Current ...............................................1.25mA

BAT Pin Current ..........................................................1A

, PV

IN1

EN1, EN2, MODE, V

FB1, SW1 ......................... –0.3V to (PV

FB2, V

SWCD2 ............................–0.3V to (V

...............................................................600mA DC

SW1

SWAB2

..................................–0.3V to (BAT + 0.3V)

IN2

.............................. –0.3V to 6V

OUT2

IN1

, SWAB2 ............. –0.3V to (PV

, I

SWCD2

, I

VOUT2

................................... 750mA DC

IN2

OUT2

Junction Temperature (Note 2) ............................. 125°C

Operating Temperature Range (Note 3).... –40°C to 85°C

Storage Temperature ..............................–65°C to 125°C

+ 0.3V)

, BAT) + 0.3V

+ 0.3V) or 6V + 0.3V) or 6V + 0.3V) or 6V

PIN CONFIGURATION

TOP VIEW

VCCCHRG

PROG

7 8

IN1PVIN2

NTC

SWAB2

20 19 18 17 16

GND

BAT

MODE

FB1

EN1

SW1

20-LEAD (3mm × 3mm) PLASTIC QFN

EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB

UD PACKAGE

= 125°C, θJA = 68°C/W

JMAX

HPWR

SWCD2

EN2

FB2

SUSP

OUT2

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC3558EUD#PBF LTC3558EUD#TRPBF LDCD

Consult LTC Marketing for parts speciﬁ ed with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based ﬁ nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel speciﬁ cations, go to: http://www.linear.com/tapeandreel/

20-Lead (3mm × 3mm) Plastic QFN

–40°C to 85°C

3558f

Page 3

LTC3558

ELECTRICAL CHARACTERISTICS

The l denotes speciﬁ cations that apply over the full operating temperature range, otherwise speciﬁ cations are at T

= 25°C. VCC = 5V, BAT = PV

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Battery Charger

VCC

FLOAT

CHG

BAT

Input Supply Voltage

Battery Charger Quiescent Current (Note 4)

Standby Mode, Charge Terminated Suspend Mode, V

BAT Regulated Output Voltage

0°C ≤ T

≤ 85°C

Constant-Current Mode Charge Current

HPWR = 1 HPWR = 0

Battery Drain Current Standby Mode, Charger Terminated, EN1 = EN2 = 0

Shutdown, V

Suspend Mode, SUSP = 5V, BAT = 4.2V, EN1 = EN2 = 0 V

= 0V, EN1 = EN2 = 1, MODE = 1,

FB1 = FB2 = 0.85V, V

UVLO

ΔV

DUVLO

UVLO

Undervoltage Lockout Threshold BAT = 3.5V, VCC Rising 3.85 4 4.125 V

Undervoltage Lockout Hysteresis BAT = 3.5V 200 mV

Differential Undervoltage Lockout

BAT = 4.05V, (VCC – BAT) Falling 30 50 70 mV

Threshold

ΔV

DUVLO

PROG

Differential Undervoltage Lockout Hysteresis

PROG Pin Servo Voltage HPWR = 1

BAT = 4.05V 130 mV

HPWR = 0 BAT < V

TRKL

PROG

TRKL

ΔV

TRKL

ΔV

RECHRG

TERM

BADBAT

C/10

ON(CHG)

LIM

Ratio of I

Trickle Charge Current BAT < V

Trickle Charge Threshold Voltage BAT Rising 2.8 2.9 3 V

Trickle Charge Hysteresis Voltage 100 mV

Recharge Battery Threshold Voltage Threshold Voltage Relative to V

Recharge Comparator Filter Time BAT Falling 1.7 ms

Safety Timer Termination Period BAT = V

Bad Battery Termination Time BAT < V

End-of-Charge Indication Current Ratio (Note 5) 0.085 0.1 0.11 mA/mA

End-of-Charge Comparator Filter Time I

Battery Charger Power FET OnResistance (Between V

Junction Temperature in Constant

to PROG Pin Current 800 mA/mA

BAT

TRKL

FLOAT

TRKL

Falling 2.2 ms

BAT

= 190mA 500

and BAT)

BAT

Temperature Mode

NTC

COLD

HOT

DIS

Cold Temperature Fault Threshold Voltage

Hot Temperature Fault Threshold Voltage

NTC Disable Threshold Voltage Falling NTC Voltage

Rising NTC Voltage Hysteresis

Falling NTC Voltage Hysteresis

Hysteresis

NTC

NTC Leakage Current V

= VCC = 5V –1 1 µA

NTC

= PV

IN1

= 5V

SUSP

< V

, BAT = 4.2V, EN1 = EN2 = 0

UVLO

OUT2

IN2

= 3.6V

= 3.6V, R

FLOAT

= 1.74k, unless otherwise noted.

PROG

4.3 5.5 V

285

8.5

4.179

4.165

4408446092500

4.200

–3.5 –2.5 –1.5

–50

1.000

0.200

0.100

36 46 56 mA

–75 –95 –115 mV

3.5 4 4.5 Hour

0.4 0.5 0.6 Hour

105 °C

75 76.5

1.6

33.4 34.9

1.6

0.7 1.7 50

400

4.221

4.235

100

–7 –4 –3

–100

78 %V

36.4 %V %V

2.7 %V

µA µA

V V

mA mA

µA µA µA

µA

V V V

mΩ

CC CC

3558f

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LTC3558

ELECTRICAL CHARACTERISTICS

The l denotes speciﬁ cations that apply over the full operating temperature range, otherwise speciﬁ cations are at T

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Logic (HPWR, SUSP, CHRG, EN1, EN2, MODE)

CHRG

Buck Switching Regulator

IN1

PVIN1

UVLO PV

IN1

OSC

LIMSW1

FB1

MAX1

PMOS1

NMOS1

SW1(PD)

Buck-Boost Switching Regulator

IN2

PVIN2

UVLO PV

IN2

OUT2(LOW)

OUT2(HIGH)

LIMF2

PEAK2(BURST)

LIMR2

ZERO2(BURST)

MAX2(BURST)

FB2

OSC

Input Low Voltage HPWR, SUSP, MODE, EN1, EN2 Pins 0.4 V

Input High Voltage HPWR, SUSP, MODE, EN1, EN2 Pins 1.2 V

Logic Pin Pull-Down Resistance HPWR, SUSP Pins

CHRG Pin Output Low Voltage I CHRG Pin Input Current BAT = 4.5V, V

Input Supply Voltage

Pulse Skip Input Current Burst Mode Current Shutdown Current Supply Current in UVLO

Falling

IN1

Rising

IN1

Switching Frequency MODE = 0 1.91 2.25 2.59 MHz

Peak PMOS Current Limit 550 800 1050 mA

Feedback Voltage MODE = 0

FB Input Current FB1 = 0.85V –50 50 nA

Maximum Duty Cycle FB1 = 0V

of PMOS I

DS(ON)

of NMOS I

DS(ON)

SW Pull-Down in Shutdown 13

Input Supply Voltage

PWM Input Current Burst Mode Input Current Shutdown Current Supply Current in UVLO

Falling

IN2

Rising

IN2

Minimum Regulated Buck-Boost V

Maximum Regulated Buck-Boost V

Forward Current Limit (Switch A) MODE = 0

Forward Current Limit (Switch A) MODE = 1

Reverse Current Limit (Switch D) MODE = 0

Reverse Current Limit (Switch D) MODE = 1

Maximum Deliverable Output Current in Burst Mode Operation

Feedback Servo Voltage

FB2 Input Current FB2 = 0.85V –50 50 nA

Switching Frequency MODE = 0 1.91 2.25 2.59 MHz

= 25°C. VCC = 5V, BAT = PV

= 5mA 100 250 mV

CHRG

FB1 = 0.85V, MODE = 0 (Note 6) FB1 = 0.85V, MODE = 1 (Note 6) EN1 = 0 PV

= PV

IN2

OUT OUT

OUT

IN2

IN2 OUT2

= 2V

= 0A

= 2V

< 4.2V

< 5.5V

IN1

= 100mA 0.65

SW1

= –100mA 0.75

SW1

MODE = 0, I MODE = 1, I EN2 = 0, I PV

IN1

2.65 2.75 V

OUT

2.7V < PV

2.75V < V

IN1

= PV

= 3.6V, R

IN2

= 1.74k, unless otherwise noted.

PROG

1.9 4 6.3

= 5V 0 1 µA

2.7 4.2 V

= 0A, FB2 = 0.85V (Note 6) = 0A, FB2 = 0.85V (Note 6)

220

2.30 2.45

780 800 820 mV

100 %

2.7 4.2 V

2.55 2.70

220

2.30 2.45

2.55 2.70

400

50 0 4

2 8

400

30 0 4

1 8

5.45 5.60 V

580 700 820 mA

180 250 320 mA

325 450 575 mA

–35 0 35 mA

50 mA

780 800 820 mV

MΩ

µA µA µA µA

V V

Ω Ω

kΩ

µA µA µA µA

V V

3558f

Page 5

LTC3558

ELECTRICAL CHARACTERISTICS

The l denotes speciﬁ cations that apply over the full operating temperature range, otherwise speciﬁ cations are at T

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

DSP(ON)

DSN(ON)

LEAK(P)

LEAK(N)

BUCK(MAX)

BOOST(MAX)

SS2

OUT(PD)

PMOS R

DS(ON)

NMOS R

DS(ON)

PMOS Switch Leakage Switches A, D –1 1 µA

NMOS Switch Leakage Switches B, C –1 1 µA

Maximum Buck Duty Cycle MODE = 0

Maximum Boost Duty Cycle MODE = 0 75 %

Soft-Start Time 0.5 ms

Pull-Down in Shutdown 10

OUT

= 25°C. VCC = 5V, BAT = PV

= 3.6V 0.6

OUT

IN1

= PV

= 3.6V, R

IN2

= 1.74k, unless otherwise noted.

PROG

0.6

100 %

Ω Ω

kΩ

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime.

Note 2: T dissipation P

T Note 3: The LTC3558E is guaranteed to meet speciﬁ cations from 0°C to

85°C. Speciﬁ cations over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls.

is calculated from the ambient temperature TA and power

according to the following formula:

= TA + (PD • θJA)

Note 4: VCC supply current does not include current through the PROG pin or any current delivered to the BAT pin. Total input current is equal to this speciﬁ cation plus 1.00125 • I

Note 5: I with indicated PROG resistor.

Note 6: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency.

is expressed as a fraction of measured full charge current

C/10

BAT

where I

is the charge current.

BAT

3558f

Page 6

LTC3558

TYPICAL PERFORMANCE CHARACTERISTICS

Suspend State Supply and BAT Currents vs Temperature

VCC = 5V

BAT = 4.2V SUSP = 5V

CURRENT (µA)

EN1 = EN2 = 0V

–55

–35

VCC

BAT

–15

TEMPERATURE (°C)

3558 G01

Battery Regulation (Float) Voltage vs Temperature

4.24 VCC = 5V

4.23

4.22

4.21

(V)

4.20

FLOAT

4.19

4.18

4.17

4.16

–35 –15 25

–55

TEMPERATURE (°C)

45 65 85

= 25°C, unless otherwise noted.

Battery Regulation (Float) Voltage vs Battery Charge Current, Constant-Voltage Charging

4.205

4.200

4.195

4.190

4.185

4.180

(V)

BAT

4.175

4.170

4.165 VCC = 5V

4.160

HPWR = 5V

= 845

PROG

4.155 EN1 = EN2 = 0V

4.150

3558 G02

100

300

2000

400

700

800

900

500

600

(mA)

BAT

1000

3558 G03

Battery Charge Current vs Supply Voltage

500

VCC = 5V

495

HPWR = 5V

490

485

480

475

(mA)

470

BAT

465

460

455

450

445

440

= 1.74k

PROG

EN1 = EN2 = 0V

4.3

4.5

4.6 4.9 5.1

4.8 5.0 5.2 5.4 5.5

4.7 VCC (V)

Battery Charger Undervoltage Lockout Threshold vs Temperature

4.2 BAT = 3.5V

(V)

4.1

4.0

3.9

3.8

3.7

3.6

RISING

FALLING

Battery Charge Current vs Battery Voltage

500

VCC = 5V

450

400

350

300

(mA)

250

BAT

200

150

100

5.34.4

3558 G04

= 1.74k

PROG

2.5

HPWR = 5V

HPWR = 0V

BAT

(V)

3.5

4.5

3558 G05

Battery Drain Current in Undervoltage Lockout vs Temperature

3.0 EN1 = EN2 = 0V

(µA)

BAT

2.5

2.0

1.5

1.0

0.5

BAT = 4.2V

BAT = 3.6V

Battery Charge Current vs Ambient Temperature in Thermal Regulation

500

450

400

350

300

(mA)

250

BAT

200

150

VCC = 5V

100

HPWR = 5V

= 1.74k

PROG

EN1 = EN2 = 0

–55

–35 5

–15

TEMPERATURE (°C)

45 125

PROG Voltage vs Battery Charge Current

1.2 VCC = 5V

HPWR = 5V

(V)

PROG

1.0

0.8

0.6

0.4

0.2

= 1.74k

PROG

EN1 = EN2 = 0V

105

3.5 –55

–35 –15

TEMPERATURE (°C)

25 65 85

545

3558 G07

–55

–35 –15

25 65 85

545

TEMPERATURE (°C)

3558 G08

100 200 300 400

(mA)

BAT

500500 150 250 350 450

3558 G09

3558f

Page 7

LTC3558

TYPICAL PERFORMANCE CHARACTERISTICS

Recharge Threshold vs Temperature

115

VCC = 5V

111

107

103

(mV)

RECHARGE

–55

–15

–35

TEMPERATURE (°C)

CHRG Pin Output Low Voltage vs Temperature

140

VCC = 5V

= 5mA

CHRG

120

100

VOLTAGE (mV)

3558 G10

Battery Charger FET On-Resistance vs Temperature

700

VCC = 4V

= 200mA

BAT

650

EN1 = EN2 = 0V

600

550

(m)

500

DS(ON)

450

400

350

300

–55

–15 5 25 45 65

–35

TEMPERATURE (°C)

CHRG Pin I-V Curve

VCC = 5V BAT = 3.8V

(mA)

CHRG

TA = 25°C, unless otherwise noted.

SUSP/HPWR Pin Rising Thresholds vs Temperature

1.2 VCC = 5V

1.1

1.0

0.9

0.8

0.7

THRESHOLD (V)

0.6

0.5

0.4

3558 G11

–35 –15 25

–55

TEMPERATURE (°C)

Timer Accuracy vs Supply Voltage

2.0

1.5

1.0

0.5

PERCENT ERROR (%)

–0.5

45 65 85

3558 G12

–55

–35 –15

TEMPERATURE (°C)

25 65 85

545

3558 G13

Timer Accuracy vs Temperature

VCC = 5V

PERCENT ERROR (%)

–1

–2

–55

–15

–35

TEMPERATURE (°C)

585

45 65

3558 G16

Complete Charge Cycle 2400mAh Battery

1000

800 600

(mA)BAT (V)CHRG (V)

400

BAT

200

5.0

4.5

4.0

3.5

3.0

5.0

4.0

3.0

2.0

1.0 0

12 4 6

TIME (HOUR)

CHRG (V)

3558 G14

VCC = 5V

= 0.845k

PROG

HPWR = 5V

3558 G17

–1.0

4.3

4.7 4.9 5.1

4.5 VCC (V)

Buck and Buck-Boost Regulator Switching Frequency vs Temperature

2.425 VCC = 0V, MODE = 0

BAT = PV

2.325

2.225

BAT = 2.7V

2.125

2.025

FREQUENCY (MHz)

1.925

1.825

1.725

–55

= PV

IN1

–35 5

–15

TEMPERATURE (°C)

IN2

BAT = 4.2V

45 125

5.3 5.5

3558 G15

BAT = 3.6V

105

3558 G18

3558f

Page 8

LTC3558

TYPICAL PERFORMANCE CHARACTERISTICS

Buck and Buck-Boost Regulator Undervoltage Thresholds vs Temperature

2.750 BAT = PV

2.700

2.650

2.600

2.550

2.500

2.450

INPUT VOLTAGE (V)

2.400

2.350

2.300

2.250

–55

IN1

–35 5

–15

= PV

IN2

RISING

FALLING

45 125

TEMPERATURE (°C)

Buck Regulator Input Current vs Temperature, Pulse Skip Mode

400

FB1 = 0.85V

350

300

= 4.2V

250

200

INPUT CURRENT (µA)

150

100

–35 5

–55

–15

IN1

45 125

TEMPERATURE (°C)

IN1

= 2.7V

105

3558 G19

105

3558 G22

Buck and Buck-Boost Regulator Enable Thresholds vs Temperature

1200

(V)

1100

1000

900

800

700

600

500

400

BAT = PV

–55

IN1

–35 5

–15

= PV

= 3.6V

IN2

RISING

FALLING

45 125

TEMPERATURE (°C)

Buck Regulator PMOS R vs Temperature

1300

1200

1100

1000

900

(m)

800

DS(ON)

700

600

500

400

–55

–35

–15

= 2.7V

IN1

TEMPERATURE (°C)

45 12

IN1

= 4.2V

DS(0N)

TA = 25°C, unless otherwise noted.

Buck Regulator Input Current vs Temperature, Burst Mode Operation

FB1 = 0.85V

INPUT CURRENT (µA)

105

3558 G20

–35 5

–55

–15

Buck Regulator NMOS R vs Temperature

1300

1200

1100

1000

900

(m)

800

DS(ON)

700

600

500

105

3558 G23

400

–55

–35

IN1

–15

= 2.7V

TEMPERATURE (°C)

= 4.2V

IN1

45 125

TEMPERATURE (°C)

IN1

45 12

IN1

DS(0N)

= 4.2V

= 2.7V

105

3558 G21

105

3558 G24

Buck Regulator Efﬁ ciency vs I

100

Burst Mode

OPERATION

EFFICIENCY (%)

0.1 10 100 1000

PULSE SKIP

MODE

(mA)

LOAD

= 1.2V

OUT

= 2.7V

IN1

= 4.2V

IN1

3558 G25

Buck Regulator Load Regulation

1.25 PV

= 3.6V

IN1

(V)

OUT

1.24

1.23

1.22

1.21

1.20

1.19

1.18

1.17

1.16

1.15

= 1.2V

OUT

Burst Mode OPERATION

PULSE SKIP

MODE

10 100 100

(mA)

LOAD

3558 G26

Buck Regulator Line Regulation

1.250 I

= 200mA

LOAD

1.240

1.230

1.220

1.210

(V)

1.200

OUT

1.190

1.180

1.170

1.160

1.150

3.000 3.600

2.700 3.300 (V)

IN1

3.900

3558 G27

4.200

3558f

Page 9

LTC3558

TYPICAL PERFORMANCE CHARACTERISTICS

Buck Regulator Pulse Skip Mode Operation

OUT

20mV/

DIV (AC)

2V/DIV

CURRENT

= 50mA/

DIV

= 3.8V

IN1

LOAD = 10mA

200ns/DIV

Buck Regulator Transient Response, Burst Mode Operation

CURRENT

= 200mA/

DIV

OUT

50mV/

DIV (AC)

5mA TO

290mA

= 3.8V 50µs/DIV

IN1

500mV/DIV

INDUCTOR

CURRENT

= 200mA/

2V/DIV

INDUCTOR

CURRENT

= 200mA/

50mV/

DIV (AC)

LOAD STEP

5mA TO

290mA

Buck Regulator Start-Up Transient

OUT

DIV

= 3.8V

IN1

PULSE SKIP MODE LOAD = 6

50µs/DIV

Buck Regulator Transient Response, Pulse Skip Mode

DIV

OUT

= 3.8V 50µs/DIV

IN1

3558 G28

3558 G31

INDUCTOR

LOAD STEP

TA = 25°C, unless otherwise noted.

Buck Regulator Burst Mode Operation

OUT

20mV/

DIV (AC)

2V/DIV

INDUCTOR

CURRENT

= 60mA/

3558 G29

DIV

PV LOAD = 60mA

IN1

= 3.8V

2µs/DIV

Buck-Boost Regulator Input Current vs Temperature

Burst Mode OPERATION FB2 = 0.85V

= 4.2V

IN2

= 2.7V

IN2

3558 G32

INPUT CURRENT (µA)

3558 G30

Buck-Boost Regulator Input Current vs Temperature

500

PWM MODE FB2 = 0.85V

450

400

350

300

250

INPUT CURRENT (µA)

200

150

100

–35 5

–55

IN2

–15

TEMPERATURE (°C)

= 4.2V

= 2.7V

45 125

105

3558 G34

Buck-Boost Regulator PMOS R

vs Temperature

DS(ON)

800

750

700

650

600

550

(m)

500

450

DS(ON)

400

350

300

250

200

–35 5

–55 –15

= 2.7V

IN2

= 4.2V

IN2

45 125

TEMPERATURE (°C)

–55

–35 5

–15

TEMPERATURE (°C)

45 125

105

3558 G33

Buck-Boost Regulator NMOS R

vs Temperature

DS(ON)

1200

1100

1000

900

800

(m)

700

DS(ON)

600

500

400

300

105

3558 G35

200

–35 5

–55

= 2.7V

IN2

= 4.2V

IN2

–15

TEMPERATURE (°C)

45 125

105

3558 G36

3558f

Page 10

LTC3558

TYPICAL PERFORMANCE CHARACTERISTICS

Buck-Boost Regulator Efﬁ ciency vs Input Voltage

100

= 3.3V

OUT

LOAD

EFFICIENCY (%)

2.700 3.300

= 10mA

LOAD

= 400mA

3.000 3.600 PV

(V)

IN2

LOAD

= 100mA

Burst Mode OPERATION PWM MODE

3.900

Buck-Boost Regulator Load Regulation

3.36 PV

= 3.6V

IN2

3.35

3.34

3.33

3.32

Burst Mode OPERATION

3.31

(V)

3.30

OUT

3.29

3.28

3.27

3.26

3.25

3.24

0.10 10 100 1000

LOAD

PWM MODE

(mA)

= 1mA

4.200

3558 G37

3558 G39

100

EFFICIENCY (%)

0.10 10 100 10001

3.36

3.35

3.34

3.33

3.32

3.31

(V)

3.30

OUT

3.29

3.28

3.27

3.26

3.25

3.24

2.700 3.300

TA = 25°C, unless otherwise noted.

Buck-Boost Efﬁ ciency vs Load Current

= 3.3V

OUT

3.6V

4.2V

2.7V

3.6V

4.2V

LOAD

OPERATION PV

(mA)

, Burst Mode

IN2

, PWM MODE

IN2

Buck-Boost Regulator Line Regulation

PWM MODE

= 100mA

LOAD

Burst Mode OPERATION

= 10mA

LOAD

3.000 3.600 PV

IN2

3.900

(V)

3558 G38

4.200

3558 G40

Buck-Boost Regulator Start-Up Transient, Burst Mode Operation

= 3.6V

IN2

= 332

LOAD

OUT

1V/DIV

INDUCTOR

CURRENT

= 200mA/DIV

EN2

1V/DIV

100µs/DIV

3558 G41

INDUCTOR

CURRENT

= 200mA/DIV

Buck-Boost Regulator Start-Up Transient, PWM Mode

= 3.6V

IN2

= 16

LOAD

OUT

1V/DIV

EN2

1V/DIV

100µs/DIV

3558 G42

3558f

Page 11

PIN FUNCTIONS

LTC3558

GND (Pin 1): Ground. Connect to Exposed Pad (Pin 21).

BAT (Pin 2): Charge Current Output. Provides charge cur-

rent to the battery and regulates ﬁ nal ﬂ oat voltage to 4.2V.

MODE (Pin 3): MODE Pin for Switching Regulators. When held high, both regulators operate in Burst Mode Operation. When held low, the buck regulator operates in pulse skip mode and the buck-boost regulator operates in PWM mode. This pin is a high impedance input; do not ﬂ oat.

FB1 (Pin 4): Buck Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output.

EN1 (Pin 5): Enable Input Pin for the Buck Regulator. This pin is a high impedance input; do not ﬂ oat. Active high.

SW1 (Pin 6): Buck Regulator Switching Node. External inductor connects to this node.

(Pin 7): Input Supply Pin for Buck Regulator. Con-

IN1

nect to BAT and PV

. A single 10µF input decoupling

IN2

capacitor to GND is required.

(Pin 8): Input Supply Pin for Buck-Boost Regulator.

IN2

Connect to BAT and PV

. A single 10µF input decoupling

IN1

capacitor to GND is required.

SWAB2 (Pin 9): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches A and B. External inductor connects between this node and SWCD2.

SWCD2 (Pin 10): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches C and D. External inductor connects between this node and SWAB2.

(Pin 11): Regulated Output Voltage for Buck-Boost

OUT2

Regulator.

SUSP (Pin 12): Suspend Battery Charging Operation. A voltage greater than 1.2V on this pin puts the battery charger in suspend mode, disables the charger and resets the termination timer. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally.

FB2 (Pin 13): Buck-Boost Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output.

(Pin 14): Output of the Error Ampliﬁ er and Voltage

Compensation Node for the Buck-Boost Regulator. External Type I or Type III compensation (to FB2) connects to this pin.

EN2 (Pin 15): Enable Input Pin for the Buck-Boost Regulator. This pin is a high impedance input; do not ﬂ oat. Active high.

HPWR (Pin 16): High Current Battery Charging Enabled. A voltage greater than 1.2V at this pin programs the BAT pin current at 100% of the maximum programmed charge current. A voltage less than 0.4V sets the BAT pin current to 20% of the maximum programmed charge current. When used with a 1.74k PROG resistor, this pin can toggle between low power and high power modes per USB speciﬁ cation. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally.

NTC (Pin 17): Input to the NTC Thermistor Monitoring Circuit. The NTC pin connects to a negative temperature coefﬁ cient thermistor which is typically co-packaged with the battery pack to determine if the battery is too hot or too cold to charge. If the battery temperature is out of range, charging is paused until the battery temperature re-enters the valid range. A low drift bias resistor is required from

to NTC and a thermistor is required from NTC to

ground. To disable the NTC function, the NTC pin should be tied to ground.

PROG (Pin 18): Charge Current Program and Charge Current Monitor Pin. Charge current is programmed by connecting a resistor from PROG to ground. When charging in constant-current mode, the PROG pin servos to 1V if the HPWR pin is pulled high, or 200mV if the HPWR pin is pulled low. The voltage on this pin always represents the BAT pin current through the following formula:

BAT

PROG

•800

CHRG (Pin 19): Open-Drain Charge Status Output. The

CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging (i.e., the charge current is less than one-tenth

3558f

Page 12

LTC3558

PIN FUNCTIONS

of the full-scale charge current), unresponsive battery (i.e., the battery voltage remains below 2.9V after one half hour of charging) and battery temperature out of range. CHRG requires a pull-up resistor and/or LED to provide indication.

BLOCK DIAGRAM

BAT

BODY

MAXER

CHRG

HPWR

SUSP

NTC

MODE

EN1

EN2

FB1

TEMPERATURE

FB2

DIE

BANDGAP

OSCILLATOR

2.25MHz

IN1

IN2

UNDERVOLTAGE

LOCKOUT

DIE

REF

= 0.8V

CLK

LOGIC

NTC REF

BUCK-BOOST REGULATOR

0.8V

NTCA

0.8V

–

ERROR

AMP

–

(Pin 20): Battery Charger Input. A 1µF decoupling

capacitor to GND is recommended.

Exposed Pad (Pin 21): Ground. The Exposed Pad must be soldered to PCB ground to provide electrical contact and rated thermal performance.

800x

BAT

PROG

SW1

OUT2

SWAB2

SWCD2

IN1

IN2

–

DIE

BATTERY CHARGER

MODEEN

LOGIC

MODEEN

CONTROL

LOGIC

BUCK REGULATOR

DCA

CLK

CONTROL

GND EXPOSED PAD

1 21

3558 BD

3558f

Page 13

OPERATION

LTC3558

The LTC3558 is a linear battery charger with a monolithic synchronous buck regulator and a monolithic synchronous buck-boost regulator. The buck regulator is internally compensated and needs no external compensation components.

The battery charger employs a constant-current, constantvoltage charging algorithm and is capable of charging a single Li-Ion battery at charging currents up to 950mA. The user can program the maximum charging current available at the BAT pin via a single PROG resistor. The actual BAT pin current is set by the status of the HPWR pin.

USB (5V)

PROG

HIGH

HIGH HIGH

LOW

PROG

SUSP

HPWR

EN1 EN2 MODE

LTC3558

BAT

IN1

IN2

SWAB2

SWCD2

OUT2

SW1

3558 F01

For proper operation, the BAT, PV

and PV

IN1

IN2

pins must be tied together, as shown in Figure 1. Current being delivered at the BAT pin is 500mA. Both switching regulators are enabled. The sum of the average input currents drawn by both switching regulators is 200mA. This makes the effective battery charging current only 300mA. If the HPWR pin were tied LO, the BAT pin current would be 100mA. With the switching regulator conditions unchanged, this would cause the battery to discharge at 100mA.

500mA

2.2µH

OUT1

200mA

300mA

10µF

SINGLE Li-lon CELL 3.6V

Figure 1. For Proper Operation, the BAT, PV

APPLICATIONS INFORMATION

Battery Charger Introduction

The LTC3558 has a linear battery charger designed to charge single-cell lithium-ion batteries. The charger uses a constant-current/constant-voltage charge algorithm with a charge current programmable up to 950mA. Additional features include automatic recharge, an internal termination timer, low-battery trickle charge conditioning, bad-battery detection, and a thermistor sensor input for out of temperature charge pausing.

Furthermore, the battery charger is capable of operating from a USB power source. In this application, charge current can be programmed to a maximum of 100mA or 500mA per USB power speciﬁ cations.

and PV

IN1

Pins Must Be Tied Together

IN2

Input Current vs Charge Current

The battery charger regulates the total current delivered to the BAT pin; this is the charge current. To calculate the total input current (i.e., the total current drawn from the

pin), it is necessary to sum the battery charge current,

charger quiescent current and PROG pin current.

Undervoltage Lockout (UVLO)

The undervoltage lockout circuit monitors the input voltage (V above V

) and disables the battery charger until VCC rises

(typically 4V). 200mV of hysteresis prevents

UVLO

oscillations around the trip point. In addition, a differential undervoltage lockout circuit disables the battery charger

3558f

Page 14

LTC3558

APPLICATIONS INFORMATION

when VCC falls to within V

(typically 50mV) of the

DUVLO

BAT voltage.

Suspend Mode

The battery charger can also be disabled by pulling the SUSP pin above 1.2V. In suspend mode, the battery drain current is reduced to 1.5µA and the input current is reduced to 8.5µA.

Charge Cycle Overview

When a battery charge cycle begins, the battery charger ﬁ rst determines if the battery is deeply discharged. If the battery voltage is below V

, typically 2.9V, an automatic

TRKL

trickle charge feature sets the battery charge current to 10% of the full-scale value.

Once the battery voltage is above 2.9V, the battery charger begins charging in constant-current mode. When the battery voltage approaches the 4.2V required to maintain a full charge, otherwise known as the ﬂ oat voltage, the charge current begins to decrease as the battery charger switches into constant-voltage mode.

Trickle Charge and Defective Battery Detection

Any time the battery voltage is below V

, the charger

TRKL

goes into trickle charge mode and reduces the charge current to 10% of the full-scale current. If the battery voltage remains below V

for more than 1/2 hour, the

TRKL

charger latches the bad-battery state, automatically terminates, and indicates via the CHRG pin that the battery was unresponsive. If for any reason the battery voltage rises above V

, the charger will resume charging. Since the

TRKL

charger has latched the bad-battery state, if the battery voltage then falls below V V

RECHRG

ﬁ rst, the charger will immediately assume that

again but without rising past

TRKL

the battery is defective. To reset the charger (i.e., when the dead battery is replaced with a new battery), simply remove the input voltage and reapply it or put the part in and out of suspend mode.

Charge Termination

The battery charger has a built-in safety timer that sets the total charge time for 4 hours. Once the battery voltage rises above V

RECHRG

(typically 4.105V) and the charger

enters constant-voltage mode, the 4-hour timer is started. After the safety timer expires, charging of the battery will discontinue and no more current will be delivered.

Automatic Recharge

After the battery charger terminates, it will remain off, drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below V

RECHRG

(typically 4.105V). In the event that the safety timer is running when the battery voltage falls below V

RECHRG

, it will reset back to zero. To prevent brief excursions below V

RECHRG

must be below V cycle and safety timer will also restart if the V DUVLO cycles low and then high (e.g., V

from resetting the safety timer, the battery voltage

RECHRG

for more than 1.7ms. The charge

UVLO or

is removed

and then replaced) or the charger enters and then exits suspend mode.

Programming Charge Current

The PROG pin serves both as a charge current program pin, and as a charge current monitor pin. By design, the PROG pin current is 1/800th of the battery charge current. Therefore, connecting a resistor from PROG to ground programs the charge current while measuring the PROG pin voltage allows the user to calculate the charge current.

Full-scale charge current is deﬁ ned as 100% of the constant-current mode charge current programmed by the PROG resistor. In constant-current mode, the PROG pin servos to 1V if HPWR is high, which corresponds to charging at the full-scale charge current, or 200mV if HPWR is low, which corresponds to charging at 20% of the fullscale charge current. Thus, the full-scale charge current and desired program resistor for a given full-scale charge current are calculated using the following equations:

800

CHG

PROG

800

CHG

3558f

Page 15

APPLICATIONS INFORMATION

LTC3558

In any mode, the actual battery current can be determined by monitoring the PROG pin voltage and using the following equation:

PROG

= •800

BAT

PROG

Thermal Regulation

To prevent thermal damage to the IC or surrounding components, an internal thermal feedback loop will automatically decrease the programmed charge current if the die temperature rises to approximately 115°C. Thermal regulation protects the battery charger from excessive temperature due to high power operation or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3558 or external components. The beneﬁ t of the LTC3558 battery charger thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions.

Charge Status Indication

The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging, unresponsive battery and battery temperature out of range.

The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. The CHRG pin, which is an open-drain output, can drive an indicator LED through a current limiting resistor for human interfacing, or simply a pull-up resistor for microprocessor interfacing.

To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either a low for charging, a high for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults: unresponsive battery and battery temperature out of range.

When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When the

charge current has dropped to below 10% of the full-scale current, the CHRG pin is released (high impedance). If a fault occurs after the CHRG pin is released, the pin remains high impedance. However, if a fault occurs before the CHRG pin is released, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate enough to make an LED appear to be on or off thus giving the appearance of “blinking”. Each of the two faults has its own unique “blink” rate for human recognition as well as two unique duty cycles for microprocessor recognition.

Table 1 illustrates the four possible states of the CHRG pin when the battery charger is active.

Table 1. CHRG Output Pin

MODULATION

STATUS FREQUENCY

Charging 0Hz 0 Hz (Lo-Z) 100%

< C/10 0Hz 0 Hz (Hi-Z) 0%

BAT

NTC Fault

Bad Battery

35kHz

(BLINK)

FREQUENCY DUTY CYCLE

1.5Hz at 50% 6.25%, 93.75%

6.1Hz at 50% 12.5%, 87.5%

An NTC fault is represented by a 35kHz pulse train whose duty cycle alternates between 6.25% and 93.75% at a

1.5Hz rate. A human will easily recognize the 1.5Hz rate as a “slow” blinking which indicates the out of range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault.

If a battery is found to be unresponsive to charging (i.e., its voltage remains below V

for over 1/2 hour), the

TRKL

CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz “fast” blinking of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault.

Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading.

3558f

Page 16

LTC3558

APPLICATIONS INFORMATION

NTC Thermistor

The battery temperature is measured by placing a negative temperature coefﬁ cient (NTC) thermistor close to the battery pack. The NTC circuitry is shown in Figure 3.

To use this feature, connect the NTC thermistor, R between the NTC pin and ground, and a bias resistor, R from V

to NTC. R

should be a 1% resistor with a

NOM

NTC

NOM

, ,

value equal to the value of the chosen NTC thermistor at 25°C (R25). A 100k thermistor is recommended since thermistor current is not measured by the battery charger and its current will have to be considered for compliance with USB speciﬁ cations.

The battery charger will pause charging when the resistance of the NTC thermistor drops to 0.54 times the

POWER

FAULT

BAT b 2.9V

DUVLO, UVLO AND SUSPEND DISABLE MODE

IF SUSP < 0.4V AND

> 4V AND

> BAT + 130mV?

BATTERY CHARGING SUSPENDED CHRG PULSES

value of R25 or approximately 54k (for a Vishay “Curve 1” thermistor, this corresponds to approximately 40°C). If the battery charger is in constant-voltage mode, the safety timer will pause until the thermistor indicates a return to a valid temperature.

As the temperature drops, the resistance of the NTC thermistor rises. The battery charger is also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For a Vishay “Curve 1” thermistor, this resistance, 325k, corresponds to approximately 0°C. The hot and cold comparators each have approximately 3°C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables all NTC functionality.

CHRG HIGH IMPEDANCE

YES

NTC FAULT

NO FAULT

2.9V < BAT < 4.105V

STANDBY MODE

NO CHARGE CURRENT CHRG HIGH IMPEDANCE

TRICKLE CHARGE MODE

1/10 FULL CHARGE CURRENT CHRG STRONG PULL-DOWN 30 MINUTE TIMER BEGINS

30 MINUTE

TIMEOUT

DEFECTIVE BATTERY

NO CHARGE CURRENT CHRG PULSES

Figure 2. State Diagram of Battery Charger Operation

BAT > 2.9V

CONSTANT CURRENT MODE

FULL CHARGE CURRENT CHRG STRONG PULL-DOWN

CONSTANT VOLTAGE MODE

4-HOUR TERMINATION TIMER BEGINS

BAT DROPS BELOW 4.105V

4-HOUR TERMINATION TIMER RESETS

4-HOUR TIMEOUT

3558 F02

3558f

Page 17

APPLICATIONS INFORMATION

LTC3558

Alternate NTC Thermistors and Biasing

The battery charger provides temperature qualiﬁ ed charging if a grounded thermistor and a bias resistor are connected to the NTC pin. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor).

The upper and lower temperature thresholds can be adjusted by either a modiﬁ cation of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modiﬁ ed but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique are given below.

NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F, used in the following examples, has a nominal value of 100k and follows the Vishay “Curve 1” resistance-temperature characteristic.

In the explanation below, the following notation is used.

R25 = Value of the thermistor at 25°C

NTC|COLD

NTC|HOT

COLD

HOT

NOM

= Value of thermistor at the cold trip point

= Value of the thermistor at the hot trip point

= Ratio of R

NTC|COLD

NTC|HOT

to R25

= Primary thermistor bias resistor (see Figure 3)

R1 = Optional temperature range adjustment resistor (see Figure 4)

The trip points for the battery charger’s temperature qualiﬁ cation are internally programmed at 0.349 • V hot threshold and 0.765 • V

for the cold threshold.

for the

Therefore, the hot trip point is set when:

NTC HOT

NOM NTCHOT

•.•

= 0 349

CC CC

and the cold trip point is set when:

NTC COLD

NOM NTC COLD

Solving these equations for R

•.•

= 0 765

CC CC

NTC|COLD

and R

NTC|HOT

results in the following:

NTC|HOT

= 0.536 • R

NOM

and

NTC|COLD

By setting R in r

HOT

= 3.25 • R

equal to R25, the above equations result

NOM

= 0.536 and r

NOM

= 3.25. Referencing these ratios

COLD

to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40°C and a cold trip point of about 0°C. The difference between the hot and cold trip points is approximately 40°C.

By using a bias resistor, R

, different in value from

NOM

R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the nonlinear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor:

HOT

and r

HOT

0 536

COLD

325

•

are the resistance ratios at the

COLD

de-

NOM

where r

hot and cold trip points. Note that these equations

sired are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60°C hot trip point is desired.

From the Vishay Curve 1 R-T characteristics, r at 60°C. Using the above equation, R

NOM

is 0.2488

HOT

should be set

3558f

Page 18

LTC3558

APPLICATIONS INFORMATION

to 46.4k. With this value of R

, the cold trip point is

NOM

about 16°C. Notice that the span is now 44°C rather than the previous 40°C.

The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 4. The following formulas can be used to compute the values of R

–

NOM

RRr

1 0 536 RR25

COLD HOT

.• – •

NOM

100k

NTC

100k

2 714

NOM HOT

0.765 • V

(NTC RISING)

NTC

0.349 • V

(NTC FALLING)

NOM

•

NTC BLOCK

and R1:

–

TOO_COLD

–

TOO_HOT

For example, to set the trip points to 0°C and 45°C with a Vishay Curve 1 thermistor choose:

3 266 0 4368

Rkk

NOM

.–.

2 714

100 104 2

•.

the nearest 1% value is 105k.

R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k

the nearest 1% value is 12.7k. The ﬁ nal solution is shown in Figure 4 and results in an upper trip point of 45°C and a lower trip point of 0°C.

0.765 • V

NTC

(NTC RISING)

0.349 • V

(NTC FALLING)

–

TOO_COLD

–

TOO_HOT

NOM

105k

12.7k

R 100k

NTC

NTC_ENABLE

0.017 • V

(NTC FALLING)

–

Figure 3. Typical NTC Thermistor Circuit

3558 F03

NTC_ENABLE

0.017 • V

(NTC FALLING)

–

Figure 4. NTC Thermistor Circuit with Additional Bias Resistor

3558f

Page 19

APPLICATIONS INFORMATION

LTC3558

USB and Wall Adapter Power

Although the battery charger is designed to draw power from a USB port to charge Li-Ion batteries, a wall adapter can also be used. Figure 5 shows an example of how to combine wall adapter and USB power inputs. A P-channel MOSFET, MP1, is used to prevent back conduction into the USB port when a wall adapter is present and Schottky diode, D1, is used to prevent USB power loss through the 1k pull-down resistor.

Typically, a wall adapter can supply signiﬁ cantly more current than the 500mA-limited USB port. Therefore, an N-channel MOSFET, MN1, and an extra program resistor are used to increase the maximum charge current to 950mA when the wall adapter is present.

5V WALL

ADAPTER

950mA I

POWER

500mA I

CHG

USB

CHG

MP1

BATTERY

CHARGER

PROG

BAT

Li-Ion BATTERY

current. It is not necessary to perform any worst-case power dissipation scenarios because the LTC3558 will automatically reduce the charge current to maintain the die temperature at approximately 105°C. However, the approximate ambient temperature at which the thermal feedback begins to protect the IC is:

TCP

=°

105

ADJA

TCVVI

=°

105––– ••

A CC BAT BAT JA

()

Example: Consider an LTC3558 operating from a USB port providing 500mA to a 3.5V Li-Ion battery. The ambient temperature above which the LTC3558 will begin to reduce the 500mA charge current is approximately:

TCVVmACW

=°

105 5 3 5 500 68

=°

105 0

=°

––.• • /

–.. • / –75 68 105 51

()()

WCW C C

°=° °

The LTC3558 can be used above 70°C, but the charge current will be reduced from 500mA. The approximate current at a given ambient temperature can be calculated:

1.65k

MN1

Figure 5. Combining Wall Adapter and USB Power

1.74k

3558 F05

Power Dissipation

The conditions that cause the LTC3558 to reduce charge current through thermal feedback can be approximated by considering the power dissipated in the IC. For high charge currents, the LTC3558 power dissipation is approximately:

PVV I

D CC BAT BAT

–•

()

where PD is the power dissipated, VCC is the input supply voltage, V

is the battery voltage, and I

BAT

is the charge

BAT

105 –

–•θ

()

CC BAT JA

Using the previous example with an ambient temperature of 88°C, the charge current will be reduced to approximately:

102

105 88

BAT

IImA

BAT

535 68

()

= 167

°°

–. • / /

VV CWCCA

–

Furthermore, the voltage at the PROG pin will change proportionally with the charge current as discussed in the Programming Charge Current section.

It is important to remember that LTC3558 applications do not need to be designed for worst-case thermal conditions since the IC will automatically reduce power dissipation when the junction temperature reaches approximately 105°C.

3558f

Page 20

LTC3558

APPLICATIONS INFORMATION

Battery Charger Stability Considerations

The LTC3558 battery charger contains two control loops: the constant-voltage and constant-current loops. The constantvoltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1.5µF from BAT to GND. Furthermore, a 4.7µF capacitor with a 0.2Ω to 1Ω series resistor from BAT to GND is required to keep ripple voltage low when the battery is disconnected.

High value capacitors with very low ESR (especially ceramic) reduce the constant-voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22µF may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2Ω to 1Ω of series resistance.

In constant-current mode, the PROG pin is in the feedback loop, not the battery. Because of the additional pole created by the PROG pin capacitance, capacitance on this pin must be kept to a minimum. With no additional capacitance on the PROG pin, the charger is stable with program resistor values as high as 25K. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin is loaded with a capacitance, C to calculate the maximum resistance value for R

PROG

≤

210

, the following equation should be used

PROG

π ••

PROG

Average, rather than instantaneous, battery current may be of interest to the user. For example, if a switching power supply operating in low-current mode is connected in parallel with the battery, the average current being pulled out of the BAT pin is typically of more interest than the instantaneous current pulses. In such a case, a simple RC ﬁ lter can be used on the PROG pin to measure the average battery current as shown in Figure 6. A 10k resistor has been added between the PROG pin and the ﬁ lter capacitor to ensure stability.

USB Inrush Limiting

When a USB cable is plugged into a portable product, the inductance of the cable and the high-Q ceramic input capacitor form an L-C resonant circuit. If there is not much impedance in the cable, it is possible for the voltage at the input of the product to reach as high as twice the USB voltage (~10V) before it settles out. In fact, due to the high voltage coefﬁ cient of many ceramic capacitors (a nonlinearity), the voltage may even exceed twice the USB voltage. To prevent excessive voltage from damaging the LTC3558 during a hot insertion, the soft connect circuit in Figure 7 can be employed.

In the circuit of Figure 7, capacitor C1 holds MP1 off when the cable is ﬁ rst connected. Eventually C1 begins to charge up to the USB input voltage applying increasing gate support to MP1. The long time constant of R1 and C1 prevents the current from building up in the cable too fast thus dampening out any resonant overshoot.

LTC3558

10k

PROG

3558 F06

FILTER

PROG

GND

Figure 6. Isolated Capacitive Load on PROG Pin and Filtering

CHARGE CURRENT MONITOR CIRCUITRY

5V USB

INPUT

MP1

Si2333

C1 100nF

USB CABLE

R1 40k

Figure 7. USB Soft Connect Circuit

C2 10µF

LTC3558

GND

3558 F07

3558f

Page 21

APPLICATIONS INFORMATION

LTC3558

Buck Switching Regulator General Information

The LTC3558 contains a 2.25MHz constant-frequency current mode buck switching regulator that can provide up to 400mA. The switcher can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory or other logic circuitry. The regulator supports 100% duty cycle operation (dropout mode) when the input voltage drops very close to the output voltage and is also capable of operating in Burst Mode operation for highest efﬁ ciencies at light loads (Burst Mode operation is pin selectable). The buck switching regulator also includes soft-start to limit inrush current when powering on, short-circuit current protection, and switch node slew limiting circuitry to reduce radiated EMI.

A MODE pin sets the buck switching regulator in Burst Mode operation or pulse skip operating mode. The regulator is enabled individually through its enable pin. The buck regulator input supply (PV battery pin (BAT) and PV

) should be connected to the

IN1

. This allows the undervoltage

IN2

lockout circuit on the BAT pin to disable the buck regulators when the BAT voltage drops below 2.45V. Do not drive the buck switching regulator from a voltage other than BAT. A 10µF decoupling capacitor from the PV

pin to GND

IN1

is recommended.

Buck Switching Regulator Output Voltage Programming

The buck switching regulator can be programmed for output voltages greater than 0.8V. The output voltage for the buck switching regulator is programmed using a resistor divider from the switching regulator output connected to its feedback pin (FB1), as shown in Figure 8, such that:

= 0.8(1 + R1/R2)

OUT

Typical values for R1 are in the range of 40k to 1M. The capacitor CFB cancels the pole created by feedback resistors and the input capacitance of the FB pin and also helps to improve transient response for output voltages much greater than 0.8V. A variety of capacitor sizes can be used for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between 2pF and 22pF may yield improved transient response if so desired by the user.

Buck Switching Regulator Operating Modes

The buck switching regulator includes two possible operating modes to meet the noise/power needs of a variety of applications.

In pulse skip mode, an internal latch is set at the start of every cycle, which turns on the main P-channel MOSFET

VIN

PWM

CONTROL

MODE

GND

Figure 8. Buck Converter Application Circuit

0.8V

OUT

3558 F08

3558f

Page 22

LTC3558

APPLICATIONS INFORMATION

switch. During each cycle, a current comparator compares the peak inductor current to the output of an error ampliﬁ er. The output of the current comparator resets the internal latch, which causes the main P-channel MOSFET switch to turn off and the N-channel MOSFET synchronous rectiﬁ er to turn on. The N-channel MOSFET synchronous rectiﬁ er turns off at the end of the 2.25MHz cycle or if the current through the N-channel MOSFET synchronous rectiﬁ er drops to zero. Using this method of operation, the error ampliﬁ er adjusts the peak inductor current to deliver the required output power. All necessary compensation is internal to the buck switching regulator requiring only a single ceramic output capacitor for stability. At light loads in pulse skip mode, the inductor current may reach zero on each pulse which will turn off the N-channel MOSFET synchronous rectiﬁ er. In this case, the switch node (SW1) goes high impedance and the switch node voltage will “ring”. This is discontinuous operation, and is normal behavior for a switching regulator. At very light loads in pulse skip mode, the buck switching regulator will automatically skip pulses as needed to maintain output regulation. At high duty cycle (V possible for the inductor current to reverse causing the buck converter to switch continuously. Regulation and low noise operation are maintained but the input supply current will increase to a couple mA due to the continuous gate switching.

During Burst Mode operation, the buck switching regulator automatically switches between ﬁ xed frequency PWM operation and hysteretic control as a function of the load current. At light loads the buck switching regulator controls the inductor current directly and use a hysteretic control loop to minimize both noise and switching losses. During Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck switching regulator then goes into sleep mode, during which the output capacitor provides the load current. In sleep mode, most of the switching regulator’s circuitry is

OUT

> PV

/2) in pulse skip mode, it is

IN1

powered down, helping conserve battery power. When the output voltage drops below a pre-determined value, the buck switching regulator circuitry is powered on and another burst cycle begins. The sleep time decreases as the load current increases. Beyond a certain load current point (about 1/4 rated output load current) the buck switching regulator will switch to a low noise constant-frequency PWM mode of operation, much the same as pulse skip operation at high loads. For applications that can tolerate some output ripple at low output currents, Burst Mode operation provides better efﬁ ciency than pulse skip at light loads.

The buck switching regulator allows mode transition onthe-ﬂ y, providing seamless transition between modes even under load. This allows the user to switch back and forth between modes to reduce output ripple or increase low current efﬁ ciency as needed. Burst Mode operation is set by driving the MODE pin high, while pulse skip mode is achieved by driving the MODE pin low.

Buck Switching Regulator in Shutdown

The buck switching regulator is in shutdown when not enabled for operation. In shutdown, all circuitry in the buck switching regulator is disconnected from the regulator input supply, leaving only a few nanoamps of leakage pulled to ground through a 13k resistor on the switch (SW1) pin when in shutdown.

Buck Switching Regulator Dropout Operation

It is possible for the buck switching regulator’s input voltage to approach its programmed output voltage (e.g., a battery voltage of 3.4V with a programmed output voltage of 3.3V). When this happens, the PMOS switch duty cycle increases until it is turned on continuously at 100%. In this dropout condition, the respective output voltage equals the regulator’s input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor.

3558f

Page 23

APPLICATIONS INFORMATION

LTC3558

Buck Switching Regulator Soft-Start Operation

Soft-start is accomplished by gradually increasing the peak inductor current for each switching regulator over a 500μs period. This allows an output to rise slowly, helping minimize the battery in-rush current required to charge up the regulator’s output capacitor. A soft-start cycle occurs when the buck switcher ﬁ rst turns on, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes using the MODE pin. This allows seamless output operation when transitioning between operating modes.

Buck Switching Regulator Switching Slew Rate Control

The buck switching regulator contains circuitry to limit the slew rate of the switch node (SW1). This circuitry is designed to transition the switch node over a period of a couple of nanoseconds, signiﬁ cantly reducing radiated EMI and conducted supply noise while maintaining high efﬁ ciency.

Buck Switching Regulator Low Supply Operation

An undervoltage lockout (UVLO) circuit on PV down the step-down switching regulators when BAT drops below 2.45V. This UVLO prevents the buck switching regulator from operating at low supply voltages where loss of regulation or other undesirable operation may occur.

shuts

IN1

Buck Switching Regulator Inductor Selection

The buck switching regulator is designed to work with inductors in the range of 2.2µH to 10µH, but for most applications a 4.7µH inductor is suggested. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current which improves transient response time. To maximize efﬁ ciency, choose an inductor with a low DC resistance. For a 1.2V output efﬁ ciency is reduced about 2% for every 100mΩ series resistance at 400mA load current, and about 2% for every 300mΩ series resistance at 100mA load current. Choose an inductor with a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition the inductor should be rated to handle the maximum peak current speciﬁ ed for the buck regulators.

Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher DCR losses, and will not give the best efﬁ ciency. The choice of which style inductor to use often depends more on the price vs size, performance, and any radiated EMI requirements than on what the buck regulator requires to operate.

The inductor value also has an effect on Burst Mode operation. Lower inductor values will cause Burst Mode switching frequency to increase.

3558f

Page 24

LTC3558

APPLICATIONS INFORMATION

Table 2 shows several inductors that work well with the LTC3558 buck switching regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors.

for most applications. For good transient response and stability the output capacitor should retain at least 4μF of capacitance over operating temperature and bias voltage. The buck switching regulator input supply should be bypassed with a 10µF capacitor. Consult manufacturer for detailed information on their selection and speciﬁ ca-

Buck Switching Regulator Input/Output Capacitor Selection

Low ESR (equivalent series resistance) ceramic capacitors should be used at switching regulator outputs as well as the switching regulator input supply. Ceramic capacitor

tions of ceramic capacitors. Many manufacturers now offer very thin (< 1mm tall) ceramic capacitors ideal for use in height-restricted designs. Table 3 shows a list of several ceramic capacitor manufacturers.

Table 3: Recommended Ceramic Capacitor Manufacturers

dielectrics are a compromise between high dielectric constant and stability versus temperature and versus DC bias voltage. The X5R/X7R dielectrics offer the best

Taiyo Yuden (408) 537-4150 www.t-yuden.com

compromise with high dielectric constant and acceptable performance over temperature and under bias. Do not use Y5V dielectrics. A 10µF output capacitor is sufﬁ cient

Table 2. Recommended Inductors for Buck Switching Regulators

INDUCTOR TYPE

DE2818C DE2812C

CDRH3D16 4.7 0.9 110

SD3118 SD3112

LPS3015 4.7 1.1 200

*Typical DCR

(μH)

4.7

MAX I

(A)

1.25

1.15

1.3

0.8

MAX DCR

(mΩ)

72*

130*

162 246

AVX (803) 448-9411 www.avxcorp.com

Murata (714) 852-2001 www.murata.com

TDK (888) 835-6646 www.tdk.com

SIZE IN mm (L × W × H) MANUFACTURER

3 × 2.8 × 1.8 3 × 2.8 × 1.2

4 × 4 × 1.8

3.1 × 3.1 × 1.8

3.1 × 3.1 × 1.2

3 × 3 × 1.5

Toko

www.toko.com

Sumida

www.sumida.com

Cooper

www.cooperet.com

Coilcraft

www.coilcraft.com

3558f

Page 25

APPLICATIONS INFORMATION

LTC3558

Buck-Boost Switching Regulator

The LTC3558 contains a 2.25MHz constant-frequency, voltage mode, buck-boost switching regulator. The regulator provides up to 400mA of output load current. The buck-boost switching regulator can be programmed for a minimum output voltage of 2.75V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or other logic circuitry. To suit a variety of applications, different mode functions allow the user to trade off noise for efﬁ ciency. Two modes are available to control the operation of the buck-boost regulator. At moderate to heavy loads, the constant-frequency PWM mode provides the least noise switching solution. At lighter loads, Burst Mode operation may be selected. Regulation is maintained by an error ampliﬁ er that compares the divided output voltage with a reference and adjusts the compensation voltage accordingly until the FB2 voltage has stabilized at 0.8V. The buck-boost switching regulator also includes soft-start to limit inrush current and voltage overshoot when powering on, short-circuit current protection, and switch node slew limiting circuitry for reduced radiated EMI.

Buck-Boost Regulator PWM Operating Mode

In PWM mode, the voltage seen at the feedback node is compared to a 0.8V reference. From the feedback voltage, an error ampliﬁ er generates an error signal seen at the V

pin. This error signal controls PWM waveforms that

modulate switches A (input PMOS), B (input NMOS), C (output NMOS), and D (output PMOS). Switches A and B operate synchronously, as do switches C and D. If the input voltage is signiﬁ cantly greater than the programmed output voltage, then the regulator will operate in buck mode. In this case, switches A and B will be modulated, with switch D always on (and switch C always off), to stepdown the input voltage to the programmed output. If the input voltage is signiﬁ cantly less than the programmed output voltage, then the converter will operate in boost mode. In this case, switches C and D are modulated, with switch A always on (and switch B always off), to step up the input voltage to the programmed output. If the input voltage is close to the programmed output voltage, then

the converter will operate in four-switch mode. While operating in four-switch mode, switches turn on as per the following sequence: switches A and D

→ switches B and D → switches A and D.

and C

Buck-Boost Regulator Burst Mode Operation

In Burst Mode operation, the switching regulator uses a hysteretic feedback voltage algorithm to control the output voltage. By limiting FET switching and using a hysteretic control loop switching losses are greatly reduced. In this mode, output current is limited to 50mA. While in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck-boost converter then goes into a SLEEP state, during which the output capacitor provides the load current. The output capacitor is charged by charging the inductor until the input current reaches 250mA typical, and then discharging the inductor until the reverse current reaches 0mA typical. This process of bursting current is repeated until the feedback voltage has charged to the reference voltage plus 6mV (806mV typical). In the SLEEP state, most of the regulator’s circuitry is powered down, helping to conserve battery power. When the feedback voltage drops below the reference voltage minus 6mV (794mV typical), the switching regulator circuitry is powered on and another burst cycle begins. The duration for which the regulator operates in SLEEP depends on the load current and output capacitor value. The SLEEP time decreases as the load current increases. The maximum deliverable load current in Burst Mode operation is 50mA typical. The buck-boost regulator may not enter SLEEP if the load current is greater than 50mA. If the load current increases beyond this point while in Burst Mode operation, the output may lose regulation. Burst Mode operation provides a signiﬁ cant improvement in efﬁ ciency at light loads at the expense of higher output ripple when compared to PWM mode. For many noise-sensitive systems, Burst Mode operation might be undesirable at certain times (i.e., during a transmit or receive cycle of a wireless device), but highly desirable at others (i.e., when the device is in low power standby mode).

→ switches A

3558f

Page 26

LTC3558

APPLICATIONS INFORMATION

Buck-Boost Switching Regulator Output Voltage Programming

The buck-boost switching regulator can be programmed for output voltages greater than 2.75V and less than 5.45V. To program the output voltage, a resistor divider is connected between V shown in Figure 9. The output voltage is given by V

and the feedback node (FB2) as

OUT2

= 0.8(1 + R1/R2).

LTC3558

OUT2

FB2

3558 F09

Figure 9. Programming the Buck-Boost Output Voltage Requires a Resistor Divider Connected Between V

OUT2

and FB2

Closing the Feedback Loop

The LTC3558 incorporates voltage mode PWM control. The control to output gain varies with operation region (buck, boost, buck-boost), but is usually no greater than 20. The output ﬁ lter exhibits a double pole response given by:

The output ﬁ lter zero is given by:

FILTER ZERO

where R

ESR

•• •

2 π

is the capacitor equivalent series resistance.

ESR OUT

A troublesome feature in boost mode is the right-half plane zero (RHP), and is given by:

RHPZ

2• • • •π

ILV

OUT OUT

The loop gain is typically rolled off before the RHP zero frequency.

A simple Type I compensation network, as shown in Figure 10, can be incorporated to stabilize the loop, but at the cost of reduced bandwidth and slower transient response. To ensure proper phase margin, the loop requires to be crossed over a decade before the LC double pole.

The unity-gain frequency of the error ampliﬁ er with the Type I compensation is given by:

•• •

FILTER POLE

where C

is the output ﬁ lter capacitor.

OUT

2 π

•• •

OUT

OUT2

0.8V

ERROR

AMP

Figure 10. Error Ampliﬁ er with Type I Compensation

FB2

–

3558 F10

3558f

Page 27

APPLICATIONS INFORMATION

LTC3558

Most applications demand an improved transient response to allow a smaller output ﬁ lter capacitor. To achieve a higher bandwidth, Type III compensation is required. Two zeros are required to compensate for the double-pole response. Type III compensation also reduces any V

overshoot

OUT2

seen during a start-up condition. A Type III compensation circuit is shown in Figure 11 and yields the following transfer function:

VRCC

OUT22

sR C s R R C

1221 133

+++

()[()]

11 2

()

• sssRCC sRC1212133+

⎡

⎣

+(|| )( )

⎤

⎦

A Type III compensation network attempts to introduce a phase bump at a higher frequency than the LC double pole. This allows the system to cross unity gain after the LC double pole, and achieve a higher bandwidth. While attempting to cross over after the LC double pole, the system must still cross over before the boost right-half plane zero. If unity gain is not reached sufﬁ ciently before the right-half plane zero, then the –180° of phase lag from the LC double pole combined with the –90° of phase lag from the right-half plane zero will result in negating the phase bump of the compensator.

The compensator zeros should be placed either before or only slightly after the LC double pole such that their positive phase contributions offset the –180° that occurs

at the ﬁ lter double pole. If they are placed at too low of a frequency, they will introduce too much gain to the system and the crossover frequency will be too high. The two high frequency poles should be placed such that the system crosses unity gain during the phase bump introduced by the zeros and before the boost right-half plane zero and such that the compensator bandwidth is less than the bandwidth of the error amp (typically 900kHz). If the gain of the compensation network is ever greater than the gain of the error ampliﬁ er, then the error ampliﬁ er no longer acts as an ideal op amp, and another pole will be introduced at the same point.

Recommended Type III compensation components for a

3.3V output are: R1: 324kΩ

: 105kΩ

C1: 10pF

R2: 15k

C2: 330pF R3: 121kΩ

C3: 33pF

OUT

: 22F

: 2.2H

Figure 11. Error Ampliﬁ er with Type III Compensation

ERROR

AMP

OUT2

0.8V

FB2

–

3558 F11

3558f

Page 28

LTC3558

APPLICATIONS INFORMATION

Input Current Limit

The input current limit comparator will shut the input PMOS switch off once current exceeds 700mA typical. Before the switch current limit, the average current limit amp (620mA typical) will source current into the feedback pin to drop the output voltage. The input current limit also protects against a short-circuit condition at the V

OUT2

pin.

Reverse Current Limit

The reverse current limit comparator will shut the output PMOS switch off once current returning from the output exceeds 450mA typical.

Output Overvoltage Protection

If the feedback node were inadvertently shorted to ground, then the output would increase indeﬁ nitely with the maximum current that could be sourced from the input supply. The buck-boost regulator protects against this by shutting off the input PMOS if the output voltage exceeds a 5.75V maximum.

Buck-Boost Regulator Soft-Start Operation

Soft-start is accomplished by gradually increasing the reference voltage over a 500µs typical period. A softstart cycle occurs whenever the buck-boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output operation when transitioning between Burst Mode operation and PWM mode operation.

Buck-Boost Switching Regulator Inductor Selection

The buck-boost switching regulator is designed to work with inductors in the range of 1µH to 5µH. For most applications, a 2.2µH inductor will sufﬁ ce. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time. To maximize efﬁ ciency, choose an inductor with a low DC resistance and a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition, the inductor current should be rated to handle up to the peak current speciﬁ ed for the buck-boost regulator.

The inductor value also affects Burst Mode operation. Lower inductor values will cause Burst Mode switching frequencies to increase.

Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses and will not give the best efﬁ ciency.

Table 4 shows some inductors that work well with the buck-boost regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors.

Table 4. Recommended Inductors for the Buck-Boost Switching Regulator.

INDUCTOR TYPE

DB3018C D312C DE2812C DE2812C

CDRH3D16 2.2 1.2 72

SD12 2.2 1.8 74

*Typical DCR

(μH)

2.4

2.2 2

2.7

MAX I

(A)

1.31

1.14

1.4

1.2

MAX DCR

(mΩ)

140

81 87

SIZE IN mm (L × W × H) MANUFACTURER

3.8 × 3.8 × 1.4

3.6 × 3.6 × 1.2 3 × 3.2 × 1.2 3 × 3.2 × 1.2

4 × 4 × 1.8

5.2 × 5.2 × 1.2

Toko

www.toko.com

Sumida

www.sumida.com

Cooper

www.cooperet.com

3558f

Page 29

APPLICATIONS INFORMATION

LTC3558

Buck-Boost Switching Regulator Input/Output Capacitor Selection

Low ESR (equivalent series resistance) ceramic capacitors should be used at both the buck-boost regulator input

) and the output (V

(PV

IN2

input be bypassed with a 10µF capacitor. The output should be bypassed with at least a 10µF capacitor if using Type I compensation and 22µF if using Type III compensation.

The same selection criteria apply for the buck-boost regulator input and output capacitors as described in the Buck Switching Regulator Input/Output Capacitor Selection section.

). It is recommended that the

OUT2

PCB Layout Considerations

In order to deliver maximum charge current under all conditions, it is critical that the backside of the LTC3558 be soldered to the PC board ground.

The LTC3558 has dual switching regulators. As with all switching regulators, care must be taken while laying out a PC board and placing components. The input decoupling capacitors, the output capacitor and the inductors must all be placed as close to the pins as possible and on the same side of the board as the LTC3558. All connections must also be made on the same layer. Place a local unbroken ground plane below these components. Avoid routing noisy high frequency lines such as those that connect to switch pins over or parallel to lines that drive high impedance inputs.

3558f

Page 30

LTC3558

TYPICAL APPLICATIONS

USB

(4.3V TO 5.5V)

OR AC ADAPTER

UP TO 500mA

510

110k

28.7K

1.74k

100k (NTC) NTH50603NO1

DIGITAL

CONTROL

10µF

NTC

CHRG

PROG

SUSP

HPWR

EN1

EN2

MODE

GND

LTC3558

GND2

(EXPOSED

PAD)

BAT

IN1

IN2

SW1

FB1

SWAB2

SWCD2

OUT2

FB2

3558 TA02

SINGLE

10µF

4.7µH

806k

649k

2.2µH

619k

200k

15k

150pF

4.7µF

1.8V AT 400mA

10pF

3.3V AT 400mA

Li-lon CELL (2.7V TO 4.2V)

10µF

Figure 12. Li-Ion to 3.3V at 400mA, 1.8V at 400mA and USB-Compatible Battery Charger

As shown in Figure 12, the LTC3558 can be operated with no battery connected to the BAT pin. A 1Ω resistor in series with a 4.7µF capacitor at the BAT pin ensures battery charger stability. 10µF V

decoupling capacitors

are required for proper operation of the DC/DC converters. A three-resistor bias network for NTC sets hot and cold trip points at approximately 55°C and 0°C.

The battery can be charged with up to 950mA of charge current when powered from a 5V wall adaptor, as shown

in Figure 13. CHRG has a LED to provide a user with a visual indication of battery charge status. The buck-boost regulator starts up only after V

is up to approximately

OUT1

0.7V. This provides a sequencing function which may be desirable in applications where a microprocessor needs to be powered up before peripherals. A Type III compensation network improves the transient response of the buck-boost switching regulator.

3558f

Page 31

PACKAGE DESCRIPTION

3.50 ± 0.05 (4 SIDES)

1.65 ± 0.05

2.10 ± 0.05

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED

3.00 ± 0.10 (4 SIDES)

PIN 1 TOP MARK (NOTE 6)

UD Package

20-Lead Plastic QFN (3mm × 3mm)

(Reference LTC DWG # 05-08-1720 Rev A)

0.70 ±0.05

PACKAGE OUTLINE

0.20 ±0.05

0.40 BSC

0.75 ± 0.05

R = 0.05

1.65 ± 0.10 (4-SIDES)

R = 0.115

TYP

BOTTOM VIEW—EXPOSED PAD

TYP

19 20

LTC3558

PIN 1 NOTCH R = 0.20 TYP OR 0.25 × 45° CHAMFER

0.40 ± 0.10

0.200 REF

NOTE:

1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE

0.00 – 0.05

(UD20) QFN 0306 REV A

0.20 ± 0.05

0.40 BSC

3558f

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

Page 32

LTC3558

TYPICAL APPLICATIONS

5V WALL

ADAPTER

510

887

100k

100k (NTC)

CONTROL

DIGITAL

1µF

NTC

CHRG

PROG

SUSP HPWR MODE EN1

EN2

GND

LTC3558

GND2

(EXPOSED

PAD)

BAT

SW1

FB1

SWAB2

SWCD2

OUT2

FB2

UP TO 950mA

IN1

IN2

4.7µH

2.2µH

324k

105k 15k

10µF

324k

649k

3.3V AT 400mA

121k

33pF

330pF

1.2V AT 400mA

10pF

22µF

10pF

3558 TA03

SINGLE Li-lon CELL (2.7V TO 4.2V)

10µF

Figure 13. Battery Charger Can Charge a Battery with Up to 950mA When Powered From a Wall Adapter

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BATT(MIN)

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BAT

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

3558f

LT 0408 • PRINTED IN USA