Linear Technology LTC3129-1 User Manual

15V, 200mA Synchronous

Buck-Boost DC/DC Converter

with 1.3µA Quiescent Current

FeaTures DescripTion

n

Regulates V

n

Wide VIN Range: 2.42V to 15V, 1.92V to 15V After

Above, Below or Equal to V

OUT

IN

Start-Up (Bootstrapped)

n

Fixed Output Voltage with Eight User-Selectable

Settings from 2.5V to 15V

n

200mA Output Current in Buck Mode

n

Single Inductor

n

1.3µA Quiescent Current

n

Programmable Maximum Power Point Control

n

1.2MHz Ultralow Noise PWM

n

Current Mode Control

n

Pin Selectable Burst Mode® Operation

n

Up to 95% Efficiency

n

Accurate RUN Pin Threshold

n

Power Good Indicator

n

10nA Shutdown Current

n

Thermally Enhanced 3mm × 3mm QFN and

16-Lead MSOP Packages

applicaTions

n

Industrial Wireless Sensor Nodes

n

Post-Regulator for Harvested Energy

n

Solar Panel Post-Regulator/Charger

n

Intrinsically Safe Power Supplies

n

Wireless Microphones

n

Avionics-Grade Wireless Headsets

The LT C®3129-1 is a high efficiency, 200mA buck-boost
DC/DC converter with a wide VIN and V
includes an accurate RUN pin threshold to allow predict­able regulator turn-on and a maximum power point control
(MPPC) capability that ensures maximum power extraction
from non-ideal power sources such as photovoltaic panels.

The LTC3129-1 employs an ultralow noise, 1.2MHz PWM
switching architecture that minimizes solution footprint by
allowing the use of tiny, low profile inductors and ceramic
capacitors. Built-in loop compensation and soft-start
simplify the design. For high efficiency operation at light
loads, automatic Burst Mode operation can be selected,
reducing the quiescent current to just 1.3µA. To further
reduce part count and improve light load efficiency, the
LTC3129-1 includes an internal voltage divider to provide
eight selectable fixed output voltages.

Additional features include a power good output, less than
10nA of shutdown current and thermal shutdown.

The LTC3129-1 is available in thermally enhanced
3mm×3mm QFN and 16-lead MSOP packages. For an
adjustable output voltage, see the functionally equivalent
LTC3129.

L, LT , LT C , LT M , Linear Technology, the Linear logo and Burst Mode are registered trademarks

Technology Corporation. All other trademarks are the property of their respective owners.

of Linear

LTC3129-1

range. It

OUT

Typical applicaTion

22nF

10µH

BST1

V

2.42V TO 15V

AA OR AAA
BATTERIES

IN

10µF

V

CC

SW1 SW2

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

LTC3129-1

GND

22nF

BST2

V

OUT

PGOOD

V

CC

PGND

31291 TA01a

For more information www.linear.com/LTC3129-1

10µF

2.2µF

V

OUT

5V AT

< V

100mA V

IN

200mA VIN > V

OUT
OUT

Efficiency and Power Loss vs Load

100

EFFICIENCY

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

V

= 5V

OUT

0

0.01

0.1 100 1000101
OUTPUT CURRENT (mA)

POWER LOSS

VIN = 2.5V
V
V
V

= 3.6V

IN

= 5V

IN

= 15V

IN

3129 TA01b

1000

100

POWER LOSS (mW)

10

1

0.1

0.01

31291fc

1

LTC3129-1

TOP VIEW

16-LEAD (3mm × 3mm) PLASTIC QFN

BST1

PGOOD

MPPC

PGOOD

TOP VIEW

16-LEAD PLASTIC MSOP

absoluTe MaxiMuM raTings

(Notes 1, 8)

VIN, V

SW1 DC Voltage .............................. –0.3V to (V

SW2 DC Voltage............................–0.3V to (V

Voltages ..................................... –0.3V to 18V

OUT

IN

OUT

+ 0.3V)

+ 0.3V)

SW1, SW2 Pulsed (<100ns) Voltage ..............–1V to 19V

BST1 Voltage ..................... (SW1 – 0.3V) to (SW1 + 6V)

BST2 Voltage .....................(SW2 – 0.3V) to (SW2 + 6V)

RUN, PGOOD Voltage ................................. –0.3V to 18V

pin conFiguraTion

SW1

PGND

SW2

BST2

16 15 14 13

V

VS3

VS2

12

OUT

11

PWM

10

VS1

9

1

V

2

IN

V

CC

RUN

T

= 125°C, θJC = 7.5°C/W, θJA = 68°C/W (NOTE 6)

JMAX

EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB

3

4

5 6 7 8

MPPC

UD PACKAGE

17

PGND

GND

, PWM, MPPC, VS1, VS2,

V

CC

VS3 Voltages ............................................... –0.3V to 6V

PGOOD Sink Current .............................................. 15mA

Operating Junction Temperature Range

(Notes 2, 5) ............................................ –40°C to 125°C

Storage Tempera ture Range .................. –65°C to 150°C

MSE Lead Temperature (Soldering, 10 sec) .......... 300°C

1

V

CC

2

RUN

3
4

GND

5

VS3

6

VS2

7

VS1

8

PWM

T

= 125°C, θJC = 10°C/W, θJA = 40°C/W (NOTE 6)

JMAX

EXPOSED PAD (PIN 17) IS PGND, MUST BE SOLDERED TO PCB

17

PGND

MSE PACKAGE

V

16

IN

15

BST1

14

SW1

13

PGND

12

SW2

11

BST2

10

V

OUT

9

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC3129EUD-1#PBF LTC3129EUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C

LTC3129IUD-1#PBF LTC3129IUD-1#TRPBF LGDS 16-Lead (3mm × 3mm) Plastic QFN –40°C to 125°C

LTC3129EMSE-1#PBF LTC3129EMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C

LTC3129IMSE-1#PBF LTC3129IMSE-1#TRPBF 31291 16-Lead Plastic MSOP –40°C to 125°C

Consult LT C Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.

2

31291fc

For more information www.linear.com/LTC3129-1

LTC3129-1

elecTrical characTerisTics

The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, V

PARAMETER CONDITIONS MIN TYP MAX UNITS

Start-Up Voltage

V

IN

Input Voltage Range V

UVLO Threshold (Rising) VCC > 2.42V (Back-Driven)

V

IN

UVLO Hysteresis

V

IN

Voltages VS1 = VS2 = VS3 = 0V

V

OUT

> 2.42V (Back-Driven)

CC

VS1 = VCC, VS2 = VS3 = 0V
VS2 = VCC, VS1 = VS3 = 0V
VS1 = VS2 = VCC, VS3 = 0V
VS1 = VS2 = 0V, VS3 = VCC
VS2 = 0V, VS1 = VS3 = VCC
VS1 = 0V, VS2 = VS3 = VCC

Quiescent Current (V

Quiescent Current (V

VS1 = VS2 = VS3 = V

) – Shutdown RUN = 0V, Including Switch Leakage 10 100 nA

IN

) UVLO Either VIN or VCC Below Their UVLO Threshold, or

IN

CC

RUN Below the Threshold to Enable Switching

, V

> V

Quiescent Current – Burst Mode Operation Measured on V

PWM = 0V, RUN = V

N-Channel Switch Leakage on V

and V

IN

OUT

SW1 = 0V, VIN = 15V
SW2 = 0V, V

OUT

IN

= 15V

OUT

IN

REG

RUN = 0V

N-Channel Switch On-Resistance V

Inductor Average Current Limit V

= 4V 0.75 Ω

CC

> UV Threshold (Note 4)

OUT

V

< UV Threshold (Note 4)

OUT

Inductor Peak Current Limit (Note 4)

< V

Maximum Boost Duty Cycle V

OUT

as Set by VS1-VS3. Percentage of

REG

Period SW2 is Low in Boost Mode (Note 7)

> V

Minimum Duty Cycle V

OUT

as Set by VS1-VS3. Percentage of

REG

Period SW1 is High in Buck Mode (Note 7)

Switching Frequency PWM = V

CC

SW1 and SW2 Minimum Low Time (Note 3) 90 ns

MPPC Voltage

MPPC Input Current MPPC = 5V 1 10 nA

RUN Threshold to Enable V

RUN Threshold to Enable Switching (Rising) V

CC

CC

> 2.4V

RUN (Switching) Threshold Hysteresis 50 80 120 mV

RUN Input Current RUN = 15V 1 10 nA

VS1, VS2, VS3 Input High

VS1, VS2, VS3 Input Low

VS1, VS2, VS3 Input Current VS1, VS2, VS3 = V

= 5V 1 10 nA

CC

PWM Input High

PWM Input Low

PWM Input Current PWM = 5V 0.1 1 µA

Soft-Start Time 3 ms

Voltage VIN > 4.85V

V

CC

Dropout Voltage (VIN – VCC) VIN = 3.0V, Switching

V

CC

VIN = 2.0V (VCC in UVLO)

l

l

1.92 15 V

l

1.8 1.9 2.0 V

l

80 100 130 mV

l

2.425

l

3.2175

l

3.998

l

4.875

l

6.727

l

7.995

l

11.64

l

14.50

2.25 2.42 V

2.5

3.3

4.1

5.0

6.9

8.2
12

15.0

1.9 3 µA

1.3 2.0 µA

10 50 nA

l

220

l

80

l

400 500 680 mA

l

85 89 95 %

l

l

1.0 1.2 1.4 MHz

l

1.12 1.175 1.22 V

l

0.5 0.9 1.15 V

l

1.16 1.22 1.28 V

l

1.2 V

l

l

1.6 V

l

l

3.4 4.1 4.7 V

275
130

35

0

= 5V.

OUT

2.575

3.383

4.203

5.125

7.073

8.405

12.40

15.50

350
200

0 %

0.4 V

0.5 V

60

2

mA
mA

mV
mV

V
V
V
V
V
V
V
V

For more information www.linear.com/LTC3129-1

31291fc

3

LTC3129-1

elecTrical characTerisTics

The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). Unless otherwise noted, VIN = 12V, V

PARAMETER CONDITIONS MIN TYP MAX UNITS

UVLO Threshold (Rising)

V

CC

UVLO Hysteresis 60 mV

V

CC

Current Limit VCC = 0V

V

CC

Back-Drive Voltage (Maximum)

V

CC

Input Current (Back-Driven) VCC = 5.5V (Switching) 2 4 mA

V

CC

Leakage to VIN if VCC>V

V

CC

UV Threshold (Rising)

V

OUT

UV Hysteresis 150 mV

V

OUT

Current – Shutdown RUN = 0V, V

V

OUT

Current – Sleep PWM = 0V, V

V

OUT

Current – Active PWM = VCC, V

V

OUT

IN

PGOOD Threshold, Falling Referenced to Programmed V

PGOOD Hysteresis Referenced to Programmed V

PGOOD Voltage Low I

VCC = 5.5V, VIN = 1.8V, Measured on V

= 15V Including Switch Leakage 10 100 nA

OUT

≥ V

OUT

REG

= 15V (Note 4) 5 9 µA

OUT

OUT

OUT

= 1mA 250 300 mV

SINK

IN

Voltage –5.5 –7.5 –10 %

Voltage 2.5 %

PGOOD Leakage PGOOD = 15V 1 50 nA

l

2.1 2.25 2.42 V

l

4 20 60 mA

l

–27 µA

l

0.95 1.15 1.35 V

V

/27 µA

OUT

= 5V.

OUT

5.5 V

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.

Note 2: The LTC3129-1 is tested under pulsed load conditions such
that T

≈ TA. The LTC3129E-1 is guaranteed to meet specifications

J

from 0°C to 85°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3129I-1 is guaranteed over the full –40°C to 125°C operating junction
temperature range. The junction temperature (T
the ambient temperature (T

, in °C) and power dissipation (PD, in watts)

A

, in °C) is calculated from

J

according to the formula:
T

= TA + (PD • θJA),

J

where θ

(in °C/W) is the package thermal impedance.

JA

Note that the maximum ambient temperature consistent with these
specifications is determined by specific operating conditions in
conjunction with board layout, the rated thermal package thermal
resistance and other environmental factors.

Note 3: Specification is guaranteed by design and not 100% tested in
production.

Note 4: Current measurements are made when the output is not switching.
Note 5: This IC includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction
temperature

will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may result in device degradation or failure.

Note 6: Failure to solder the exposed backside of the package to the PC
board ground plane will result in a much higher thermal resistance.

Note 7: Switch timing measurements are made in an open-loop test
configuration. Timing in the application may vary somewhat from these
values due to differences in the switch pin voltage during non-overlap
durations when switch pin voltage is influenced by the magnitude and
duration of the inductor current.

Note 8: Voltage transients on the switch pin(s) beyond the DC limits
specified in the Absolute Maximum Ratings are non-disruptive to normal
operation when using good layout practices as described elsewhere in the
data sheet and Application Notes and as seen on the product demo board.

4

31291fc

For more information www.linear.com/LTC3129-1

Typical perForMance characTerisTics

31291 G01

31291 G02

31291 G03

31291 G04

31291 G04b

31291 G04a

LTC3129-1

= 25°C, unless otherwise noted.

T

A

Efficiency, V

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

Power Loss, V

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

0.01

= 2.5V

OUT

BURST

PWM

OUT

BURST

100.1

= 3.3V

100.1

OUTPUT CURRENT (mA)

PWM

OUTPUT CURRENT (mA)

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

Power Loss, V

1000

100

10

1

POWER LOSS (mW)

0.1

10001 100

0.01

0.01

PWM

OUTPUT CURRENT (mA)

Efficiency, V

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

10001 100

0.01

BURST

OUTPUT CURRENT (mA)

= 2.5V Efficiency, V

OUT

BURST

OUT

100.1

= 4.1V

100.1

PWM

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

10001 100

10001 100

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

0.01

OUT

BURST

OUTPUT CURRENT (mA)

Power Loss, V

PWM

OUTPUT CURRENT (mA)

= 3.3V

= 4.1V

OUT

100.1

BURST

100.1

PWM

VIN = 2.5V
V
V
V
V

VIN = 2.5V
V
V
V
V

= 3.6V

IN

= 5V

IN

= 10V

IN

= 15V

IN

= 3.6V

IN

= 5V

IN

= 10V

IN

= 15V

IN

10001 100

10001 100

Efficiency, V

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

= 5V Power Loss, V

OUT

BURST

PWM

OUTPUT CURRENT (mA)

100.1

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

31291 G05

= 5V

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

10001 100

0.01

OUT

PWM

BURST

100.1

OUTPUT CURRENT (mA)

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

31291 G06

10001 100

For more information www.linear.com/LTC3129-1

Efficiency, V

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

= 6.9V

OUT

BURST

PWM

OUTPUT CURRENT (mA)

100.1

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

31291 G06a

31291fc

10001 100

5

LTC3129-1

31291 G06b

31291 G06c

31291 G06d

31291 G07

31291 G09

31291 G08

31291 G12

Typical perForMance characTerisTics

Power Loss, V

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

0.01

Efficiency, V

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

PWM

OUTPUT CURRENT (mA)

BURST

OUTPUT CURRENT (mA)

= 6.9V Efficiency, V

OUT

100

90

80

70

60

50

BURST

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

10001 100

10001 100

100.1

= 12V Power Loss, V

OUT

PWM

100.1

40

EFFICIENCY (%)

30

20

10

0

0.01

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

0.01

= 8.2V Power Loss, V

OUT

BURST

PWM

OUTPUT CURRENT (mA)

PWM

OUTPUT CURRENT (mA)

100.1

= 12V Efficiency, V

OUT

BURST

100.1

TA = 25°C, unless otherwise noted.

1000

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

100

10

1

POWER LOSS (mW)

0.1

0.01

10001 100

10001 100

0.01

100

90

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

0.01

PWM

OUTPUT CURRENT (mA)

BURST

OUTPUT CURRENT (mA)

OUT

OUT

BURST

= 15V

= 8.2V

100.1

PWM

100.1

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

10001 100

10001 100

1000

100

10

1

POWER LOSS (mW)

0.1

0.01

0.01

6

= 15V

OUT

PWM

BURST

OUTPUT CURRENT (mA)

100.1

VIN = 2.5V

= 3.6V

V

IN

= 5V

V

IN

= 10V

V

IN

= 15V

V

IN

31291 G10

Maximum Output Current
vs V

and V

250

IN

200

150

(mA)

OUT

I

100

50

0

10001 100

2

OUT

VIN (V)

V

= 2.5V

OUT

= 3.3V

V

OUT

= 4.1V

V

OUT

= 5V

V

OUT

= 6.9V

V

OUT

= 8.2V

V

OUT

= 12V

V

OUT

= 15V

V

OUT

1510 1411 1285 96 7

133 4

31291 G11

No Load Input Current
vs VIN and V

5

4

3

(µA)

IN

I

2

1

0

2

OUT

(PWM = 0V)Power Loss, V

VIN (V)

V

= 2.5V

OUT

V

= 3.3V

OUT

V

= 4.1V

OUT

V

= 5V

OUT

V

= 6.9V

OUT

V

= 8.2V

OUT

V

= 12V

OUT

V

= 15V

OUT

12 14

16104 86

31291fc

For more information www.linear.com/LTC3129-1

LTC3129-1

R

(Ω)

1.3

31291 G14

130

CHANGE IN V

(%)

1.0

31291 G15

130

DROPOUT (mV)

60

31291 G20

130

DROPOUT (mV)

60

31291 G21

4

31291 G17

31291 G18

Typical perForMance characTerisTics

Burst Mode Threshold
vs V

and V

80

70

60

50

40

LOAD (mA)

30

20

10

0

2

IN

OUT

4

VIN (V)

Accurate RUN Threshold
vs Temperature (Normalized to 25°C)

2

1

0

–1

CHANGE IN RUN THRESHOLD (%)

–2

–45

–20

TEMPERATURE (°C)

V

= 2.5V

OUT

V

= 3.3V

OUT

V

= 4.1V

OUT

V

= 5V

OUT

V

= 6.9V

OUT

V

= 8.2V

OUT

V

= 12V

OUT

V

= 15V

OUT

1610 141286

3129 G13

13055 10580305

1.2

1.1

1.0

0.9

0.8

DS(ON)

0.7

0.6

0.5

0.4

Switch R

–45

–20

VCC = 2.5V
V

CC

V

CC

V

CC

vs Temperature

DS(ON)

= 3V
= 4V
= 5V

TEMPERATURE (°C)

55 10580305

Average Input Current Limit
vs MPPC Voltage

100

90

80

70

60

50

40

30

20

10

PERCENTAGE OF FULL INPUT CURRENT (%)

0

1.13

1.135
MPPC PIN VOLTAGE (V)

TA = 25°C, unless otherwise noted.

Output Voltage vs Temperature
(Normalized to 25°C)

0.5

OUT

0

–0.5

–1.0

–20

–45

TEMPERATURE (°C)

Maximum Output vs Temperature
(Normalized to 25°C)

15

10

5

0

–5

–10

CHANGE IN MAXIMUM OUTPUT CURRENT (%)

1.171.1651.161.1551.145 1.151.14

–15

–45

–20

TEMPERATURE (°C)

55 10580305

13055 10580305

31291 G19

VCC Dropout Voltage vs Temperature
(PWM Mode, Switching)

50

40

30

20

10

0

–45

–20

TEMPERATURE (°C)

55 10580305

VCC Dropout Voltage vs VIN
(PWM Mode, Switching)

50

40

30

20

10

0

2

2.25

For more information www.linear.com/LTC3129-1

3 3.5 3.753.252.752.5

VIN (V)

SW2

5V/DIV

SW1

5V/DIV

200mA/DIV

Fixed Frequency PWM
Waveforms

I

L

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 200mA

I

OUT

500ns/DIV

31291 G22

31291fc

7

LTC3129-1

Typical perForMance characTerisTics

Burst Mode Waveforms

SW1

5V/DIV

SW2

5V/DIV

I

L

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 5mA

I

OUT

= 22µF

C

OUT

50µs/DIV

Step Load Transient Response in
Fixed Frequency

V

OUT

V

OUT

20mV/DIV

200mA/DIV

V

OUT

5V/DIV

V

5V/DIV

Fixed Frequency Ripple on V

I

L

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 200mA

I

OUT

= 10µF

C

OUT

200ns/DIV

Start-Up Waveforms

CC

OUT

200mA/DIV

31291 G23

100mV/DIV

TA = 25°C, unless otherwise noted.

Burst Mode Ripple on V

V

OUT

100mV/DIV

I

L

100mA/DIV

31291 G24

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 5mA

I

OUT

= 22µF

C

OUT

100µs/DIV

Step Load Transient Response in
Burst Mode Operation

V

OUT

100mV/DIV

OUT

31291 G25

RUN

5V/DIV

I

VIN

200mA/DIV

VIN = 7V

= 5V

V

OUT

= 50mA

I

OUT

= 22µA

C

OUT

1ms/DIV

PGOOD Response to a Drop
On V

OUT

PGOOD

2V/DIV

V

OUT

2V/DIV

V

= 5V

OUT

1ms/DIV

31291 G26

I

VOUT

100mA/DIV

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 10µF

C

OUT

= 50mA to 150mA STEP

I

OUT

31291 G29

500µs/DIV

V

OUT

2V/DIV

V

2V/DIV

I

VOUT

100mA/DIV

I

VOUT

100mA/DIV

31291 G27

L = 10µH

= 7V

V

IN

= 5V

V

OUT

= 22µF

C

OUT

= 5mA to 125mA STEP

I

OUT

MPPC Response to a Step Load

IN

OUT

2ms/DIV

= 22µF

VIN = 5V

OC

V

SET TO 3.5V

MPPC

= 22µF, RIN = 10Ω,

C

IN

= 5V, C

V

OUT

= 25mA to 125mA STEP

I

OUT

500µs/DIV

31291 G30

31291 G28

8

31291fc

For more information www.linear.com/LTC3129-1

LTC3129-1

pin FuncTions

(QFN/MSOP)

BST1 (Pin 1/Pin 15): Boot-Strapped Floating Supply for

High Side NMOS Gate Drive. Connect to SW1 through a
22nF capacitor, as close to the part as possible. The value

is not critical. Any value from 4.7nF to 47nF may be used.

(Pin 2/Pin 16): Input Voltage for the Converter. Connect

V

IN

a minimum of 4.7µF ceramic decoupling capacitor from
this pin to the ground plane, as close to the pin as possible.

(Pin 3/Pin 1): Output Voltage of the Internal Voltage

V

CC

Regulator. This is the supply pin for the internal circuitry.
Bypass this output with a minimum of 2.2µF ceramic ca
pacitor close to the pin. This pin may be back-driven by
an external supply, up to a maximum of 5.5V.

RUN (Pin 4/Pin 2): Input to the Run Comparator. Pull
this pin above 1.1V to enable the V

regulator and above

CC

1.28V to enable the converter. Connecting this pin to a
resistor divider from V

start threshold higher than the 1.8V (typical) VIN UVLO

V

IN

threshold. In this case, the typical V
determined by V

MPPC (Pin 5/

IN

Pin 3): Maximum Power Point Control

to ground allows programming a

IN

turn-on threshold is

IN

= 1.22V • [1+(R3/Pin R4)] (see Figure 2).

Programming Pin. Connect this pin to a resistor divider
from V
If the V

to ground to enable the MPPC functionality.

IN

load is greater than what the power source

OUT

can provide, the MPPC will reduce the inductor current
to regulate V

• [1 + (R5/R6)] (see Figure 3). By setting the V

to a voltage determined by: VIN = 1.175V

IN

regula-

IN

tion voltage appropriately, maximum power transfer from
the limited source is assured. Note this pin is very noise
sensitive, therefore minimize trace length and stray capaci
tance. Please refer to the Applications Information section
for more detail on programming the MPPC for different
sources. If this function is not needed, tie the pin to V

CC

GND (Pin 6/Pin 4): Signal Ground. Provide a short direct
PCB path between GND and the ground plane where the
exposed pad is soldered.

VS3 (Pin 7/Pin 5): Output Voltage Select Pin. Connect this
pin to ground or V

to program the output voltage (see

CC

Table 1). This pin should not float or go below ground.
If this pin is externally driven above V

, a 1M resistor

CC

should be added in series.

VS2 (Pin 8/Pin 6): Output Voltage Select Pin.
pin to ground or V

to program the output voltage (see

CC

Connect this

Table 1). This pin should not float or go below ground.

VS1 (Pin 9/Pin 7): Output Voltage Select Pin. Connect this

-

pin to ground or V

to program the output voltage (see

CC

Table 1). This pin should not float or go below ground.

PWM (Pin 10/Pin 8): Mode Select Pin.

PWM = Low (ground): Enables automatic Burst Mode

operation.

PWM = High (tie to V

): Fixed frequency PWM

CC

operation.

This pin should not be allowed to float. It has internal 5M
pull-down resistor.

PGOOD (Pin 11/Pin 9): Open drain output that pulls to
ground when FB drops too far below its regulated voltage.
Connect a pull-up resistor from this pin to a positive sup
ply. This pin can sink up to the absolute maximum rating
of 15mA when low. Refer to the Operation section of the
data sheet for more detail.

(Pin 12/Pin 10): Output voltage of the converter, set

V

OUT

by the VS1-VS3 programming pins according to Table 1.
Connect a minimum value of 4.7µF ceramic capacitor from

­this pin to the ground plane, as close to the pin as possible.

BST2 (Pin 13/Pin 11): Boot-Strapped

.

High Side NMOS Gate Drive. Connect to SW2 through a

Floating Supply for

22nF capacitor, as close to the part as possible. The value

is not critical. Any value from 4.7nF to 47nF may be used

SW2 (Pin 14/Pin 12): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.

-

For more information www.linear.com/LTC3129-1

31291fc

9

LTC3129-1

pin FuncTions

(QFN/MSOP)

PGND (Pin 15/Pin 13, Exposed Pad Pin 17/Pin 17): Power

Ground. Provide a short direct PCB path between PGND
and the ground plane. The exposed pad must also be
soldered to the PCB ground plane. It serves as a power
ground connection, and as a means of conducting heat
away from the die.

SW1 (Pin 16/Pin 14): Switch Pin. Connect to one side of
the inductor. Keep PCB trace lengths as short and wide
as possible to reduce EMI.

block DiagraM

V

V

IN

IN

START

V

CC

START

RUN

V

V

REF

1.22V

IN

0.9V

1.175V

4.1V

V

CC

LDO

V

CC_GD

V

CC

V

1.175V

REF

V

REF

V

REF_GD

+

START

–

+

SD

–

–

UVLO

I

SENSE

500mA

DRV_B

DRV_A

+

I

SENSE

20mA

+
–

SOFT-START

100mV

–
+

RESET

MPPC

1.175V

PWM

5M

BST1 SW1 SW2

DRIVER

DRIVER

A

B

+

I

–

LIM

ENABLE

–

I

ZERO

+

SHUTDOWN

SLEEP

SLEEP

THERMAL

GND

Table 1. V

Program Settings

OUT

VS3 PIN VS2 PIN VS1 PIN V

0 0 0 2.5V

I

SENSE

I

SENSE

0 0 V

0 V

0 V

V

CC

V

CC

V

CC

V

CC

LOGIC

PWM

D

C

+
–

OSC

CC

CC

0 0 6.9V

0 V

V

CC

V

CC

BST2

DRIVER

DRIVER

DRV_C

UV

–
+

V

–
+

1.1V

CC

0 4.1V

V

CC

CC

0 12V

V

CC

CC

I

SENSE

–

V

C

+

FB

1.175V

–

–7.5%

–
+

600mV

PGND

+

CLAMP

V

OUT

VS1

VS2

VS3DRV_D

PGOOD

31291 BD

OUT

3.3V

5V

8.2V

15V

V
SELECT
INPUTS

OUT

V

OUT

10

31291fc

For more information www.linear.com/LTC3129-1

operaTion

C

C

31291 F01

LTC3129-1

INTRODUCTION

The LTC3129-1 is a 1.3µA quiescent current, monolithic,
current mode, buck-boost DC/DC converter that can operate
over a wide input voltage range of 1.92V to 15V and provide
up to 200mA to the load. Eight fixed, user-programmable
output voltages can be selected using the three digital
programming pins. Internal, low R

N-channel power

DS(ON)

switches reduce solution complexity and maximize effi
ciency. A proprietary switch control algorithm allows the
buck-boost converter to maintain output voltage regulation
with input voltages that are above, below or equal to the
output voltage. Transitions between the step-up or step­down operating modes are seamless and free of transients
and sub-harmonic switching, making this product ideal
for noise sensitive applications. The LTC3129-1 operates
at a fixed nominal switching frequency of 1.2MHz, which
provides an ideal trade-off between small solution size and
high efficiency. Current mode control provides inherent
input line voltage rejection, simplified compensation and
rapid response to load transients.

Burst Mode capability is also included in the LTC3129-1
and is user-selected via the PWM input pin. In Burst Mode
operation, the LTC3129-1 provides exceptional efficiency at
light output loading conditions

by operating the converter
only when necessary to maintain voltage regulation. The
Burst Mode quiescent current is a miserly 1.3µA. At higher
loads, the LTC3129-1 automatically switches to fixed fre
quency PWM mode when Burst Mode operation is selected.
(Please refer to the Typical Performance Characteristic
curves for the mode transition point at different input and
output voltages). If the application requires extremely low
noise, continuous PWM operation can also be selected
via the PWM pin.

A MPPC (maximum power point control) function is also
provided that allows the input voltage to the converter to
be servo’d to a programmable point for maximum power
when operating from various non-ideal power sources
such as photovoltaic cells. The LTC3129-1 also features
an accurate RUN comparator threshold with hysteresis,
allowing the buck-boost DC/DC converter to turn on and
off at user-selected V

voltage thresholds. With a wide

IN

voltage range, 1.3µA Burst Mode current and program
mable RUN and MPPC pins, the LTC3129-1 is well suited

for many diverse applications.

PWM MODE OPERATION

If the PWM pin is high or if the load current on the con

verter is high enough to command PWM mode operation

-

PWM low, the LTC3129-1 operates in a fixed 1.2MHz

with
PWM mode using an internally compensated average
current mode control loop. PWM mode minimizes output
voltage ripple and yields a low noise switching frequency
spectrum. A proprietary switching algorithm provides
seamless transitions between operating modes and
eliminates discontinuities in the average inductor cur
rent, inductor ripple current and loop transfer function

throughout all modes of operation. These advantages

result in increased efficiency, improved loop stability and
lower output voltage ripple in comparison to the traditional
buck-boost converter.

Figure 1 shows the topology of the LTC3129-1 power stage
which is comprised of four N-channel DMOS switches and
their associated gate drivers. In PWM mode operation

both switch pins transition on every cycle independent of
the input and output voltages. In response to the internal
control loop command, an internal pulse width modulator
generates the appropriate switch duty cycle to maintain

-

regulation of the output voltage.

BST1

BST1

V

CC

A

V

CC

B

PGND PGND

Figure 1. Power Stage Schematic

SW1 SW2

IN

L

BST2

OUT

BST2V

V

CC

V

D

C

V

CC

LTC3129-1

-

-

-

For more information www.linear.com/LTC3129-1

31291fc

11

LTC3129-1

operaTion

When stepping down from a high input voltage to a lower
output voltage, the converter operates in buck mode and
switch D remains on for the entire switching cycle except
for the minimum switch low duration (typically 90ns). Dur
ing the switch low duration, switch C is turned on which
forces SW2 low and charges the flying capacitor, C
This ensures that the switch D gate driver power supply
rail on BST2 is maintained. The duty cycle of switches A
and B are adjusted to maintain output voltage regulation
in buck mode.

If the input voltage is lower than the output voltage, the
converter operates in boost mode. Switch A remains on
for the entire switching cycle except for the minimum
switch low duration (typically 90ns). During the switch
low duration, switch B is turned on which forces SW1
low and charges the flying capacitor, C
that the switch A gate driver power supply rail on BST1 is
maintained. The duty cycle of switches C and D are adjusted
to maintain output voltage regulation in boost mode.

Oscillator

The LTC3129-1 operates from an internal oscillator with a
nominal fixed frequency of 1.2MHz. This allows the
converter efficiency to be maximized while still using small
external components.

Current Mode Control

The LTC3129-1 utilizes average current mode control for
the pulse width modulator. Current mode control, both
average and the better known peak method, enjoy some
benefits compared to other control methods including:
simplified loop compensation, rapid response to load
transients and inherent line voltage rejection.

Referring to the Block Diagram, a high gain, internally
compensated transconductance amplifier monitors V
through an internal voltage divider. The error amplifier out
put is used by the current mode control loop to command
the appropriate inductor current level. The inverting input
of the internally compensated average current amplifier is
connected to the inductor current sense circuit. The aver
age current amplifier’s output is compared to the oscillator

. This ensures

BST1

BST2

DC/DC

OUT

ramps, and the comparator outputs are used to control
the duty cycle of the switch pins on a cycle-by-cycle basis.

The voltage error amplifier monitors the output voltage,

-

.

-

-

through the internal voltage divider and makes adjust-

V

OUT

ments to the current command as necessary to maintain

regulation. The voltage error amplifier therefore controls

the outer voltage regulation loop. The average current

amplifier makes adjustments to the

directed by the voltage error amplifier output via V

commonly referred to as the inner current loop amplifier.

The average current mode control technique is similar to

peak current mode control except that the average current

amplifier, by virtue of its configuration as an integrator,

controls average current instead of the peak current. This

difference eliminates the peak to average current error

inherent to peak current mode control, while maintaining

most of the advantages inherent to peak current mode

control.

Average current mode control requires appropriate com

pensation for the inner current loop, unlike peak current

mode control. The compensation network must have high

DC gain to minimize errors between the actual and com

manded average current level, high bandwidth to quickly
change the commanded current level following transient
load steps and a controlled mid-band gain to provide a
form of slope compensation unique to average current
mode control. The compensation components required
to ensure proper operation have been carefully selected
and are integrated within the LTC3129-1.

Inductor Current Sense and Maximum Output Current

As part of the current control loop required for current
mode control, the LTC3129-1 includes a pair of current

sensing circuits that measure the buck-boost converter

inductor current.

The voltage error amplifier output, V

to a nominal level of 0.6V. Since the average inductor

current is proportional to V

the maximum average inductor current that can be pro

grammed by the inner current loop. Taking into account

the current sense amplifier’s gain, the maximum average

, the 0.6V clamp level sets

C

inductor current as

, is internally clamped

C

and is

C

-

-

-

12

31291fc

For more information www.linear.com/LTC3129-1

operaTion

LTC3129-1

inductor current is approximately 275mA (typical). In
buck mode, the output current is approximately equal to
the inductor current I

I

OUT(BUCK)

≈ IL • 0.89

.

L

The 90ns SW1/SW2 forced low time on each switching
cycle briefly disconnects the inductor from V

and VIN

OUT

resulting in about 11% less output current in either buck
or Boost mode for a given inductor current. In boost mode,
the output current is related to average inductor current
and duty cycle by:

I

OUT(BOOST)

≈ IL • (1 – D) • Efficiency

where D is the converter duty cycle.

Since the output current in boost mode is reduced by the
duty cycle (D), the output current rating in buck mode is
always greater than in boost mode. Also, because boost
mode operation requires a higher inductor current for a
given output current compared to buck mode, the efficiency

2

in boost mode will be lower due to higher I

L

• R

DS(ON)

losses in the power switches. This will further reduce the
output current capability in boost mode. In either operating
mode, however, the inductor peak-to-peak ripple current
does not play a major role in determining the output cur
rent capability, unlike peak current mode control.

Overload Current Limit and I

Comparator

ZERO

The internal current sense waveform is also used by the

peak overload current (I

parators. The I

current comparator monitors I

PEAK

) and zero current (I

PEAK

ZERO

and turns off switch A if the inductor current level exceeds

its maximum internal threshold, which

is approximately
500mA. An inductor current level of this magnitude will
occur during a fault, such as an output short-circuit, or
during large load or input voltage transients.

The LTC3129-1 features near discontinuous inductor
current operation at light output loads by virtue of the

comparator circuit. By limiting the reverse current

I

ZERO

magnitude in PWM mode, a balance between low noise
operation and improved efficiency at light loads is achieved.
The I

comparator threshold is set near the zero current

ZERO

level in PWM mode, and as a result, the reverse current

magnitude will be a function of inductance value and out
put voltage due to the comparator's propagation delay. In
general, higher output voltages and lower inductor values
will result in increased reverse current magnitude.

In automatic Burst Mode operation (PWM pin low), the

comparator threshold is increased so that reverse

I

ZERO

-

inductor current does not normally occur. This maximizes
efficiency at very light loads.

) com-

SENSE

-

With peak current mode control, the maximum output
current capability is reduced by the magnitude of inductor
ripple current because the peak inductor current level is the
control variable, but the average inductor current is what
determines the output current. The LTC3129-1 measures
and controls average inductor current, and therefore, the
inductor ripple current magnitude has little effect on the
maximum current capability in contrast to an equivalent
peak current mode converter. Under most conditions in
buck mode, the LTC3129-1 is capable of providing a mini
mum of 200mA to the load. In boost mode, as described
previously, the output current capability is related to the
boost ratio or duty cycle (D). For example, for a 3.6V V

IN

to 5V output application, the LTC3129-1 can provide up
to 150mA to the load. Refer to the Typical Performance
Characteristics section for more detail on output current
capability.

Burst Mode OPERATION

When the PWM pin is held low, the LTC3129-1 is con

figured for automatic Burst Mode operation. As a result,
the buck-boost DC/DC converter will operate with normal
continuous PWM switching above a predetermined mini
mum output load and will automatically transition to power

saving Burst Mode operation below this output load level.
Note that if the PWM pin is low, reverse inductor current is

­not allowed at any load. Refer to the Typical Performance

Characteristics section of this data sheet to determine the
Burst Mode transition threshold for various combinations

of V

and V

IN

. If PWM is low, at light output loads, the

OUT

-

-

31291fc

For more information www.linear.com/LTC3129-1

13

LTC3129-1

operaTion

LTC3129-1 will go into a standby or sleep state when the
output voltage achieves its nominal regulation level. The
sleep state halts PWM switching and powers down all
nonessential functions of the IC, significantly reducing the
quiescent current of the LTC3129-1 to just 1.3µA typical.
This greatly improves overall power conversion efficiency
when the output load is light. Since the converter is not
operating in sleep, the output voltage will slowly decay
at a rate determined by the output load resistance and
the output capacitor value. When the output voltage has
decayed by a small amount, the LTC3129-1 will wake and
resume normal PWM switching operation until the volt
age on V

is restored to the previous level. If the load

OUT

is very light, the LTC3129-1 may only need to switch for
a few cycles to restore V

and may sleep for extended

OUT

periods of time, significantly improving efficiency. If the
load is suddenly increased above the burst transition
threshold, the part will automatically resume continuous
PWM operation until the load is once again reduced.

Note that Burst Mode operation is inhibited until soft-start
is done, the MPPC pin is greater than 1.175V and V

OUT

has reached regulation.

Soft-Start

The LTC3129-1 soft-start circuit minimizes input current
transients and output voltage overshoot on initial power up.
The required timing components for soft-start are internal
to the LTC3129-1 and produce a nominal soft-start dura
tion of approximately 3ms. The internal soft-start circuit
slowly ramps the error amplifier output, V

. In doing so,

C

the current command of the IC is also slowly increased,
starting from zero. It is unaffected by output loading or
output capacitor value. Soft-start is reset by the UVLO on
both V

V

CC

and VCC, the RUN pin and thermal shutdown.

IN

Regulator

An internal low dropout regulator (LDO) generates a nomi
nal 4.1V VCC rail from VIN. The VCC rail powers the internal
control circuitry and the gate drivers of the LTC3129-1. The

regulator is disabled in shutdown to reduce quiescent

V

CC

current and is enabled by raising the RUN pin above its
logic threshold. The V

regulator includes current-limit

CC

protection to safeguard against accidental short-circuiting
of the V

CC

rail.

Undervoltage Lockout (UVLO)

There are two undervoltage lockout (UVLO) circuits within
the LTC3129-1 that inhibit switching; one that monitors V
and another that monitors V

. Either UVLO will disable

CC

operation of the internal power switches and keep other
IC functions in a reset state if either V

or VCC are below

IN

their respective UVLO thresholds.

The V

-

of 1.8V (typical). If V
is disabled until V
the V

The V
(typical). If the V
operation is disabled until V

as long as V

UVLO comparator has a falling voltage threshold

IN

falls below this level, IC operation

IN

rises above 1.9V (typical), as long as

IN

voltage is above its UVLO threshold.

CC

UVLO has a falling voltage threshold of 2.19V

CC

voltage falls below this threshold, IC

CC

rises above 2.25V (typical)

CC

is above its nominal UVLO threshold level.

IN

Depending on the particular application, either of these

UVLO thresholds could be the limiting factor affecting the

minimum input voltage required for operation. Because the

regulator uses VIN for its power input, the minimum

V

CC

input voltage required for operation is determined by the

minimum voltage, as input voltage (VIN) will always

V

CC

be higher than V

configuration. Therefore, the minimum V
to start up is 2.25V (typical).

­ applications where VCC is bootstrapped (powered

In

through a Schottky diode by either V

in the normal (non-bootstrapped)

CC

for the part

IN

or an auxiliary

OUT

power rail), the minimum input voltage for operation will
be limited only by the V

UVLO threshold (1.8V typical).

IN

Please note that if the bootstrap voltage is derived from
the LTC3129-1 V

and not an independent power rail,

OUT

then the minimum input voltage required for initial start-up
is still 2.25V (typical).

-

Note that if either V

or VCC are below their UVLO

IN

thresholds, or if RUN is below its accurate threshold of

1.22V (typical), then the LTC3129-1 will remain in a soft
shutdown state, where the V

quiescent current will be

IN

only 1.9µA typical.

IN

14

31291fc

For more information www.linear.com/LTC3129-1

operaTion

31291 F02

V

Undervoltage

OUT

There is also an undervoltage comparator that monitors
the output voltage. Until V
average current limit is reduced by a factor of two. This
reduces power dissipation in the device in the event of a
shorted output. In addition, N-channel switch D, which
feeds V

, will be disabled until V

OUT

RUN Pin Comparator

reaches 1.15V (typical), the

OUT

exceeds 1.15V.

OUT

LTC3129-1

LTC3129-1

V

IN

R3

R4

1.22V

RUN

0.9V

ACCURATE THRESHOLD

–
+

+
–

LOGIC THRESHOLD

ENABLE SWITCHING

ENABLE LDO AND
CONTROL CIRCUITS

In addition to serving as a logic level input to enable cer
tain functions of the IC, the RUN pin includes an accurate
internal comparator that allows it to be used to set custom
rising and falling ON/OFF thresholds with the addition of
an optional external resistor divider. When RUN is driven
above its logic threshold (0.9V typical), the V

regulator

CC

is enabled, which provides power to the internal control
circuitry of the IC. If the voltage on RUN is increased
further so that it exceeds the RUN comparator’s accurate
analog threshold (1.22V typical), all functions of the buck­boost converter will be enabled and a start-up sequence
will ensue (assuming the V

and V

IN

UVLO thresholds

CC

are satisfied).

If RUN is brought below the accurate comparator threshold,
the buck-boost

converter will inhibit switching, but the VCC
regulator and control circuitry will remain powered unless
RUN is brought below its logic threshold. Therefore, in
order to completely shut down the IC and reduce the V

IN

current to 10nA (typical), it is necessary to ensure that
RUN is brought below its worst case low logic threshold of

0.5V. RUN is a high voltage input and can be tied directly

to continuously enable the IC when the input supply

to V

IN

is present. Also note that RUN can be driven above V
or V

as long as it stays within the operating range of

OUT

IN

the IC (up to 15V).

With the addition of an optional resistor divider as shown
in Figure 2, the RUN pin can be used to establish a user­programmable turn-on and turn-off threshold. This feature
can be utilized to minimize battery drain below a certain
input voltage, or to operate the converter in a hiccup mode
from very low current sources.

-

Figure 2. Accurate RUN Pin Comparator

Note that once RUN is above 0.9V typical, the quiescent
input current on V
about 1.9µA typical until the V

(or VCC if back-driven) will increase to

IN

and VCC UVLO thresholds

IN

are satisfied.

The converter is enabled when the voltage on RUN exceeds

1.22V (nominal). Therefore, the turn on voltage threshold
is given by:

on V

IN

V

IN(TURN-ON)

= 1.22V • (1 + R3/R4)

The RUN comparator includes a built-in hysteresis of
approximately 80mV, so that the turn off threshold will
be 1.14V.

There may be cases due to PCB layout, very large value
resistors for R3 and R4, or proximity to noisy components
where noise pickup may cause the turn-on or turn-off of the
IC to be intermittent. In these cases, a small filter capaci
tor can be added across R4 to ensure proper operation.

PGOOD Comparator

The LTC3129-1 provides an open-drain PGOOD output that
pulls low if V
programmed value. When V

falls more than 7.5% (typical) below its

OUT

rises to within 5% (typical)

OUT

of its programmed value, the internal PGOOD pull-down
will turn off and PGOOD will go high if an external pull­up resistor has been provided. An internal filter prevents
nuisance trips of PGOOD due to short transients on V
Note that PGOOD can be pulled up to any voltage, as long

as the absolute maximum rating of 18V is

not exceeded,

and as long as the maximum sink current rating is not
exceeded when PGOOD is low. Note that PGOOD will
also be driven low if V

is below its UVLO threshold or

CC

OUT

-

.

For more information www.linear.com/LTC3129-1

31291fc

15

LTC3129-1

31291 F03

+

–

operaTion

if the part is in shutdown (RUN below its logic threshold)
while V
not affected by V

In cases where V

is being held up (or back-driven). PGOOD is

CC

UVLO or the accurate RUN threshold.

IN

is not being back-driven in shutdown,

CC

PGOOD will not be held low indefinitely. The internal PGOOD
pull-down will be disabled as the V

voltage decays below

CC

approximately 1V.

Maximum Power-Point Control (MPPC)

The MPPC input of the LTC3129-1 can be used with an
optional external voltage divider to dynamically adjust
the commanded inductor current in order to maintain
a minimum input voltage when using high resistance
sources, such as photovoltaic panels, so as to maximize
input power transfer and prevent V

from dropping too

IN

low under load. Referring to Figure 3, the MPPC pin is
internally connected to the noninverting input of a g

m

amplifier, whose inverting input is connected to the 1.175V
reference. If the voltage at MPPC, using the external volt
age divider, falls below the reference voltage, the output of
the amplifier pulls the internal V

node low. This reduces

C

the commanded average inductor current so as to reduce
the input current and regulate

VIN to the programmed

minimum voltage, as given by:

V

IN(MPPC)

= 1.175V • (1 + R5/R6)

The MPPC feature provides capabilities to the LTC3129-1

that can ease the design of intrinsically safe power sup

plies. For an example of an application that must operate

from a supply with intentional series resistance, refer to

the application example on the bottom of page 25.

Note that external compensation should not be required

for MPPC loop stability if the input filter capacitor, C

at least 22µF. See Typical Applications for an example of

external compensation that can be added in applications

where C

must be less than the recommended minimum

IN

value.

The divider resistor values can be in the megohm range to

minimize the input current in very low power applications.

However, stray capacitance and noise pickup on the MPPC

pin must also be minimized.

The MPPC pin controls the converter in a linear fashion

when using sources that can provide a minimum of 5mA

to 10mA of continuous input current. For operation from

­weaker input sources, refer to the Application Information

section to see how the programmable RUN pin can be used
to control the converter in a hysteretic manner
an effective MPPC function for sources that can provide
as little as 5µA or less.

If the MPPC function is not required, the MPPC pin should
be tied to V

CC

.

, is

IN

to provide

-

Programming Pins

V

IN

*C

IN

R

S

V

SOURCE

* CIN SHOULD BE AT
LEAST 22µF FOR
MPPC APPLICATIONS

Figure 3. MPPC Amplifier with External Resistor Divider

16

R5

MPPC

R6

1.175V

V

IN

+
–

+
–

VOLTAGE
ERROR AMP

LTC3129-1

V

C

CURRENT
COMMAND

For more information www.linear.com/LTC3129-1

V

OUT

The LTC3129-1 has a precision internal voltage divider on

, eliminating the need for high-value external feedback

V

OUT

resistors. This not only eliminates two external compo
nents, it minimizes no-load quiescent current by using very

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operaTion

LTC3129-1

high resistance values that would not be practical due to the
effects of noise and board leakages that would cause V

OUT

regulation errors. The tap point on this divider is digitally
selected by using the VS1, VS2 and VS3 pins to program
one of eight fixed output voltages. The VS pins should be
grounded or connected to V

to select the desired output

CC

voltage, according to the following table. The VS1, VS2
and VS3 pins can also be driven by external logic signals
as long as the absolute maximum voltage ratings are not
exceeded. Note however that driving any of the voltage
select pins high to a voltage less than the V

operating

CC

voltage will result in increased quiescent current. Also
note that if the VS3 pin is driven above V

, an external

CC

1M resistor should be added in series. For other output
voltages, refer to the LTC3129 which has a feedback pin,
allowing any output voltage from 1.4V to 15.75V.

V

Program Settings for the LTC3129-1

OUT

VS3 PIN VS2 PIN VS1 PIN V

0 0 0 2.5V

0 0 V

0 V

0 V

V

CC

V

CC

V

CC

V

CC

CC

CC

0 0 6.9V

0 V

V

CC

V

CC

CC

0 4.1V

V

CC

CC

0 12V

V

CC

OUT

3.3V

5.0V

8.2V

15V

Note that in shutdown, or if VCC is below its UVLO thresh­old, the internal voltage divider on V
disconnected to eliminate any current draw on V

is automatically

OUT

OUT

.

dissipation of the IC. As described elsewhere in this data
sheet, bootstrapping of the V
can essentially eliminate the V

for 5V output applications

CC

power dissipation term

CC

and significantly improve efficiency. As a result, careful
consideration must be given to the thermal environment
of the IC in order to provide a means to remove heat from
the IC and ensure that the

LTC3129-1 is able to provide

its full rated output current. Specifically, the exposed die

attach pad of both the QFN and MSE packages must be
soldered to a copper layer on the PCB to maximize the
conduction of heat out of the IC package. This can be ac

complished by utilizing multiple vias from the die attach
pad connection underneath the IC package to other PCB
layer(s) containing a large copper plane. A typical board
layout incorporating these concepts is shown in Figure 4.

If the IC die temperature exceeds approximately 180°C,
overtemperature shutdown will be invoked and all switching
will be inhibited. The part will remain disabled until the die
temperature cools by approximately 10°C. The soft-start
circuit is re-initialized in over temperature shutdown to
provide a smooth recovery when the IC die temperature
cools enough to resume operation.

GND V

V

CC

IN

C

IN

C

BST1

C

BST2

L

-

Thermal Considerations

The power switches of the LTC3129-1 are designed to op
erate continuously with currents up to the internal current
limit thresholds. However, when operating at high current
levels, there may be significant heat generated within the
IC. In addition, the V
heat when V

is very high, adding to the total power

IN

regulator can also generate wasted

CC

For more information www.linear.com/LTC3129-1

C

-

GND

Figure 4. Typical 2-Layer PC Board Layout (MSE Package)

OUT

31291 F04

V

OUT

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17

LTC3129-1

applicaTions inForMaTion

A standard application circuit for the LTC3129-1 is shown
on the front page of this data sheet. The appropriate selec
tion of external components is dependent upon the required
performance of the IC in each particular application given
considerations and trade-offs such as PCB area, input
and output voltage range, output voltage ripple, transient
response, required efficiency, thermal considerations and
cost. This section of the data sheet provides some basic
guidelines and considerations to aid in the selection of
external components and the design of the applications
circuit, as well as more application circuit examples.

Capacitor Selection

V

CC

The V
by a low dropout linear regulator. The V

output of the LTC3129-1 is generated from VIN

CC

regulator has

CC

been designed for stable operation with a wide range
of output capacitors. For most applications, a low ESR
capacitor of at least 2.2µF should be used. The capacitor
should be located as close to the V
connected to the V
traces possible. V

pin and ground through the shortest

CC

is the regulator output and is also the

CC

pin as possible and

CC

internal supply pin for the LTC3129-1 control circuitry as
well as the gate
The V

pin is not intended to supply current to other

CC

drivers and boost rail charging diodes.

external circuitry.

Inductor Selection

The choice of inductor used in LTC3129-1 application cir
cuits influences the maximum deliverable output current,
the converter bandwidth, the magnitude of the inductor
current ripple and the overall converter efficiency. The
inductor must have a low DC series resistance, when
compared to the internal switch resistance, or output
current capability and efficiency will be compromised.
Larger inductor values reduce inductor current ripple
but may not increase output current capability as is the
case with peak current mode control as described in the
Maximum Output Current section. Larger value inductors
also tend to have a higher DC series resistance for a given
case size, which will have a negative impact on efficiency.
Larger values of inductance will also lower the right half
plane (RHP) zero frequency when operating in boost mode,
which can compromise loop stability. Nearly all LTC3129-1
application circuits deliver the best performance with
an inductor value between 3.3µH and 10µH. Buck mode

only applications can use the larger inductor values as
they are unaffected by the RHP zero, while mostly boost

­applications generally require inductance

on the low end

of this range depending on how large the step-up ratio is.

Regardless of inductor value, the saturation current rating
should be selected such that it is greater than the worst
case average inductor current plus half of the ripple cur
rent. The peak-to-peak inductor current ripple for each
operational mode can be calculated from the following
formula, where f is the switching frequency (1.2MHz), L
is the inductance in µH and t

is the switch pin mini-

LOW

mum low time in µs. The switch pin minimum low time
is typically 0.09µs.

ΔI

L(P−P)(BUCK)

ΔI

L(P−P)(BOOST)

⎛

OUT

L

V

IN

L

VIN– V

⎜
⎝

⎛

V

⎜
⎝

OUT

V

V

OUT

V

=

=

IN

– V

OUT

IN

⎞

⎛

1

– t

⎜

⎟
⎠

⎞
⎟
⎠

LOW

⎝

f

⎛

1

– t

⎜

LOW

⎝

f

It should be noted that the worst-case peak-to-peak in­ductor ripple current occurs when the duty cycle in buck
mode is minimum (highest V
the duty cycle is 50% (V

(minimum) = 2.5V and VIN (maximum) = 15V, V

V

IN

) and in boost mode when

IN

= 2 • VIN). As an example, if

OUT

= 5V and L = 10µH, the peak-to-peak inductor ripples at
the voltage extremes (15V V

for buck and 2.5V VIN for

IN

boost) are:

­ BUCK = 248mA peak-to-peak

BOOST = 93mA peak-to-peak

One half of this inductor ripple current must be added to
the highest expected average inductor current in order to
select the proper saturation current rating for the inductor.

To avoid the possibility of inductor saturation during load
transients, an inductor with a saturation current rating of
at least 600mA is recommended for all applications.

In addition to its influence on power conversion efficiency,
the inductor DC resistance can also impact the maximum
output current capability of the buck-boost converter
particularly at low input voltages. In buck mode, the
output current of the buck-boost converter is primarily
limited by the inductor current reaching the average cur
rent limit threshold. However, in boost mode, especially

-

⎞

A

⎟
⎠

⎞

A

⎟
⎠

OUT

-

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applicaTions inForMaTion

LTC3129-1

at large step-up ratios, the output current capability can
also be limited by the total resistive losses in the power
stage. These losses include, switch resistances, inductor
DC resistance and PCB trace resistance. Avoid inductors
with a high DC resistance (DCR) as they can degrade the
maximum output current capability from what is shown
in the Typical Performance Characteristics section and
from the Typical Application circuits.

As a guideline, the inductor DCR should be significantly
less than the typical power switch resistance of 750mΩ
each. The only exceptions are applications that have a
maximum output current requirement much less than
what the LTC3129-1 is capable of delivering. Generally
speaking, inductors with a DCR in the range of 0.15Ω to

0.3Ω are recommended. Lower values of DCR will improve
the efficiency at the expense of size, while higher DCR
values will reduce efficiency (typically by a few percent)
while allowing the use of a physically smaller inductor.

Different inductor core materials and styles have an impact
on the size and price of an inductor at any given current
rating. Shielded construction is generally preferred as it
minimizes the chances of interference with other circuitry.
The choice of inductor style depends

upon the price, sizing,
and EMI requirements of a particular application. Table 2
provides a wide sampling of inductors that are well suited
to many LTC3129-1 applications.

Table 2. Recommended Inductors

VENDOR PART

Coilcraft
www.coilcraft.com

Coiltronics
www.cooperindustries.com

Murata
www.murata.com

Sumida
www.sumida.com

Taiyo-Yuden
www.t-yuden.com

TDK
www.tdk.com

Toko
www.tokoam.com

Würth
www.we-online.com

EPL2014, EPL3012, EPL3015, XFL3012
LPS3015, LPS3314

SDH3812, SD3814
SD3114, SD3118

LQH3NP
LQH32P
LQH44P

CDRH2D16, CDRH2D18
CDRH3D14, CDRH3D16

NR3012T, NR3015T, NRS4012T
BRC2518

VLS3012, VLS3015
VLF302510MT, VLF302512MT

DB3015C, DB3018C, DB3020C
DP418C, DP420C, DEM2815C,
DFE322512C, DFE252012C

WE-TPC 2813, WE-TPC 3816,
WE-TPC 2828

Recommended inductor values for different operating
voltage ranges are given in Table 3. These values were
chosen to minimize inductor size while maintaining an
acceptable amount of inductor ripple current for a given

and V

V

IN

Table 3. Recommended Inductor and Output Capacitor Values

VIN AND V

and V

V

IN

and V

V

IN

and V

V

IN

and V

V

IN

range.

OUT

RANGE RECOMMENDED

OUT

Both < 4.5V 3.3µH to 4.7µH 10µF

OUT

Both < 8V 4.7µH to 6.8µH 10µF

OUT

Both < 11V 6.8µH to 8.2µH 10µF

OUT

Up to 15V 8.2µH to 10µH 10µF

OUT

INDUCTOR

VALUES

MAXIMUM RECOMMENDED
TOTAL OUTPUT CAPACITOR

VALUE FOR PWM MODE

OPERATION AT LIGHT LOAD

(<15mA, PWM PIN HIGH)

Due to the fixed, internal loop compensation and feedback
divider provided by the LTC3129-1, there are limitations to
the maximum recommended total output capacitor value in
applications that must operate in PWM mode at light load
(PWM pin pulled high with minimum load currents less
than ~15mA). In these applications, a maximum output
capacitor value, shown in Table 3, is recommended. For
applications that must operate in PWM mode at light load
with higher values of output capacitance, the LTC3129 is
recommended. Its external feedback pin allows the use
of additional feedforward compensation for improved
light-load stability under these conditions.

Note that for applications where Burst Mode operation
is enabled (PWM pin grounded), the output capacitor
value can be increased without limitation regardless of
the minimum load current or

inductor value.

Output Capacitor Selection

A low effective series resistance (ESR) output capacitor
of 4.7µF minimum should be connected at the output of
the buck-boost converter in order to minimize output volt
age ripple. Multilayer ceramic capacitors are an excellent
option as they have low ESR and are available in small
footprints. The capacitor value should be chosen large
enough to reduce the output voltage ripple to acceptable
levels. Neglecting the capacitor’s ESR and ESL (effec
tive series inductance), the peak-to-peak output voltage
ripple in PWM mode can be calculated by the following

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19

LTC3129-1

I

I

applicaTions inForMaTion

formula, where f is the frequency in MHz (1.2MHz), C
is the capacitance in µF, t
low time in µs (0.09µs typical) and I

is the switch pin minimum

LOW

is the output

LOAD

OUT

current in amperes.

ΔV

ΔV

P−P(BUCK)

P−P(BOOST)

LOADtLOW

=

C

I

LOAD

=

fC

OUT

OUT

V

⎛

V

– VIN+ t

OUT

⎜
⎝

V

LOW

OUT

fV

IN

⎞

V

⎟
⎠

Examining the previous equations reveals that the output
voltage ripple increases with load current and is gener
ally higher in boost mode than in buck mode. Note that
these equations only take into account the voltage ripple
that occurs from the inductor current to the output being
discontinuous. They provide a good approximation to the
ripple at any significant load current but underestimate the
output voltage ripple at very light loads where the output
voltage ripple is dominated by the inductor current ripple.

In addition to the output voltage ripple generated across
the output capacitance, there is also output voltage ripple
produced across the internal resistance of the output
capacitor. The ESR-generated output voltage ripple is
proportional to the series resistance of the output capacitor
and is given by the following expressions where R

ESR

is
the series resistance of the output capacitor and all other
terms as previously defined.

ΔV

ΔV

P−P(BUCK)

P−P(BOOST)

LOADRESR

=

1– t

LOW

I

LOADRESRVOUT

=

VIN1– t

( )

≅I

LOADRESR

≅I

f

LOW

⎛
⎜

⎝

LOADRESR

f

⎞

V

OUT

V

⎟

V

⎠

IN

V

In most LTC3129-1 applications, an output capacitor be­tween 10µF and 22µF will work well. To minimize output
ripple in Burst Mode operation, values of 22µF operation
or larger are recommended.

Input Capacitor Selection

The V

pin carries the full inductor current and provides

IN

power to internal control circuits in the IC. To minimize
input voltage ripple and ensure proper operation of the IC,
a low ESR bypass capacitor with a value of at least 4.7µF
should be located as close to the V
traces connecting this capacitor to V

pin as possible. The

IN

and the ground

IN

plane should be made as short as possible.

When powered through long leads or from a power source
with significant resistance, a larger value bulk input ca
pacitor may be required and is generally recommended.

­In such applications, a 47µF to 100µF low-ESR electrolytic

capacitor in parallel with a 1µF ceramic capacitor generally
yields a high performance, low cost solution.

Note that applications using the MPPC feature should
use a minimum C

of 22µF. Larger values can be used

IN

without limitation.

Recommended Input and Output Capacitor Types

The capacitors used to filter the input and output of the
LTC3129-1 must

have low ESR and must be rated to handle
the AC currents generated by the switching converter.
This is important to maintain proper functioning of the
IC and to reduce output voltage ripple. There are many
capacitor types that are well suited to these applications
including multilayer ceramic, low ESR tantalum, OS-CON
and POSCAP technologies. In addition, there are certain
types of electrolytic capacitors such as solid aluminum
organic polymer capacitors that are designed for low ESR
and high AC currents and these are also well suited to
some LTC3129-1 applications. The choice of capacitor
technology is primarily dictated by a trade-off between
size, leakage current and cost. In backup power applica
tions, the input or output capacitor might be a super or
ultra capacitor with a capacitance value measuring in the
farad range. The selection criteria in these applications
are generally similar except that voltage ripple is generally
not a concern. Some capacitors exhibit a high DC leak
age current which may preclude their consideration for
applications that require a very low quiescent current in
Burst Mode operation. Note that ultra capacitors may have

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20

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applicaTions inForMaTion

LTC3129-1

a rather high ESR, therefore a 4.7µF (minimum) ceramic
capacitor is recommended in parallel, close to the IC pins.

Ceramic capacitors are often utilized in switching con

­verter applications due to their small size, low ESR and
low leakage currents. However, many ceramic capacitors
intended for power applications experience a significant
loss in capacitance from their rated value as the DC bias
voltage on the capacitor increases. It is not uncommon for
a small surface mount capacitor to lose more than 50%
of its rated capacitance when operated at even half of its
maximum rated voltage. This effect is generally reduced
as the case size is increased for the same nominal value
capacitor. As a result, it is often necessary to use a larger
value capacitance or a higher voltage rated capacitor than
would ordinarily be required to actually realize the intended
capacitance at the operating voltage of the application. X5R
and X7R dielectric types are recommended as they exhibit
the best performance over the wide operating range and
temperature of the LTC3129-1. To verify that the intended
capacitance is achieved in the application circuit, be sure
to consult the capacitor vendor’s curve of capacitance
versus DC bias voltage.

Using the Programmable RUN Function to Operate
from Extremely Weak Input Sources

Another application of the programmable RUN pin is that
it can be used to operate the converter in a hiccup mode
from extremely low current sources. This allows opera

­tion from sources that can only generate microamps of
output current, and would be far too weak to sustain
normal steady-state operation, even with the use of the
MPPC pin. Because the LTC3129-1 draws only 1.9µA
typical from V
programmed to keep the IC disabled until V

until it is enabled, the RUN pin can be

IN

reaches the

IN

programmed voltage level. In this manner, the input source
can trickle-charge an input storage capacitor, even if it
can only supply microamps of current, until V

reaches

IN

the turn-on threshold set by the RUN pin divider. The
converter will then be enabled, using the stored charge
in the input capacitor, until V

drops below the turn-off

IN

threshold, at which point the converter will turn off and
the process will repeat.

This approach allows the converter to run from weak
sources such as thin-film solar cells using indoor lighting.

Although the

converter will be operating in bursts, it is
enough to charge an output capacitor to power low duty
cycle loads, such as wireless sensor applications, or to
trickle charge a battery. In addition, note that the input
voltage will be cycling (with a small ripple as set by the
RUN hysteresis) about a fixed voltage, as determined by
the divider. This allows the high impedance source to
operate at the programmed optimal voltage for maximum
power transfer.

When using high value divider resistors (in the MΩ range)
to minimize current draw on V

, a small noise filter ca-

IN

pacitor may be necessary across the lower divider resis­tor to prevent noise from erroneously tripping the RUN
comparator. The capacitor value should be minimized
so as not to introduce a time delay long enough for the
input voltage to drop significantly below the desired V

IN

threshold before the converter is turned off. Note that
larger V
effect by providing more holdup time on V

decoupling capacitor values will minimize this

IN

.

IN

Programming the MPPC Voltage

As discussed in the previous section, the LTC3129-1 in
cludes an MPPC function to optimize performance when
operating from voltage sources with relatively high source
resistance. Using an external voltage divider from V

IN

, the
MPPC function takes control of the average inductor current
when necessary to maintain a minimum input voltage, as
programmed by the user. Referring to Figure 3:

V

IN(MPPC)

= 1.175V • (1 + R5/R6)

This is useful for such applications as photovoltaic pow
ered converters, since the maximum power transfer point
occurs when the photovoltaic panel is operated at about
75% of its open-circuit voltage. For example, when operat
ing from a photovoltaic panel with an open-circuit voltage
of 5V, the maximum power transfer point will be when
the panel is loaded such that its output voltage is about

3.75V. Choosing values of 2MΩ for R5 and 909k for R6
will program the MPPC function to regulate the maximum
input current so as to maintain V

at a minimum of 3.74V

IN

(typical). Note that if the panel can provide more power
than the LTC3129-1 can draw, the input voltage will rise
above the programmed MPPC point. This is fine as long
as the input voltage doesn't exceed 15V.

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21

LTC3129-1

applicaTions inForMaTion

For weak input sources with very high resistance (hun­dreds of Ohms or more), the LTC3129-1 may still draw
more current than the source can provide, causing V

IN

to
drop below the UVLO threshold. For these applications, it
is recommended that the programmable RUN feature be
used, as described in the previous section.

MPPC Compensation and Gain

When using MPPC, there are a number of variables that
affect the gain and phase of the input voltage control
loop. Primarily these are the input capacitance, the MPPC
divider ratio and the V

source resistance (or current). To

IN

simplify the design of the application circuit, the MPPC
control loop in the LTC3129 is designed with a relatively
low gain, such that external MPPC loop compensation is
generally not required when using a V

capacitor value

IN

of at least 22µF. The gain from the MPPC pin to the in
ternal VC control voltage is about 12, so a drop of 50mV
on the MPPC pin (below the 1.175V MPPC threshold),
corresponds to a 600mV drop on the internal VC voltage,
which reduces the average inductor current all the way
to zero. Therefore, the programmed input MPPC

voltage

will be maintained within about 4% over the load range.

Note that if large-value V

capacitors are used (which may

IN

have a relatively high ESR) a small ceramic capacitor of
at least 4.7µF should be placed in parallel across the V
input, near the V

Bootstrapping the V

pin of the IC.

IN

Regulator

CC

IN

The high and low side gate drivers are powered through
the V

rail, which is generated from the input voltage, VIN,

CC

through an internal linear regulator. In some applications,
especially at high input voltages, the power dissipation
in the linear regulator can become a major contributor to
thermal heating of the IC and overall efficiency. The Typical
Performance Characteristics section provides data on the

current and resulting power loss versus VIN and V

V

CC

OUT

A significant performance advantage can be attained in high

applications where converter output voltage (V

V

IN

programmed to 5V, if V

is used to power the VCC rail.

OUT

OUT

) is

Powering V

in this manner is referred to as bootstrap-

CC

ping. This can be done by connecting a Schottky diode
(such as a BAT54) from V

to VCC as shown in Figure 5.

OUT

With the bootstrap diode installed, the gate driver currents
are supplied by the buck-boost converter at high efficiency
rather than through the internal linear regulator. The in
ternal linear regulator contains reverse blocking circuitry
that allows V

to be driven above its nominal regulation

CC

level with only a very slight amount of reverse current.
Please note that the bootstrapping supply (either V
a separate regulator) must be limited to less than 5.7V so
as not to exceed the maximum V

voltage of 5.5V after

CC

the diode drop.

By maintaining V

above its UVLO threshold, bootstrap-

CC

ping, even to a 3.3V output, also allows operation down

to the V

UVLO threshold of 1.8V (typical).

IN

-

V

OUT

LTC3129-1

V

CC

31291 F05

Figure 5. Example of VCC Bootstrap

C

OUT

2.2µF

V

OUT

BAT54

Sources of Small Photovoltaic Panels

A list of companies that manufacture small solar panels
(sometimes referred to as modules or solar cell arrays)
suitable for use with the LTC3129-1 is provided in Table 4.

Table 4. Small Photovoltaic Panel Manufacturers

Sanyo http://panasonic.net/energy/amorton/en/

PowerFilm http://www.powerfilmsolar.com/

.

IXYS
Corporation

G24
Innovations

http://www.ixys.com/ProductPortfolio/GreenEnergy.aspx

http://www.g24i.com/

OUT

-

or

22

31291fc

For more information www.linear.com/LTC3129-1

Typical applicaTions

LTC3129-1

V

2.42V TO 15V

Low Noise, Fixed Frequency, Wide V

22nF

6.8µH

BST1

IN

4.7µF

V

CC

SW1 SW2

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

LTC3129-1

GND

Range 12V Converter

IN

22nF

VIN < 12V, I
V

OUT

V

IN

CC

31291 TA02

BST2

V

PGOOD

PGND

> 12V, I

10µF
16V

2.2µF

OUT
OUT

1M

= 30mA
= 200mA

V

OUT

12V

PGOOD

3.3V Converter Provides Extremely Long Run Time in Low Drain Applications Using Lithium Thionyl Chloride Battery

22nF

BST1

V

IN

V

CC

Li-SoCl

2

AA
SAFT LS14500
TADIRAN TL-4903

RUN TIME
> 100,000 HRS (11.4 YEARS) AT 10µA (33µW) AVERAGE LOAD
> 34,000 HRS (3.9 YEARS) AT 50µA (165µW) AVERAGE LOAD

SW1 SW2

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

GND

4.2µH

LTC3129-1

PGND

22nF

BST2

PGOOD

V

V

OUT

V

CC

22µF47µF

1M

PGOOD

2.2µF

31291 TA03

OUT

3.3V

For more information www.linear.com/LTC3129-1

31291fc

23

LTC3129-1

Typical applicaTions

15V Converter Powered from Flexible Solar Panel

22nF

BST1

V

IN

PowerFilm
MPT6-150

SOLAR

MODULE

11.4cm × 15cm

V

47µF

MPPC

= 6V

1M

V

CC

243k

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

SW1 SW2

10µH

LTC3129-1

GND

PGND

22nF

BST2

V

PGOOD

OUT

V

CC

31291 TA04a

I

OUT

= 32mA IN FULL SUN

V

OUT

15V

10µF

2.2µF

vs Light Level (Daylight)

I

OUT

100

(mA)

10

OUT

I

1

10000 100000 1000000

LIGHT LEVEL (Lx)

31291 TA04b

PV PANEL

SANYO

AM-1815

4.9cm × 5.8cm

Hiccup Converter Keeps Li-Ion Battery Charged with Indoor Lighting

22nF

BST1

V

IN

+

470µF

6.3V

UVLO = 3.5V

10pF

4.42M

2.37M

4.7µF

V

CC

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

SW1 SW2

LTC3129-1

GND

3.3µH

PGND

22nF

BST2

V

PGOOD

OUT

V

CC

4.7µF

2.2µF

31291 TA05a

V

4.1V

Li-Ion

OUT

Average I

vs Light Level

OUT

(Indoors)

1000

(µA)

100

OUT

I

10

100 1000 10000

LIGHT LEVEL (Lx)

31291 TA05b

24

31291fc

For more information www.linear.com/LTC3129-1

Typical applicaTions

5V Converter Operates from Tw o to Eight AA or AAA Cells Using Bootstrap Diode to Increase Efficiency

at High VIN and Extend Operation at Low V

LTC3129-1

IN

V

1.92V TO 15V

AFTER STARTUP

TWO TO EIGHT

AA OR AAA
BATTERIES

22nF

BST1

IN

V

CC

10µF

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

SW1 SW2

LTC3129-1

GND

8.2µH

PGND

22nF

BST2

PGOOD

VIN < 5V, I
V

IN

V

OUT

V

CC

31291 TA06

> 5V, I

22µF

2.2µF

OUT
OUT

= 100mA
= 200mA

V

OUT

5V

BAT54

3.3V Converter Uses MPPC Function to Work with High Resistance Battery Pack

10Ω

22nF

BST1

V

= 2.9V

10µF

MPPC

1.5M

V

IN

V

RUN

IN

SW1 SW2

LTC3129-1

3.3µH

22nF

BST2

V

OUT

I

OUT

= 100mA

10µF

V

OUT

3.3V

PGND

PGOOD

V

CC

31291 TA07

MPPC

1.5V

1.5V

1.5V

NOTE: R
CAPACITOR VALUE LESS THAN THE RECOMMENDED MINIMUM OF 22µF

R

C

V

150k

C
33pF

AND CC HAVE BEEN ADDED FOR IMPROVED MPPC LOOP STABILITY WHEN USING AN INPUT

C

CC

C

1M

PWM

VS1

VS2

VS3

GND

For more information www.linear.com/LTC3129-1

2.2µF

31291fc

25

LTC3129-1

Typical applicaTions

Solar Powered Converter Extends Battery Life in Low Power 3V Primary Battery Applications

PV PANEL

SANYO AM-1815

OR

PowerFilm SP4.2-37

22nF

BST1

SW1 SW2

V

IN

RUN

MPPC

PWM

VS1

VS2

VS3

LTC3129-1

GND

470µF

6.3V

UVLO = 3.7V

4.7µF

10pF

4.99M

V

2.43M

CC

V

IN

+

3.3µH

PGND

22nF

BST2

PGOOD

FDC6312P

DUAL PMOS

V

OUT

V

CC

3.30V

22µF

BAT54

2.2µF

2.43M

D1 D2

G1

74LVC2G04

S2S1

G2

V

OUT

31291 TA09

V
3V TO 3.3V

2.2µF

CR2032
3V COIN CELL

OUT

Percentage of Added Battery Life vs Light Level and Load

(PowerFilm SP4.2-37, 30sq cm Panel)

1000

100

10

ADDED BATTERY LIFE (%)

1
100

AVERAGE LOAD = 165µW
AVERAGE LOAD = 330µW
AVERAGE LOAD = 660µW
AVERAGE LOAD = 1650µW
AVERAGE LOAD = 3300µW

LIGHT LEVEL (Lx)

10,0001,000

31291 TA09b

31291fc

26

For more information www.linear.com/LTC3129-1

package DescripTion

UD Package

16-Lead Plastic QFN (3mm × 3mm)

(Reference LTC DWG # 05-08-1700 Rev A)

Exposed Pad Variation AA

Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings.

UD Package

16-Lead Plastic QFN (3mm × 3mm)

(Reference LTC DWG # 05-08-1700 Rev A)

Exposed Pad Variation AA

0.70 ±0.05

LTC3129-1

3.50 ±0.05

2.10 ±0.05

1.65 ±0.05
(4 SIDES)

PACKAGE OUTLINE

0.25 ±0.05

0.50 BSC

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

3.00 ±0.10
(4 SIDES)

PIN 1
TOP MARK
(NOTE 6)

NOTE:

1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WEED-4)

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE

0.75 ±0.05

1.65 ±0.10
(4-SIDES)

0.200 REF

0.00 – 0.05

BOTTOM VIEW—EXPOSED PAD

R = 0.115

TYP

15 16

0.25 ±0.05

0.50 BSC

PIN 1 NOTCH R = 0.20 TYP
OR 0.25 × 45° CHAMFER

0.40 ±0.10

1

2

(UD16 VAR A) QFN 1207 REV A

For more information www.linear.com/LTC3129-1

31291fc

27

LTC3129-1

package DescripTion

Please refer to http://www.linear.com/product/LTC3129-1#packaging for the most recent package drawings.

MSE Package

16-Lead Plastic MSOP, Exposed Die Pad

(Reference LTC DWG # 05-08-1667 Rev F)

BOTTOM VIEW OF

EXPOSED PAD OPTION

2.845 ±0.102
(.112 ±.004)

0.889 ±0.127
(.035 ±.005)

2.845 ±0.102
(.112 ±.004)

1

8

0.35
REF

5.10

(.201)

MIN

0.305 ±0.038

(.0120 ±.0015)

TYP

RECOMMENDED SOLDER PAD LAYOUT

0.254

(.010)

GAUGE PLANE

0.18

(.007)

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.

1.651 ±0.102
(.065 ±.004)

(.0197)

DETAIL “A”

DETAIL “A”

0.50

BSC

0° – 6° TYP

(.021 ±.006)

3.20 – 3.45

(.126 – .136)

0.53 ±0.152

SEATING

PLANE

4.90 ±0.152

(.193 ±.006)

(.043)

0.17 –0.27

(.007 – .011)

TYP

1.10

MAX

(.0197)

16

4.039 ±0.102
(.159 ±.004)

(NOTE 3)

1615 1413 121110

1 2 3 4 5 6 7 8

0.50

BSC

1.651 ±0.102
(.065 ±.004)

DETAIL “B”

9

9

3.00 ±0.102

(.118 ±.004)

0.12 REF

DETAIL “B”

CORNER TAIL IS PART OF

THE LEADFRAME FEATURE.

FOR REFERENCE ONLY

NO MEASUREMENT PURPOSE

0.280 ±0.076
(.011 ±.003)

REF

(NOTE 4)

0.86

(.034)

REF

0.1016 ±0.0508
(.004 ±.002)

MSOP (MSE16) 0213 REV F

28

31291fc

For more information www.linear.com/LTC3129-1

LTC3129-1

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

A 5/14 Clarified V

B 10/14 Clarified PGOOD Pin Description

Clarified Operation Paragraph

C 10/15 Changed MAX V

Modified MPPC section
Modified Table 4

Leakage to VIN if VCC > VIN: from –7µA to –27µA 4

CC

Current Limit

CC

9

16

4
16
22

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa­tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

For more information www.linear.com/LTC3129-1

31291fc

29

LTC3129-1

31291 TA08

+

Typical applicaTion

TEG Powered Converter Operates from a 10°C Temperature Differential and Provides 3.3V at 25mA

+

MARLOW NL1025T
TEG MOUNTED TO
A HEAT SINK WITH
LESS THAN 15°C/W
THERMAL RESISTANCE

for 50ms Every 15 Seconds for a Wireless Sensor

COILCRAFT

LPR6235-123QML

220µF

1:50

33nF

C1A

1nF

••

330k

C1B

C2A

C2B

SWA

SWB

V

INA

V

INB

VS2

VS1

V

AUX

1µF

LTC3109

V

V

OUT2_EN

PGOOD

STORE

V

OUT2

V

OUT

VLDO

V

V

470µF

6.3V

AUX

AUX

1N4148

1M

1µF

10pF

3.01M

1M

22nF

BST1

SW1

V

IN

RUN

V

CC

MPPC

PWM

VS1

VS2

VS3

GND

4.7µH

LTC3129-1

22nF

SW2

BST2

V

OUT

PGOOD PGOOD

V

CC

PGND

1M

2.2µF

BAT54

10µF

V

3.3V

OUT

relaTeD parTs

PART NUMBER DESCRIPTION COMMENTS

LTC3103 15V, 300mA Synchronous Step-Down DC/DC Converter with

Ultralow Quiescent Current

LTC3104 15V, 300mA Synchronous Step-Down DC/DC Converter with

Ultralow Quiescent Current and 10mA LDO

LTC3105 400mA Step-up Converter with MPPC and 250mV Start-Up V

LTC3112 15V, 2.5A, 750kHz Monolithic Synch Buck/Boost V

= 2.2V, V

V

IN(MIN)

ISD <1µA, 3mm × 3mm DFN-10, MSOP-10 Packages

= 2.2V, V

V

IN(MIN)

I

<1µA, 4mm × 3mm DFN-14, MSOP-16 Packages

SD

= 0.2V, V

IN(MIN)

I

<1µA, 3mm × 3mm DFN-10/MSOP-12 Packages

SD

= 2.7V, V

IN(MIN)

ISD <1µA, 4mm × 5mm DFN-16 TSSOP-20E Packages

LTC3115-1 40V, 2A, 2MHz Monolithic Synch Buck/Boost V

LTC3531 5.5V, 200mA, 600kHz Monolithic Synch Buck/Boost V

LTC3388-1/

20V, 50mA High Efficiency Nano Power Step-Down Regulator V

LTC3388-3
LTC3108/

Ultralow Voltage Step-Up Converter and Power Manager V

LTC3108-1
LTC3109 Auto-Polarity, Ultralow Voltage Step-Up Converter and Power

Manager

LTC3588-1 Piezo Electric Energy Harvesting Power Supply V

= 2.7V, V

IN(MIN)

I

<1µA, 4mm × 5mm DFN-16 and TSSOP-20E Packages

SD

= 1.8V, V

IN(MIN)

I

<1µA, 3mm × 3mm DFN-8 and ThinSOT Packages

SD

= 2.7V, V

IN(MIN)

IQ = 720nA, ISD = 400nA, 3mm × 3mm DFN-10, MSOP-10 Packages

= 0.02V, V

IN(MIN)

IQ = 6µA, ISD <1µA, 3mm × 4mm DFN-12, SSOP-16 Packages

= 0.03V, V

V

IN(MIN)

IQ = 7µA, ISD <1µA, 4mm × 4mm QFN-20, SSOP-20 Packages

= 2.7V, V

IN(MIN)

IQ = 950nA, ISD 450nA, 3mm × 3mm DFN-10, MSOP-10E Packages

LTC4070 Li-Ion/Polymer Low Current Shunt Battery Charger System V

= 450nA to 50mA, V

IN(MIN)

2mm × 3mm DFN-8, MSOP-8 Packages

Linear Technology Corporation

30

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

For more information www.linear.com/LTC3129-1

●

www.linear.com/3129-1

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

IN(MAX)

= 15V, V

= 15V, V

= 5V, V

OUT(MIN)

= 15V, V

= 40V, V

= 5.5V, V

=20V, V

OUT(MIN)

= 1V, V

= 1V, V

= 20V, V

FLOAT

 LINEAR TECHNOLOGY CORPORATION 2013

= 0.8V, IQ = 1.8µA,

OUT(MIN)

= 0.8V, IQ = 2.8µA,

OUT(MIN)

= 0 5.25V

= 2.7V to 14V, IQ = 50µA,

OUT(MIN)

= 2.7V to 40V, IQ = 50µA,

OUT(MIN)

OUT(MIN)

MAX

= 2V to 5V, IQ = 16µA,

= Fixed 1.1V to 5.5V,

= Fixed 2.35V to 5V,

OUT(MIN)

= Fixed 2.35V to 5V,

OUT(MIN)

= Fixed 1.8V to 3.6V,

OUT(MIN)

+ 4.0V, 4.1V, 4.2V, IQ = 300nA,

LT 1015 REV C • PRINTED IN USA

, IQ = 22µA,

31291fc