Linear Technology LTC1872 User Manual

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Page 1

Constant Frequency

Current Mode Step-Up

DC/DC Controller in SOT-23

FeaTures DescripTion

LTC1872

High Efﬁciency: Over 90%

High Output Currents Easily Achieved

Wide VIN Range: 2.5V to 9.8V

Limited Only by External Components

OUT

Constant Frequency 550kHz Operation

Burst Mode™ Operation at Light Load

Current Mode Operation for Excellent Line and Load Transient Response

Low Quiescent Current: 270µA

Shutdown Mode Draws Only 8µA Supply Current

±2.5% Reference Accuracy

Tiny 6-Lead SOT-23 Package

applicaTions

Lithium-Ion-Powered Applications

Cellular Telephones

Wireless Modems

Portable Computers

Scanners

The LTC®1872 is a constant frequency current mode step- up DC/DC controller providing excellent AC and DC load and line regulation. The device incorporates an accurate undervoltage lockout feature that shuts down the LTC1872 when the input voltage falls below 2.0V.

The LTC1872 boasts a ±2.5% output voltage accuracy and consumes only 270µA of quiescent current. For ap

plications where efficiency is a prime consideration, the LTC1872

is configured for Burst Mode operation, which

enhances efficiency at low output current.

In shutdown, the device draws a mere 8µA. The high 550kHz constant operating frequency allows the use of a small external inductor.

The LTC1872 is available in a small footprint 6-lead SOT-23.

L, LT , LT C , LT M, Burst Mode, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

Typical applicaTion

147k

220pF

80.6k

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: MURATA GRM42-2X5R226K6.3 D1: IR10BQ015 L1: MURATA LQN6C4R7M04 M1: IRLMS2002 R1: DALE 0.25W

ITH/RUN

LTC1872

GND

422k

Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter

SENSE

NGATE

–

0.03Ω

4.7µH

For more information www.linear.com/LTC1872

C1 10µF 10V

C2 2× 22F

6.3V

3.3V

OUT

5V 1A

1872 TA01

Efficiency vs Load Current

100

VIN = 3.3V

= 5V

OUT

EFFICIENCY (%)

10 100 1000

LOAD CURRENT (mA)

1872 TA01b

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Page 2

LTC1872

–

TOP VIEW

6-LEAD PLASTIC SOT-23

pin conFiguraTionabsoluTe MaxiMuM raTings

(Note 1)

Input Supply Voltage (VIN) ......................... –0.3V to 10V

–

SENSE V

NGATE Peak Output Current (< 10µs) ........................ 1A

Storage Ambient Temperature Range ....– 65°C to 150°C

Operating Temper ature Range (Note 2)....– 40°C to 85°C

Junction Tempera ture (Note 3) ............................ 150°C

, NGATE Voltages ............. –0.3V to (VIN + 0.3V)

, ITH/RUN Voltages .............................. –0.3V to 2.4V

/RUN 1

GND 2

JMAX

6 NGATE

5 V

4 SENSE

S6 PACKAGE

= 150°C, θJA = 230°C/W

Lead Temperature (Soldering, 10 sec) ...................300°C

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC1872ES6#PBF LTC1872ES6#TRPBF LTMK 6-Lead Plastic SOT-23 –40°C to 85°C

Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on nonstandard lead based finish parts.

For more information on lead free part marking, go to: For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

http://www.linear.com/leadfree/

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)

PARAMETER CONDITIONS MIN TYP MAX UNITS

Input DC Supply Current Normal Operation Sleep Mode Shutdown UVLO

Undervoltage Lockout Threshold V

Shutdown Threshold (at I

Start-Up Current Source V

Regulated Feedback Voltage 0°C to 70°C(Note 5)

Input Current (Note 5) 10 50 nA

Oscillator Frequency V

Gate Drive Rise Time C

Gate Drive Fall Time C

Peak Current Sense Voltage (Note 6) 114 120 mV

Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired.

Note 2: The LTC1872E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls.

Note 3: T dissipation P

is calculated from the ambient temperature TA and power

according to the following formula:

= TA + (PD • θJA°C/W)

/RUN)

Typicals at V

2.4V ≤ VIN ≤ 9.8V

2.4V ≤ VIN ≤ 9.8V, V VIN < UVLO Threshold

Falling

Rising

/RUN = 0V 0.25 0.5 0.85 µA

ITH

–40°C to 85°C(Note 5)

= 0.8V 500 550 650 kHz

LOAD

= 4.2V (Note 4)

/RUN = 0V

ITH

1.55

1.85

0.15 0.35 0.55 V

0.780

0.770

= 3000pF 40 ns

Dynamic supply current is higher due to the gate charge being

Note 4:

delivered at the switching frequency. Note 5: The LTC1872 is tested in a feedback loop that servos V

output of the error amplifier. Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense

voltage is V increases due to slope compensation as shown in Figure 2.

/6.67 at duty cycle <40%, and decreases as duty cycle

REF

270 230

8 6

2.00

2.10

0.800

420 370

22 10

2.35

2.40

0.820

0.830

to the

µA µA µA µA

1872fa

V V

For more information www.linear.com/LTC1872

Page 3

Typical perForMance characTerisTics

LTC1872

Reference Voltage vs Temperature

825

VIN = 4.2V

820

815

810

805

800

795

VOLTAGE (mV)

790

785

780

775

–35 5

–15

–55

45 125

TEMPERATURE (°C)

105

1872 G01

Maximum Current Sense Trip Voltage vs Duty Cycle

130

120

110

100

– (mV)

SENSE

– V

20 30

40 50

DUTY CYCLE (%)

60 70

Normalized Oscillator Frequency vs Temperature

VIN = 4.2V

–2

–4

–6

NORMALIZED FREQUENCY (%)

–8

–10

–35 5

–55

VIN = 4.2V

= 25°C

80 90

–15

TEMPERATURE (°C)

100

1872 G04

45 125

105

1872 G02

Shutdown Threshold vs Temperature

600

VIN = 4.2V

560

520

480

440

400

360

/RUN VOLTAGE (mV)

320

280

240

200

–35 5

–15

–55

TEMPERATURE (°C)

Undervoltage Lockout Trip Voltage vs Temperature

2.24 VIN FALLING

2.20

2.16

2.12

2.08

2.04

2.00

1.96

UVLO TRIP VOLTAGE (V)

1.92

1.88

1.84

–35 5

–15

–55

45 125

TEMPERATURE (°C)

105

1872 G05

45 125

105

1872 G03

pin FuncTions

ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run control input. Nominal voltage range for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes the device to be shut down. In shutdown all functions are disabled and the NGATE pin is held low.

GND (Pin 2): Ground Pin.

(Pin 3): Receives the feedback voltage from an external

resistive divider across the output.

–

SENSE

(Pin 4): The Negative Input to the Current Com-

parator.

(Pin 5): Supply Pin. Must be closely decoupled to

GND Pin 2.

NGATE (Pin 6): Gate Drive for the External N-Channel MOSFET. This pin swings from 0V to V

For more information www.linear.com/LTC1872

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LTC1872

FuncTional DiagraM

–

SENSE

ICMP

–

OSC

FREQ

FOLDBACK

0.3V

GND

–

SLOPE

COMP

–

0.5µA

VOLTAGE

REFERENCE

UNDERVOLTAGE

LOCKOUT

0.3V

0.15V

0.8V

REF

0.35V

–

/RUN

–

BURST

CMP

SHDN

CMP

SWITCHING

LOGIC AND BLANKING

CIRCUIT

OVP

SLEEP

EAMP

SHDN

NGATE

REF

–

60mV

REF

0.8V V

–

1.2V

1872FD

operaTion

(Refer to Functional Diagram)

Main Control Loop

The LTC1872 is a constant frequency current mode switching regulator. During normal operation, the external N-channel power MOSFET is turned on each cycle by the oscillator and turned off when the current comparator (ICMP) resets the RS latch. The peak inductor current at which ICMP resets the RS latch is controlled by the voltage on the I

/RUN pin, which is the output of the

error amplifier EAMP. An external resistive divider connected between V

and ground allows the EAMP to

OUT

receive an output feedback voltage V current increases, it causes a slight decrease in V

relative to the 0.8V reference, which in turn causes the

/RUN voltage to increase until the average inductor

current matches the new load current.

The main control loop is shut down by pulling the ITH/ RUN pin low. Releasing I current source to charge up the external compensation network. When the I control loop is enabled with the I pulled up to its zero current level of approximately 0.7V. As the external compensation network continues to charge

. When the load

For more information www.linear.com/LTC1872

up, the corre allowing normal operation.

/RUN allows an internal 0.5µA

/RUN pin reaches 0.35V, the main

/RUN voltage then

sponding output current trip level follows,

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Page 5

operaTion

0.7

( )

LTC1872

Comparator OVP guards against transient overshoots > 7.5% by turning off the external N-channel power MOSFET and keeping it off until the fault is removed.

Burst Mode Operation

The LTC1872

enters Burst Mode operation at low load currents. In this mode, the peak current of the inductor is set as if V the voltage at the I

/RUN = 1V (at low duty cycles) even though

ITH

/RUN pin is at a lower value. If the

inductor’s average current is greater than the load requirement, the voltage at the I

/RUN voltage goes below 0.85V, the sleep signal goes

/RUN pin will drop. When the

high, turning off the external MOSFET. The sleep signal goes low when the I

/RUN voltage goes above 0.925V

and the LTC1872 resumes normal operation. The next oscillator cycle will turn the external MOSFET on and the switching cycle repeats.

Undervoltage Lockout

To prevent operation of the N-channel MOSFET below safe input voltage levels, an undervoltage lockout is incorpo

rated into the LTC1872. When the input supply voltage drops below approximately 2.0V,

the N-channel MOSFET and all circuitry is turned off except the undervoltage block, which draws only several microamperes.

Overvoltage Protection

The overvoltage comparator in the LTC1872 will turn the external MOSFET off when the feedback voltage has risen

7.5% above the reference voltage of 0.8V. This comparator has a typical hysteresis of 20mV.

Slope Compensation and Inductor’s Peak Current

The inductor’s peak current is determined by:

−

IPK=

ITH

10 R

SENSE

when the LTC1872 is operating below 40% duty cycle. However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor

current

The amount of reduction is given by the curves

in Figure 2.

Short-Circuit Protection

Since the power switch in a boost converter is not in series with the power path from input to load, turning off the switch provides no protection from a short-circuit at the output. External means such as a fuse in series with the boost inductor must be employed to handle this fault condition.

110

100

(%)

OUT(MAX)

OUT

SF = I

Figure 2. Maximum Output Current vs Duty Cycle

For more information www.linear.com/LTC1872

= 0.4I

RIPPLE

AT 5% DUTY CYCLE

VIN = 4.2V

0 70 80 90 1006010 20 30 40 50

= 0.2I

RIPPLE

AT 5% DUTY CYCLE

DUTY CYCLE (%)

1872 F02

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Page 6

LTC1872

0.03

SENSE

applicaTions inForMaTion

The basic LTC1872 application circuit is shown in Figure1. External component selection is driven by the

OUT

and

(= C2).

load requirement and begins with the selection of L1 R diode D1 is selected followed by C

(= R1). Next, the power MOSFET and the output

SENSE

(= C1) and C

Selection for Output Current

SENSE

is chosen based on the required output current.

SENSE

With the current comparator monitoring the voltage developed across R

, the threshold of the comparator

SENSE

determines the inductor’s peak current. The output current the LTC1872 can provide is given by:



OUT

where I

0.12





SENSE

is the inductor peak-to-peak ripple current

RIPPLE

−

RIPPLE

(see Inductor Value Calculation section) and V





OUT

+ V

is the



forward drop of the output diode at the full rated output current.

A reasonable starting point for setting ripple current is:

Inductor Value Calculation

The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use of a smaller inductor for the same amount of inductor ripple current.

However, this is at the expense of efficiency due

to an increase in MOSFET gate charge losses.

The inductance value also has a direct effect on ripple current. The ripple current, I inductance or frequency and increases with higher V

, decreases with higher

RIPPLE

OUT

The inductor’s peak-to-peak ripple current is given by:

RIPPLE





f L

( )



OUT

+ VD−V

+ V

OUT



 

where f is the operating frequency. Accepting larger values of I

allows the use of low inductances, but results

RIPPLE

in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is:

RIPPLE

= 0.4 I

( )

OUT MAX

( )



OUT





+ V



 

RIPPLE

= O.4

( )

OUT

Rearranging the above equation, it becomes:

for Duty Cycle <40%

SENSE

( )

OUT







OUT

+ V

  

However, for operation that is above 40% duty cycle, slope compensation’s effect has to be taken into consideration to select the appropriate value to provide the required amount of current. Using the scaling factor (SF, in %) in Figure 2, the value of R

SENSE

( )

OUT

SENSE

100

( )

is:







OUT

+ V

  

In Burst Mode operation, the ripple current is normally set such that the inductor current is continuous during the burst periods. Therefore, the peak-to-peak ripple current must not exceed:

RIPPLE

≤

This implies a minimum inductance of:

 





 

MIN



0.03





SENSE

A smaller value than L

+ VD−V

OUT

+ V

OUT

could be used in the circuit;

MIN



 

however, the inductor current will not be continuous during burst periods.

1872fa

For more information www.linear.com/LTC1872

Page 7

applicaTions inForMaTion

CIN Required I

0.3

( )

LTC1872

Inductor Selection

When selecting the inductor, keep in mind that inductor saturation current has to be greater than the current limit set by the current sense resistor. Also, keep in mind that the DC resistance of the inductor will affect the efficiency. Off the shelf inductors are available from Murata, Coilcraft, Toko, Panasonic, Coiltronics and many other suppliers.

Power MOSFET Selection

The main selection criteria for the power MOSFET are the threshold voltage V

, the “on” resistance R

GS(TH)

reverse transfer capacitance C

and total gate charge.

RSS

DS(ON)

Since the LTC1872 is designed for operation down to low input voltages, a logic level threshold MOSFET (R guaranteed at V

= 2.5V) is required for applications

DS(ON)

that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1872 is less than the absolute maximum V

The required minimum R

DS(ON)

rating, typically 8V.

of the MOSFET is governed

by its allowable power dissipation given by:

1+δp

( )

DS(ON)

≅

( )

where PP is the allowable power dissipation and δp is the temperature dependency of R given for a MOSFET in the form of a normalized R

. (1 + δp) is generally

DS(ON)

vs temperature curve, but δp = 0.005/°C can be used as an approximation for low voltage MOSFETs. DC is the maximum operating duty cycle of the LTC1872.

Output Diode Selection

It is important to adequately specify the diode peak current and average power dissipation so as not to exceed the diode ratings.

Schottky diodes are recommended for low for

ward drop and fast switching times. Remember to keep lead length short and observe proper grounding (see Board Layout Checklist) to avoid ringing and increased dissipation.

and C

Selection

OUT

To prevent large input voltage ripple, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current for a boost converter is approximately equal to:

where I

≈

RMS

is as defined in the Inductor Value Calcula-

RIPPLE

tion section.

Note that capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may be paralleled to meet the size or height requirements in the design. Due to the high operating frequency of the LTC1872, ceramic capacitors can also be used for C

Always consult the manufacturer if there is any question.

The selection of C

is driven by the required effective

OUT

series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆V

) is approximated by:

OUT

Under normal load conditions, the average current con ducted by the diode in a boost converter is equal to the output load current:

D(avg)

OUT



≈ IO•

 

ΔV

For more information www.linear.com/LTC1872

OUT

 

ESR

 

OUT



•

 







 







1872fa

+ V

RIPPLE





2πfC



Page 8

LTC1872

applicaTions inForMaTion

where f is the operating frequency, C capacitance and I

is the ripple current in the inductor.

RIPPLE

is the output

OUT

Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR (size) product of any aluminum electrolytic at a somewhat higher price. The output capacitor RMS current is approximately equal to:

IPK• DC−DC

where IPK is the peak inductor current and DC is the switch duty cycle.

When using electrolytic output capacitors, if the ripple and ESR requirements are met, there is likely to be far more capacitance than required.

In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. An excellent choice of tantalum capacitors is the AVX TPS and KEMET T510 series of surface mount tantalum capacitors. Also, ceramic capacitors in X5R pr X7R dielectrics offer excellent performance.

Low Supply Operation

Setting Output Voltage

The LTC1872 develops a 0.8V reference voltage between the feedback (Pin 3) terminal and ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set the overall output voltage. The regulated output voltage is determined by:

= 0.8V 1+

OUT

 

105

100

NORMALIZED VOLTAGE (%)

2.0

Figure 3. Line Regulation of V

LTC1872





 

REF

ITH

2.2 2.4 2.6 2.8 INPUT VOLTAGE (V)

REF

1872 F03

and V

OUT

3.0

ITH

Although the LTC1872 can function down to approximately 2.0V, the maximum allowable output current is reduced when

VIN decreases below 3V. Figure 3 shows the amount of change as the supply is reduced down to 2V. Also shown in Figure 3 is the effect of VIN on V VIN goes below 2.3V.

REF

For more information www.linear.com/LTC1872

1872 F04

Figure 4. Setting Output Voltage

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Page 9

applicaTions inForMaTion

LTC1872

For most applications, an 80k resistor is suggested for R1. To prevent stray pickup, locate resistors R1 and R2 close to LTC1872.

Efficiency Considerations

The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as:

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percentage of input power.

Although all

dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1872 circuits: 1) LTC1872 DC bias current, 2) MOSFET gate charge current, 3) I2R losses and 4) voltage drop of the output diode.

1. The VIN current is the DC supply current, given in the

electrical characteristics, that excludes MOSFET driver and control currents. VIN current results in a small loss which increases with VIN.

2. MOSFET gate charge current results from switching the gate capacitance of the power MOSFET. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from V to ground. The resulting dQ/dt is a current out of V

IN IN

which is typically much larger than the contoller’s DC supply current. In continuous mode, I

R losses are predicted from the DC resistances of

3. I

GATECHG

= f(Qp).

the MOSFET, inductor and current sense resistor. The MOSFET R

multiplied by duty cycle times

DS(ON)

the average output current squared can be summed with I2R losses in the inductor ESR in series with the current sense resistor.

4. The output diode is a major source of power loss at high currents. The diode loss is calculated by multiplying the forward voltage by the load current.

5. Transition losses apply to the external MOSFET and increase at higher operating frequencies and input voltages. Transition losses can be estimated from:

Transition Loss = 2(VIN)2I

IN(MAX)CRSS

Other losses, including CIN and C

(f)

ESR dissipative

OUT

losses, and inductor core losses, generally account for less than 2% total additional loss.

For more information www.linear.com/LTC1872

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Page 10

LTC1872

applicaTions inForMaTion

PC Board Layout Checklist

When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1872. These items are illustrated graphically in the layout diagram in Figure 5. Check the following in your layout:

1. The Schottky diode should be closely connected between the output capacitor and the drain of the external MOSFET.

The (+) plate of C

should connect to the sense resis-

tor as closely as possible. This capacitor provides AC current to the inductor.

3. The input decoupling capacitor (0.1µF) should be connected closely between VIN (Pin 5) and ground (Pin 2).

ITH/RUN

LTC1872

ITH

GND

NGATE

SENSE

0.1µF

–

4. Connect the end of R

as close to VIN (Pin 5) as

SENSE

possible. The VIN pin is the SENSE+ of the current comparator.

5. The trace from SENSE– (Pin 4) to the Sense resistor should be kept short. The trace should connect close to R

SENSE

6. Keep the switching node NGATE away from sensitive small signal nodes.

7. The VFB pin should connect directly to the feedback resistors. The resistive divider R1 and R2 must be connected between the (+) plate of C

and signal

OUT

ground.

OUT

BOLD LINES INDICATE HIGH CURRENT PATHS

Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)

1872 F05

1872fa

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Page 11

Typical applicaTion

10k

220pF

LTC1872 12V/500mA Boost Converter

0.033Ω

SENSE

NGATE

–

L1 10µH

78.7k

ITH/RUN

LTC1872

GND

1.1M

LTC1872

1872 TA02

3V TO 9.8V

OUT

12V

C1 10µF 10V

C2 47µF 16V

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSE476M016R0150 D1: IR10BQ015

= 3 AA CELLS ≈ 2.7V TO 4.8V

10k

220pF

L1: COILTRONICS UP2B-100 M1: Si9804DV R1: DALE 0.25W

LTC1872 Three-Cell White LED Driver

0.27Ω

SENSE

NGATE

–

L1 150µH

ITH/RUN

LTC1872

GND

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT C2: AVX TPSD156M035R0300 D0: MOTOROLA MBR0540 D1-D7: CMD333UWC

C1 10µF 10V

L1: COILCRAFT DO1608C-154 M1: Si9804 R1: DALE 0.25W

C2 15µF 35V

0.1µF CERAMIC

1 TO 8

WHITE

LEDs

≈ 28.8V

OUT

(WITH 8 LEDs)

15mA

53.6Ω

•

1872 TA04

For more information www.linear.com/LTC1872

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Page 12

LTC1872

package DescripTion

3.85 MAX

0.62

MAX

2.62 REF

RECOMMENDED SOLDER PAD LAYOUT

PER IPC CALCULATOR

0.95 REF

1.22 REF

1.4 MIN

Dimensions in inches (millimeters) unless otherwise noted.

S6 Package

6-Lead Plastic TSOT-23

(Reference LTC DWG # 05-08-1636)

2.90 BSC (NOTE 4)

2.80 BSC

1.50 – 1.75 (NOTE 4)

PIN ONE ID

0.95 BSC

0.80 – 0.90

0.30 – 0.45 6 PLCS (NOTE 3)

0.20 BSC

DATUM ‘A’

0.30 – 0.50 REF

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DRAWING NOT TO SCALE

3. DIMENSIONS ARE INCLUSIVE OF PLATING

4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

5. MOLD FLASH SHALL NOT EXCEED 0.254mm

6. JEDEC PACKAGE REFERENCE IS MO-193

0.09 – 0.20 (NOTE 3)

1.00 MAX

0.01 – 0.10

1.90 BSC

S6 TSOT-23 0302

1872fa

For more information www.linear.com/LTC1872

Page 13

LTC1872

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

A 09/15 Revised package drawing 12

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

For more information www.linear.com/LTC1872

1872fa

Page 14

LTC1872

Typical applicaTion

LTC1872 –2.5V to 3.3V/0.5A Boost Converter

LTC1872 2.7V to 9.8V Input

to 3.3V/1.2A Output SEPIC Converter

0.1µF

CERAMIC

–2.5V

220pF

C1 100µF 10V

C1, C2: AVX TPSE107M010R0100 D1: MOTOROLA MBR2045CT L1: COILTRONICS UP2B-4R7

10k

ITH/RUN

LTC1872

GND

–

SENSE

NGATE

M1: Si9804DV R1: DALE 0.25W U1: PANASONIC 2SB709A

0.034Ω

4.7µH

C2 2¥ 100F

10V

332k

80.6k 180k

1872 TA03

3.3V

0.5A

OUT

220pF

10k

ITH/RUN

GND

80.6k R

252k

, CS; TOKO, MURATA OR TAIYO YUDEN

: PANASONIC EEFUE0G181R

L1: BH ELECTRONICS 511-1012 M1: IRLMS2002

: DALE OR IRC

LTC1872

SENSE

NGATE

2.7V TO 9.8V

= 5V CHANGE

10µF 10V, X5R

C01 180µF 4V, SP

OUT

3.3V/1.2A

1872 TA05

0.03Ω

L1A L1B

–

MBRM120

4.7µF 10V

FOR V

OUT

TO 427kΩ AND

TO 150µF, 6V PANASONIC

SP TYPE CAPACITOR

relaTeD parTs

PART NUMBER DESCRIPTION COMMENTS

LT1304 Micropower DC/DC Converter with Low-Battery Detector 120µA Quiescent Current, 1.5V ≤ V

LT1610 1.7MHz, Single Cell Micropower DC/DC Converter 30µA Quiescent Current, V

LT1613 1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23 Internally Compensated, V

LT1619 Low Voltage Current Mode PWM Controller 8-Lead MSOP Package, 1.9V ≤ V

LT1680 High Power DC/DC Step-Up Controller Operation Up to 60V, Fixed Frequency Current Mode

LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 8-Pin N-Channel Drive, 3.5V ≤ V

LT1615 Micropower Step-Up DC/DC Converter in SOT-23 20µA Quiescent Current, V

LTC1700 No RSENSE Synchronous Current Mode DC/DC Step-Up Controller 95% Efficient, 0.9V ≤ V

LTC1772 Constant Frequency Current Mode Step-Down DC/DC Controller V

2.5V to 9.8V, I

≤ 5V, 550kHz Operation

up to 4A, SOT-23 Package

OUT

LTC3401/LTC3402 1A/2A, 3MHz Micropower Synchronous Boost Converter 10-Lead MSOP Package, 0.5V ≤ V

≤ 8V

Down to 1V

≤ 18V

≤ 36V

Down to 1V

≤ 5V

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507

For more information www.linear.com/LTC1872

●

www.linear.com/LTC1872

1872fa

1872f LT/TP 0915 4K REV A • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 2000