Linear Technology LTC1702 Demo Manual

DEMO MANUAL DC275

DC/DC CONVERTER

LTC1702

Dual 550kHz Synchronous

U

2-Phase 15A DC/DC Converter

DESCRIPTIO

Demonstration circuit DC275 is a dual, high efficiency
regulator using the LTC®1702 switching regulator con­troller. The LTC1702 is optimized for high efficiency with
low input voltages. Typical applications are power for a
digital signal processor (DSP), microprocessor and/or
an application specific integrated circuit (ASIC). The
input voltage of the LTC1702 can range from 3V to 7V.
One of the output voltages (V
other (V

) is programmable from 1.6V to 2.5V by

OUT1

) is fixed at 3.3V and the

OUT2

means of a jumper. The LTC1702 includes two complete,
on-chip, independent switching regulator controllers,
each designed to drive a pair of external N-channel
MOSFET devices in a voltage mode control, synchronous

buck configuration. The LTC1702 also provides open­drain logic outputs (PGOOD1 and PGOOD2) that indicate
whether either output has risen to within 5% of the final
output voltage. An optional latching fault mode protects
the load if the output rises 15% above the intended
voltage. The LTC1702 uses a constant 550kHz switching
frequency, minimizing external component size and maxi­mizing load transient performance. Operating efficien­cies exceeding 90% are obtained for load current currents
from 1A to 14A. Additionally, the supply current in
shutdown is less than 100µA. Gerber files for this circuit

board are available. Call the LTC factory.

, LTC and LT are registered trademarks of Linear Technology Corporation.

UWWW

PERFOR A CE SU ARY

PARAMETER CONDITIONS VALUE

V

IN

V

OUT2

I

OUT2

Typical Output Ripple I
V

OUT1

I

OUT1

Typical Output Ripple I
I

Q

Input Voltage Range 4.75V to 7V
Fixed Output Voltage 3.3V
Maximum Output Load Current 15A

= 15A 18mV

OUT

Jumper Selectable Output Voltage 1.6V, 1.8V, 2V or 2.5V
Maximum Output Load Current 15A

= 15A 17mV

OUT

Supply Current in Shutdown 100µA

U

W

TYPICAL PERFOR A CE CHARACTERISTICS A D BOARD PHOTO

LTC1702 Efficiency

100

V

= 5V

IN

90

EFFICIENCY (%)

80

70

0

= 3.3V

V

OUT

= 2.5V

V

OUT

= 1.6V

V

OUT

510

LOAD CURRENT (A)

15

1702 G01

U

1

DEMO MANUAL DC275

123456789

101112

TOP VIEW

GN PACKAGE

24-LEAD NARROW PLASTIC SSOP

2423222120191817161514

13

PV

CC

BOOST1

BG1

TG1

SW1

I

MAX1

PGOOD1

FCB

RUN/SS1

COMP1

SGND

FB1

I

MAX2

BOOST2

BG2

TG2

SW2

PGND

PGOOD2

FAULT

RUN/SS2

COMP2

FB2

V

CC

DC/DC CONVERTER

WUW

PACKAGE A D SCHE ATIC DIAGRA SM

E10

IN

V

E9

GND

1µF

C24

C3

330µF

+

+

C2

C1

330µF

330µF

OUT2

3.3V/15A

E4

V

L2

Q6

Q5

R11

1µH

D4

Q8

Q7

+

D1

C4

10µF

R1

10Ω

C16 1µF

R10 27k

2423222120191817161514

5%

TG2

BG2

MAX2

I

BOOST2

SW2

PGND

+

+

1.6k

PGOOD2

C26

C19

C20

R12

1µF

+

180µF

+

180µF

15.8k

C18

C17 1µF

FAULT

C27

180µF

C25

180µF

1%

680pF

RUN/SS2

E11

R13

4.99k

COMP2

GND

1%

FB2

E6

FAULTE7SD2

DC175 F01

R15 68k

C22

C21

3300pF

27pF

E5

PWRGD2

R14

10k

IN

V

13

CC

V

CC

PV

BOOST1

1

2

C5

1µF

D2

C6

1µF

1µF

C23

NOTES: UNLESS OTHERWISE SPECIFIED

D1, D2: ON SEMICONDUCTOR MBR0520LT1

D3, D4: ON SEMICONDUCTOR MBRS340T3

Q1 TO Q8: FAIRCHILD FDS6670A

3

BG1

Q2Q1

LTC1702

MAX1

TG1

SW1

I

PGOOD1

FCB

RUN/SS1

COMP1

SGND

4

5

6

7

8

9

R2 27k

Q4Q3

D3

L1

1µH

C7 1µF

IN

V

C13

820pF

R3

1.2k

R4

10k

1%

101112

BURST

R16

8.87k

2.5V

1%

R7

20k2V1%

R6

40.2k

1.8V

1%

R5

10k

1.6V

1%

JP4

FB1

CONT

JP3

JP2

JP1

C15

27pF

R9 47k

C14

680pF

JP3

OUT

OUT

OUT

JP2

OUT

OUTINOUT

JP1

OUTINOUT

JUMPER SELECT

OUT1

1.6V

1.8V

2.0V

V

Figure 1. Dual 550kHz Synchronous 2-Phase 15A DC/DC Converter

IN

OUT

2.5V

LTC1702CGN

+

C11

180µF

C12

180µF

+

+

C9

180µF

C10

AT 15A

180µF

+

C8

1µF

E1

OUT1

V

1.6V, 1.8V,

2V OR 2.5V

R8

10k

IN

V

E2

E8

GND

E3

PWRGD1

SD1

2

DEMO MANUAL DC275

DC/DC CONVERTER

PARTS LIST

REFERENCE
DESIGNATOR QUANTITY PART NUMBER DESCRIPTION VENDOR TELEPHONE

C1 to C3 3 T510X337K010AS 330µF 10V 10% Tantalum Capacitor Kemet [phone]
C4 1 1206ZG106ZAT1A 10µF 10V Y5V Capacitor AVX [phone]
C5 to C8, C16, 9 06036D105MAT1A 1µF 6V X5R Capacitor AVX [phone]

C17, C20, C23
C24

C9 to C12, 8 EEFUEOG181R 180µF 4V SP Capacitor Panasonic [phone]
C19, C25 to
C27

C13 1 06035C821MAT1A 820pF 50V X7R Capacitor AVX [phone]
C14, C18 2 06035C681MAT1A 680pF 50V X7R Capacitor AVX [phone]
C15, C21 2 06035A270MAT1A 27pF 50V NPO Capacitor AVX [phone]
C22 1 06035C332MAT1A 3300pF 50V X7R Capacitor AVX [phone]
D1, D2 2 MBR0520LT1 Schottky Diode ON Semiconductor [phone]
D3, D4 2 MBRS340T3 Schottky Diode ON Semiconductor [phone]
E1, E4, E8 to E11 6 2501-2 1-Pin Terminal Mill-Max [phone]
E2, E3, E5 to E7 5 2308-2 1-Pin Terminal Mill-Max [phone]
JP1 to JP3 3 3801S-02G2 0.100"CC 2-Pin Jumper Comm Con [phone]
JP4 1 3801S-03G2 0.100"CC 3-Pin Jumper Comm Con [phone]
JP1, JP4 2 CCIJ230-G 0.100"CC Shunt Comm Con [phone]
L1, L2 2 CEP125-1R0MC-H or 1µH 20A SMT Inductor Sumida [phone]

ETQP6F1R0SSP Panasonic [phone]

Q1 to Q8 8 FDS6670A SO-8 N-Channel MOSFET Fairchild [phone]
R1 1 CR16-100JM 10Ω 1/16W 5% Chip Resistor Tad [phone]
R2, R10 2 CR16-273JM 27k 1/16W 5% Chip Resistor Tad [phone]
R3 1 CR16-122JM 1.2k 1/16W 5% Chip Resistor Tad [phone]
R4, R5 2 CR16-1002FM 10k 1/16W 1% Chip Resistor Tad [phone]
R6 1 CR16-4022FM 40.2k 1/16W 1% Chip Resistor Tad [phone]
R7 1 CR16-2002FM 20k 1/16W 1% Chip Resistor Tad [phone]
R8, R14 2 CR16-103JM 10k 1/16W 5% Chip Resistor Tad [phone]
R9 1 CR16-473JM 47k 1/16W 5% Chip Resistor Tad [phone]
R11 1 CR16-162JM 1.6k 1/16W 5% Chip Resistor Tad [phone]
R12 1 CR16-1582FM 15.8k 1/16W 1% Chip Resistor Tad [phone]
R13 1 CR16-4991FM 4.99k 1/16W 1% Chip Resistor Tad [phone]
R15 1 CR16-683JM 68k 1/16W 5% Chip Resistor Tad [phone]
R16 1 CR16-8871FM 8.87k 1/16W 1% Chip Resistor Tad [phone]
U1 1 LTC1702CGN 24-Lead SSOP IC LTC [phone]

3

DEMO MANUAL DC275

DC/DC CONVERTER

QUICK START GUIDE

Refer to Figure 2 for proper measurement equipment
setup and follow the procedure outlined below:

1. Connect the input power to the VIN and GND terminals
on the board using 12-gauge or heavier wire soldered
to the terminals. The input voltage is limited to
between 4.75V and 7V.

2. Connect an ammeter in series with the input supply to
measure input current.

3. Since this demo board operates from a low input
voltage and supplies high output current, it is essen­tial that the input supply voltage be well regulated. If
the input power supply is equipped with remote
sense lines, connect SENSE+ to the VIN terminal and
SENSE– to GND terminal on the board.

4. Connect either power resistors or an electronic load
to the V

OUT1

, V

and GND terminals using

OUT2

12-gauge or heavier wire, soldered to the terminals.

5. Connect an ammeter in series with each of the output
loads to measure output currents.

6. The SD1 and SD2 pins should be left floating for
normal operation and tied to GND for shutdown.

7. Connect a voltmeter across the VIN and GND termi­nals to measure input voltage.

8. Connect a voltmeter across the V
terminals and another across the V

and GND

OUT1

and GND

OUT2

terminals to measure the output voltages.

9. For applications where the minimum load current is
greater than 1A, set jumper JP4 to the “Continuous”
position.

10. Set the desired output voltage (V

) with jumpers

OUT1

JP1 to JP3, as shown in Table 1.

11. After all connections are made, turn on the power and
verify that V

Table 1

POSITION OUTPUT VOLTAGE

No Jumper 1.6V

JP1 1.8V
JP2 2.0V
JP3 2.5V

OUT1

and V

are correct.

OUT2

LOAD

INPUT

SUPPLY

PWRGD2

FAULT

SD2

GND

V

A

V

OUT2

Figure 2. Proper Measurement Setup

GND

DUAL OUTPUT BUCK REGULATORS

A

V

V

BURST CONT

DC275A-LTC1702CGN

PWRGD1

IN

SD1

V

GND

LOAD

V

A

OUT1

JP2
JP1
JP3

4

OPERATIO

DEMO MANUAL DC275

DC/DC CONVERTER

U

The circuit in Figure 1 highlights the capabilities of the
LTC1702. This design provides one fixed 3.3V output
(V
from 1.6V to 2.5V. The LTC1702 is a voltage mode
controller, designed to drive a pair of external N-channel
MOSFETs using a fixed 550kHz switching frequency. The
synchronous buck architecture automatically shifts to
discontinuous operation and then to Burst ModeTM opera­tion as the output load decreases, ensuring maximum
efficiency over a wide range of load currents. This mode is
recommended for load currents less than 1A and can be
implemented on the demo board by moving jumper JP4 to
the “Burst” position.

Theory of Operation

The LTC1702 has two independent switching regulators.
For the sake of simplicity and to minimize repetition, only
side “1” will be discussed. The divided output (V
compared to the 0.8V reference. The difference voltage is
multiplied by the error amplifier’s (FB) gain. The resulting
error signal is then compared to an internally generated,
fixed frequency sawtooth waveform by the PWM com­parator, which generates a pulse width modulated signal.
This PWM signal drives the external MOSFETs through
TG1 and BG1. The output of this chopper circuit is then
filtered by L1 and C9 to C12 to produce the desired DC
output voltage.

) and one output (V

OUT2

) that is jumper selectable

OUT1

OUT1

) is

Capacitor Considerations

The input capacitors are Kemet T510X337K010AS, 330µF,
10V tantalums. The input capacitors must be rated for the
RMS input ripple. A good rule of thumb is that the input
ripple current will be 50% of the output current. Since the
LTC1702 uses 2-phase switching, the input bulk capaci­tors should be able to fully handle the RMS ripple current
of just one load. As the load current increases on the other
side, it tends to cancel, rather than to add to, the ripple
current requirements for the input capacitors. For a con­tinuous output current of 15A, the ripple current rating of
the input capacitors should be 7.5A. The capacitors cho­sen are rated at 2.5A each, so three are adequate. Without
the 2-phase operation, six capacitors would be required to
handle two 15A loads.

Output capacitors need to have a ripple current rating
greater than the RMS value of the inductor ripple current.
This is a function of the operating frequency and inductor
value, as well as input and output voltages. Because the
ripple current is relatively small, the controlling parameter
is generally the capacitor’s ESR (equivalent series resis­tance). The maximum allowable ESR is equal to the
maximum allowable peak-to-peak output ripple voltage
divided by the peak-to-peak inductor ripple current. In
general, if the ESR is low enough for the ripple voltage and
transient requirements, the capacitors will have more than
adequate ripple current capability.

2-Phase Operation

The LTC1702 dual switching regulator controller also
features the considerable benefits of 2-phase operation.
The LTC1702 includes a single master clock that drives the
two sides such that side 1 is 180° out of phase with side

2. This technique, known as 2-phase switching, has the
effect of doubling the frequency of the switching pulses
seen by the input capacitor and significantly reduces their
RMS value. With 2-phase switching, the input capacitor is
sized as required to support the larger of the two sides at
maximum load current. As the load current increases on
the lower current side, it tends to cancel, rather than add
to, the RMS current seen by the input capacitor; thus no
additional capacitance is needed.

Inductor Selection

Inductor selection is not extremely critical. The inductor
used here was chosen for fairly low cost and ready
availability. The main concerns in choosing an appropri­ate inductor are the inductance value required, the satu­ration current rating and the temperature rise. Most
manufacturers specify a DC current rating that produces
a temperature rise of 40°C. If a design will not see high
ambient temperatures, a larger temperature rise can
usually be tolerated. Another maximum current specifica­tion is related to core saturation. A manufacturer may
specify that maximum rated current is the point at which
inductance is down by 10% (some specify 25%). Since
most core materials and structures will result in a gentle,

Burst Mode is a trademark of Linear Technology Corporation.

5

DEMO MANUAL DC275

DC/DC CONVERTER

U

OPERATIO

controlled
magical point where the inductor is no longer useful. Look
at what the inductance will be at the maximum load current
expected and determine if the output ripple will remain
within specified limits. If it will, the inductor will most likely
work correctly. Ripple current is generally designed for
between 10% and 40% of output current.

MOSFET Selection

The main concern with FET selection in very low voltage
applications is thermal management. At high current lev­els, power devices will get hot. The trick is to keep the
temperature rise within acceptable limits. Most of the
FETs’ power dissipation will be due to conduction losses.
Therefore, by choosing a FET with a sufficiently low R
the power dissipation, and therefore, the temperature rise,
can be made arbitrarily low. The price paid for very low
temperature rise is more expensive FETs. Switching losses
are a concern only for the high side FET. The low side FET
turns on and off into a forward-biased diode, so its tran­sition losses are very small. The high side FET, in contrast,
must provide all of the reverse recovery charge that the
low side FETs body diode will demand. This can result in
a significant amount of switching loss in this device.

Although it may seem that a lower on-resistance FET is
always desirable from an efficiency perspective, this is not
necessarily true. A smaller device will have a lower gate­charge power requirement and will also exhibit faster
switching transition times. The resulting reduction in AC
losses may more than offset the increase in conduction
losses. A smaller, higher on-resistance FET may prove the
more efficient, as well as the lower cost solution. As the
load current increases, gate-drive losses become less of a
concern. At output currents on the order of 15A, lower
resistance FETs will probably be better in terms of overall
efficiency, but not necessarily the most cost effective
choice. Each application will place a different value on a
few points of efficiency.

Shutdown/Soft-Start

Each half or the LTC1702 has a RUN/SS pin. This pin
performs two functions: when pulled to ground, each
shuts down its half of the LTC1702, and each acts as a

roll off of inductance with DC bias, there is no

,

DS(ON)

conventional soft-start pin, enforcing a duty cycle limit
proportional to the voltage at RUN/SS. An internal 4µA
current source pull-up is connected to each RUN/SS pin,
allowing a soft-start ramp to be generated with a single
external capacitor (C7 for side 1 and C17 for side 2) to
ground.

Current Limit

The I
maximum allowable voltage drop across the bottom
MOSFET before the current limit circuit engages. The
voltage across the bottom MOSFET is determined by its
on-resistance and by the current flowing in the inductor,
which is the same as the output current. To set the current
limit, connect an R
value of R

I

LIM

operating load current to account for MOSFET R
variations with temperature.

How to Measure Voltage Regulation and Efficiency

When trying to measure load regulation or efficiency,
voltage measurements should be made directly across the
V

OUT

of test leads at the load. Similarly, input voltage should be
measured directly on the VIN and GND terminals of the
LTC1702 demo board. Input and output current should be
measured by placing an ammeter in series with the input
supply and load. Refer to Figure 2 for the proper test
equipment setup. Refer to page one for typical efficiency
curves for VIN = 5V, V
IL = 1A to 15A.

How to Measure Output Voltage Ripple

In order to measure output voltage ripple, care must be
taken to avoid a long ground lead on the oscilloscope
probe. Therefore, a sturdy wire should be soldered on the
output side of the GND terminal. The other end of the wire
is looped around the ground side of the probe and should
be kept as short as possible. The tip of the probe is touched
directly to V

resistor, R2, sets the current limit by setting the

MAX

resistor from I

IMAX

is calculated as follows:

IMAX

R

= [(I

IMAX

should be chosen to be 150% of the maximum

and GND terminals and should not be taken at the end

• R

LIM

(see Figure 3). Bandwidth is generally

OUT

) + 100mV]/10µA

DS(ON)

= 3.3V, 2.5V and 1.8V, for

OUT

to GND. The

MAX

DS(ON)

6

OPERATIO

DEMO MANUAL DC275

DC/DC CONVERTER

U

limited to 20MHz for ripple measurements. Also, if mul­tiple pieces of line-powered test equipment are used, be
sure to use isolation transformers on their power lines to
prevent ground loops, which can cause erroneous results.
Figures 4 and 5 show the output voltage ripple for the 3.3V
and the 2.5V supplies for a 15A load.

GND V

Figure 3. Measuring Output Voltage Ripple

OUT

Transient Response

The LTC1702 uses true 25MHz gain bandwidth op amps
as the feedback amplifiers. This allows the use of an
OPTI-LOOPTM compensation scheme that can precisely
tailor the loop response. The high gain-bandwidth prod­uct allows the loop to be crossed over beyond 50kHz
while maintaining good stability, and significantly
enhances load transient response. Figures 6 and 7 show
the transient response of the 3.3V and the 2.5V output
supplies for a 0A to 10A load step. For more information
about loop compensation and stability analysis, consult
the LTC1702 data sheet.

OPTI-LOOP is a trademark of Linear Technology Corporation.

Figure 4. 3.3V Output Voltage Ripple, IL = 15A

Figure 5. 2.5V Output Voltage Ripple, IL = 15A

Figure 6. 3.3V Transient Response, IL = 0A to 10A

Figure 7. 2.5V Transient Response, IL = 0A to 10A

7

DEMO MANUAL DC275

DC/DC CONVERTER

U

OPERATIO

Heat Dissipation Issues

Since each side of the LTC1702 demo board can supply
15A of continuous load current, care must be taken not to
exceed the maximum junction temperature for the power
MOSFETs. A few possibilities for dissipating the power are
to use heat sinks and/or forced air cooling. Another
possibility is to use the PC board as a heat sink. On the
LTC1702 demo board, power MOSFETS Q1 to Q8 are
surrounded by ground and power planes on both sides of
the PC board. Also, there is metal on the inner layers
directly underneath the power MOSFETs. This helps in
spreading the heat and improves the power dissipation
capability of the PCB.

Layout Guidelines

Since the LTC1702 is a switching regulator, a good layout
is essential for good load regulation and minimizing radi­ated/conducted noise. If you want a layout that is guaran­teed to work, copy the LTC1702 Gerber files provided with
this demo board; otherwise, be sure to follow the layout
guidelines below:

1. The inductor L1, MOSFETs Q1 to Q4 and the Schottky
diode (D3) should be placed as close as possible to
each other; similarly, L2, Q5 to Q8 and D4 should be
placed as close together as possible. This junction
forms the switch node and should be kept as small as
possible to minimize radiated emissions. It must also
be large enough to carry the full rated output current.

4. C5 (1µF) should be as close as possible to Pin 1 on the

LTC1702.

5. R2 should be connected directly to the sources of Q3
and Q4.

6. R10 should be connected directly to the sources of Q7
and Q8.

7. Keep the trace from the FB1 pin to the junction of R4
and R5 short and use a long trace from the top of
resistor R4 to the output terminal, rather than vice
versa.

8. Keep the trace from the FB2 pin to the junction of R12
and R13 short and use a long trace from the top of
resistor R12 to the output terminal, rather than vice
versa.

9. The sources of the bottom MOSFETs Q3, Q4, Q7 and
Q8 should be tied back to the ground of input capaci­tors C1 to C3 by means of a wide trace, not by the
ground plane.

10. The grounds of the output capacitors C19–C20,
C25–C27 and C8–C12 should be tied directly to the
input capacitor’s ground by means of a wide trace or
by the ground plane.

11. The grounds of the feedback resistors, soft-start ca­pacitors and C4 should be referenced to the chip SGND
pin, which is then tied to the input bulk capacitors’
grounds.

2. The SW1 and the SW2 pins should be connected
directly to the respective switch nodes with a short
trace.

3. C4 (10µF) should be as close as possible to Pin 13 on
the LTC1702.

8

12. PGND, Pin 19, should connect directly to the ground
plane.

UW

PCB LAYOUT A D FIL

DEMO MANUAL DC275

DC/DC CONVERTER

Top Silkscreen

Top Solder Mask

Top Pastemask

9

DEMO MANUAL DC275

DC/DC CONVERTER

UW

PCB LAYOUT A D FIL

Layer 1, Top Layer Layer 2, VIN Plane

Layer 3, GND Plane Layer 4, Bottom Layer

10

UW

PCB LAYOUT A D FIL

DEMO MANUAL DC275

DC/DC CONVERTER

Bottom Silkscreen

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

Bottom Solder Mask

Bottom Pastemask

11

DEMO MANUAL DC275

DC/DC CONVERTER

U

PC FAB DRAWI G

2.50"

AA

F

C

E

EE

F

A

A

D

DD

F

EE

C

A

B

E

AA

A

F

NOTES: UNLESS OTHERWISE SPECIFIED

1. MATERIAL: FR4 OR EQUIVALENT EPOXY,

2 OZ COPPER CLAD, THICKNESS 0.062 ±0.006
TOTAL OF 4 LAYERS

2. FINISH: ALL PLATED HOLES 0.001 MIN/0.0015 MAX
COPPER PLATE, ELECTRODEPOSITED TIN-LEAD COMPOSITION
BEFORE REFLOW, SOLDER MASK OVER BARE COPPER (SMOBC)

3. SOLDER MASK: BOTH SIDES USING LPI OR EQUIVALENT

4. SILKSCREEN: USING WHITE NONCONDUCTIVE EPOXY INK

5. UNUSED SMD COMPONENTS SHOULD BE FREE OF SOLDER

2.10"

6. FILL UP ALL VIAS WITH SOLDER

7. SCORING

0.017

A
B
C
D
E
F

DIAMETER

0.015

0.035

0.064

0.070

0.094

0.125

TOTAL HOLES

OF HOLES

SYMBOL

NUMBER

70

5
5
3
6
4

93

Linear Technology Corporation

12

1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● [phone]
FAX: [phone]

●

TELEX: 499-3977 ● www.linear-tech.com

dc275f LT/TP 0100 500 • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 2000