Datasheet LTC1649 Datasheet (Linear Technology)

FEATURES

■

High Power 3.3V to 1.xV-2.xV Switching Regulator
Controller: Up to 20A Output

■

All N-Channel External MOSFETs

■

Provides 5V MOSFET Gate Drive with 3.3V Input

■

Constant Frequency Operation Minimizes
Inductor Size

■

Excellent Output Regulation: ±1% Over Line, Load
and Temperature Variations

■

High Efficiency: Over 90% Possible

■

No Low-Value Sense Resistor Needed

■

Available in 16-Lead SO Package

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APPLICATIONS

■

3.3V Input Power Supply for Low Voltage
Microprocessors and Logic

■

Low Input Voltage Power Supplies

■

High Power, Low Voltage Regulators

■

Local Regulation for Multiple Voltage Distributed
Power Systems

LTC1649

3.3V Input High Power
Step-Down Switching

Regulator Controller

U

DESCRIPTION

The LTC®1649 is a high power, high efficiency switching
regulator controller optimized for use with very low supply
voltages. It operates from 2.7V to 5V input, and provides
a regulated output voltage from 1.26V to 2.5V at up to 20A
load current. A typical 3.3V to 2.5V application features
efficiency above 90% from 1A to 10A load. The LTC1649
uses a pair of standard 5V logic-level N-channel external
MOSFETs, eliminating the need for expensive P-channel
or super-low-threshold devices.

The LTC1649 shares its internal switching architecture
with the LTC1430, and features the same ±1% line, load
and temperature regulation characteristics. Current limit
is user-adjustable without requiring an external low-value
sense resistor. The LTC1649 uses a 200kHz switching
frequency and voltage mode control, minimizing external
component count and size. Shutdown mode drops the
quiescent current to below 10µA.

The LTC1649 is available in the 16-pin narrow SO package.

, LTC and LT are registered trademarks of Linear Technology Corporation.

TYPICAL APPLICATION

3.3V to 2.5V, 15A Converter LTC1649 Efficiency

V

IN

3.3V

MBR0530

22Ω 1k

SHDN

+

10µF

R

7.5k

C1
220pF

IRF7801 = INTERNATIONAL RECTIFIER
MBR0530 = MOTOROLA
*12TS-1R2HL = PANASONIC

C

C

C

0.01µF

R

IMAX

50k

0.1µF

P

VCC1

P

VCC2

V

CC

LTC1649

I

MAX

SHDN

COMP
SS C
GND CP

MBR0530

OUT

G1

I

FB

G2

FB

V

IN

+

C

–

U

+

1µF

10µF

1µF

0.33µF

Q1, Q2
IRF7801
TWO IN
PARALLEL

Q3
IRF7801

+

C

IN

3300µF

*

L

EXT

1.2µH

V

OUT

2.5V

12.4k

12.7k

R1

R2

@15A

+

C

OUT

4400µF

1649 TA01

100

90

80

70

EFFICIENCY (%)

60

50

40

0.1 1 10
LOAD CURRENT (A)

1649 TA02

1

LTC1649

WW

W

ABSOLUTE MAXIMUM RATINGS

U

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PACKAGE/ORDER INFORMATION

(Note 1)

Supply Voltage

V

...........................................................................................

IN

V

...........................................................................................

CC

P

VCC1, 2

................................................................................

6V
9V

13V

Input Voltage

I

.......................................................................

FB

–

C+, C

................................................

–0.3V to (VIN + 0.3V)

–0.3V to 18V

All Other Inputs ....................... –0.3V to (VCC + 0.3V)

Operating Temperature Range ..................... 0°C to 70°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)..................300°C

P

VCC1

GND

SHDN

V

G1

FB

SS

C

TOP VIEW

1
2
3
4
5
6
7

IN

–

8

S PACKAGE

16-LEAD PLASTIC SO

T

= 150°C, θJA = 110°C/W

JMAX

16

G2

15

P

VCC2

14

V

CC

13

I

FB

12

I

MAX

11

COMP

10

CP

OUT

+

9

C

Consult factory for Industrial and Military grade parts.

ELECTRICAL CHARACTERISTICS

VIN = 3.3V, TA = 25°C unless otherwise noted. (Note 2)

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

V

IN

V

FB

V

CPOUT

I

IN

I

PVCC1, 2

f

CP

f

OSC

V

IH

V

IL

I

IN

gm

V

gm

I

I

IMAX

I

SS

tr, t

f

t

NOV

DC

MAX

The ● denotes specifications which apply over the full operating
temperature range.

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a part may be impaired.

Note 2: All currents into device pins are positive; all currents out of device
pins are negative. All voltages are referenced to ground unless otherwise
specified.

Note 3: Maximum Duty Cycle limitations will limit the output voltage

Minimum Supply Voltage Figure 1 (Note 3) ● 2.7 V
Feedback Voltage Figure 1 ● 1.25 1.265 1.28 V
Charge Pump Output Voltage Figure 1 ● 4.8 5 5.2 V
Supply Current (VIN)V

Supply Current (P

)P

VCC1, 2

Internal Charge Pump Frequency I

= VCC, I

SHDN

= 0V 10 25 µA

V

SHDN

= 5V, V

VCC

= 0V 0.1 µA

V

SHDN

= 20mA (Note 5) 700 kHz

CPOUT

= 0 ● 35 mA

LOAD

= VCC (Note 4) 1.5 mA

SHDN

Internal PWM Oscillator Frequency ● 140 200 260 kHz
SHDN Input High Voltage ● 2.4 V
SHDN Input Low Voltage ● 0.8 V
SHDN Input Current ● ±0.01 ±1 µA
Error Amplifier Transconductance 650 µMho
I

Amplifier Transconductance (Note 6) 1300 µMho

LIM

I

Sink Current V

MAX

= VCC ● 81216 µA

IMAX

Soft Start Source Current VSS = 0V ● –8 –12 –16 µA
Driver Rise/Fall Time P
Driver Non-Overlap Time P
Maximum Duty Cycle V

VCC1

VCC1

COMP

= P

= 5V 80 250 ns

VCC2

= P

= 5V 25 130 250 ns

VCC2

= V

CC

Note 4: Supply current at P

VCC1

90.5 93 %

and P

VCC2

needed to charge and discharge the external MOSFET gates. This current
will vary with the operating voltage and the external MOSFETs used.

Note 5: Under normal operating conditions, the charge pump will skip
cycles to maintain regulation and the apparent frequency will be lower than
700kHz.

Note 6: The I
(not current limited) operation, the I

amplifier can sink but not source current. Under normal

LIM

output current will be zero.

LIM

obtainable at very low supply voltages.

ORDER PART

NUMBER

LTC1649CS

is dominated by the current

2

UW

TYPICAL PERFOR A CE CHARACTERISTICS

I

Pin Current vs Temperature

MAX

14.0
VCC = 5V

13.5

LTC1649

Oscillator Frequency
vs Temperature

240

VCC = 5V

230

13.0

12.5

12.0

CURRENT (µA)

MAX

I

11.5

11.0

10.5

–40

–20 0

Maximum Duty Cycle
vs Temperature

100

V

= V

COMP

VFB = 1.265V

95

V

= 5V

CC

90

85

80

DUTY CYCLE (%)

75

70

–20 20

–40

0

40 80 100

20 60

TEMPERATURE (°C)

CC

TEMPERATURE (°C)

40

220

210

200

190

OSCILLATOR FREQUENCY (kHz)

180

1649 G01

170

–40

–20 0

TEMPERATURE (°C)

40 80 100

20 60

1649 G02

Error Amplifier Transconductance
vs Temperature

850

∆I

800
750
700
650
600
550
500
450

TRANSCONDUCTANCE (µmho)

400

80

1649 G03

100

60

350

–40

–20 20

0

TEMPERATURE (°C)

40

COMP

gm =

∆V

FB

VCC = 5V

60 100

80

1649 G04

Load Regulation

0.4
TA = 25°C
V

OUT

0.2

–0.2

(mV)

OUT

–0.4

∆V

–0.6

–0.8

–1.0

0

V
FIGURE 1

0

= 5V

CC

1

= 3.3V

3

4

2

LOAD CURRENT (A)

7

68

5

9

1649 G06

Output Voltage vs Load Current
with Current Limit

4.0

3.5

3.0

2.5

2.0

1.5

OUTPUT VOLTAGE (V)

1.0

0.5

10

R

= 16k R

IMAX

TA = 25°C

= 5V

V

CC

FIGURE 1

0

24 8

0

LOAD CURRENT (A)

= 33k

IMAX

12

6

10

1649 G07

3

LTC1649

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PIN FUNCTIONS

G1 (Pin 1): Driver Output 1. Connect this pin to the gate of
the upper N-channel MOSFET, Q1. This output will swing
from P

to GND. G1 will always be low when G2 is high.

VCC1

In shutdown, G1 and G2 go low.

P

(Pin 2): Power VCC for Driver 1. This is the power

VCC1

supply input for G1. G1 will swing from P
P

must be connected to a potential of at least VIN +

VCC1

V

(Q1). This potential can be generated using a

GS(ON)

VCC1

to GND.

simple charge pump connected to the switching node
between the two external MOSFETs as shown in Figure 1.

GND (Pin 3): System Ground. Connect to a low impedance
ground in close proximity to the source of Q2. The system
signal and power grounds should meet at only one point,
at the GND pin of the LTC1649.

FB (Pin 4): Feedback. The FB pin is connected to the output
through a resistor divider to set the output voltage.
V

OUT

= V

[1 + (R1/R2)].

REF

SHDN (Pin 5): Shutdown, Active Low. A TTL compatible
LOW level at SHDN for more than 50µs puts the LTC1649
into shutdown mode. In shutdown, G1, G2, COMP and SS
go low, and the quiescent current drops to 25µA max.
CP

remains at 5V in shutdown mode. A TTL compatible

OUT

HIGH level at SHDN allows the LTC1649 to operate nor­mally.

C+ (Pin 9): Flying Capacitor, Positive Terminal.
CP

(Pin 10): Charge Pump Output. CP

OUT

provides a

OUT

regulated 5V output to provide power for the internal
switching circuitry and gate drive for the external MOSFETs.
CP

should be connected directly to P

OUT

VCC2

in most

applications. At least 10µF of reservoir capacitance to
ground is required at CP
be met by the bypass capacitor at P

. This requirement can usually

OUT

.

VCC2

COMP (Pin 11): External Compensation. The COMP pin is
connected directly to the output of the internal error
amplifier and the input of the PWM generator. An RC
network is used at this node to compensate the feedback
loop to provide optimum transient response.

I

(Pin 12): Current Limit Set. I

MAX

sets the threshold

MAX

for the internal current limit comparator. If IFB drops below
I

with G1 on, the LTC1649 will go into current limit.

MAX

I

has an internal 12µA pull-down to GND. The voltage

MAX

at I

can be set with an external resistor to the drain of

MAX

Q1 or with an external voltage source.
IFB (Pin 13): Current Limit Sense. Connect to the switched

node at the source of Q1 and the drain of Q2 through a 1kΩ
resistor. The resistor is required to prevent voltage tran­sients at the switched node from damaging the IFB pin. I

FB

can be taken up to 18V above GND without damage.

SS (Pin 6): Soft Start. An external capacitor from SS to
GND controls the startup time and also compensates the
current limit loop, allowing the LTC1649 to enter and exit
current limit cleanly.

VIN (Pin 7): Charge Pump Input. This is the main low
voltage power supply input. VIN requires an input voltage
between 3V and 5V. Bypass VIN to ground with a 1µF
ceramic capacitor located close to the LTC1649.

C– (Pin 8): Flying Capacitor, Negative Terminal. Connect
a 1µF ceramic capacitor from C– to C+.

4

VCC (Pin 14): Internal Power Supply. VCC provides power
to the feedback amplifier and switching control circuits.
VCC is designed to run from the 5V supply provided by
CP

. VCC requires a 10µF bypass capacitor to GND.

OUT

P

(Pin 15): Power VCC for Driver 2. This is the power

VCC2

supply input for G2. G2 will swing from P
P

must be connected to a potential of at least

VCC2

V
pin. P

(Q2). This voltage is usually supplied by the CP

GS(ON)

requires a bypass capacitor to GND; this

VCC2

VCC2

to GND.

OUT

capacitor also provides the reservoir capacitance required
by the CP

OUT

pin.

G2 (Pin 16): Driver Output 2. Connect this pin to the gate
of the lower N-channel MOSFET, Q2. This output will
swing from P

to GND. G2 will always be low when G1

VCC2

is high. In shutdown, G1 and G2 go low.

LTC1649

BLOCK DIAGRA

V

IN

SHDN

COMP

V

CC

12µA

SS

I

MAX

12µA

W

DELAY

50µs

I

LIM

+

–

+

INTERNAL
SHUTDOWN

PWM

FB MIN

–

40mV

+

+

C

CHARGE

PUMP

C

+

–

MAX

40mV

CP

OUT

PV

CC1

G1

PV

CC2

G2

I

FB

FB

TEST CIRCUIT

V

IN

3.3V

SHDN

+

10µF

+

1.26V

R

IMAX

MBR0530

22Ω 1k

R

C

7.5k
C

C

0.01µF

C1
220pF

50k

0.1µF

P

VCC1

P

VCC2

V

CC

LTC1649

I

MAX

SHDN

COMP
SS C
GND CP

MBR0530

G1

I

G2

V

OUT

FB

FB

IN

+

C

–

1µF

1µF

+

10µF

0.33µF

Q1, Q2
IRF7801
TWO IN
PARALLEL

Q3
IRF7801

1649 BD

+

C

IN

3300µF

L

EXT

1.2µH

V

OUT

2.5V

R1

12.4k

+

C

OUT

4400µF

R2

12.7k

Figure 1

1649 TA03

5

LTC1649

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APPLICATIONS INFORMATION

OVERVIEW

T

he LTC1649 is a voltage feedback PWM switching regu­lator controller (see Block Diagram) designed for use in
high power, low input voltage step-down (buck) convert­ers. It includes an onboard PWM generator, a precision
reference trimmed to ±0.5%, two high power MOSFET
gate drivers and all necessary feedback and control cir­cuitry to form a complete switching regulator circuit. Also
included is an internal charge pump which provides 5V
gate drive to the external MOSFETs with input supply
voltage as low as 2.7V. The LTC1649 runs at an internally
fixed 200kHz clock frequency and requires an external
resistor divider to set the output voltage.

The LTC1649 includes a current limit sensing circuit that
uses the upper external power MOSFET as a current
sensing element, eliminating the need for an external
sense resistor. Also included is an internal soft start
feature that requires only a single external capacitor to
operate.

THEORY OF OPERATION

Primary Feedback Loop

The LTC1649 senses the output voltage of the circuit at the
output capacitor through a resistor divider connected to
the FB pin and feeds this voltage back to the internal
transconductance amplifier FB. FB compares the resistor­divided output voltage to the internal 1.26V reference and
outputs an error signal to the PWM comparator. This is
then compared to a fixed frequency sawtooth waveform
generated by the internal oscillator to generate a pulse
width modulated signal. This PWM signal is fed back to the
external MOSFETs through G1 and G2, closing the loop.
Loop compensation is achieved with an external compen­sation network at COMP, the output node of the FB
transconductance amplifier.

MIN, MAX Feedback Loops

Two additional comparators in the feedback loop provide
high speed fault correction in situations where the FB
amplifier may not respond quickly enough. MIN compares
the feedback signal to a voltage 40mV (3%) below the

internal reference. At this point, the MIN comparator
overrides the FB amplifier and forces the loop to full duty
cycle, set by the internal oscillator at about 93%. Similarly,
the MAX comparator monitors the output voltage at 3%
above the internal reference and forces the output to 0%
duty cycle when tripped. These two comparators prevent
extreme output perturbations with fast output transients,
while allowing the main feedback loop to be optimally
compensated for stability.

Current Limit Loop

The LTC1649 includes yet another feedback loop to con­trol operation in current limit. The I
the voltage drop across external MOSFET Q1 with the I
pin during the portion of the cycle when G1 is high. It
compares this voltage to the voltage at the I
peak current rises, the drop across Q1 due to its R
increases. When IFB drops below I
drain current has exceeded the maximum level, I
to pull current out of the external soft start capacitor,
cutting the duty cycle and controlling the output current
level. At the same time, the I
signal to disable the MIN comparator to prevent it from
conflicting with the current limit circuit. If the internal
feedback node drops below about 0.8V, indicating a se­vere output overload, the circuitry will force the internal
oscillator to slow down by a factor of as much as 100. If
desired, the turn on time of the current limit loop can be
controlled by adjusting the size of the soft start capacitor,
allowing the LTC1649 to withstand brief overcurrent con­ditions without limiting.

By using the R
the current limit circuit eliminates the sense resistor that
would otherwise be required and minimizes the number of
components in the external high current path. Because
power MOSFET R
with temperature, the LTC1649 current limit is not de­signed to be accurate; it is meant to prevent damage to the
power supply circuitry during fault conditions. The actual
current level where the limiting circuit begins to take effect
may vary from unit to unit, depending on the power
MOSFETs used. See Soft Start and Current Limit for more
details on current limit operation.

of Q1 to measure the output current,

DS(ON)

DS(ON)

LIM

is not tightly controlled and varies

amplifier monitors

LIM

FB

pin. As the

MAX

DS(ON)

, indicating that Q1’s

MAX

starts

LIM

comparator generates a

6

LTC1649

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APPLICATIONS INFORMATION

MOSFET Gate Drive

The LTC1649 is designed to operate from supplies as low
as 2.7V while using standard 5V logic-level N-channel
external MOSFETs. This poses somewhat of a challenge—
from as little as 2.7V, the LTC1649 must provide a 0V to
5V signal to the lower MOSFET, Q2, while the upper
MOSFET, Q1, requires a gate drive signal that swings from
0V to (VIN + 5V). The LTC1649 addresses this situation
with two specialized circuits. An onboard charge pump
boosts the input voltage at VIN to a regulated 5V at CP
This 5V supply is used to power the PV

pin, which in

CC2

turn supplies 5V gate drive to Q2. This 5V supply is also
used to power the VCC pin, which allows the internal drive
circuitry to interface to the boosted driver supplies.

Gate drive for the top N-channel MOSFET, Q1, is supplied
by PV

. This supply must reach VIN + 5V while Q1 is on.

CC1

Conveniently, the switching node at the source of Q1 rises
to VIN whenever Q1 is on. The LTC1649 uses this fact to
generate the required voltage at PV

with a simple

CC1

external charge pump as shown in Figure 2. This circuit
charges the flying capacitor C2 to the 5V level at CP
when the switching node is low. As the top MOSFET turns
on, the switching node begins to rise to VIN, and the PV
is pulled up to VIN + 5V by C2. The 93% maximum duty
cycle (typical) means the switching node at the source of
Q1 will return to ground during at least 7% of each cycle,
ensuring that the charge pump will always provide ad­equate gate drive to Q1.

OUT

OUT

CC1

.

Synchronous Operation

The LTC1649 uses a synchronous switching architecture,
with MOSFET Q2 taking the place of the diode in a classic
buck circuit (Figure 3). This improves efficiency by reduc­ing the voltage drop and the resultant power dissipation
across Q2 to VON = (I)(R

DS(ON)(Q2)

), usually much lower
than VF of the diode in the classical circuit. This more than
offsets the additional gate drive required by the second
MOSFET, allowing the LTC1649 to achieve efficiencies in
the mid-90% range for a wide range of load currents.

Another feature of the synchronous architecture is that
unlike a diode, Q2 can conduct current in either direction.
This allows the output of a typical LTC1649 circuit to sink
current as well as sourcing it while remaining in regula­tion. The ability to sink current at the output allows the
LTC1649 to be used with reactive or other nonconventional
loads that may supply current to the regulator as well as
drawing current from it.

V

IN

CONTROLLER

Q1

V

OUT

D1

1649 F03a

D

CP

+

10µF

CP

OUTPVCC2

LTC1649

Figure 2. PV

PV

CC1

G1

G2

Charge Pump

CC1

1µF

V

IN

Q1

L1

V

OUT

+

Q2

C

OUT

1649 F02

Figure 3a. Classical Buck Architecture

V

IN

Q1

CONTROLLER

Q2

Figure 3b. Synchronous Buck Architecture

V

OUT

1649 F03b

7

LTC1649

)

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APPLICATIONS INFORMATION

EXTERNAL COMPONENT SELECTION

Power MOSFETs

Two N-channel power MOSFETs are required for most
LTC1649 circuits. These should be selected primarily by
on-resistance considerations; thermal dissipation is often
a secondary concern in high efficiency designs. The
LTC1649 is designed to be used with 5V logic-level MOS­FETs; “standard” threshold MOSFETs with R
fied at 10V only will not provide satisfactory performance.

MOSFET R

should be chosen based on input and

DS(ON)

output voltage, allowable power dissipation and maxi­mum required output current. In a typical LTC1649 buck
converter circuit operating in continuous mode, the aver­age inductor current is equal to the output load current.
This current is always flowing through either Q1 or Q2 with
the power dissipation split up according to the duty cycle:

V

DC (Q1) =

OUT

V

IN

DS(ON)

speci-

efficiency loss at full load for each MOSFET. Assuming
roughly 90% efficiency at this current level, this gives a
P

value of (2.5V)(10A/0.9)(0.03) = 833mW per FET

MAX

and a required R

R

DS(ON)

R

DS(ON)

(Q1) = = 0.011Ω

(Q2) = = 0.034Ω

Note that while the required R

of:

DSON

(3.3V)(833mW)

(2.5V)(10A

(3.3V)(833mW)

(3.3V – 2.5V)(10A

DS(ON)

2

)

2

)

values suggest large
MOSFETs, the dissipation numbers are less than a watt per
device— large TO-220 packages and heat sinks are not
necessarily required in high efficiency applications. Siliconix
Si4410DY and International Rectifier IRF7801 are two
small, surface mount devices with RON values of 0.03Ω or
below with 5V of gate drive; both work well in LTC1649
circuits. A higher P

value will generally decrease

MAX

MOSFET cost and circuit efficiency and increase MOSFET
heat sink requirements.

V

(V

IN

OUT

V

– V

V

IN

IN

OUT

)

DC (Q2) = 1 –

=

The RON required for a given conduction loss can now be
calculated by rearranging the relation P = I2R:

(Q1)

P

DC(Q1)(I

VIN(P

=

V

OUT(IMAX

P

DC(Q2)(I

VIN(P

=

(V

IN

MAX

MAX

MAX

– V

MAX

)(Q1)

(Q2)

MAX

MAX

OUT

2

)

2

)

2

)

)(Q2)

)(I

MAX

2

P

R

R

MAX

DS(ON)

DS(ON)

(Q1) =

(Q2) =

should be calculated based primarily on required

efficiency. A typical high efficiency circuit designed for

3.3V in, 2.5V at 10A out might require no more than 3%

Inductor

The inductor is often the largest component in an LTC1649
design and should be chosen carefully. Inductor value and
type should be chosen based on output slew rate require­ments and expected peak current. Inductor value is prima­rily controlled by the required current slew rate. The
maximum rate of rise of the current in the inductor is set
by its value, the input-to-output voltage differential and the
maximum duty cycle of the LTC1649. In a typical 3.3V to

2.5V application, the maximum rise time will be:

– V

(V

93% =

IN

)

OUT

L

AMPS

SECOND

0.744AµsI
L

where L is the inductor value in µH. A 2µH inductor would
have a 0.37A/µs rise time in this application, resulting in a
14µs delay in responding to a 5A load current step. During
this 14µs, the difference between the inductor current and
the output current must be made up by the output capaci­tor, causing a temporary droop at the output. To minimize
this effect, the inductor value should usually be in the 1µH
to 5µH range for most typical 3.3V to 2.xV LTC1649
circuits. Different combinations of input and output volt-

8

LTC1649

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APPLICATIONS INFORMATION

ages and expected loads may require different values.
Once the required value is known, the inductor core type

can be chosen based on peak current and efficiency
requirements. Peak current in the inductor will be equal to
the maximum output load current added to half the peak­to- peak inductor ripple current. Ripple current is set by the
inductor value, the input and output voltage and the
operating frequency. If the efficiency is high and can be
considered approximately equal to 1, the ripple current is
approximately equal to:

(V

∆I = DC

DC =

f

OSC

L = inductor value

Solving this equation with our typical 3.3V to 2.5V appli­cation, we get:

(200kHz)(2µH)

Peak inductor current at 10A load:

The inductor core must be adequate to withstand this peak
current without saturating, and the copper resistance in
the winding should be kept as low as possible to minimize
resistive power loss. Note that the current may rise above
this maximum level in circuits under current limit or under
fault conditions in unlimited circuits; the inductor should
be sized to withstand this additional current.

Input and Output Capacitors

A typical LTC1649 design puts significant demands on
both the input and output capacitors. Under normal steady
load operation, a buck converter like the LTC1649 draws
square waves of current from the input supply at the
switching frequency, with the peak value equal to the
output current and the minimum value near zero. Most of
this current must come from the input bypass capacitor,
since few raw supplies can provide the current slew rate to

IN – VOUT

(f

OSC

V

OUT

V

IN

= LTC1649 oscillator frequency = 200kHz

(0.8)(0.76)

1.5A
2

)(L)

= 10.8A10A +

)

= 1.5A

P–P

feed such a load directly. The resulting RMS current flow
in the input capacitor will heat it up, causing premature
capacitor failure in extreme cases. Maximum RMS current
occurs with 50% PWM duty cycle, giving an RMS current
value equal to I
adequate ripple current rating must be used to ensure
reliable operation. Note that capacitor manufacturers’
ripple current ratings are often based on only 2000 hours
(3 months) lifetime; further derating of the input capacitor
ripple current beyond the manufacturer’s specification is
recommended to extend the useful life of the circuit.

The output capacitor in a buck converter sees much less
ripple current under steady-state conditions than the input
capacitor. Peak-to-peak current is equal to that in the
inductor, usually a fraction of the total load current. Output
capacitor duty places a premium not on power dissipation
but on low ESR. During an output load transient, the
output capacitor must supply all of the additional load
current demanded by the load until the LTC1649 can
adjust the inductor current to the new value. ESR in the
output capacitor results in a step in the output voltage
equal to the ESR value multiplied by the change in load
current. A 5A load step with a 0.05Ω ESR output capacitor
will result in a 250mV output voltage shift; this is a 10%
output voltage shift for a 2.5V supply! Because of the
strong relationship between output capacitor ESR and
output load transient response, the output capacitor is
usually chosen for ESR, not for capacitance value; a
capacitor with suitable ESR will usually have a larger
capacitance value than is needed to control steady-state
output ripple.

Electrolytic capacitors rated for use in switching power
supplies with specified ripple current ratings and ESR can
be used effectively in LTC1649 applications. OS-CON
electrolytic capacitors from Sanyo give excellent perfor­mance and have a very high performance/size ratio for an
electrolytic capacitor. Surface mount applications can use
either electrolytic or dry tantalum capacitors. Tantalum
capacitors must be surge tested and specified for use in
switching power supplies; low cost, generic tantalums are
known to have very short lives followed by explosive
deaths in switching power supply applications. AVX TPS
series surface mount devices are popular tantalum capaci-

/2. A low ESR input capacitor with an

OUT

9

LTC1649

U

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APPLICATIONS INFORMATION

tors that work well in LTC1649 applications. A common
way to lower ESR and raise ripple current capability is to
parallel several capacitors. A typical LTC1649 application
might require an input capacitor with a 5A ripple current
capacity and 2% output shift with a 10A output load step,
which requires a 0.005Ω output capacitor ESR. Sanyo OS­CON part number 10SA220M (220µF/10V) capacitors
feature 2.3A allowable ripple current at 85°C and 0.035Ω
ESR; three in parallel at the input and seven at the output
will meet the above requirements.

Input Supply Considerations/Charge Pump

The LTC1649 requires four supply voltages to operate:
VIN, VCC, PV
input, supplying current to the drain of Q1 and the input to
the internal charge pump at the VIN pin. This supply must
be between 2.7V and 6V for the LTC1649 to operate
properly. An internal charge pump uses the voltage at V
to generate a regulated 5V output at CP
pump requires an external 1µF capacitor connected be-
tween the C+ and C– pins, and an external 10µF reservoir
capacitor connected from CP
always be greater than or equal to VIN. If VIN is expected to
rise above 5V, an additional Schottky diode (DS) should be
added from VIN to CP

CP

is typically connected to PV

OUT

the 5V supply that the G2 driver output uses to drive Q2.

CC1

and PV

. VIN is the primary high power

CC2

. This charge

OUT

to ground. CP

OUT

.

OUT

directly, providing

CC2

OUT

IN

must

PV

requires a 10µF bypass to ground; this capacitor

CC2

can double as the CP
typical application with CP

reservoir capacitor, allowing a

OUT

OUT

and PV

connected to-

CC2

gether to get away with only a single 10µF capacitor at this
node, located close to the PV
powered from CP
PV

happens to be a significant noisemaker, so most

CC2

, but is somewhat sensitive to noise.

OUT

applications require an RC filter from CP

pin. VCC can also be

CC2

/PV

OUT

CC2

to VCC.

22Ω and 10µF are typical filter values that work well in
most applications.

PV

needs to be boosted to a level higher than CP

CC1

OUT

to
provide gate drive to Q1. The LTC1649 initially used a
charge pump from VIN to create CP
cation uses a second charge pump to generate the PV

; the typical appli-

OUT

CC1

supply. This second charge pump consists of a Schottky
diode (DCP) from CP
from PV

to the source of Q1. While Q2 is on, the diode

CC1

charges the capacitor to CP
source rises to VIN, and the cap hauls PV

OUT

to PV

, and a 1µF capacitor

CC1

. When Q1 comes on, its

OUT

CC1

up to (CP

OUT

+ VIN), adequate to fully turn on Q1. When Q1 turns back
off, PV

drops back down to CP

CC1

; fortunately, we’re

OUT

not interested in turning Q1 on at this point, so the lower
voltage doesn’t cause problems. The next time Q1 comes
on, PV

bounces back up to (CP

CC1

+ VIN), keeping Q1

OUT

happy. Figure 4 shows a complete power supply circuit for
the LTC1649.

10

*OPTIONAL
FOR V

IN

+

C

1µF

–

C

≥ 5V

LTC1649

DS*

V

IN

CHARGE

PUMP

22Ω

+

10µF

CP

OUT

V

CC

DRIVE

CIRCUITRY

Figure 4. LTC1649 Power Supplies

+

10µF

PV

CC2

V

IN

D

CP

+

C

1µF

PV

CC1

G1

G2

Q1

Q2

IN

L1

V

OUT

+

C

OUT

1649 F04

LTC1649

1649 F06

C

SS

COMP

SS

LTC1649

12µA

V

CC

FB

U

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APPLICATIONS INFORMATION

The CP
to power the VCC and PVCC pins. This supply can also be
used to power external circuitry, but any additional current
drawn from CP
drive the external MOSFETs. Circuits with small external
MOSFETs can draw as much as 20mA or 30mA from
CP

OUT

rent circuits with large or multiple external MOSFETs may
need every milliamp they can get from CP
loads should be minimized. The charge pump at PV
more limited in its abilities, and should not be connected
to anything except PV
bypass capacitor from PV
charge from the charge pump and actually degrade perfor­mance.

Compensation and Transient Response

The LTC1649 voltage feedback loop is compensated at the
COMP pin; this is the output node of the internal gm error
amplifier. The loop can generally be compensated prop­erly with an RC network from COMP to GND and an
additional small C from COMP to GND (Figure 5). Loop
stability is affected by inductor and output capacitor
values and by other factors. Optimum loop response can
be obtained by using a network analyzer to find the loop
poles and zeros; nearly as effective and a lot easier is to
empirically tweak the RC values until the transient recovery
looks right with an output load step.

pin can typically supply 50mA at 5V, adequate

OUT

subtracts from the current available to

OUT

without hindering performance. High output cur-

, and external

OUT

is

CC1

. In particular, do not connect a

CC1

to ground—it will steal

CC1

Output transient response is set by three major factors: the
time constant of the inductor and the output capacitor, the
ESR of the output capacitor, and the loop compensation
components. The first two factors usually have much
more impact on overall transient recovery time than the
third; unless the loop compensation is way off, more
improvement can be had by optimizing the inductor and
the output capacitor than by fiddling with the loop com­pensation components. In general, a smaller value induc­tor will improve transient response at the expense of ripple
and inductor core saturation rating. Minimizing output
capacitor ESR will also help optimize output transient
response. See Input and Output Capacitors for more
information.

Soft Start and Current Limit

The LTC1649 includes a soft start circuit at the SS pin; this
circuit is used both for initial start-up and during current
limit operation. SS requires an external capacitor to GND
with the value determined by the required soft start time.
An internal 12µA current source is included to charge the
external capacitor. Soft start functions by clamping the
maximum voltage that the COMP pin can swing to, thereby
controlling the duty cycle (Figure 6). The LTC1649 will
begin to operate at low duty cycle as the SS pin rises to
about 2V below the VCC pin. As SS continues to rise, the
duty cycle will increase until the error amplifier takes over
and begins to regulate the output. When SS reaches 1V
below VCC the LTC1649 will be in full operation. An internal
switch shorts the SS pin to GND during shutdown.

LTC1649

COMP

R

C

C

Figure 5. Compensation Pin Hook-Up

C1C

1659 F05

Figure 6. Soft Start Clamps COMP Pin

11

LTC1649

U

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APPLICATIONS INFORMATION

The LTC1649 detects the output current by watching the
voltage at IFB while Q1 is ON. The I
this voltage to the voltage at I
state, Q1 has a known resistance; by calculating back­wards, the voltage generated at IFB by the maximum
output current in Q1 can be determined. As IFB falls below
I

, I

MAX

will begin to sink current from the soft start pin,

LIM

causing the voltage at SS to fall. As SS falls, it will limit the
output duty cycle, limiting the current at the output.
Eventually the system will reach equilibrium, where the
pull-up current at the SS pin matches the pull-down
current in the I

amplifier; the LTC1649 will stay in this

LIM

state until the overcurrent condition disappears. At this
time IFB will rise, I

will stop sinking current and the

LIM

internal pull-up will recharge the soft start capacitor,
restoring normal operation. Note that the IFB pin requires
an external 1k series resistor to prevent voltage transients
at the drain of Q2 from damaging internal structures.

The I

amplifier pulls current out of SS in proportion to

LIM

the difference between IFB and I
conditions, the SS pin will fall gradually, creating a time
delay before current limit takes effect. Very short, mild
overloads may not trip the current limit circuit at all.
Longer overload conditions will allow the SS pin to reach
a steady level, and the output will remain at a reduced
voltage until the overload is removed. Serious overloads

0.1µF

R

IMAX

amplifier compares

LIM

(Figure 7). In the ON

MAX

. Under mild overload

MAX

V

IN

will generate a larger overdrive at I

, allowing it to pull SS

LIM

down more quickly and preventing damage to the output
components.

The I

amplifier output is disabled when Q1 is OFF to

LIM

prevent the low IFB voltage in this condition from activating
the current limit. It is re-enabled a fixed 170ns after Q1
turns on; this allows for the IFB node to slew back high and
the I

amplifier to settle to the correct value. As the

LIM

LTC1649 goes deeper into current limit, it will reach a point
where the Q1 on-time needs to be cut to below 170ns to
control the output current. This conflicts with the mini­mum settling time needed for proper operation of the I

LIM

amplifier. At this point, a secondary current limit circuit
begins to reduce the internal oscillator frequency, length­ening the off-time of Q1 while the on-time remains con­stant at 170ns. This further reduces the duty cycle, allow­ing the LTC1649 to maintain control over the output
current.

Under extreme output overloads or short circuits, the I

LIM

amplifier will pull the SS pin more than 2V below VCC in a
single switching cycle, cutting the duty cycle to zero. At
this point all switching stops, the output current decays
through Q2 and the LTC1649 runs a partial soft start cycle
and restarts. If the short is still present the cycle will
repeat. Peak currents can be quite high in this condition,
but the average current is controlled and a properly
designed circuit can withstand short circuits indefinitely
with only moderate heat rise in the output FETs. In addi­tion, the soft start cycle repeat frequency can drop into the
low kHz range, causing vibrations in the inductor which
provide an audible alarm that something is wrong.

12

I

MAX

+

–

12µA

SS

C

SS

Figure 7. Current Limit Operation

I

LIM

V

CC

12µA

LTC1649

1649 F07

1k

I

FB

Q1

Shutdown

The LTC1649 includes a low power shutdown mode,

Q2

controlled by the logic at the SHDN pin. A high at SHDN
allows the part to operate normally. A low level at SHDN
stops all internal switching, pulls COMP and SS to ground
internally and turns Q1 and Q2 off. In shutdown, the
LTC1649 itself will drop below 25µA quiescent current
typically, although off-state leakage in the external MOS­FETs may cause the total VIN current to be somewhat
higher, especially at elevated temperatures. When SHDN
rises again, the LTC1649 will rerun a soft start cycle and

LTC1649

V

OUT

=

R

1

10kΩ

+ 1.265V

U

WUU

APPLICATIONS INFORMATION

resume normal operation. The CP
lated at 5V in shutdown, and can be used as a keep-alive
supply for external circuitry if desired. Note that any
current drawn from the CP

pin adds to the quiescent

OUT

current in shutdown, and subtracts from the current
available to drive the external MOSFETs if the load remains
connected while the LTC1649 is active.

External Clock Synchronization

The LTC1649 SHDN pin can double as an external clock
input for applications that require a synchronized clock or
a faster switching speed. The SHDN pin terminates the
internal sawtooth wave and resets the oscillator immedi­ately when it goes low, but waits 50µs before shutting
down the rest of the internal circuitry. A clock signal
applied directly to the SHDN pin will force the LTC1649
internal oscillator to lock to its frequency as long as the
external clock runs faster than the internal oscillator
frequency. The LTC1649 can be synchronized to frequen­cies between 250kHz and about 350kHz.

Frequencies above 350kHz can cause erratic current limit
operation and are not recommended.

Setting the Output Voltage

The LTC1649 feedback loop senses the output voltage at
the FB pin. The loop regulates FB to 1.265V; to set the
output voltage, FB should be connected to the output node
through a resistor divider, set up so the voltage at FB is

1.265V when the output is at the desired voltage (see
Figure 8). The upper end of R1 should be connected to the
output voltage as close to the load as possible, to minimize
errors caused by resistance in the output leads. The
bottom of R2 should be connected to the high power
ground node, at the GND pin of the LTC1649.

R1 and R2 should be chosen so that:

V

= V

OUT

R

+ R

1

2

R2

REF

=

R

pin remains regu-

OUT

+ R

1

2

R2

(1.265V)

An easy way to simplify the math is to choose
R2 = 12.65kΩ. This simplifies the equation to:

A typical 2.5V output application might use R1 = 12.35kΩ,
R2 = 12.65kΩ. The nearest standard 1% values are
R1 = 12.4kΩ, R2 = 12.7kΩ, which gives an output voltage
of 2.5001V—pretty close to 2.5V.

Note that using 1% resistors can cause as much as 1%
error in the output voltage in a typical LTC1649 applica­tion—a significant fraction of the total output error. 0.1%
or 0.25% feedback resistors are recommended for appli­cations which require the output voltage to be controlled
to better than 3%.

V

OUT

LTC1649

FB

GND

Figure 8. Resistor Divider at FB Pin

R1

+

C

OUT

R2

1659 F08

LAYOUT CONSIDERATIONS

Grounding

Proper grounding is critical for the LTC1649 to obtain
specified output regulation. Extremely high peak currents
(as high as several amps) can flow between the bypass
capacitors and the PV

CC1

, PV

and GND pins. These

CC2

currents can generate significant voltage differences be­tween two points that are nominally both “ground.” As a
general rule, power and signal grounds should be totally
separated on the layout, and should be brought together
at only one point, right at the LTC1649 GND pin. This helps
minimize internal ground disturbances in the LTC1649,
while preventing excessive current flow from disrupting
the operation of the circuits connected to GND. The high
power GND node should be as compact and low imped­ance as possible, with the negative terminals of the input

13

LTC1649

U

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APPLICATIONS INFORMATION

and output capacitors, the source of Q2, the LTC1649 GND
pin, the output return and the input supply return all
clustered at one point. Figure 9 is a modified schematic
showing the common connections in a proper layout. Note
that at 10A current levels or above, current density in the
PC board itself can become a concern; traces carrying high
currents should be as wide as possible.

Power Component Hook-Up/Heat Sinking

As current levels rise much above 1A, the power compo­nents supporting the LTC1649 start to become physically
large (relative to the LTC1649, at least) and can require
special mounting considerations. Input and output ca­pacitors need to carry high peak currents and must have
low ESR; this mandates that the leads be clipped as short
as possible and PC traces be kept wide and short. The
power inductor will generally be the most massive single
component on the board; it can require a mechanical hold­down in addition to the solder on its leads, especially if it
is a surface mount type.

The power MOSFETs used require some care to ensure
proper operation and reliability. Depending on the current
levels and required efficiency, the MOSFETs chosen may
be as large as TO-220s or as small as SO-8s. High
efficiency circuits may be able to avoid heat sinking the
power devices, especially with TO-220 type MOSFETs. As
an example, a 90% efficient converter working at a steady

2.5V/10A output will dissipate only (25W/90%)10% =

2.8W. The power MOSFETs generally account for the
majority of the power lost in the converter; even assuming
that they consume 100% of the power used by the
converter, that’s only 2.8W spread over two or three
devices. A typical SO-8 MOSFET with a RON suitable to
provide 90% efficiency in this design can commonly
dissipate 2W when soldered to an appropriately sized
piece of copper trace on a PC board. Slightly less efficient
or higher output current designs can often get by with
standing a TO-220 MOSFET straight up in an area with
some airflow; such an arrangement can dissipate as much
as 3W without a heat sink. Designs which must work in
high ambient temperatures or which will be routinely
overloaded will generally fare best with a heat sink.

V

IN

D

CP

PV

V

IN

CP

OUT

22Ω

V

CC

+

10µF

C1

R

C

C

C

SHDN

1µF

C

C

C

COMP
SS

SS

+

–

LTC1649

GND

PV

I

MAX

SHDN

CC2
CC1

G1

I

+

10µF

1k

FB

G2

FB

0.1µF

R

IMAX

1µF

+

C

IN

Q1

L1

V

OUT

R1

Q2

R2

1649 F09

+

C

OUT

Figure 9. Typical Schematic Showing Layout Considerations

14

PACKAGE DESCRIPTIO

U

Dimensions in inches (millimeters) unless otherwise noted.

S Package

16-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.386 – 0.394*
(9.804 – 10.008)

13

16

14

15

12

11

LTC1649

10

9

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

×

45

0.016 – 0.050

0.406 – 1.270

°

0° – 8° TYP

0.228 – 0.244

(5.791 – 6.197)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

0.150 – 0.157**
(3.810 – 3.988)

4

5

0.050

(1.270)

TYP

3

2

1

7

6

8

0.004 – 0.010

(0.101 – 0.254)

S16 0695

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

15

LTC1649

TYPICAL APPLICATIO

U

SSTL Logic Termination Supply

V

IN

3.3V

SHDN

10µF

CERAMIC

MBR0530

22Ω

220pF

Si9802 = SILICONIX

7.5k

0.01µF

CERAMIC

0.1µF

0.1µF

1µF

P

VCC1

P

VCC2

V

CC

LTC1649

I

MAX

SHDN

COMP
SS C
GND CP

4.7k

–
+

1/2 LT1211

V

OUT

2200pF

Q1
1/2
Si9802

G1

I

FB

G2

FBNC

IN

+

C

–

1µF
CERAMIC

+

MBR0530

10k

10k

10µF

–
+

Q2
1/2
Si9802

50pF

10k

1/2 LT1211

R1

18.2k

R2
15k

L

EXT

1.5µH

1649 TA04

+

1500µF

+

1500µF

V

OUT

= 0.45VIN AT ±5A
= 1.48V AT V

= 3.3V

IN

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Pentium is a registered trademark of Intel Corporation.

1649fs sn1649 LT/TP 1098 4K • PRINTED IN USA

 LINEAR TE CHNOLOGY CORPORATION 1998

16

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com