Linear Technology LTC1628 User Manual

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

LTC1628

2-Phase Constant-Frequency

Synchronous, Dual-Output

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DESCRIPTIO

This demonstration board provides 3.3V/4A and 5V/4A
outputs using a low EMI, 2-phase, adjustable, dual switch­ing regulator controller. This design is ideally suited for
notebook computer system power supply applications.
Operating the two high side MOSFETs 180° out of phase
significantly reduces peak input ripple current, thereby
reducing radiated and conducted EMI. External parts
count, cost and size are minimized in this design. Output
voltages can be externally set to as low as 0.8V. The
controllers have overcurrent latch-off, which can be exter­nally defeated, as well as internal current foldback for
overload situations. The overcurrent latch-off on one
controller can be configured to shut off the other output.
A soft latch for overvoltage conditions is also provided. In
addition to the two high current outputs, on-chip 5V/50mA

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PERFOR A CE SU ARY

Operating Temperature Range: 0°C to 50°C (continued on Page 2)

DC/DC Converter

and 3.3V/25mA linear regulators are also included. In the
optional standby mode, these internal regulators are ca­pable of powering external system wake-up circuitry when
both high current controllers are shut down. Two low
current modes of operation are available: Burst Mode
operation offers highest efficiency while Burst Disable
mode provides constant-frequency operation down to 1%
of maximum designed load. The frequency is externally
DC-controlled over a 130kHz to 300kHz range. The con­troller can operate at up to 99% duty cycle for very low
dropout conditions. The demonstration board operates on
an input supply of from 5.2V to 30V. Refer to the LTC®1628
data sheet for other possible configurations. Gerber files
for this circuit board are available. Call the LTC factory.

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

TM

Input Voltage Range Input Voltage Limited by External MOSFET Drive and Breakdown Requirements 5.2V to 30V
Outputs Output Voltage: Controller 1; Externally Adjustable; 0 to 3A, 4A Pk 5V ± 0.10V

Output Voltage: Controller 2; Externally Adjustable; 0 to 3A, 4A Pk 3.33V ± 0.067V
5V Linear Regulator 5V ± 4%

3.3V Linear Regulator 3.3V ± 4%
Typical Output Ripple at 10MHz BW; 300kHz; IO = 1A; 3.3V and 5V Outputs; VIN = 15V 20mV

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P-P

TYPICAL PERFOR A CE CHARACTERISTICS A D BOARD PHOTO

Demo Circuit 236A

100

90

80

70

EFFICIENCY (%)

60

50

0.001

Efficiency vs Load

5V OUTPUT

3.3V OUTPUT

0.01

0.1

OUTPUT CURRENT (A)

VIN = 15V

1

DC236TP01

10

DC236 BP

1

DEMO MANUAL DC236

A

LOAD

LOAD

5V

V

OUT1

V

OUT2

V

V

A

A

V

V

IN

1

RUN/SS1

FLTCPL

FREQ

STBYMD

FCB

RUN/SS2

0

GND 5V

GND

GND

LTC1628

MULTI-PHASE SYSTEM POWER

DEMO CIRCUIT DC236

V

IN

3.3V

3.3V

DC236 F01

L1

L2

+

DESIGN-READY SWITCHERS

UWWW

PERFOR A CE SU ARY

Frequency FREQSET Pin Tied to INTVCC Pin 300kHz
Line Regulation VIN = 7V to 20V; 3.3V and 5V Outputs ±1mV
Load Regulation IO = 0 to 3A; 3.3V and 5V Outputs –20mV
Supply Current VIN = 15V, 5V and 3.3V On, EXTVCC = V
Shutdown Current VIN = 15V, STBYMD = 0V 20µA
Standby Current 5V INTVCC and 3.3V LDO On; VIN = 15V, RUN/SS1 and RUN/SS2 = 0V, 1MΩ STBYMD to V
Efficiency VIN = 15V, 5V at 3A and 3.3V at 3A 94%

Operating Temperature Range: 0°C to 50°C (continued from Page 1)

OUT1

IN

390µA

125µA

QUICK START GUIDE

This demonstration board is easily set up to evaluate the
performance of the LTC1628. Please follow the proce­dure outlined below for proper operation.

1. Refer to Figure 1 for board orientation and proper
measurement equipment setup.

2. Place the jumpers as shown in the diagram. Tempo­rarily leave off the STBYMD and FCB jumpers.

3. Connect the desired loads between V

OUT1

, V

OUT2

their closest PGND terminals on the board. The loads
can be up to 4A for V

and 4A for V

OUT1

OUT2

. Soldered
wires should be used when the load current exceeds
1A in order to achieve optimum performance.

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EASURE E T SETUP

The circuit shown in Figure 2 provides fixed voltages of 5V
and 3.3V at currents of up to 4A. Figure 1 illustrates the
correct measurement setup in order to verify the typical
numbers found in the Performance Summary table. Small
spring clip leads are very convenient for small-signal
bench testing but should not be used at the current and
impedance levels associated with this switching regulator.
Soldered wire connections are required to properly ascer­tain the performance of this demonstration PC board. Do
not tie the grounds together off the test board.

The six jumpers on the left side of the board are settable
as follows: the center pin is connected to ground when the
jumper is in the rightmost position. The center pin is
connected to a positive bias source when the jumper is in
the leftmost position.

and

4. Connect the input power supply to the VIN and GND
terminals on the right, center of the board. Do
increase VIN over 30V or the

aged

. The recommended VIN to start is < 7V.

MOSFETs may be dam-

not

5. Switch on the desired channel(s) by removing the
RUN/SS1 or RUN/SS2 jumper.

6. Measure V

OUT1

and V

to verify output voltages of

OUT2

5V ± 0.1V and 3.3V ± 0.067V, respectively, at load
currents of up to 3A each.

7. Active loads can cause confusing results. Refer to

the active load discussion in the Operation section.

Figure 1. Proper Measurement Setup

2

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

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PACKAGE A D SCHE ATIC DIAGRA S

TOP VIEW

RUN/SS1
SENSE1
SENSE1
V

OSENSE1

FREQSET

STBYMD

FCB
I

SGND

3.3V

OUT

I

V

OSENSE2

SENSE2
SENSE2

+

–

TH1

TH2

–

+

1
2
3
4
5
6
7
8

9
10
11
12
13
14

G PACKAGE

28-LEAD PLASTIC SSOP

28
27
26
25
24
23
22
21
20
19
18
17
16
15

FLTCPL
TG1
SW1
BOOST1
V

IN

BG1
EXTV

CC

INTV

CC

PGND
BG2
BOOST2
SW2
TG2
RUN/SS2

LTC1628CG

TP1

3.3V
LDO

TP2

C5,0.1µF

R5

20k,1%

C13,180pF

R3,105k,1%

C9
33pF

C10
33pF

R4,63.4k,1%

C14,180pF

C7,1000pF

C8,1000pF

R9
1M

R12

1M

R10

1M

0.01µF × 3

5V

R7,15k

C19C21C20

R8,15k

R6,20k,1%

C6,0.1µF

C11
1000pF

C12
1000pF

10

11

12

13

14

1

2

3

4

5

6

7

8

9

RUN/SS1

SENSE

SENSE1

V

OSENSE1

FREQSET

STBYMD

FCB

I

TH1

SGND

3.3V

OUT

I

TH2

V

OSENSE2

SENSE2

SENSE2

+

–

U1

LTC1628

–

+

FLTCPL

TG1

SW1

BOOST1

V

BG1

EXTV

INTV

PGND

BG2

BOOST2

SW2

TG2

RUN/SS2

28

27

26

25

24

IN

23

22

CC

21

CC

20

19

18

17

0.1µF

16

15

C15
1µF

C3

0.1µF

R11,10Ω

D3

+

C16

D4

C4

4.7µF

SWITCHING FREQUENCY = 300kHz
D3, D4 = CMDSH-3TR
M1, M2 = FDS8936A
L1, L2 = PANASONIC N6 SERIES 10.1µH

M1a

C18

0.1µF

M2a

L1,10µH

+

C17

22µF

50V

L2,10µH

M1b

M2b

R1,0.015Ω

D1
MBRM
140T3

C1,150µF,6V

+

SP CAP

C2,180µF,4V

+

SP CAP

D2
MBRM
140T3

R2,0.015Ω

GND

V

OUT1

5V
4A PK

V

IN

5V TO
30V

V

OUT2

3.3V
4A PK

DC236 F02

Figure 2. LTC1628 Fixed 5V/4A, 3.3V/4A, High Efficiency Dual Regulator

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DEMO MANUAL DC236

DESIGN-READY SWITCHERS

PARTS LIST

REFERENCE
DESIGNATOR QUANTITY PART NUMBER DESCRIPTION VENDOR TELEPHONE

C1 1 EEFUE0J151R 150µF 6.3V 20% Capacitor Panasonic [phone]
C2 1 EEFUE0G181R 180µF 4V 20% Capacitor Panasonic [phone]
C3 to C6, C18 5 0603ZC104MAT1A 0.1µF 10V 20% X7R Capacitor AVX [phone]
C7, C8, C11, C12 4 06033A102JAT1A 1000pF 25V 5% NPO Capacitor AVX [phone]
C9, C10 2 06035A330JAT1A 33pF 50V 5% NPO Capacitor AVX [phone]
C13, C14 2 06035C181JAT1A 180pF 50V 5% NPO Capacitor AVX [phone]
C15 1 0805ZC105MAT1A 1µF 10V 20% X7R Capacitor AVX [phone]
C16 1 TACR475M010R 4.7µF 10V 20% Tantalum Capacitor AVX [phone]
C17 1 THCR70E1H226ZT 22µF 50V Y5U Capacitor Marcon [phone]
C19 to C21 3 0603ZC103KAT1A 0.01µF 10V 10% X7R Capacitor AVX [phone]
D1, D2 2 MBRM140T3 40V 1A Schottky Diode ON Semiconductor [phone]
D3, D4 2 CMDSH-3TR 30V 0.1A Schottky Diode Central [phone]
L1, L2 2 CEP123-8R0MC or 8µH Low Profile Inductor Sumida [phone]

M1, M2 2 FDS8936A Dual N-Channel MOSFET Fairchild [phone]
R1, R2 2 LR1206-01-R015-F 0.015Ω 1/4W 1% Chip Resistor IRC [phone]
R3 1 CR16-1053FM 105k 1/16W 1% Chip Resistor TAD [phone]
R4 1 CR16-6342FM 63.4k 1/16W 1% Chip Resistor TAD [phone]
R5, R6 2 CR16-2002FM 20k 1/16W 1% Chip Resistor TAD [phone]
R7, R8 2 CR16-153JM 15k 1/16W 5% Chip Resistor TAD [phone]
R9, R10, R12 3 CR16-105JM 1M 1/16W 5% Chip Resistor TAD [phone]
R11 1 CR16-100JM 10Ω 1/16W 5% Chip Resistor TAD [phone]
U1 1 LTC1628CG28 Multiphase Dual DC/DC Controller IC LTC [phone]

CDRH125-100MC or 10µH Inductor
ETQP6F102HFA 10µH Inductor Panasonic [phone]

UW

A UFACTURER TELEPHO E DIRECTORY

MANUFACTURER USA EUROPE JAPAN HONG KONG SINGAPORE TAIWAN/KOREA

AVX [phone] 44-1252-770-000 81-751-592-3897 852-2-363-3303 65-258-2833 886-2-516-7010
BH Elect. [phone]
Central [phone] 49-0816-143-963 822-2-268-9795
Coilcraft [phone] 886-2-264-3646 65-296-6933 886-2-264-3646
Fairchild [phone] 44-1793-856-856 81-3-5620-6175 852-2-273-7200 65-252-5077 886-2-712-0500
Gowanda [phone]
IR [phone] 44-1883-713-215 81-3-3983-0086 852-2-803-7380 65-221-8371 822-2-858-8773
IRC [phone] 852-2-388-0629 65-280-0200 0342-43-2822
Kemet [phone] 44-1279-757-343 852-2-305-1168 65-484-2220 886-2-752-8585
Linear Technology [phone] 44-1276-677-676 81-3-3267-7891 852-2-803-7380 65-753-2692 886-2-521-7575
Midcom [phone]
Murata [phone]
Marcon [phone]
ON Semiconductor [phone] 81-3-3521-8315 852-2-662-9298 65-481-8188
Panasonic [phone]

4

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DEMO MANUAL DC236

DESIGN-READY SWITCHERS

UW

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A UFACTURER TELEPHO E DIRECTORY

MANUFACTURER USA EUROPE JAPAN HONG KONG SINGAPORE TAIWAN/KOREA

Sanyo [phone] 49-06102-7154-17 81-3-0720-70-1005 852-2-887-2109 65-747-9755
Sumida [phone] 81-3-3607-5111 852-2-880-6688 65-296-3388 886-2-726-2177
TAD [phone]
Taiyo Yuden [phone] 44-1494-464-642 81-3-3833-5441 852-2-736-3803 65-861-4400 886-2-797-2155
Temic [phone] 44-1344-707-300 81-3-5562-3321 852-2-378-9789 65-788-6668 886-2-755-6108
Toko [phone]
Tokin [phone] 44-1236-780-850 852-2-730-0028 886-2-521-3998

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OPERATIO

Theory and Benefits of 2-Phase Operation

The LTC1628 dual high efficiency DC/DC controller brings
the considerable benefits of 2-phase operation to portable
applications for the first time. Notebook computers, PDAs,
handheld terminals and automotive electronics will all
benefit from the lower input filtering requirement, reduced
electromagnetic interference (EMI) and increased effi­ciency associated with 2-phase operation.

Why the need for 2-phase operation? Before the LTC1628,
constant-frequency dual switching regulators operated
both channels in phase (i.e., 1-phase operation). This
means that both switches turned on at the same time,
causing current pulses of up to twice the amplitude of
those for one regulator to be drawn from the input capaci­tor and battery. These large amplitude current pulses
increased the total RMS current flowing from the input

capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.

With 2-phase operation, the two channels of the dual
switching regulator are operated 180 degrees out of
phase. This effectively interleaves the current pulses com­ing from the switches, greatly reducing the overlap time
where they add together. The result is a significant reduc­tion in total RMS input current, which, in turn, allows less
expensive input capacitors to be used, reduces shielding
requirements for EMI and improves real world operating
efficiency.

Figure 3 compares the input waveforms for a representa­tive 1-phase dual switching regulator to the new LTC1628
2-phase dual switching regulator. An actual measurement
of the RMS input current under these conditions shows

Typical Single-Phase

I

= 2.53A

IN(MEAS)

Figure 3. Input Waveforms Comparing Single-Phase and 2-Phase Operation for Dual
Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple
with the LTC1628 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces
Shielding Requirements for EMI and Improves Efficiency

RMS

(a)

DC236 F03a

5V SWITCH: 20V/DIV

3.3V SWITCH: 20V/DIV

INPUT CURRENT: 5A/DIV

INPUT VOLTAGE: 500mV/DIV

LTC1628 2-Phase

I

= 1.55A

IN(MEAS)

RMS

(b)

DC236 F03b

5

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

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OPERATIO

that 2-phase operation lowers the input current from

2.53A

to 1.55A

RMS

RMS

.

Although this is an impressive reduction in itself, remem­ber that the power losses are proportional to I

RMS

2

,
meaning that the actual power wasted is reduced by a
factor of 2.66. The reduced input ripple voltage also means
less power lost in the input power path, which could
include batteries, switches, trace/connector resistances
and protection circuitry. Improvements in both conducted
and radiated EMI also directly accrue as a result of the
reduced RMS input current and voltage.

Of course, the improvement afforded by 2-phase opera­tion is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = V

OUT/VIN

). Figure 4 shows how
the RMS input current varies for 1-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.

It can be readily seen that the advantages of 2-phase
operation are not limited to a narrow operating range, but
in fact extend over a wide region. A good rule of thumb for
most applications is that 2-phase operation will reduce the
input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.

A final question: If 2-phase operation offers such an
advantage over 1-phase operation for dual switching
regulators, why hasn’t it been done before? The answer is

3.0
SINGLE-PHASE

2.5

2.0

1.5

1.0

INPUT RMS CURRENT (A)

0.5

VO1 = 5V/3A

= 3.3V/3A

V

O2

0

0

Figure 4. RMS Input Current Comparison

DUAL CONTROLLER

2-PHASE

DUAL CONTROLLER

10 20 30 40

INPUT VOLTAGE (V)

DC236 F04

that, while simple in concept, it is hard to implement.
Constant-frequency, current mode switching regulators
require an oscillator-derived “slope compensation” signal
to allow stable operation of each regulator at over 50%
duty cycle. This signal is relatively easy to derive in
1-phase dual switching regulators, but required the devel­opment of a new and proprietary technique to allow
2-phase operation. In addition, isolation between the two
channels becomes more critical with 2-phase operation
because switch transitions in one channel could poten­tially disrupt the operation of the other channel.

The LTC1628 is proof that these hurdles have been sur­mounted. The new device offers unique advantages for the
ever expanding number of high efficiency power supplies
required in portable electronics.

DC236 Operation

The LTC1628 switching regulator performs high effi­ciency DC/DC voltage conversion while maintaining con­stant frequency over a wide range of load current, using a
2-phase current mode architecture. The 2-phase approach
results in 75% less power loss (and heat generated) in the
input source resistance because dissipated power is
proportional to the square of the RMS current. The input
ripple frequency is also double the individual controller’s
switching frequency, further reducing the input capaci­tance requirement. Reducing peak currents and doubling
the radiated frequency significantly reduces EMI related
problems.

The internal oscillator frequency is set by the voltage
applied to the FREQSET pin. The FREQ jumper on the
demonstration board allows selection of three different
voltages: 0V, 1.2V when the jumper is left off, and 5V. The
internal oscillator will run at 130kHz, 200kHz and 300kHz
respectively. The frequency can be continuously varied
over a 130kHz to 300kHz range by applying an external 0V
to 2.4V to the FREQSET pin.

High efficiency is made possible by selecting either of two
low current modes: 1) Burst Mode operation for maximum
efficiency and 2) constant frequency, burst disable mode
for slightly less efficiency. Constant frequency is desirable
in applications requiring minimal electrical noise.

6

OPERATIO

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

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Burst Mode operation allows the output MOSFETs to
“sleep” between several PWM switching cycle periods of
normal MOSFET activity. The current loss due to charging
the MOSFETs is not present during these “sleeping”
periods. Hysteretic output voltage detection results in a
slight increase of output voltage ripple during Burst Mode
operation. Bursting starts at approximately 20% of maxi­mum designed load current.

The burst disable mode allows heavily discontinuous,
constant-frequency operation down to approximately 1%
of maximum designed load current. This mode results in
the elimination of switching frequency subharmonics over
99% of the output load range. Switching cycles start to be
dropped at approximately 1% of maximum designed load
current in order to maintain proper output voltage.

The FCB input pin allows the selection of the low current
operating mode of both switching regulator controllers.
Burst disable mode is enabled when the FCB pin is tied to
INTVCC.

Tying the FCB pin to ground potential forces the controller
into PWM or forced continuous mode. In forced continu­ous mode, the output MOSFETs are always driven, regard­less of output loading conditions. Operating in this mode
allows the switching regulator to source or sink current—
but be careful: when the output stage sinks current, power
is transferred back into the input supply terminals and the
input voltage rises.

Burst Mode operation is enabled when the voltage applied
to the FCB pin is less than (INTVCC – 0.8V) or if the pin is
left open. A comparator, having a precision 0.8V thresh­old, allows the pin to be used to regulate a secondary
winding on the switching regulator’s output. A small
amount of hysteresis is included in the design of the
comparator to facilitate clean secondary operation. When
the resistively divided secondary output voltage falls
below the 0.8V threshold, the controller operates in the
forced continuous operating mode for as long as it takes
to bring the secondary voltage above the 0.8V + hysteresis
level.

The FLTCPL pin allows coupling between the two con­trollers in several situations. The controllers will act

independently when FLTCPL is grounded. When the pin
is tied to INTVCC the following operations result:

1. When the FCB input voltage falls below its 0.8V thresh­old, both controllers go into a forced continuous oper­ating mode.

2. When either controller latches off due to an overload
condition (or short circuit), the other channel will be
latched off as well. Either the STBYMD mode pin or both
RUN/SS1 and RUN/SS2 pins need to be pulled to
ground in order to unlatch this condition. The STBYMD
mode pin internally pulls down both RUN/SS pins when
grounded. If the latches are defeated through the use of
an external pull-up current, neither latch will be acti­vated.

The STBYMD PC board input is tied to the STBYMD IC pin.
Pulling the STBYMD IC pin up with greater than 5µA to
the input supply turns on the internal 5V INTVCC and the

3.3V LDO regulators when neither of the two switching

regulator controllers is turned on. The 5V INTVCC regula­tor will supply up to 50mA
supply up to 25mA
higher but internal power dissipation must be calculated to
guarantee that die temperature does not exceed the data
sheet specifications.

The demonstration board is shipped in a standard configu­ration of 5V/3.3V but may be modified to produce output
voltages as low as 0.8V. Modifications will require changes
to the resistive voltage feedback divider and, in some
cases, the ITH pin compensation components.

Efficiency measurement depends on the operating condi­tions of both regulators and must be performed thought­fully and carefully. The maximum efficiency will occur with
the minimum required circuitry operating on an individual
regulator. Since there is much common circuitry operat­ing in the IC when both regulators are running, overall
efficiency numbers will actually increase when the two
switching regulators are active. The increase is not signifi­cant at high output currents but can become very signifi­cant at low output currents, when the IC supply current
becomes an appreciable part of the total input supply
current.

. Peak currents may be significantly

RMS

and the 3.3V LDO will

RMS

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DEMO MANUAL DC236

DESIGN-READY SWITCHERS

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OPERATIO

Refer to the LTC1628 data sheet for further information on
the internal operation and functionality descriptions of
the IC.

Overcurrent and Overvoltage Protection

The RUN/SS capacitor, CSS, is used initially to turn on and
limit the inrush current of the controller. After the control­ler has been started and given adequate time to charge the
output capacitor and provide full load current, CSS is used
as a short-circuit time-out circuit. If the output voltage falls
to less than 70% of its nominal output voltage, CSS begins
discharging on the assumption that the output is in an
overcurrent and/or short-circuit condition. If the condition
lasts for a long enough period as determined by the size of
CSS, the controller will be shut down until the RUN/SS pin
voltage is recycled. This built-in latch-off can be overrid­den by providing >5µA pull-up at a compliance of 4V to the
RUN/SS pin. This current shortens the soft start period but
also prevents net discharge of the RUN/SS capacitor
during an overcurrent and/or short-circuit condition. Fold­back current limiting is activated when the output voltage
falls below 70% of its nominal level, whether or not the
short-circuit latch-off circuit is enabled.

The output is protected from overvoltage by a “soft latch.”
When the output voltage exceeds the regulation value by
more than 7.5%, the synchronous MOSFET turns on and
remains on for as long as the overvoltage condition is
present. If the output voltage returns to a safe level, normal
operation resumes. This self-resetting action prevents
“nuisance trips” due to momentary transients and elimi­nates the need for the Schottky diode that is necessary
with conventional OVP to prevent V

DC236 Physical Design

The demonstration board is manufactured using a typical
4-layer copper PC board. The outside layers are 2 oz
copper and the inside layers are 1oz copper. The board is
designed to use the minimum number of external compo­nents but has a few components added to facilitate
optional IC configurations. These added components will
not be required in a final design. These components
include R9, R10, R12 and C19 to C21. Other components
that may not be necessary depending upon the particular

reversal.

OUT

design include C9, C10, C13, C14, C18 and R11. Certain
components may be larger than specific applications will
require. The output capacitance and the inductance values
selected are larger than may be required in order to
accommodate the very wide operating frequency range
(130kHz to 300kHz) capability of the demonstration board.
Output capacitance as low as 47µF and inductance values
as low as several microhenries will work well at the higher
frequencies. The 2-phase controller technique signifi­cantly reduces the capacity and ESR requirements of the
input capacitor when compared to a 1-phase approach.
The dual output MOSFETs used in the design reduce the
overall size of the design and take advantage of an
extended copper foil trace to help dissipate power on the
board. The Schottky diodes, D1 and D2, can also be
removed to reduce system cost but will decrease effi­ciency slightly.

Active Loads—Beware!

Beware of Active Loads. They are convenient but problem­atic. Some active loads do not turn on until the applied
voltage rises above 0.1V to 0.8V. The turn-on may be
delayed as well. Under these conditions, a switching
regulator with soft start may appear to start up and then
shut down before eventually reaching the correct output
voltage. What actually happens is as follows: at switching
regulator turn-on, the output voltage is below the active
load’s turn-on requirements. The switching regulator’s
output rises to the correct output voltage level due to the
inherent delay in the active load. The active load turns on
after its internal delay and now pulls down the switching
regulator’s output because the switcher is in its soft start
interval. The switching regulator’s output may come up at
some later time when the soft start interval has passed.

A switching regulator with foldback current limit will also
have difficulty with the unrealistic I-V characteristic of the
active load. Foldback current limiting will reduce the
output current available as the output voltage drops below
a threshold level (this level is 70% of nominal V
LTC1628). This reduction in available output current will
result in the active load immediately pulling down the
output because the active load’s current demand remains
constant as the output voltage decreases. Most actual

OUT

for the

8

OPERATIO

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

U

loads do not behave like the active load’s I-V characteris­tics. Actual loads normally have VIN • C • f dependency
where C is internal chip capacitance and f is the frequency
of operation. To alleviate the active load problem during
testing, the active load should be initially programmed to
a much lower current value until the switching regulator’s
soft start inverval has passed and then reprogrammed to
the higher level. The switching regulator will supply the
increased current required according to the transient
response behavior of the design. Sufficient output capaci­tance is needed to accommodate the current step during
the transient period, keeping the output voltage at or above
the foldback threshold of 70%.

PC Board Layout Hints

Switching power supply printed circuit layouts are cer­tainly among the most difficult analog circuits to design.
The following suggestions will help.

The input circuit, including the external switching MOS­FETs, input capacitor(s) and Schottky diode(s) all have
fast voltage and current transitions associated with them.
These components and the radiated fields (electrostatic
and/or electromagnetic)
sensitive control circuitry and loop compensation compo­nents required for a current mode switching regulator.

The electrostatic or capacitive coupling problems can be
reduced by increasing the distance from the very large or
very fast moving voltage signals. The signal points that
cause problems generally include the switch node, any
secondary flyback winding voltage and any nodes that also
move with these nodes. The switch, MOSFET gate and
boost nodes move between VIN and PGND during each
cycle, with less than a 50ns transition time. Secondary
flyback windings produce an AC signal component of – V
times the turns ratio of the transformer and also have a
similar <50ns transition time. The control input signals
need to have less than a few millivolts of noise in order for
the regulator to perform properly. A rough calculation
shows that 80dB of isolation at 2MHz is required from the
switch node for low noise switcher operation. The situa­tion is worsened by a factor of the turns ratio for any

must

be kept away from the very

IN

secondary flyback winding. Keep these switch node­related PC traces small and away from the “quiet” side of
the IC (not just above and below each other on the opposite
side of the board).

The electromagnetic or current loop-induced feedback
problems can be minimized by keeping the high AC
current (transmitter) paths
(receiver) path small and/or short. Maxwell’s equations
are at work here, trying to disrupt our clean flow of current
and voltage information from the output back to the
controller input. It is crucial to understand and minimize
the susceptibility of the control input stage as well as the
more obvious reduction of radiation from the high current
output stage(s). An inductive transmitter depends upon
the frequency, current amplitude and the size of the
current loop to determine the radiation characteristic of
the generated field. The current levels are set in the output
stage once the input voltage, output voltage and inductor
value(s) have been selected. The frequency is set by the
output stage transition times. The only parameter over
which we have some control is the size of the antenna we
create on the PC board, i.e., the loop. A loop is formed with
the input capacitance, the top MOSFET, the Schottky diode
and the path from the Schottky diode’s ground connection
to the input capacitor’s ground connection. A second path
is formed when a secondary winding is used, comprising
the secondary output capacitor, the secondary winding
and the rectifier diode or switching MOSFET (in the case
of a synchronous approach). These loops should be kept
as small and tightly packed as possible in order to mini­mize their “far field” radiation effects. The radiated field
produced is picked up by the current comparator input
filter circuit(s), as well as by the voltage feedback circuit(s).
The current comparator’s filter capacitor, placed across
the sense pins, attenuates the radiated current signal. It is
important to place this capacitor immediately adjacent to
the IC SENSE pins. The voltage sensing input(s) minimize
the inductive pickup component by using an input capaci­tance filter to SGND. The capacitors in both cases serve to
integrate the induced current, reducing the susceptibility
to both the loop radiated magnetic fields and the trans­former or inductor leakage fields.

and

the feedback circuit

9

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

U

OPERATIO

The PGND-SGND tie point for the LTC1628 switching
regulator controllers is optimized by connecting the
grounds directly under the IC, creating a close surface
grounding plane.

The capacitor on INTVCC acts as a reservoir to supply the
high transient currents to the bottom gates
recharge the boost capacitor. This capacitor should be a
10µF ceramic capacitor or a 1µF ceramic capacitor in
parallel with a 4.7µF tantalum capacitor. The ceramic
capacitor must be placed as close as possible to the
INTVCC and PGND pins of the IC. Peak currents exceed 1A
when charging the gates of the bottom MOSFETs.

and

to

UW

PCB LAYOUT A D FIL

The traces that sense the voltage across the current­sensing resistor can be long but should run parallel to each
other and be spaced with the minimum separation allowed
in order to experience the same electrostatic and electro­magnetic fields from radiating sources. The traces should
be wider than minimum if they are long in order to
minimize self-inductance. Keep these traces on a PC board
plane farthest from the high current and large switching
voltage plane. Any filtering resistors in series with these
traces should be placed close to the IC rather than close to
radiating nodes, such as the switch and boost nodes.

Component Side Silkscreen Copper Layer 1 Top

10

UW

PCB LAYOUT A D FIL

Copper Layer 2 Copper Layer 3

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

Copper Layer 4

Solder Mask Top

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

Paste Mask

Solder Mask Bottom

11

DEMO MANUAL DC236

DESIGN-READY SWITCHERS

U

PC FAB DRAWI G

3000.0

A

E

G

E

E

E

G

E

G

E

A

HOLE CHART

SYMBOL DIAMETER OF HOLES PLATED

TOTAL 107

E

G

D

D

A 120 4 YES
B 94 6 YES
C70 2 NO
D 64 2 YES
E 30 24 YES

G 10 33 YES

G

F

G

F

F

G

F

G

C

E

F 15 36 YES

E

BB

G

G

F

F

F

G

F

G

F

BB

F

G

E

G

G

NUMBER

200.0

A

C

F

F

B

B

A

200.0

E

2000.0

E

NOTES: UNLESS OTHERWISE SPECIFIED

1. ALL DIMENSIONS ARE IN MILS, ±3

2. FINISHED HOLE SIZES ARE ±3

3. FINISHED MATERIAL IS FR4, 62-THICK, 2 OZ Cu, 4-LAYERS

4. PLATED HOLE WALL THICKNESS IS 1MIL MINIMUM

5. INTERNAL LAYERS ARE 1 OZ Cu

Linear Technology Corporation

12

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com

dc236fa LT/TP 0200 1K REV A • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1998