Linear Technology LT8302 Operation

42VIN Micropower No-Opto

8302 TA01b

Isolated Flyback Converter

with 65V/3.6A Switch

FeaTures DescripTion

LT8302

n

2.8V to 42V Input Voltage Range

n

3.6A, 65V Internal DMOS Power Switch

n

Low Quiescent Current:
106µA in Sleep Mode
380µA in Active Mode

n

Quasi-Resonant Boundary Mode Operation at

Heavy Load

n

Low Ripple Burst Mode® Operation at Light Load

n

Minimum Load < 0.5% (Typ) of Full Output

n

No Transformer Third Winding or Opto-Isolator

Required for Output Voltage Regulation

n

Accurate EN/UVLO Threshold and Hysteresis

n

Internal Compensation and Soft-Start

n

Temperature Compensation for Output Diode

n

Output Short-Circuit Protection

n

Thermally Enhanced 8-Lead SO Package

applicaTions

n

Isolated Automotive, Industrial, Medical Power

Supplies

n

Isolated Auxiliary/Housekeeping Power Supplies

The LT®8302 is a monolithic micropower isolated flyback
converter. By sampling the isolated output voltage directly
from the primary-side flyback waveform, the part requires
no third winding or opto-isolator for regulation. The output
voltage is programmed with two external resistors and a
third optional temperature compensation resistor. Bound

­ary mode operation provides a small magnetic solution with
excellent load regulation. Low ripple Burst Mode operation
maintains high efficiency at light load while minimizing the
output voltage ripple. A 3.6A, 65V DMOS power switch
is integrated along with all the high voltage circuitry and
control logic into a thermally enhanced 8-lead SO package.

The LT8302 operates from an input voltage range of 2.8V
to 42V and delivers up to 18W of isolated output power.
The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
component count, and high efficiency application solution
for isolated power delivery.

L, LT, LTC , LT M, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522.

Typical applicaTion

2.8V to 32VIN/5V

V

IN

2.8V TO 32V

10µF

EN/UVLO

LT8302

GND

INTV

CC

1µF

Isolated Flyback Converter

OUT

V

IN

SW

R

FB

R

REF

115k 10k

TC

+

V

220µF

= 5V)

IN

= 12V)

IN

= 24V)

IN

OUT

5V

–

V

OUT

3:1

470pF

39Ω

154k

8302 TA01a

For more information www.linear.com/LT8302

•

1µH

9µH

•

10mA TO 1.1A (V
10mA TO 2.0A (V
10mA TO 2.9A (V

Efficiency vs Load Current

90

FRONT PAGE APPLICATION

85

80

75

EFFICIENCY (%)

70

65

60

0

0.5 1.0

2.0 3.0

1.5 2.5

LOAD CURRENT (A)

VIN = 5V

= 12V

V

IN

= 24V

V

IN

8302fb

1

LT8302

(Note 1)

pin conFiguraTionabsoluTe MaxiMuM raTings

SW (Note 2) ..............................................................65V

............................................................................42V

V

IN

EN/UVLO ....................................................................V

RFB ........................................................V

– 0.5V to V

IN

IN
IN

Current Into RFB ....................................................200µA

, R

INTV

CC

, TC ......................................................... 4V

REF

Operating Junction Temperature Range (Notes 3, 4)

LT8302E, LT8302I.............................. –40°C to 125°C

LT8302H ............................................ –40°C to 150°C

EN/UVLO

INTV

CC

V

GND

EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB

TOP VIEW

1

2

3

IN

4

S8E PACKAGE

8-LEAD PLASTIC SO

= 33°C/W

θ

JA

9

GND

TC

8

R

7

REF

R

6

FB

SW

5

LT8302MP ......................................... –55°C to 150°C

Storage Temperature Range .................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec) ...................300°C

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT8302ES8E#PBF LT8302ES8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302IS8E#PBF LT8302IS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 125°C
LT8302HS8E#PBF LT8302HS8E#TRPBF 8302 8-Lead Plastic SO –40°C to 150°C
LT8302MPS8E#PBF LT8302MPS8E#TRPBF 8302 8-Lead Plastic SO –55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.

2

8302fb

For more information www.linear.com/LT8302

LT8302

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT

V

IN

I

Q

I

HYS

V

INTVCC

I

INTVCC

V

TC

I

TC

f

MIN

t

ON(MIN)

t

OFF(MAX)

I

SW(MAX)

I

SW(MIN)

R

DS(ON)

I

LKG

t

SS

VIN Voltage Range
VIN Quiescent Current V

V

EN/UVLO
EN/UVLO

= 0.3V

= 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)

EN/UVLO Shutdown Threshold For Lowest Off I

Q

EN/UVLO Enable Threshold Falling
EN/UVLO Enable Hysteresis 14 mV
EN/UVLO Hysteresis Current V

V
V

INTVCC Regulation Voltage I
INTVCC Current Limit V

UVLO Threshold Falling 2.39 2.47 2.55 V

INTV

CC

UVLO Hysteresis 105 mV

INTV

CC

– VIN) Voltage I

(R

FB

Regulation Voltage

R

REF

Regulation Voltage Line Regulation 2.8V ≤ VIN ≤ 42V –0.01 0 0.01 %/V

R

REF

= 0.3V

EN/UVLO

= 1.1V

EN/UVLO

= 1.3V

EN/UVLO

= 0mA to 10mA 2.85 3 3.1 V

INTVCC

= 2.8V 10 13 16 mA

INTVCC

= 75µA to 125µA –50 50 mV

RFB

TC Pin Voltage 1.00 V
TC Pin Current VTC = 1.2V

V

= 0.8V

TC

Minimum Switching Frequency 11.3 12 12.7 kHz
Minimum Switch-On Time 160 ns
Maximum Switch-Off Time Backup Timer 170 µs
Maximum Switch Current Limit 3.6 4.5 5.4 A
Minimum Switch Current Limit 0.78 0.87 0.96 A
Switch On-Resistance ISW = 1.5A 80 mΩ
Switch Leakage Current VSW = 65V 0.1 0.5 µA
Soft-Start Timer 11 ms

EN/UVLO

= VIN, C

l

l

l

l

= 1µF to GND, unless otherwise noted.

INTVCC

2.8 42 V

0.5

2 µA

53
106
380

0.3 0.75 V

1.178 1.214 1.250 V

–0.1

2.3

–0.1

0

2.5
0

0.1

2.7

0.1

0.98 1.00 1.02 V

12 15

18 µA

–200

µA
µA
µA

µA
µA
µA

µA

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.

Note 2: The SW pin is rated to 65V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 65V as shown
in Figure 5.

Note 3: The LT8302E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The

LT8302I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8302H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT8302MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.

Note 4: The LT8302 includes overtemperature protection that is intended
to protect the device during momentary
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.

For more information www.linear.com/LT8302

overload conditions. Junction

8302fb

3

LT8302

8302 G07

Typical perForMance characTerisTics

Output Load and Line Regulation Output Temperature Variation

5.20

5.15

5.10

5.05

5.00

4.95

OUTPUT VOLTAGE (V)

4.90

4.85

4.80

0.5 1.0 2.0

0

LOAD CURRENT (A)

1.5

VIN = 5V

= 12V

V

IN

= 24V

V

IN

2.5 3.0

8302 G01

Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms

5.3
FRONT PAGE APPLICATION

= 12V

V

IN

5.2

= 1A

I

OUT

5.1

RTC = 115k

5.0

4.9

OUTPUT VOLTAGE (V)

4.8

4.7

–50

RTC = OPEN

–25 0

50 100 150125

25 75

TEMPERATURE (°C)

= 25°C, unless otherwise noted.

T

A

Switching Frequency
vs Load Current

500

FRONT PAGE APPLICATION

400

300

200

FREQUENCY (kHz)

100

8302 G02

0

0 0.5

1.0

LOAD CURRENT (A)

1.5

2.0

VIN = 5V

= 12V

V

IN

= 24V

V

IN

2.5

3.0

8302 G03

V

SW

20V/DIV

V

OUT

50mV/DIV

FRONT PAGE APPLICATION

= 12V

V

IN

= 2A

I

OUT

VIN Shutdown Current

10

8

6

(µA)

Q

I

4

2

0

0

V

SW

20V/DIV

V

OUT

50mV/DIV

2µs/DIV

8302 G04

FRONT PAGE APPLICATION

= 12V

V

IN

= 0.5A

I

OUT

2µs/DIV

8302 G05

VIN Quiescent Current,
Sleep Mode

TJ = 150°C

= 25°C

T

J

= –55°C

T

J

10

20

VIN (V)

30

40

50

140

130

120

110

(µA)

Q

I

100

TJ = 150°C

TJ = 25°C

TJ = –55°C

90

80

0

10 20 30 40

VIN (V)

50

8302 G08

V

20V/DIV

V

OUT

50mV/DIV

420

400

380

(µA)

Q

I

360

340

320

SW

FRONT PAGE APPLICATION

= 12V

V

IN

= 10mA

I

OUT

20µs/DIV

VIN Quiescent Current,
Active Mode

TJ = 150°C

TJ = 25°C

TJ = –55°C

10

0

20

VIN (V)

8302 G06

30

40

50

8302 G09

4

8302fb

For more information www.linear.com/LT8302

= 25°C, unless otherwise noted.

8302 G12

8302 G13

8302 G14

8302 G16

8302 G17

Typical perForMance characTerisTics

T

A

EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature

1.240

1.235

1.230

1.225

(V)

1.220

EN/UVLO

V

1.215

1.210

1.205

1.200
–50

–25

0

RISING

FALLING

50

25

TEMPERATURE (°C)

75

100

125

150

8302 G10

(µA)

HYST

I

5

4

3

2

1

0

–50

–25 0

50

25 75 150

TEMPERATURE (°C)

100 125

8302 G11

3.10

3.05

3.00

(V)

2.95

INTVCC

V

2.90

2.85

2.80
–50

I

0 50

–25 25

TEMPERATURE (°C)

I

INTVCC

INTVCC

LT8302

= 0mA

= 10mA

100

75

125

150

INTVCC Voltage vs V

3.10

3.05

3.00

(V)

2.95

INTVCC

V

2.90

2.85

1.010

1.008

1.006

1.004

1.002

(V)

1.000

RREF

V

0.998

0.996

0.994

0.992

0.990

2.80
10 15 45

5

R

Regulation Voltage R

REF

–50

–25 25

I

= 0mA

INTVCC

I

= 10mA

INTVCC

VIN (V)

0

50

TEMPERATURE (°C)

IN

35 4020 25 30

125

100

75

150

INTVCC UVLO Threshold (RFB-VIN) Voltage

2.8

2.7

2.6

2.5

INTVCC (V)

V

2.4

2.3

2.2

1.010

1.008

1.006

1.004

1.002

(V)

1.000

RREF

V

0.998

0.996

0.994

0.992

0.990

RISING

FALLING

100

20

VIN (V)

75

30

150

125

40

50

–50

REF

0

TEMPERATURE (°C)

Line Regulation TC Pin Voltage

10

0 50

–25 25

40

30

20

10

0

–10

VOLTAGE (mV)

–20

–30

–40

–25

–50

1.5

1.4

1.3

1.2

(V)

1.1

TC

V

1.0

0.9

0.8

0.7
–25

–50

I

= 125µA

RFB

I

= 100µA

RFB

I

= 75µA

RFB

0

0

50

25

TEMPERATURE (°C)

50

25

TEMPERATURE (°C)

75

100

75

100

125

125

150

8302 G15

150

8302 G18

For more information www.linear.com/LT8302

8302fb

5

LT8302

Typical perForMance characTerisTics

R

DS(ON)

200

160

120

80

RESISTANCE (mΩ)

40

0

–50

–25 0

50

25 75 150

TEMPERATURE (°C)

100 125

8302 G19

Switch Current Limit Maximum Switching Frequency

5

MAXIMUM CURRENT LIMIT

MINIMUM CURRENT LIMIT

50

–25 0

25 75 150

TEMPERATURE (°C)

(A)

SW

I

4

3

2

1

0

–50

TA = 25°C, unless otherwise noted.

500

400

300

200

FREQUENCY (kHz)

100

0

–50

100 125

8302 G20

–25 0

Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time

20

16

12

8

FREQUENCY (kHz)

4

400

300

200

TIME (ns)

100

400

300

200

TIME (ns)

100

50

25 75 150

TEMPERATURE (°C)

100 125

8302 G21

0

–50

–25 0

50

25 75 150

TEMPERATURE (°C)

100 125

8302 G22

0

–50

–25 0 25 50

TEMPERATURE (°C)

75 100 125 150

8302 G23

0

–50

–25 0 25 50

TEMPERATURE (°C)

75 100 125 150

8302 G24

8302fb

6

For more information www.linear.com/LT8302

pin FuncTions

LT8302

EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin

is used to enable the LT8302. Pull the pin
below 0.3V to shut down the LT8302. This pin has an ac­curate 1.214V threshold

and can be used to program a VIN
undervoltage lockout (UVLO) threshold using a resistor
divider from V
allows the programming of V
function is used, tie this pin directly to V

INTV

INTV

(Pin 2): Internal 3V Linear Regulator Output. The

CC

pin is supplied from VIN and powers the internal

CC

to ground. A 2.5µA current hysteresis

IN

UVLO hysteresis. If neither

IN

.

IN

control circuitry and gate driver. Do not overdrive the
INTV

pin with any external supply, such as a third winding

CC

supply. Locally bypass this pin to ground with a minimum
1µF ceramic capacitor.

(Pin 3): Input Supply. The VIN pin supplies current to

V

IN

the internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the R

pin. Locally

FB

bypass this pin to ground with a capacitor.
GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed

pad provides both electrical contact to ground and good
thermal contact to the printed circuit board

. Solder the

exposed pad directly to the ground plane.

(Pin 5): Drain of the Internal DMOS Power Switch.

SW

Minimize trace area at this pin to reduce EMI and voltage
spikes.

(Pin 6): Input Pin for External Feedback Resistor.

R

FB

Connect a resistor from this pin to the transformer primary
SW pin. The ratio of the R

resistor to the R

FB

resistor,

REF

times the internal voltage reference, determines the output
voltage (plus the effect of any non-unity transformer turns
ratio). Minimize trace area at this pin.

(Pin 7): Input Pin for External Ground Referred Ref-

R

REF

erence Resistor.

The resistor at this pin should be in the
range of 10k, but for convenience in selecting a resistor
divider ratio, the value may range from 9.09k to 11.0k.

TC (Pin 8): Output Voltage Temperature Compensation. The
voltage at this pin is proportional to absolute temperature
(PTAT) with temperature coefficient equal to 3.35mV/°K,
i.e., equal to 1V at room temperature 25°C. The TC pin

REF

-

pin

voltage can be used to estimate the LT8302 junction tem
perature. Connect

a resistor from this pin to the R

to compensate the output diode temperature coefficient.

For more information www.linear.com/LT8302

8302fb

7

LT8302

block DiagraM

V

IN

INTV

CC

C

INTVCC

R

R

EN1

EN2

2

1

EN/UVLO

LDO

2.5µA

M4

D

OUT

T1

C

IN

R

FB

3

V

IN

M3

25µA

1.214V

–

A1

6

R

FB

1:4

M2

BOUNDARY
DETECTOR

OSCILLATOR

–

–

A3

+

+

1V

+

VOLTAGE

g

PTAT

m

S

R M1

Q

V

IN

START-UP,

REFERENCE,

CONTROL

INTV

CC

DRIVER

A2

+

N:1

•

L1A L1B C

•

5

SW

R

SENSE

+

V

OUT

OUT

–

V

OUT

–

R

REF

7

R

R

REF

TC

8

TC

4, EXPOSED PAD PIN 9

GND

8302 BD

8

8302fb

For more information www.linear.com/LT8302

operaTion

LT8302

The LT8302 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
the output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation bound
ary. Opto-isolator circuits waste output power, and the

components increase the cost and physical size of

extra
the power supply. Opto-isolators can also cause system
issues due to limited dynamic response, nonlinearity, unit­to-unit variation and aging over lifetime. Circuits employing
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.

The LT8302 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is re
quired for
boundar
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method im
proves load regulation without the need of external load
compensation components.

The
back converter housed in a thermally enhanced 8-lead
SO package. The output voltage is programmed with two
external resistors. An optional TC resistor provides easy

regulation. Since the LT8302 operates in either

y conduction mode or discontinuous conduction

LT8302 is a simple to use micropower isolated fly

-

-

-

-

output diode temperature compensation. By integrating
the loop compensation and soft-start inside, the part
reduces the number of external components. As shown
in the Block Diagram, many of the blocks are similar to
those found in traditional switching regulators including
reference, regulators, oscillator, logic, current amplifier,
current comparator, driver, and power switch. The novel
sections include a flyback pulse sense circuit, a sample­and-hold error amplifier, and a boundary mode detector,
as well as the additional logic for boundary conduction
mode, discontinuous conduction mode, and low ripple
Burst Mode operation.

Quasi-Resonant Boundary Mode Operation

The LT8302 features quasi-resonant boundary conduction
mode operation at heavy load, where the chip turns on the
primary power switch when the secondary current is zero
and the SW rings to its valley. Boundary conduction mode
is a variable frequency, variable peak-current switching
scheme. The power switch turns on and the transformer
primary current increases until an internally controlled peak
current limit. After

the SW pin rises to the output voltage multiplied by

on
the primary-to-secondary transformer turns ratio plus the
input voltage. When the secondary current through the
output diode falls to zero, the SW pin voltage collapses
and rings around V
this event and turns the power switch back on at its valley.

the power switch turns off, the voltage

. A boundary mode detector senses

IN

For more information www.linear.com/LT8302

8302fb

9

LT8302

operaTion

Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit subharmonic oscillation.

Discontinuous Conduction Mode Operation

As the load gets lighter, boundary conduction mode in
creases the
peak
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8302 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 380kHz. Once the
switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates
in discontinuous conduction mode.

switching frequency and decreases the switch

current at the same ratio. Running at a higher switching

-

-

Low Ripple Burst Mode Operation

Unlike traditional flyback converters, the LT8302 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch cur
rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.

As the load gets very light
the

switching frequency while keeping the minimum switch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sample-and­hold error amplifier. Meanwhile, the part switches between
sleep mode and active mode, thereby reducing the effec
tive quiescent current to improve light load efficiency. In
this condition, the LT8302 runs in low ripple Burst Mode
operation. The typical 12kHz minimum switching frequency
determines how often the output voltage is sampled and
also the minimum load requirement.

, the LT8302 starts to fold back

-

-

10

8302fb

For more information www.linear.com/LT8302

applicaTions inForMaTion

( )

LT8302

Output Voltage

The R

and R

FB

resistors as depicted in the Block Diagram

REF

are external resistors used to program the output voltage.
The LT8302 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.

Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the V

supply. The

IN

amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and V

V

FLBK

= (V

OUT

+ VF + I

supply, is given as:

IN

• ESR) • N

SEC

PS

VF = Output diode forward voltage
I

= Transformer secondary current

SEC

ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary

turns ratio

The flyback voltage is then converted to a current, I
by the R
(M2 and M3). This current, I
R

REF

resistor and the flyback pulse sense circuit

FB

, also flows through the

RFB

resistor to generate a ground-referred voltage. The

RFB

resulting voltage feeds to the inverting input of the sample­and-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (I

• ESR) term in the V

SEC

equation can be

FLBK

assumed to be zero.
The internal reference voltage, V

, 1.00V, feeds to the

REF

noninverting input of the sample-and-hold error ampli­fier. The
voltage
reference voltage V
V

FLBK

V

relatively high gain in the overall loop causes the

at the R

and V

REF

⎛
⎜

⎝

V

⎞

V

FLBK

⎟

R

⎠

FB

= V

FLBK

= Internal reference voltage 1.00V

REF

pin to be nearly equal to the internal

REF

. The resulting relationship between

REF

can be expressed as:

•R

REF

REF

•

⎛
⎜

⎝

= V

R

R

FB

REF

REF

or

⎞
⎟

⎠

Combination with the previous V
equation for V

, in terms of the RFB and R

OUT

equation yields an

FLBK

resistors,

REF

transformer turns ratio, and diode forward voltage:

⎛

= V

OUT

REF

•

⎜

⎝

V

⎛

R

R

FB

REF

⎞

•

⎟

⎠

⎞

1

– V

⎜

N

⎝

PS

F

⎟

⎠

Output Temperature Compensation

The first term in the V
ture dependence,

equation does not have tempera-

OUT

but the output diode forward voltage, VF,
has a significant negative temperature coefficient (–1mV/°C
to –2mV/°C). Such a negative temperature coefficient pro
duces approximately 200

mV to 300mV voltage variation

on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the

output diode temperature coefficient has a negligible ef
fect on the output voltage regulation. For lower voltage
outputs,

such as 3.3V and 5V, however, the output diode
temperature coefficient does count for an extra 2% to 5%
output voltage regulation.

,

The LT8302 junction temperature usually tracks the output
diode junction temperature to the first order. To compensate
the negative temperature coefficient of the output diode,
a resistor, R

, connected between the TC and R

TC

generates a proportional-to-absolute-temperature (PTAT)
current. The PTAT current is zero at 25°C, flows into the

pin at hot temperature, and flows out of the R

R

REF

at cold temperature. With the R

resistor in place, the

TC

output voltage equation is revised as follows:

V

OUT

= V

REF

T –TO

( )

R

FB

•

R

REF

R

•

R

FB

TC

1

•

•

N

PS

N

1

PS

– VFTO

( )

– VF/ T

( )

– VTC/ T

( )

• T–TO

TO=Room temperature 25°°C

VF/ T

( )

= Output diode forward voltage

temperature coefficient

VTC/ T

= 3.35mV/ C

pins

REF

pin

REF

( )

-

-

•

For more information www.linear.com/LT8302

8302fb

11

LT8302

( )

( )

REF

V

V

T1

( )

– V

T2

( )

T1– T2

( )

applicaTions inForMaTion

To cancel the output diode temperature coefficient, the
following two equations should be satisfied:

V

OUT

VTC/ T

( )

= V

REF

•

R

FB

•

R

TC

Selecting Actual R

R

FB

R

REF

•

, RFB, RTC Resistor Values

REF

•

N

1

N

PS

1

– VFTO

PS

= – VF/ T

( )

( )

The LT8302 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evaluation
of the R

and RTC resistor values. Therefore, a simple

FB

2-step sequential process is recommended for selecting
resistor values.

Rearrangement of the expression for V
sections yields the starting value for R

R

RFB=

V

REF

= Output voltage

OUT

•NPS• V

V

OUT

+ VFTO

in the previous

OUT

:

FB

VF (TO) = Output diode forward voltage at 25°C = ~0.3V
NPS = Transformer effective primary-to-secondary

turns ratio

The equation shows that the R
dent of the R
between the TC and R

resistor value. Any RTC resistor connected

TC

pins has no effect on the output

REF

resistor value is indepen-

FB

voltage setting at 25°C because the TC pin voltage is equal
to the R

The R

regulation voltage at 25°C.

REF

resistor value should be approximately 10k

REF

because the LT8302 is trimmed and specified using this
value. If the R

resistor value varies considerably from

REF

10k, additional errors will result. However, a variation in

up to 10% is acceptable. This yields a bit of freedom

R

REF

in selecting standard 1% resistor values to yield nominal
R

FB/RREF

ratios.

First, build and power up the application with the starting

, RFB values (no RTC resistor yet) and other compo-

R

REF

nents connected,
age, V

OUT(MEAS)

R

FB(NEW)

and measure the regulated output volt-

. The new RFB value can be adjusted to:

=

OUT

V

OUT(MEAS)

•R

FB

Second, with a new RFB resistor value selected, the output
diode temperature coefficient in the application can be
tested to determine the R
resistor, the V

should be measured over temperature

OUT

value. Still without the RTC

TC

at a desired target output load. It is very important for
this evaluation that uniform temperature be applied to
both the output diode and the LT8302. If freeze spray or
a heat gun is used, there can be a significant mismatch
in temperature between the two devices that causes sig
nificant error.

Attempting to extrapolate the data from a

-

diode data sheet is another option if there is no method
to apply uniform heating or cooling such as an oven. With
at least two data points spreading across the operating
temperature range, the output diode temperature coef

-

ficient can be determined by:

– δVF/δT

( )

OUT

=

OUT

Using the measured output diode temperature coefficient,
an exact R

value can be selected with the following

TC

equation:

⎛

RTC=

δV

– δVF/δT

Once the R

( )

REF

tion accuracy

/δT

TC

•

, RFB, and RTC values are selected, the regula-

from board to board for a given application

⎞

R

FB

⎜

⎟

N

⎝

⎠

PS

will be very consistent, typically under ±5% when includ­ing device
(assuming

variation of all the components in the system

resistor tolerances and transformer windings
matching within ±1%). However, if the transformer or
the output diode is changed, or the layout is dramatically
altered, there may be some change in V

OUT

.

12

8302fb

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applicaTions inForMaTion

LT8302

Output Power

A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a nonisolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In ad

­dition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.

The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,

20

MAXIMUM

OUTPUT POWER

15

10

N = 6:1

N = 4:1

N = 3:1

12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 50V dur
ing the
inductance

switch-off time. 15V of margin is left for leakage

voltage spike. To achieve this power level at

-

a given input, a winding ratio value must be calculated
to stress the switch to

50V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.

One design example would be a 5V output converter with
a minimum input voltage of 8V and a maximum input volt

­age of 32V. A three-to-one winding ratio fits this design
example perfectly and outputs equal to 15.3W at 32V but
lowers to 7.7W at 8V.

20

15

10

MAXIMUM

OUTPUT POWER

N = 3:2

N = 2:1

N = 1:1

OUTPUT POWER (W)

5

0

0

10

20

INPUT VOLTAGE (V)

N = 2:1

30

8302 F01

Figure 1. Output Power for 3.3V Output

20

15

10

OUTPUT POWER (W)

OUTPUT POWER

5

0

0

MAXIMUM

10

INPUT VOLTAGE (V)

N = 4:1

N = 3:1

N = 2:1

N = 1:1

20

30

8302 F02

Figure 2. Output Power for 5V Output

OUTPUT POWER (W)

5

40

0

0

10

20

INPUT VOLTAGE (V)

N = 1:2

30

40

8302 F03

Figure 3. Output Power for 12V Output

20

15

10

OUTPUT POWER (W)

40

OUTPUT POWER

5

0

0

MAXIMUM

10

INPUT VOLTAGE (V)

20

N = 1:1

N = 1:2

N = 1:3

N = 2:3

30

40

8302 F04

Figure 4. Output Power for 24V Output

8302fb

For more information www.linear.com/LT8302

13

LT8302

( )

( )

t

• V

applicaTions inForMaTion

The equations below calculate output power:
P

= η • VIN • D • I

OUT

SW(MAX)

• 0.5

η = Efficiency = ~85%

+ V

D= Duty Cycle=

I

SW(MAX)

= Maximum switch current limit = 3.6A (MIN)

V

OUT

V

+ V

( )

OUT

F

•NPS+ V

F

•N

PS

IN

Primary Inductance Requirement

The LT8302 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the second
ary winding
350ns.

needs to conduct current for a minimum of

The following equation gives the minimum value

-

for primary-side magnetizing inductance:

L

t
I

≥

PRI

OFF(MIN)

SW(MIN)

I

SW(MIN)

• V

t

OFF(MIN)•NPS

= Minimum switch-off time = 350ns (TYP)

= Minimum switch current limit = 0.87A (TYP)

OUT

+ V

F

In addition to the primary inductance requirement for
the minimum switch-off time, the LT8302 has minimum
switch-on time that prevents the chip from turning on

the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank

­ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:

t

L

≥

PRI

ON(MIN)

ON(MIN)

= Minimum switch-on time = 160ns (TYP)

I

SW(MIN)

IN(MAX)

In general, choose a transformer with its primary mag­netizing inductance

about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.

Selecting a Transformer

Transformer specification and design is perhaps the most
critical part of successfully applying the LT8302. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.

Linear Technology has worked with several leading mag
netic component

manufacturers to produce pre-designed

-

flyback transformers for use with the LT8302. Table 1
shows the details of these transformers.

Table 1. Predesigned Transformers–Typical Specifications

TRANSFORMER
PART NUMBER

750311625 17.75 × 13.46 × 12.70 9 0.35 4:1 43 6 Würth Elektronik 8 to 32 3.3 2.1
750311564 17.75 × 13.46 × 12.70 9 0.12 3:1 36 7 Würth Elektronik 8 to 32 5 1.5
750313441 15.24 × 13.34 x 11.43 9 0.6 2:1 75 18 Würth Elektronik 8 to 32 5 1.3
750311624 17.75 × 13.46 × 12.70 9 0.18 3:2 34 21 Würth Elektronik 8 to 32 8 0.9
750313443 15.24 × 13.34 × 11.43 9 0.3 1:1:1 85 100 Würth Elektronik 8 to 36 ±12 0.3
750313445 15.24 × 13.34 × 11.43 9 0.25 1:2 85 190 Würth Elektronik 8 to 36 24 0.3
750313457 15.24 × 13.34 × 11.43 9 0.25 1:4 85 770 Würth Elektronik 8 to 36 48 0.15
750313460 15.24 × 13.34 × 11.43 12 0.7 4:1 85 11 Würth Elektronik 4 to 18 5 0.9
750311342 15.24 × 13.34 × 11.43 15 0.44 2:1 85 22 Würth Elektronik 4 to 18 12 0.4
750313439 15.24 × 13.34 × 11.43 12 0.6 2:1 115 28 Würth Elektronik 18 to 42 3.3 2.1
750313442 15.24 × 13.34 × 11.43 12 0.75 3:2 150 53 Würth Elektronik 18 to 42 5 1.6

14

DIMENSIONS

(W × L × H) (mm)

L
(µH)

L

PRI

LKG

(µH) N

For more information www.linear.com/LT8302

P:NS

R

PRI

(mΩ)

R

SEC

(mΩ) VENDOR

TARGET APPLICATION

V

(V) V

IN

OUT

(V) I

OUT

(A)

8302fb

applicaTions inForMaTion

65V – V

– V

OUT

F

LT8302

Turns Ratio

Note that when choosing an R

FB/RREF

resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
3:1, 2:1, 1:1, etc., provides more freedom in settling total
turns and mutual inductance.

Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V
or 5V), a N:1 turns ratio can be used with multiple pri

­mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (V

LEAKAGE

) on top of
this reflected voltage. This total quantity needs to remain
below the 65V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,

, for a given application. Choose a turns ratio low

N

PS

enough to ensure

NPS<

IN(MAX)

V

LEAKAGE

+ V

For larger N:1 values, choose a transformer with a larger
physical size to deliver additional current. In addition,
choose a large enough inductance value to ensure that
the switch-off time is long enough to accurately sample
the output voltage.

For lower output power levels, choose a 1:1 or 1:N trans

­former for the absolute smallest transformer size. A 1:N
transformer will minimize the magnetizing inductance
(and minimize size), but will also limit the available output
power. A higher 1:N turns ratio makes it possible to have
very high output voltages without exceeding the breakdown
voltage of the internal power switch.

The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci

-

fies turns ratio accuracy within ±1%.

Saturation

The

current in the transformer windings should not exceed

Current

its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8302, the saturation
current should always be specified by the transformer
manufacturers.

Winding Resistance

Resistance in either the primary or secondary windings
will reduce

overall power

efficiency. Good output voltage
regulation will be maintained independent of winding re­sistance due

to the boundary/discontinuous conduction

mode operation of the LT8302.

Leakage Inductance and Snubbers

T

ransformer leakage inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored en

­ergy must be dissipated. It is very important to minimize
transformer leakage inductance.

designing an application, adequate margin should

When
be kept for the worst-case leakage voltage spikes even

IN

-

under overload conditions. In most cases shown in Fig
ure5, the

reflected output voltage on the primary plus V
should be kept below 50V. This leaves at least 15V margin
for the leakage spike across line and load conditions. A
larger voltage margin will be required for poorly wound
transformers or for excessive leakage inductance.

For more information www.linear.com/LT8302

8302fb

15

LT8302

C

1

applicaTions inForMaTion

<65V

V

t

OFF

> 350ns

TIME

LEAKAGE

8302 F05

<50V

V

SW

tSP < 250ns

then add capacitance until the period of the ringing is 1.5
to 2 times longer. The change in period determines the
value of the parasitic capacitance, from which the para

­sitic inductance can be also determined from the initial
period. Once the value of the SW node capacitance and
inductance is known, a series resistor can be added to
the snubber capacitance to dissipate power and critically
damp the ringing. The equation for deriving the optimal
series resistance using the observed periods ( t
t

PERIOD(SNUBBED)

) and snubber capacitance (C

SNUBBER

PERIOD

and

) is:

Figure 5. Maximum Voltages for SW Pin Flyback Waveform

In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig

­ger boundary mode detector, the LT8302 internally blanks
the boundary mode detector for approximately 250ns.
Any remaining voltage ringing after 250ns may turn the
power switch back on again before the secondary current
falls to zero. In this case, the LT8302 enters continuous
conduction mode. So the leakage inductance spike ringing
should be limited to less than 250ns.

To clamp and damp the leakage voltage spikes, a
(RC + DZ) snubber

circuit in Figure6 is recommended.
The RC (resistor-capacitor) snubber quickly damps the
voltage spike ringing and provides great load regulation
and EMI performance. And the DZ (diode-Zener) ensures
well defined and consistent clamping voltage to protect
SW pin from exceeding its 65V absolute maximum rating.

L

ℓ

CZ

R

D

Figure 6. (RC + DZ) Snubber Circuit

•

•

8302 F06

The recommended approach for designing an RC snub­ber is to measure the period of the ringing on the SW pin

the power switch turns off without the snubber and

when

SNUBBER

PERIOD(SNUBBED)

t

PERIOD

2

2

• 4π

L

PAR

=

C

PAR

2

⎞

–

⎟

⎠

C

PAR

L

PAR

R

SNUBBER

=

=

t

⎛
⎜

⎝

t

PERIOD

C

PAR

Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
needs to be sized for thermal dissipation. A 470pF capaci
tor in series with a 39Ω resistor is a good starting point.

the DZ snubber, proper care should be taken when

For
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leak
age inductance

spike. Choose a diode that has a reverse­voltage rating higher than the maximum SW pin voltage.
The Zener diode breakdown voltage should be chosen to
balance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown
with 5V margin. Use the following equation to make the
proper choice:

V

ZENNER(MAX)

≤ 60V – V

IN(MAX)

For an application with a maximum input voltage of 32V,
choose a 24V Zener diode, the V

ZENER(MAX)

of which is
around 26V and below the 28V maximum. The power loss
in the DZ snubber determines the power rating of the Zener
diode. A 1.5W Zener diode is typically recommended.

-

-

16

8302fb

For more information www.linear.com/LT8302

applicaTions inForMaTion

( )

1

R2

L

•I

• f

LT8302

Undervoltage Lockout (UVLO)

A resistive divider from V

to the EN/UVLO pin imple-

IN

ments undervoltage lockout (UVLO). The EN/UVLO enable

threshold is set at 1.214V with 14mV hysteresis. In

falling
addition, the EN/UVLO pin sinks 2.5µA when the voltage
on the pin is below 1.214V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:

V

V

IN(UVLO+)

IN(UVLO–)

1.228V • R1+R2

=

R2

1.214V • R1+R2

=

( )

+ 2.5µA •R

Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8302 in shutdown with quiescent current less than 2µA.

V

IN

R1

EN/UVLO

LT8302

GND

Figure 7. Undervoltage Lockout (UVLO)

R2

RUN/STOP
CONTROL
(OPTIONAL)

8302 F07

Minimum Load Requirement

The LT8302 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8302 has to turn on and off for a
minimum amount of time and with a minimum frequency.
The LT8302 delivers a minimum amount of energy even

during light load conditions to ensure accurate output volt

.

age information

The minimum energy delivery creates a

-

minimum load requirement, which can be approximately
estimated as:

I

LOAD(MIN)

L

PRI

I

SW(MIN)

f

MIN

P2RI

=

SW(MIN)

2• V

= Transformer primary inductance

= Minimum switch current limit = 0.96A (MAX)

= Minimum switching frequency = 12.7kHz (MAX)

MIN

OUT

The LT8302 typically needs less than 0.5% of its full output
power as minimum load. Alternatively, a Zener diode with
its breakdown of 10% higher than the output voltage can
serve as a minimum load if pre-loading is not acceptable.
For a 5V output, use a 5.6V Zener with cathode connected
to the output.

Output Short Protection

When the output is heavily overloaded or shorted to ground,
the reflected SW pin waveform rings longer than the in
ternal blanking

time. After the 350ns minimum switch-off

-

time, the excessive ringing falsely triggers the boundary
mode detector and turns the power switch back on again
before the secondary current falls to zero. Under this
condition, the LT8302 runs into continuous conduction
mode at 380kHz maximum switching frequency. If the
sampled R

voltage is still less than 0.6V after 11ms

REF

(typ) soft-start timer, the LT8302 initiates a new soft-start
cycle. If the sampled R

voltage is larger than 0.6V after

REF

11ms, the switch current may run away and exceed the

4.5A maximum current limit. Once the

switch current

hits

7.2A over current limit, the LT8302 also initiates a new
soft-start cycle. Under either condition, the new soft-start
cycle throttles back both the switch current limit and switch
frequency. The output short-circuit protection prevents the
switch current from running away and limits the average
output diode current.

For more information www.linear.com/LT8302

8302fb

17

LT8302

65V – V

– V

65V – 32V –15V

5V + 0.3V

( )

( )

0.87A

1

1

IN

applicaTions inForMaTion

Design Example

Use the following design example as a guide to designing
applications for the LT8302. The design example involves
designing a 5V output with a 1.5A load current and an
input range from 8V to 32V.

V

IN(MIN)

V

OUT

= 8V, V

= 5V, I

OUT

IN(NOM)

= 1.5A

= 12V, V

IN(MAX)

= 32V,

Step 1: Select the transformer turns ratio.

NPS<

V

LEAKAGE

= Output diode forward voltage = ~0.3V

V

F

IN(MAX)

V

OUT

= Margin for transformer leakage spike = 15V

+ V

LEAKAGE

F

Example:

NPS<

= 3.4

The choice of transformer turns ratio is critical in determin­ing output current

capability of the converter. Table2 shows
the switch voltage stress and output current capability at
different transformer turns ratio.

Table 2. Switch Voltage Stress and Output Current Capability vs
Turns Ratio

SW(MAX)
IN(MAX)

at

(V)

I

OUT(MAX)

V

V

NPS

1:1 37.3 0.92 14-40
2:1 42.6 1.31 25-57
3:1 47.9 1.53 33-67

V

at

(A) DUTY CYCLE (%)

IN(MIN)

Clearly, only NPS = 3 can meet the 1.5A output current
requirement, so N

= 3 is chosen as the turns ratio in

PS

this example.

Step 2: Determine the primary inductance.

Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:

L

L

t
t
I

≥

PRI

≥

PRI

OFF(MIN)
ON(MIN)
SW(MIN)

t

OFF(MIN)

t

ON(MIN)

I

SW(MIN)

= 350ns

= 160ns

= 0.87A

• NPS• V

I

SW(MIN)

• V

IN(MAX)

OUT

+ V

F

Example:

L

L

350ns • 3 • 5V + 0.3V

≥

PRI

PRI

≥

160ns • 32V

0.87A

= 6.4µH

= 5.9µH

Most transformers specify primary inductance with a toler­ance of ±20%.

With other component tolerance considered,
choose a transformer with its primary inductance 40% to
60% larger than the minimum values calculated above.

= 9µH is then chosen in this example.

L

PRI

Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:

fSW=

ISW=

tON+ t

V

OUT

η • V

• I

OFF

OUT

• D

=

• 2

L

• I

PRI

SW

V

IN

+

NPS• V

L

• I

PRI

SW

+ V

( )

OUT

F

18

8302fb

For more information www.linear.com/LT8302

applicaTions inForMaTion

5V + 0.3V

( )

• 3

V

PS

32V

3

L

• I

2

2

2 • 5V • 0.1V

LT8302

Example:

D =

5V + 0.3V

( )

=

I

SW

fSW= 277kHz

• 3 + 12V

5V •1.5A • 2

0.8 • 12V • 0.57

= 0.57

The transformer also needs to be rated for the correct
saturation current level across line and load conditions.
A saturation current rating larger than 7A is necessary
to work with the LT8302. The 750311564 from Würth is
chosen as the flyback transformer.

Step 3: Choose the output diode.

Tw o main criteria for choosing the output diode include
forward current rating and reverse-voltage rating. The
maximum load requirement is a good first-order guess
at the average current requirement for the output diode.
Under output short-circuit condition, the output diode
needs to conduct much higher current. Therefore, a con
servative metric

is 60% of the maximum switch current

limit multiplied by the turns ratio:
I

DIODE(MAX)

= 0.6 • I

SW(MAX)

• N

PS

Example:
I

DIODE(MAX)

= 8.1A

Next calculate reverse voltage requirement using maxi­mum V

IN

V

REVERSE

:

= V

OUT

+

IN(MAX)

N

Example:

V

REVERSE

= 5V +

= 15.7V

The PDS835L (8A, 35V diode) from Diodes Inc. is chosen.

Step 4: Choose the output capacitor.

The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the following equation

to calculate the output capacitance:

PRI

OUT

SW

• ΔV

OUT

C

=

OUT

2 • V

Example:
Design for output voltage ripple less than ±1% of V

i.e., 100mV.

9µH • 4.5A

C

OUT

=

( )

= 182µF

Remember ceramic capacitors lose capacitance with ap­plied voltage. The capacitance can drop to 40% of quoted
capacitance

at the maximum voltage rating. So a 220µF,

6.3V rating ceramic capacitor is chosen.

Step 5: Design snubber circuit.

The snubber circuit protects the power switch from leak
age inductance

voltage spike. A (RC + DZ) snubber is

recommended for this application. A 470pF capacitor in

-

series with a 39Ω resistor is chosen as the RC snubber.
The maximum Zener breakdown voltage is set according

to the maximum V
V

ZENNER(MAX)

:

IN

≤ 60V – V

IN(MAX)

Example:
V

ZENNER(MAX)

≤ 60V – 32V = 28V

A 24V Zener with a maximum of 26V will provide optimal
protection and minimize power loss. So a 24V, 1.5W Zener
from Central Semiconductor (CMZ5934B) is chosen.

Choose a diode that is fast and has sufficient reverse
voltage breakdown:

V
V

REVERSE

SW(MAX)

> V

= V

SW(MAX)

IN(MAX)

+ V

ZENNER(MAX)

Example:
V

REVERSE

> 60V

A 100V, 1A diode from Diodes Inc. (DFLS1100) is chosen.

OUT

,

-

For more information www.linear.com/LT8302

8302fb

19

LT8302

( )

( )

REF

10k • 3 • 5V + 0.3V

( )

1.00V

V

5V

5.14V

V

T1

( )

– V

T2

( )

5.189V – 5.041V

1.228V • R1+ R2

( )

R2

1

2 • 5V

applicaTions inForMaTion

Step 6: Select the R

Use the following equation to calculate the starting values

REF

RFB=

R

RFB=

R

FB(NEW)

RFB=

and RFB:

R

• NPS• V

REF

= 10k

FB

measure the regulated output voltage. Adjust

=

V

OUT(MEASURED)

• 158k = 154k

for R

Example:

For 1% standard values, a 158k resistor is chosen.

Step 7: Adjust R

Build and power up the application with application com­ponents and
RFB resistor based on the measured output voltage:

Example:

Step 8: Select RTC resistor based on output voltage
temperature variation.

Measure output voltage in a controlled temperature envi
ronment like an oven to determine the output temperature
coefficient. Measure output voltage at a consistent load
current and input voltage, across the operating tempera
ture range.

and RFB resistors.

REF

+ VFTO

OUT

V

REF

= 159k

resistor based on output voltage.

OUT

• R

FB

-

-

Example:

– δVF/δT

( )

R

=

TC

Step 9: Select the EN/UVLO resistors.

Determine the amount of hysteresis required and calculate
R1 resistor value:

V

IN(HYS)

Example:
Choose 2V of hysteresis, R1 = 806k
Determine the UVLO thresholds and calculate R2 resistor

value:

V

IN(UVLO+)

Example:
Set V
R2 = 232k

V
V

Step 10: Ensure minimum load.

The theoretical minimum load can be approximately
estimated as:

UVLO rising threshold to 7.5V:

IN

IN(UVLO+)
IN(UNLO–)

I

LOAD(MIN)

=

100°C – 0°C

3.35mV/°C

1.48mV/°C

= 2.5µA • R1

=

= 7.5V
= 5.5V

9µH • 0.96A

=

( )

154

⎛

•

⎜

⎝

3

( )

= 1.48mV / °C

⎞

= 115k

⎟

⎠

+ 2.5µA • R

2

• 12.7kHz
=10.5mA

Calculate the temperature coefficient of V

– δVF/δT

( )

RTC=

3.35mV/°C

– δV

20

OUT

=

/δT

( )

F

•

T1– T2

⎛

R

FB

⎜

N

⎝

PS

OUT

⎞
⎟

⎠

:

F

For more information www.linear.com/LT8302

Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con
verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 10mA. In this example, a 500Ω resistor is selected
as the minimum load.

-

8302fb

Typical applicaTions

8302 TA02c

LT8302

V

8V TO 32V

8V to 32VIN/12V

IN

C1
10µF

R1
806k

R2
232k

C2
1µF

EN/UVLO

LT8302

GND

INTV

CC

V

IN

SW

R

R

REF

Isolated Flyback Converter

OUT

C3

Z1

470pF

9µH

R3

D1

39Ω

R4

R6
OPEN

121k

R5
10k

8302 TA02a

FB

TC

D2
T1
1:1

•

9µH

•

D1: DIODES DFLS1100
D2: DIODES PDS540
T1: WURTH 750313443
Z1: CENTRAL CMZ5934B

C4
47µF

+

V

OUT

12V
5mA TO 0.8A (V
5mA TO 1.1A (V

–

V

OUT

Efficiency vs Load Current Load and Line Regulation

95

90

85

80

12.4

12.2

12.0

11.8

= 12V)

IN

= 24V)

IN

V

8V TO 32V

10µF

IN

C1

EFFICIENCY (%)

75

70

65

0

200 400 600 800 1000

R1
806k

R2
232k

C2
1µF

LOAD CURRENT (mA)

8V to 32VIN/3.3V

V

IN

LT8302

CC

SW

R

FB

R

REF

TC

EN/UVLO

GND

INTV

VIN = 12V

= 24V

V

IN

1200

8302 TA02b

Isolated Flyback Converter

OUT

D2

T1

4:1

C3

Z1

R6
105k

470pF

R3

D1

39Ω

R4
140k

R5
10k

8302 TA03

•

9µH

0.56µH

•

D1: DIODES DFLS1100
D2: DIODES PDS1040L
T1: WURTH 750311625
Z1: CENTRAL CMZ5934B

+

V

OUT

3.3V
20mA TO 2.7A (V
20mA TO 3.8A (V

C4
470µF

–

V

OUT

11.6

OUTPUT VOLTAGE (V)

11.4

11.2
0

200 400 600 800 1000 1200

LOAD CURRENT (mA)

= 12V)

IN

= 24V)

IN

OUTPUT VOLTAGE (V)

VIN = 12V

= 24V

V

IN

Output Temperature Variation

3.50
VIN = 12V

= 1A

I

OUT

3.45

3.40

3.35

RTC = 105k

3.30

3.25

3.20

3.15

3.10

–50

RTC = OPEN

–25

0

AMBIENT TEMPERATURE (°C)

50

25

75

100

125

8302 TA03b

150

For more information www.linear.com/LT8302

8302fb

21

LT8302

Typical applicaTions

8V TO 36V

V

8V TO 36V

8V to 36VIN/±12V

V

IN

C1
10µF

R1
806k

R2
232k

C2
1µF

EN/UVLO

LT8302

GND

INTV

CC

V

IN

SW

R

FB

R

REF

TC

8V to 36VIN/24V

IN

C1
10µF

R1
806k

R2
232k

C2
1µF

EN/UVLO

LT8302

GND

INTV

CC

V

IN

SW

R

FB

R

REF

TC

Isolated Flyback Converter

OUT

T1

1:1:1

C3

Z1

470pF

R3

D1

39Ω

R4
121k

9µH

•

9µH

•

•

R5

R6

10k

OPEN

8302 TA04

Isolated Flyback Converter

OUT

C3

Z1

470pF

9µH

R3

D1

39Ω

R4
121k

R5

R6

10k

OPEN

8302 TA05

9µH

D1: DIODES DFLS1100
D2, D3: DIODES PDS360
T1: WURTH 750313443
Z1: CENTRAL CMZ5934B

D2

T1

1:2

•

36µH

•

D1: DIODES DFLS1100
D2: DIODES SBR2U150SA
T1: WURTH 750313445
Z1: CENTRAL CMZ5934B

+

D2

D3

V

OUT1

12V
5mA TO 0.4A (V
5mA TO 0.55A (V

C4
22µF

V

OUT2

V

OUT2

12V
5mA TO 0.4A (V
5mA TO 0.55A (V

C5
22µF

V

OUT2

+

V

OUT

24V

2.5mA TO 0.4A (V

2.5mA TO 0.55A (V

C4
10µF

–

V

OUT

= 12V)

IN

= 24V)

IN

–
+

= 12V)

IN

= 24V)

IN

–

= 12V)

IN

= 24V)

IN

V

IN

8V TO 36V

R1

C1

806k

10µF

R2
232k

C2
1µF

22

8V to 36VIN/48V

V

IN

EN/UVLO

GND

INTV

LT8302

CC

SW

R

FB

R

REF

TC

Isolated Flyback Converter

OUT

C3

Z1

470pF

9µH

R3

D1

39Ω

R4
121k

R5

R6

10k

OPEN

8302 TA06

For more information www.linear.com/LT8302

D2

T1

1:4

•

144µH

•

D1: DIODES DFLS1100
D2: DIODES SBR1U200P1
T1: WURTH 750313457
Z1: CENTRAL CMZ5934B

V
48V

1.2mA TO 0.2A (V

1.2mA TO 0.27A (V

C4

2.2µF

V

OUT

OUT

+

= 12V)

IN

= 24V)

IN

–

8302fb

Typical applicaTions

8302 TA07b

LT8302

V

8V TO 32V

–4V TO –42V

8V to 32VIN/5V

IN

C1
10µF

R1
806k

R2
232k

C2
1µF

EN/UVLO

LT8302

GND

INTV

CC

V

–4V to –42VIN/12V

C1
10µF

C2

V

IN

1µF

Isolated Flyback Converter with LT8309

OUT

T1

3:1

C3

Z1

IN

SW

R

FB

R

REF

R6
OPEN

TC

D1: DIODES DFLS1100
D2: CENTRAL CMMSH1-60
M1: INFINEON BSC059N04LS
T1: WURTH 750311564
Z1: CENTRAL CMZ5934B

OUT

L1

12µH

V

EN/UVLO

INTV

IN

LT8302

CC

GND

SW

R

470pF

R3

D1

39Ω

R4
154k

R5
10k

Buck-Boost Converter

R4
118k

R

FB

REF

R5
10k

8302 TA08a

•

9µH

1µH

•

M1

D1

Z1

D1: DIODES PMEG6030EP
L1: WÜRTH 744770112
Z1: CENTRAL CMHZ5243B

R8

2.1k

D2

C3
47µF

R7
5Ω

V

CC

DRAIN

LT8309

GATE INTV

CC

GND

V

OUT

12V/0.45A (VIN = –5V)
12V/0.8A (V
12V/1.1A (V
12V/1.3A (V

= –12V)

IN

= –24V)

IN

= –42V)

IN

C4
10µF

C5

4.7µF

+

V

OUT

5V/1.1A (VIN = 5V)
5V/2.0A (V
5V/2.9A (V

C4
220µF

–

V

OUT

8302 TA07

Efficiency vs Load Current

= 12V)

IN

= 24V)

IN

95

90

85

80

EFFICIENCY (%)

75

70

65

0

0.5 1.0

Efficiency vs Load Current

95

90

85

80

EFFICIENCY (%)

75

70

65

0

200 400 800600 1000 1200 1400

LOAD CURRENT (mA)

2.0 3.0

1.5 2.5

LOAD CURRENT (A)

VIN = –5V

= –12V

V

IN

= –24V

V

IN

= –42V

V

IN

8302 TA08b

–18V to –42VIN/–12V

R1
806k

C1

R2

10µF

232k

C2

V

–18V TO –42V

IN

1µF

Negative Buck Converter

OUT

V

IN

EN/UVLO

EN/UVLO
INTV

LT8302

CC

SW

R

R

REF

C3
47µF

D1

R4
118k

FB

R5
10k

8302 TA09a

Z1

V

OUT

–12V

L1
12µH

D1: DIODES PMEG6030EP
L1: WÜRTH 744770112
Z1: CENTRAL CMHZ5243B

1.8A

For more information www.linear.com/LT8302

Efficiency vs Load Current

100

95

90

85

EFFICIENCY (%)

80

75

70

0

500 1000 1500 2000

LOAD CURRENT (mA)

VIN = –18V

= –24V

V

IN

= –42V

V

IN

8302 TA09b

8302fb

23

LT8302

S8E Package

package DescripTion

Please refer to http://www.linear.com/product/LT8302#packaging for the most recent package drawings.

8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad

(Reference LTC DWG # 05-08-1857 Rev C)

.189 – .197

.050

(1.27)

BSC

.045 ±.005

(1.143 ±0.127)

(4.801 – 5.004)

8

NOTE 3

.005 (0.13) MAX

7

6

5

.245

(6.22)

MIN

.030 ±.005

(0.76 ±0.127)

TYP

RECOMMENDED SOLDER PAD LAYOUT

.010 – .020

(0.254 – 0.508)

.008 – .010

(0.203 – 0.254)

NOTE:

1. DIMENSIONS IN

2. DRAWING NOT TO SCALE

3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm)

4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542)

5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542)

.118

(2.99)

REF

× 45°

.016 – .050

(0.406 – 1.270)

INCHES

(MILLIMETERS)

.089

(2.26)

REF

0°– 8° TYP

.160 ±.005

(4.06 ±0.127)

(5.791 – 6.197)

.228 – .244

.053 – .069

(1.346 – 1.752)

.014 – .019

(0.355 – 0.483)

TYP

1

(2.997 – 3.550)

2

.118 – .139

4

.150 – .157

(3.810 – 3.988)

NOTE 3

5

0.0 – 0.005

(0.0 – 0.130)

.080 – .099

(2.032 – 2.530)

3

4

.004 – .010

(0.101 – 0.254)

.050

(1.270)

BSC

S8E 1015 REV C

24

8302fb

For more information www.linear.com/LT8302

LT8302

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

A 11/14 Modified I

Modified L
Modified Schematic
Updated Related Parts

B 11/15 Revised Package Drawing 24

and I

Q

PRI

Conditions

HYS

Equation

3
14
23
26

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa­tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

For more information www.linear.com/LT8302

8302fb

25

LT8302

Typical applicaTion

V

4V TO 42V

4V to 42VIN/48V

L1

V

IN

EN/UVLO

LT8302

INTV

CC

22µH

GND

SW

R

R

REF

FB

IN

C1
10µF

C2
1µF

Boost Converter

OUT

D1

R3

R4
464k

R5
10k

Z1

1M

8302 TA10a

V

OUT

48V/1.4A (VIN = 42V)
48V/0.8A (V
48V/0.4A (V
48V/0.15A (V

C3
10µF

D1: DIODES PDS560
L1: WÜRTH 7443551221
Z1: CENTRAL CMHZ5262B

= 24V)

IN

= 12V)

IN

IN

= 5V)

Efficiency vs Load Current

100

95

90

85

EFFICIENCY (%)

80

VIN = 5V

= 12V

V

75

70

0

250 500

LOAD CURRENT (mA)

V
V

IN
IN
IN

= 24V
= 42V

1500750 1000 1250

8302 TA10b

relaTeD parTs

PART NUMBER DESCRIPTION COMMENTS

LT8301 42V

LT8300 100V

LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ V
LT3573/LT3574

LT3575
LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA

LT3748 100V Isolated Flyback Controller 5V ≤ V
LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC V
LT3757A/LT3759

LT3758
LT3957/LT3958 40V/80V Boost/Flyback Converters Monolithic with Integrated 5A/3.3A Switch

Linear Technology Corporation

26

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

Micropower Isolated Flyback Converter with 65V/1.2A

IN

Switch

Micropower Isolated Flyback Converter with

IN

150V/260mA Switch

40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A

40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive

For more information www.linear.com/LT8302

●

www.linear.com/LT8302

Low IQ Monolithic No-Opto Flyback 5-Lead TSOT-23

Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23

≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23

CC

Switch

Switch, MSOP-16(12)

≤ 100V, No-Opto Flyback, MSOP-16(12)

IN

and V

IN

Limited Only by External Components

OUT

LT 1115 REV B • PRINTED IN USA

LINEAR TECHNOLOGY CORPORATION 2013

8302fb