LINEAR TECHNOLOGY LT3757 Technical data

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LINEAR TECHNOLOGY

JUNE 2009 VOLUME XIX NUMBER 2

SENSE

LT3757

10V TO 30V

4.7µF 50V X5R

48V 1A

0.01Ω

42.2k

GATE

FBX

GND

INTV

SHDN/UVLO

SYNC

RT SS

200k

43.2k

0.1µF

8.45k

10nF

10µH

IHLP-5050EZ-01

MBRM360

590k 1%

M1 Si7850

20.0k 1%

4.7µF 10V X5R

4.7µF 50V X5R s2

IN THIS ISSUE…

COVER ARTICLE Two New Controllers for Boost,

Flyback, SEPIC and Inverting DC/DC Converters Accept Inputs up to 100V

...........................................................1

Wei Gu

Linear in the News… ...........................2

DESIGN FEATURES Charge Li-Ion Batteries Directly

from High Voltage Automotive and Industrial Supplies Using Standalone Charger in a

3mm × 3mm DFN .................................5

Jay Celani

Power Management IC Combines USB On-The-Go and USB Charging

in Compact Easy-to-Use Solution .........8

George H. Barbehenn and Sauparna Das

Power Management IC with Pushbutton Control Generates Six Voltage Rails from USB or 2 AA Cells Via Low Loss

PowerPath™ Topology .......................12

John Canﬁeld

Improve Hot Swap Performance and Save Design Time with Hot Swap™ Controller that Integrates

2A MOSFET and Sense Resistor .........16

David Soo

Compact No R Feature Fast Transient Response and Regulate to Low V Wide Ranging V

.........................................................18

Terry J. Groom

™ Controllers

SENSE

OUT

from

Two New Controllers for Boost, Flyback, SEPIC and Inverting DC/DC Converters Accept Inputs up to 100V

Introduction

Two new versatile DC/DC controller ICs, the LT®3757 and LT3758, are optimized for boost, ﬂyback, SEPIC and inverting converter applications. The LT3757 operates over an input range of 2.9V to 40V, suitable for applications from single-cell lithium-ion battery portable electronics up to high voltage automotive and industrial power supplies. The LT3758 extends the input voltage to 100V, providing ﬂexible, high performance operation in high voltage, high power telecommunications equipment. Both ICs exhibit low shutdown quiescent cur -

rent of 1µA, making them an ideal ﬁt for battery-operated systems.

Both integrate a high voltage, low dropout linear (LDO) regulator. Thanks to a novel FBX pin architecture, the LT3757 and LT3758 can be connected directly to a divider from either the positive output or the negative output to ground. They also pack many popular features such as soft-start, input undervoltage lockout, adjustable frequency and synchronization in a small 10-lead MSOP package or a 3mm × 3mm QFN package.

by Wei Gu

continued on page 3

Space-Saving, Dual Output DC/DC Converter Yields Plus/Minus Voltage Outputs

with (Optional) I2C Programming .......22

Mathew Wich

DESIGN IDEAS

....................................................26–36

(complete list on page 26)

New Device Cameos ...........................37

Design Tools ......................................39

Sales Ofﬁces .....................................40

, Li nea r E xpr ess , L ine ar Te chn olo gy, L T, LTC , L T M, Bo de CAD , B urs t M ode , F ilt erC AD, L T spi ce, OPTI-LOOP, Over-The-Top, PolyPhase, SwitcherCAD, µModule and the Linear logo are registered trademarks of Linear Technology Corporation. Adaptive Power, Bat-Track, C-Load, DirectSense, Easy Drive, FilterView, Hot Swap, LTBiCMOS, LTCMOS, LinearView, Micropower SwitcherCAD, Multimode Dimming, No Latency ∆Σ, No Latency Delta-Sigma, No R Operational Filter, PanelProtect, PowerPath, PowerSOT, SafeSlot, SmartStart, SNEAK-A-BIT, SoftSpan, Stage Shedding, Super Burst, ThinSOT, TimerBlox, Triple Mode, True Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

Figure 1. A 10V–30V input, 48V at 1A output boost converter

SENSE

L LINEAR IN THE NEWS

Linear in the News…

EDN Innovation Award Winners

EDN magazine on March 30 announced the winners of their annual Innovation Awards. Linear Technology’s LTM®4606 Ultralow EMI, 6A DC/DC µModule® Regulator was selected as the winner in the Power ICs: Modules category. This innovative device signiﬁcantly reduces switching regulator noise by attenuating conducted and radiated energy at the source. The µModule device is a complete DC/DC system-in-a-package, including the inductor, controller IC, MOSFETs, input and output capacitors and the compensation circuitry—all in a surface mount plastic package in an IC form factor.

The other Innovation Award winner, in the category Best Contributed Article, was Jim Williams for his article, “High Voltage, Low-Noise DC/DC Converters,” which you can read at www.edn.com/jimwilliams.

In addition to these winners, two other Linear Technology products were ﬁnalists for Innovation Awards:

LTC®6802 Battery Stack Monitor in the Battery ICs

DESIGN FEATURES L

7.5V TO 32V (40 MAX)

CLP

RNG/SS

BOOST

SENSE

BAT

NTC

TIMER

CMPSH1-4

CMSH3-40MA

1µF

10µF

6.8µH

0.05Ω

10µF

LT3650-4.2

Li-Ion CELL

SHDN

CHRG

FAULT

GND

Charge Li-Ion Batteries Directly from High Voltage Automotive and Industrial Supplies Using Standalone Charger in a 3mm × 3mm DFN

Introduction

Growth of the portable electronics market is in no small part due to the continued evolution of battery capacities. For many portable devices, rechargeable Li-Ion batteries are the power source of choice because of their high energy density, light weight, low internal resistance, and fast charge times. Charging these batteries safely and efficiently, however, requires a relatively sophisticated charging system.

One additional problem faced by battery charger designers is how to deal with relatively high voltage sources, such as those found in industrial and automotive applications. In these environments, system supply voltages exceed the input ranges of most charger ICs, so a DC/DC step-down converter is required to provide a local low voltage supply for the charger IC. The LT3650 standalone monolithic switching battery charger does not need this additional DC/DC converter. It directly accepts input voltages up to 40V and provides charge currents as high as 2A. It also includes a wealth of advanced features that assure safe battery charging and expand its applicability.

The LT3650 includes features that minimize the overall solution size, requiring only a few external components to complete a charger circuit. A fast 1MHz switching frequency allows the use of small inductors, and the IC is housed inside a tiny 3mm × 3mm DFN12-pin package. The IC has builtin reverse current protection, which blocks current ﬂow from the battery back to the input supply if that supply is disabled or discharged to ground, so a single-cell LT3650 charger does not require an external blocking diode on the input supply.

A Charger Designed for Lithium-Ion Batteries

A Li-Ion battery requires constantcurrent/constant-voltage (CC/CV) charging system. A Li-Ion battery is initially charged with a constant current, generally between 0.5C and 1C, where C is the battery capacity in ampere-hours. As it is charged, the battery voltage increases until it approaches the full-charge ﬂoat voltage. The charger then transitions into constant voltage operation as the charge current is slowly reduced. The LT3650-4.1 and LT3650-4.2 are designed to charge single-cell Li-Ion

by Jay Celani

batteries to ﬂoat voltages of 4.1V and

4.2V, respectively. The LT3650-8.2 and LT3650-8.4 are designed to charge 2-cell battery stacks to ﬂoat voltages of 8.2V and 8.4V.

Once the charge current falls below one tenth of the maximum constant charge current, or 0.1C, the battery is considered charged and the charging cycle is terminated. The charger must be completely disabled after terminating charging, since indeﬁnite trickle charging of Li-Ion cells, even at miniscule currents, can cause battery damage and compromise battery stability. A charger can top-off a battery by continuing to operate as the current falls lower than the 0.1C charge current threshold to make full use of battery capacity, but in such cases a backup timer is used to disable the charger after a controlled period of time. Most Li-Ion batteries charge fully in three hours.

The LT3650 addresses all of the charging requirements for a Li-Ion battery. The IC provides a CC/CV charging characteristic, transitioning automatically as the requirements of the battery change during a charging cycle. During constant-current operation, the maximum charge current

Figure 1. An LT3650 standalone battery charger is small and efﬁcient.

Linear Technology Magazine • June 2009

Figure 2. A single-cell 2A Li-Ion battery charger conﬁgured for C/10 charge termination

L DESIGN FEATURES

BAT

(A)

EFFICIENCY (%)

100

0.5

1.5

VIN = 12V

VIN = 20V

BAT

(V)

CHARGE CURRENT (A)

1.0

1.2

1.4

1.6

1.8

2.0

0.6

0.8

0.2

0.4

3.0 3.22.6 2.8

3.4

3.8 4.0 4.23.6

FAULT

CHG

10k

LT3650

Figure 3. Battery charge current vs BAT pin voltage for the charger shown in Figure 2

provided to the battery is programmable via a sense resistor, up to a maximum of 2A. Maximum charge current can also be adjusted using the RNG/SS pin. The charger transitions to constant-voltage mode operation as the battery approaches the full-charge ﬂoat voltage. Power is transferred through an internal NPN switch element, driven by a boosted drive to maximize efﬁciency. A precision SHDN pin threshold allows incorporation of accurate UVLO functions using a simple resistor divider.

Charge Cycle Termination and Automatic Restart

A LT3650 charger can be conﬁgured to terminate a battery charge cycle using one of two methods: it can use low charge current (C/10) detection, enabled by connecting the TIMER pin to ground, or terminate based on the onboard safety timer, enabled by connecting a capacitor to the TIMER pin. After termination, a new charge cycle automatically restarts should the battery voltage fall to 97.5% of the ﬂoat voltage.

is selected, the LT3650 terminates a charging cycle when the output current has dropped to 1/10 of the

When C/10 ter mination mode

Figure 5. Visual charger status is easily implemented using LEDs

A Basic Charger

Figure 2 shows a basic 2A single-cell Li-Ion battery charger that operates from a 7.5V to 32V input. Charging is suspended if the input supply voltage exceeds 32V, but the IC can withstand input voltages as high as 40V without damage. The 2A maximum charge current corresponds to 100mV across the 0.05Ω external sense resistor. This basic design does not take advantage of the status pins, battery temperature

Figure 4. Power conversion efﬁciency vs charger output current (I charger shown in Figure 2

) for the battery

BAT

programmed maximum. In a 2A charger, for example, the charge cycle terminates when the battery charge current falls to 200mA.

Timer termination, or top-of f charging, is enabled when a capacitor is connected to the TIMER pin. The value of the capacitor sets the safety timer duration—0.68µF corresponds to a 3-hour cycle time. When timer termination is implemented, the charger continues to operate in constant-voltage mode when charge currents fall below C/10, allowing additional low current charging to occur until the timer cycle has elapsed, thus maximizing use of the battery capacity. During top-off charging, the CHRG and FAULT status pins communicate “charge complete.” At the end of the timer cycle, the LT3650 terminates the charging cycle.

After charge cycle termination, the LT3650 enters standby mode where the IC draws 85µA from the input supply and less than 1µA from the battery. Both the CHRG and FAULT pins are high impedance during standby mode. Should the battery voltage drop to

97.5% of the ﬂoat voltage, the LT3650 automatically restarts and initializes a new charging cycle.

Table 1. Status pin state and corresponding operating states

CHRG FAULT Charger Status

High Impedance High Impedance Standby/Shutdown/Top-off

Low High Impedance CV/CC Charging (>C/10)

High Impedance Low Bad Battery Detected

Low Low Temperature Fault

monitoring, or a safety timer features. The battery charging cycle terminates when the battery voltage approaches

4.2V and the charge current falls to 200mA. A new charge cycle is automatically initiated when the battery voltage falls to 4.1V.

Safety Features: Preconditioning, Bad Battery Detection, and Temperature Monitor

Li-Ion batteries can sustain irreversible damage when deeply discharged, so care must be taken when charging such a battery. A gentle preconditioning charge current is recommended to activate any safety circuitry in a battery pack and to re-energize deeply discharged cells, followed by a full charge cycle. If a battery has sustained damage from excessive discharge, however, the battery should not be recharged. Deeply discharged cells can form copper shunts that create resistive shorts, and charging such a damaged battery could cause an unsafe condition due to excessive heat generation. Should a deeply discharged battery be encountered, a battery charger must be intelligent enough to determine whether or not the battery has sustained deep-discharge damage, and avoid initiating a full charge cycle on such a damaged battery.

Linear Technology Magazine • June 2009

DESIGN FEATURES L

CLP

SYSTEM LOAD

INPUT SUPPLY

CLP

LT3650

CLP

RUN/SS

BOOST

SENSE

BAT

NTC

TIMER

1N914

CMSH3-40MA

SYSTEM LOAD

1µF

6.8µH

0.057

LT3650-X

GND

10µF

Li-Ion

CELL

10k

INPUT SUPPLY

12V TO 32V

(40V MAX)

BZX384-C9V1

(9.1V)

10µF

0.05Ω

SHDN

CHRG

FAULT

10k

36k

10k

0.68µF

0.1µF

The LT3650 employs an automatic precondition mode, which gracefully initiates a charging cycle into a deeply discharged battery. If the battery voltage is below the precondition threshold of 70% of the ﬂoat voltage, the maximum charge current is reduced to 15% of the programmed maximum (0.15C) until the battery voltage rises past the precondition threshold.

If the battery does not respond to the precondition current and the battery voltage does not rise past the

temperature, and suspends charging should the temperature fall outside of

the safe charging range. precondition threshold, a full-current charge cycle does not initiate.

If the safety timer is used for termination, the LT3650 also enables deep-discharge damage detection and incorporates a “bad battery” detection fault. Should the battery voltage remain below the precondition threshold for 1/8 of the charge cycle time (typically 22.5 minutes), the charger suspends the charging cycle and signals a “bad battery” fault on the status pins. The LT3650 maintains this fault state indeﬁnitely, but automatically resets itself and starts a new charging cycle if the damaged battery is removed and another battery is connected.

Li-Ion batteries have a relatively narrow temperature range where they can be safely charged. The LT3650 has a provision for monitoring battery

Figure 7. A single cell Li-Ion 2A battery charger with 3 hour safety timer termination, LED status indicators, temperature sensing, low input voltage charge current foldback, and input supply current limit

Linear Technology Magazine • June 2009

is enabled by connecting a 10k (B =

3380) NTC thermistor from the IC’s NTC pin to ground. This thermistor must be in close proximity to the battery, and is generally housed in the battery case. This function suspends a charging cycle if the temperature of the thermistor is greater than 40°C or less than 0°C. Hysteresis corresponding to 5°C on both thresholds prevents mode glitching. Both the CHRG and FAULT status output pins are pulled low during a temperature fault, signaling that the charging cycle is suspended. If the safety timer is used for termination, the timer is paused for the duration of a temperature fault, so a battery receives a full-duration charging cycle, even if that cycle is interrupted by the battery being out of the allowed temperature range.

Figure 6. R supply current limit

Under/overtemperature protection

sets the input

CLP

Status Indicator Pins

The status of a LT3650 charger is communicated via the state of two pins: CHRG and FAULT. These status pins are open-collector pull-down, reporting the operational and fault status of the battery charger. CC/CV charging is indicated while charge currents are greater than 1/10 the programmed maximum charge current. The status pins also communicate bad battery and battery temperature fault states. Table 1 shows a fault-state matrix for these two pins.

The status outputs can be used as digital status signals in processorcontrolled systems, and/or connected to pull current through an LED for visual status display. The status pins can sink currents up to 10mA and can handle voltages as high as 40V, so a visual display can be implemented by simply connecting an LED and series resistor to VIN.

Maximum Charging Current Programming and Adjustment

Maximum charge current is set using an external sense resistor placed between the BAT and SENSE pins of the LT3650. Maximum charge current corresponds to 100mV across this resistor. The LT3650 supports maximum charge currents up to 2A, corresponding to a 0.05Ω sense resistor.

The LT3650 includes two control pins that allow reduction of the programmed maximum charge current. The RNG/SS pin voltage directly affects the maximum charge current such that the maximum voltage allowed across the sense resistor is 1/10 the voltage on RNG/SS for RNG/SS < 1V. This pin sources a constant 50µA, so the voltage on the pin can be programmed by simply connecting a resistor from the pin to ground. A capacitor tied to this pin generates a voltage ramp at start-up, creating a soft-start function. The pin voltage can be forced externally for direct control over charge current.

The IC includes a PowerPath™ control feature, activated via the CLP pin, which acts to reduce battery charge current should the load on a

continued on page 38

–

ENABLE

IN3

SW3

FB3

GND

LIM0

CHRG

CLPROG

NTCBIAS

NTC

OVSENS

OVGATE

LIM1

ENOTG

EN1

EN2

EN3

SDA

SCL

1A 2.25MHz

BUCK

REGULATOR

ENABLE

IN2

SW2

FB2

400mA 2.25MHz

BUCK

REGULATOR

ENABLE

IN1

PROG

BAT

15mV

0.3V

3.6V

IDEAL

1.18V

OR 1.15V

–

5.1V

SW1

FB1

RST3

400mA 2.25MHz

BUCK

REGULATOR

2.25MHz

BIDIRECTIONAL

PowerPath SWITCHING REGULATOR

D/A

I2C PORT

LIM

DECODE

LOGIC

CC/CV

CHARGER

3.3V LDO

CHARGE

STATUS

OVP

WALL

DETECT

CONTROL

IDGATE

OUT

ACPR

WALL

–

BATTERY

TEMPERATURE

MONITOR

SUSPEND LDO

500µA/2.5mA

LDO3V3

BUS

L DESIGN FEATURES

Power Management IC Combines USB On-The-Go and USB Charging in Compact Easy-to-Use Solution

by George H. Barbehenn and Sauparna Das

Introduction

The USB interface was originally designed so that the device providing power (an “A” device) would act as the host and the device receiving power (a “B” device) was the peripheral. The A plug of the USB cable would always connect to the host device and the B plug would connect to the peripheral. The USB On-The-Go (OTG) standard, however, removes that restriction, so that the B device can now become a host and the A device can act as a peripheral.

In the USB speciﬁcation, standard hosts and hubs are limited to providing 500mA to each downstream device, but if a device is designated as a USB charger, it can supply up to 1.5A. USB chargers come in two ﬂavors. A “dedicated charger” is a charger that is not capable of data communication with the attached B device. A ”host/hub charger” is a charger that is capable of data communications with attached B devices.

When USB OTG functionality is combined with a USB battery charger in an end-user product, power can ﬂow in both directions, with relatively complicated logic and handshaking steering the ﬂow. To implement a robust solution, an integrated USB battery charger and power manager is a necessity. This article shows how to use the LTC3576 USB power management IC to easily combine USB On-The-Go functionality and battery charger capability into a single portable product.

Overview of the LTC3576

The LTC3576 provides the power resources needed to implement a portable device with USB OTG and USB battery charger detection capabilities (see block diagram in Figure 1). The USB input block contains a bidirec-

Figure 1. The LTC3576 combines USB charging and USB On-The-Go by using bidirectional DC/DC conversion from V

BUS

to V

OUT

Linear Technology Magazine • June 2009

DESIGN FEATURES L

< 6.5µF

ENOTG

MINI/MICRO A/B

MINI/MICRO A PLUG

OTG

COMPATIBLE

DEVICE SUCH

AS LTC3576

OTG

COMPATIBLE

BUSS

TRANSCEIVER

A DEVICE

B DEVICE

MINI/MICRO A/B

MINI/MICRO A PLUG

ENOTG

OTG

COMPATIBLE

DEVICE SUCH

AS LTC3576

OTG

COMPATIBLE

BUSS

TRANSCEIVER

VBUS

D+

D–

GND

VBUS

D+

D–

GND

3.3V

FOR LOW SPEED

ONLY

FOR FULL/HIGH SPEED ONLY

3.3V

< 6.5µF

ATTACH PHASE

PHYSICAL CONNECTION

OF DEVICES

CONNECT PHASE

DETECT VOLTAGE LEVELS ON

D+/D– TO DETERMINE DATA SPEED AND POWER LEVELS

ENUMERATION PHASE

SOFTWARE

HANDSHAKE

Figure 2. USB On-The-Go system diagram

Figure 3. USB sequence of events at start-up

tional switching regulator between V

BUS

from the USB input, this regulator operates as a step-down converter. Using the Bat-Track™ charging technique, the switching regulator sets the voltage at V very efﬁcient charging solution. When operating as an OTG A device, the regulator acts as a step-up converter by taking power from V 5V on V

The LTC3576 also has overvoltage protection and can be used with an external HV Buck regulator to provide V

OUT

switching regulator can take power from the HV buck regulator to supply power to the USB connection.

In addition, the LTC3576 provides two 400mA and one 1A step-down

1.5kΩ to 3.3V on D– during Connect for

1.5kΩ to 3.3V on D+ during Connect for

Full/High Speed, measure voltage on D–

Linear Technology Magazine • June 2009

and V

. When power is coming

OUT

BUS

to V

+ 0.3V, providing a

BAT

OUT

to produce

. In OTG mode, the bidirectional

Table 1. Load power signaling during Attach and Connect

Voltage on D–

with VDAT_SRC on D+ during Attach

Low Speed, measure voltage on D+

switching regulators for generating three independent voltage rails for the portable device. The LTC3576 allows all three step-down switching regulator output voltages to be enabled/disabled and adjusted over a 2:1 range via I2C. All three step-down regulators feature pulse-skipping mode, Burst Mode

operation and LDO mode, which can also be adjusted on-the-ﬂy via I

Mode Detection

The USB speciﬁcation allows for a number of different modes of operation for products supporting both the USB OTG speciﬁcation1 and the battery charger speciﬁcation2. Figure 2 shows a typical OTG system and Figure 3 shows the sequence of events that occur when the USB cable is plugged in. The product can be a B device

BUS

Host/Hub

< 500mA

0V 0.5V–0.7V 0.5V–0.7V

— > 2V < 0.8V

Dedicated Charger

BUS

< 1.5A

and can draw up to 100mA, 500mA, 900mA or 1.5A, depending on the type of A device powering V

, as shown

BUS

in the Table 1.

When an OTG device has a micro/ mini-A plug connected to its micro/ mini-AB connector, the OTG device becomes the A device and starts off as the host. The OTG A device supplies power to V

, as any other host A

BUS

device would, when requested by an attached peripheral or OTG B Device. As an A device, the LTC3576 can supply up to 500mA

The USB OTG speciﬁcation provides two means for a B device to signal to the A device that it wants power. The B device may drive the V

line above

BUS

2.1V, momentarily, or it may signal by driving the D+ or D– signal lines. The D+/D– signaling method could be

Host/Hub Charger

< (LS,FS < 1.5A/HS < 0.9A)

BUS

L DESIGN FEATURES

7.68k

1500pF

PROCESSOR

UNLESS NOTED, RESISTORS: OHMS, 0402 1% 1/16 WATT

* THREE 1Ω, 5% RESISTORS IN PARALLEL

CAPACTORS: µF, 0402, 10% 25V

D2, D3: 1N4148

L1: 1098AS-2R0M

L2, L3: 1098AS-4R7M

L4: LPS4018-3R3MLC

M2, M3, M6: NDS0610

M4, M5, M7: 2N7002L

Q1, Q2, Q3: MMBT3904LT1

4.35V TO 5.5V

NON-OPERATING FAULT TOLERANCE

TO 30V CONTINUOUS

47k

10k

100k 100k

VBUS

D–

D+

GNDSHGND

IDPUEN

FSPUEN

VBATVEN

VBATV

1ACHARGEEN

BAT

DF3-3P-2DSA

GND

NTC-EXT

PROCESSOR

3.6V AT

400mA

BUS

47k 15k 15k

BAT

47k

2.00k 2.00k

10k

BATTERY CHARGER HANDSHAKE

U2B

LTC202

44.2k

100k

LEAKAGE

CURRENT

MUST BE

<400nA

6.2k

PROGV

CLPROGV

SCL

SDA

RST3

CHRG

VDAT_SRCEN

IDAT_SINKEN

D-V

VBUSV

D–

D+

HUBEN

IDV

DAT_SINK

DAT_SRC

4.7k

100k

1.5k

3.01k

4.7µF

50V

22µF

6.3V

1µF

10V

22µF

6.3V

10µF

6.3V

0.1µF

16V

3.3V

U2A

LTC202

“V” SUF FIX

INDICATES

A/D INP UT

WALL

ACPR

LD03V3

OVSENS

OVGATE

ILM0

ILM1

BUS

ENOTG

EN1

EN2

EN3

OUT

BAT

IDGATE

BAT

IN1

SW1

FB1

CHRG

RST3

DVCC

SDA

SCL

CLPROG

PROG

NTCBIAS

NTC

Si2306BDS

Si2333DS

100µF

6.3V

0.337*

LTC3576EUFE

3.3µH LEAKAGE CURRENT MUST BE < 50nA

0.1µF

16V

4.7µH

2.2µF

6.3V

1.02M

324k

12pF

50V 5%

3.3V AT

400mA

10µF

6.3V

IN2

SW2

FB2

4.7µH

2.2µF

6.3V

1.02M

R23

324k

18pF

50V 5%

1.8V

AT1A

22µF

6.3V

IN1

SW3

FB3

GND

2.0µH

2.2µF

6.3V 402k

324k

27pF

50V 5%

USBMICRO-AB

µC

Figure 4. Portable system with OTG and battery charger support

Linear Technology Magazine • June 2009

DESIGN FEATURES L

(D+ or D–)

BUS

2.1V

100ms

7.5ms

4.9s

B DEVICE SIGNALING

A DEVICE DELIVERING V

USB

detected by an OTG compatible USB module on the system microcontroller (µC ). The V

signaling method could

BUS

be detected via an A/D on the µC. The LTC3576 bidirectional switching regulator is then enabled as a step-up converter (OTG mode) by setting the appropriate bit in the control registers via I2C.

Implementing a System for USB OTG and Battery Charging

Figure 4 shows an application for a portable device that supports both USB battery charging and USB OTG.

When IDPUEN is low, the ID pin is pulled up via R5, and if IDV is > 3V then it is conﬁgured to be a B device. If IDV is < 0.5V then it is conﬁgured to be an A device. The components enclosed in the box labeled “battery charger handshake” enable communication of the power capabilities depending on whether the portable device is conﬁgured as an A device or a B device. During the Attach phase, if the portable device is a B device, it can apply V

DAT_SRC

D+ line, load the D– line with I (50µA~150µA), and measure the resultant voltage on D– via D–V. If the voltage is 0, the A device is a Host/Hub, if the voltage is V device is a USB Charger.

During the Connect phase, FSPUEN is pulled low to apply 3.3V to D+, indicating a full/high speed device. At the same time the voltage on the D– line is read again via D–V. If it is less than 0.8V, then the A device is a Host/Hub Charger. If the voltage on D–V is above 2V, then the A device is a Dedicated Charger.

Linear Technology Magazine • June 2009

(0.5V~0.7V) to the

DAT_SRC

Figure 5. Session Request Protocol timing reference

then the A

DAT_SINK

For OTG functionality, if the portable device is conﬁgured as an A device, then it must drive V V

, which in this case is powered

OUT

BUS

from

from the battery. Since the LTC3576 is capable of supplying 500mA as an A device, the µC asserts HUBEN to indi- cate it is a Host/Hub. The bidirectional switching regulator in the LTC3576 is enabled by setting the appropriate bit in the control registers via the I2C port. If the B device drawing current from the V

line goes idle, then the

BUS

OTG A device may turn off the V voltage to conserve the battery. When the B device needs the V

voltage

BUS

to be present at some later time, it can request that the A device again drive V

by turning the bidirectional

BUS

switching regulator back on. It can do this by signaling on the D+ or D– lines or by driving the V

line to > 2.1V

BUS

(see Figure 5).

The Host A device only needs to respond to one of two SRP signaling methods. However, since not all USB engines respond to the D+/D– signaling, the V

line is sensed to check if

BUS

it is higher than 2.1V via the VBUSV A/D input.

When the portable device’s µC detects that the B device is requesting power on V

, either by sensing the

BUS

D+/D– signaling or by sensing that V

has been driven higher than 2.1V,

BUS

it should again turn on the OTG stepup converter in the LTC3576.

The PROG (PROGV) and BAT (VBATV) pins allow a Coulomb counter to be implemented in the µC. Reading the BAT voltage requires that the sensing divider be enabled by setting VBATVEN low. This ensures that the sense divider network does not dis-

charge the battery when the battery voltage isn’t being measured.

The default battery charge current has been set to 500mA, but can be increased to 1A by asserting the 1AchargeEN signal. This turns on M7, halving the PROG resistance and increasing the charge current. The input current limit will need to be set to 10X mode (1A) using the I2C port.

The optional network of C14 and R27/R28/R29 suppresses ripple on the BAT pin (and consequently on the V

BUS

pin) if there is no battery present.

BUS

This ripple can be in the tens of mV. While this will not damage anything, it may be desirable to suppress this signal.

The CLPROG (CLPROGV) and CHRG signals are often useful for housekeeping tasks in the µC.

The LTC3576 has an overvoltage protection function that controls M1, and protects the system from excessive voltages on the USB (J1) connector. Because the A/D is conﬁgured to monitor V

, it must also be protected by

BUS

D1 from excessive voltages.

The LDO3V3 regulator is conﬁgured to power the µC in low power mode (<20mA). When the µC needs to leave low power mode it ﬁrst enables Buck Regulator 2, which will provide up to 400mA.

Conclusion

The LTC3576 is a versatile PMIC consisting of a bidirectional power manager, overvoltage protection, three step-down switching regulators and a controller for an external high voltage step-down switching regulator. In conjunction with a few support components, the LTC3576 allows the implementation of a complete power management system for portable devices that support both USB OTG and USB battery charging.

Bibliography

”On-The-Go Supplement to the USB Speciﬁcation”,

Revision 1.3

“Battery Charging Speciﬁcation”, Revision 1.0

www.usb,org/developers/docs

L DESIGN FEATURES

LTC3101

BUCK-BOOST

LDO

ON/OFF

ADAPTER

USB

HOT SWAP OUT

3.xV AT 300 to 800mA

1.xV AT 350mA

1.8V AT 50mA

3.xV AT 100mA

x.xV AT 200mA

TRACKING OUT

LI-ION

USB

BAT

Power Management IC with Pushbutton Control Generates Six Voltage Rails from USB or 2 AA Cells Via Low Loss PowerPath Topology

Introduction

As the complexity of portable electronic devices continues to increase, the demands placed on power supplies, and their designers, expand dramatically. Not only must typical power systems accommodate multiple input sources, with voltages as low as 1.8V for two AA cells, but they must also provide an increasing number of independent output rails to support a wide range of requirements—for memory, microprocessors, backlights, audio and RF components. To further complicate matters, expanding feature sets add up to increased power dissipation, making it important to optimize overall power system efﬁciency. This is particularly challenging given that the constant drive to minimize the required board area and proﬁle height of the power system is at direct odds with improving efﬁciency.

The LTC3101 addresses all of these challenges with a single-IC power management solution that allows a designer to easily maximize overall power system efﬁciency while minimizing space requirements. The LTC3101 can generate six power rails by integrating three synchronous switching converters, two protected switched

Figure 2. Complete portable power

solution with a 16mm × 19mm footprint

by John Canﬁeld

Figure 1. Six output rails, a low loss PowerPath and integrated pushbutton control

power outputs, and an LDO. Its integrated low loss PowerPath™ topology allows each switching converter to run directly from either of two input power sources.

Two 350mA, high efﬁciency low voltage rails, typically used to power processors and memory, are generated by synchronous buck converters. Each converter is able to operate down to an input voltage of 1.8V thereby enabling single stage conversion from any input power source.

A single inductor buck-boost converter generates a high efﬁciency intermediate output rail, typically at 3V or 3.3V, and is able to operate from either input power source and with input voltages that are above, below, or even equal to the output regulation voltage. The buck-boost converter can supply a 300mA load at 3.3V for battery voltages down to 1.8V and an 800mA load for input voltages of 3.0V and greater.

Two always-alive outputs—MAX, which tracks the higher voltage input power source and LDO, a ﬁxed 1.8V

output—provide power to critical functions that must remain powered under all conditions. An integrated pushbutton controller with programmable µP reset generator provides complete ON/OFF control using only a minimal number of external components while independent enables allow total power-up sequencing ﬂexibility. This complete portable power solution is packaged in a single low proﬁle 24lead 4mm × 4mm QFN package and the entire power supply, including all external components, occupies a PCB area of less than 3cm2 as shown in Figure 2.

Zero Loss PowerPath Topology Maximizes Efﬁciency

Although rechargeable Li-Ion and Li-Polymer batteries are the leading chemistries for powering portable devices due to their high energy density and long cycle life, many portable devices continue to be powered by alkaline and NiMH cells. This allows indeﬁnite periods of use away from a

Linear Technology Magazine • June 2009

BAT1 USB1

ENA1 ENA2 ENA3

PWRKEY

PWRON

PWM

PBSTAT

RESET

USB2

FB3

LDO

SW2

FB2

HSO MAX

10µF

2 AA

CELLS

USB/WALL

ADAPTER

1.8V TO 5.5V

10µF

OUT3

= 3.3V

300mA FOR VIN ≥ 1.8V

800mA FOR VIN ≥ 3V

221k

Hot Swap OUTPUT: 3.3V AT 100mA TRACKING OUTPUT: 200mA

4.7µH

4.7µF

1.8V AT 50mA

OUT2

1.8V 350mA

OUT1

1.5V 350mA

10µF

221k

110k

4.7µH

SW1

FB1

10µF

221k

147k

4.7µH

0.1µF

ON/OFF

BAT2 SW3A

LTC3101

GND

µP

SW3B OUT3

DIS ENA

USB

BAT

BUCK V

OUT

USB

BAT

BUCK-BOOST

OUT

charging socket—which is particularly important for devices intended for use in remote locales such as handheld personal navigation devices or portable medical devices. Voice recorders, digital still cameras and ultra-small video recorders are additional examples of devices that beneﬁt from the ability to operate from a pair of commonly available batteries, rather than requiring the lengthy recharging cycle needed for an internal Li-Ion battery.

Even in portable devices where the primary power source is restricted to AA or AAA form factor cells, there still exist a wide variety of compatible chemistries including alkaline, rechargeable alkaline, NiMH and single-use lithium. As a result, the AA/AAA powered device must accommodate a wide range of input voltages, from 1.8V for two series alkaline cells near end of life, to approximately 3.7V for a pair of fresh non-rechargeable lithium cells. With its wide 1.8V to

5.5V input voltage range, the LTC3101 can easily support all of these battery chemistries. In addition, the LTC3101 is able to operate from a single standard Li-Ion/Polymer cell in cases where recharging is performed independently.

Although rechargeable cells are usually charged outside these types of devices, the power supply must accommodate a secondary tethered power source such as USB or a regulated wall adapter. Consequently, the power supply must include a means to generate every power rail from either of two input sources, and the ubiquitous

3.3V rail must be generated from input power sources that can be higher or lower voltage.

In many devices, the capability to handle dual power sources is provided by using discrete power MOSFETs to switch regulator inputs between the two input power sources or by utilizing two regulators for generation of each rail (for example, a buck converter that generates a 3.3V rail from the USB input in conjunction with a boost converter that generates the 3.3V rail from the battery input).

Both of these approaches suffer from signiﬁcant drawbacks. The par-

Linear Technology Magazine • June 2009

allel converter approach increases system cost and size given that only one converter is ever active at any given time and often suffers from glitches and disruptions to the output rails during the transition between the two input power sources. Similarly, the discrete power switch technique reduces efﬁciency due to the addition of extra series elements in the power path, increases component count, and can also lead to disruptions in the output rails unless the supply crossover is carefully controlled.

The LTC3101 avoids these problems by using a low loss PowerPath topology as shown in Figure 4, where each converter is able to operate directly from either input power source. In this architecture, each switching converter utilizes an additional power switch, which is connected to the alternate power input. As a result, each converter is able to run with maximum efﬁciency from either input power source so no efﬁciency penalty

DESIGN FEATURES L

Figure 3. Typical application

is incurred in supporting dual input power sources.

The total solution area is minimized by the fact that the same inductor is used in either case. In addition, the automatic transition between the two input power sources is seamless—there is no interruption to any of the output rails. Figure 5 shows the transient response of the buck-boost converter as the input power source transitions from battery power to USB power in response to a live cable plug into a USB port.

Integrated Buck-Boost Provides High Efﬁciency 3V/3.3V Rail from Any Power Source

In many portable devices an intermediate supply rail, typically regulated to

3.3V, is required to power an RF stage or audio ampliﬁers. Often this rail is generated from the two series AA cells using a boost converter. However, the higher cell voltage of single-use lithium

Figure 4. The low loss PowerPath architecture

L DESIGN FEATURES

OUTPUT

VOLTAGE

200mV/DIV

INDUCTOR

CURRENT

200mA/DIV

USB

2V/DIV

100µs/DIV

batteries such as the Energizer e2 brand can cause problems when the battery voltage is signiﬁcantly higher than the output voltage. Depending on the boost converter utilized, this can result in low efﬁciency operation or even loss of regulation on the 3.3V rail.

To avoid this problem, the LTC3101 generates the 3.3V rail utilizing a buck-boost converter, which accepts any input voltage in the range 1.8V to

5.5V without sacriﬁcing efﬁciency. In fact, when operating with a fresh pair of single-use lithium batteries at 3.7V, the LTC3101 buck-boost efﬁciency is greater than 94% at 150mA load current. In addition, the same buck-boost converter is able to operate directly from the USB input, so generation of the 3.3V rail requires only a single inductor.

Reverse Blocking LDO Enables Data Retention During Battery Swaps

Many portable electronic devices contain critical circuitry such as real time clocks, which must remain powered under all conditions. The MAX and LDO outputs of the LTC3101 are alive as long as either input power source is present, regardless of the state of the pushbutton interface or enable inputs. It is also possible to connect a large capacitor directly to the LDO output to serve as a charge reservoir for powering critical functions during times, such as battery swaps, when both input power sources are temporarily removed. In its reverse blocking state, the maximum reverse current through the LDO is limited to under 1µA in order to preserve charge in the reservoir capacitor.

Figure 5. Buck-boost output voltage transient on USB hot plug

MAX and Hot Swap Outputs Power Additional Regulators and Flash Memory Cards

Portable electronic devices often require additional miscellaneous power supplies, such as current regulated drivers for LED backlighting and LDOs for low power rails. Typically these secondary supplies must be functional whenever either input power source is present, so they also require power path control to switch between the two input power sources.

External supplies can take advantage of the LTC3101’s PowerPath control circuit via the MAX output, which continuously tracks the higher voltage input power source. Additional regulators can be directly connected to this output, thus freeing the designer from the need to implement an additional switched power path. The MAX output is able to support a 200mA load and is current limited to protect against overload conditions and short circuits.

Many portable electronic devices provide ﬂash memory card interfaces for use as bulk storage memory. Typical ﬂash memory cards such as Compact Flash (CF) and Secure Digital (SD) formats require a regulated

3.3V supply that is typically capable of providing tens of milliamps. However, many ﬂash memory cards have a signiﬁcant amount of supply bypass capacitance installed on the card and when hot plugged into a live 3.3V rail, the inrush current required to charge these supply bypass capacitors on the memory card can momentarily drag down the host’s supply, causing disruption to other ICs powered by that rail.

The LTC3101’s dedicated 100mA hot swap output (powered from the buck-boost converter rail) does not have this problem. The independent current limit of the hot swap switch allows ﬂash memory cards to be hot plugged without disruption to the primary 3.3V rail. In addition, the hot swap output is fully short circuit protected to safeguard against accidental shorts at the memory card interface port.

Low Quiescent Current Minimizes Battery Drain

Most portable electronic devices spend signiﬁcant, if not the majority, of their time in sleep or standby modes. In fact, even when an appliance is off, there is often circuitry that must remain powered, including real time clocks and volatile memory storing conﬁguration settings. The always-alive 1.8V LDO and tracking MAX outputs remain powered whenever either input power source is present allowing them to be utilized for supplying such critical functions. In order to minimize battery discharge during this time, the total quiescent current draw of the LTC3101 with both the MAX and LDO outputs active is reduced to 15µA.

Many portable electronic devices also support a standby mode in which several of the system’s voltage rails must be kept in regulation. Typically, in standby the microprocessor and memory remain powered and the processor is placed in a low current sleep mode enabling the device to return to an active operating state with minimal delay.

In order to minimize battery drain in such modes of operation, all three switching converters in the LTC3101 feature Burst Mode operation, which can be enabled via a dedicated pin. With Burst Mode operation enabled, the buck converters automatically transition from PWM to Burst Mode operation at sufﬁciently light load (typically 10mA) while the buck-boost converter uses Burst Mode operation at all load currents. In Burst Mode operation with all six output rails maintained in regulation the total quiescent current draw of the LTC3101

Linear Technology Magazine • June 2009

is reduced to only 38µA. In addition,

HSO

(OPTIONAL)

ENA1 ENA2

FILT

500µs/DIV

OUT

BUCK 1

(1V/DIV)

OUT

BUCK 2

(1V/DIV)

OUT

BUCK-BOOST

(2V/DIV)

HOT SWAP

(2V/DIV)

500µs/DIV

OUT

BUCK 1

(1V/DIV)

OUT

BUCK 2

(1V/DIV)

OUT

BUCK-BOOST

(2V/DIV)

HOT SWAP

(2V/DIV)

to ensure low supply rail noise, the Burst Mode operation output voltage ripple is typically less than 1% of the regulation voltage of each output rail. All three switching converters can be forced into ﬁxed frequency PWM mode operation to ensure low noise operation while critical system functions are underway.

DESIGN FEATURES L

Figure 6. Default power-up sequencing

Flexible Power-Up Sequencing Options

The LTC3101 provides a variety of sequencing options. Most systems that incorporate multiple power supply rails require that they come into regulation in a certain sequence with speciﬁc timing. This is because individual ICs and modules that are powered from multiple rails need particular sequencing to minimize start-up current and ensure predictable power-up behavior.

Common examples include microprocessors and FPGAs, which often require that the peripheral supply powering the I/O buffers is made available only after the lower voltage core is in regulation. In addition, at the board level, many systems bring up the supplies for peripheral devices only after the processor is powered up to avoid erratic behavior from peripherals lacking processor oversight.

Each switching converter in the LTC3101 has an internal power-good comparator, which is used internally to sense when that rail is in regulation. The default power-up sequence enables the individual outputs in the following order: buck converter 1, buck converter 2, buck-boost converter, and ﬁnally the hot swap output. Each converter is enabled once the preceding converter in the sequence reaches

regulation (typically 94% of the target output voltage). The default power-up sequence using all converter channels is shown in Figure 6.

If the dedicated enable pin for any switching converter is held low during the pushbutton triggered initiation, that converter is simply skipped in the default power-up sequence, but that channel can still be enabled at a later time. This functionality allows the LTC3101 to implement any arbitrary power-up sequence using few if any external components.

can be accomplished by adding a simple RC ﬁlter with the desired time constant between the hot swap output and the buck enables. Notice however, that if the hot swap output is forced to ground, the buck converters will be disabled. If there is a potential for the hot swap output to fall below the enable threshold (typically 0.7V) during normal operation, then the buck enables can instead be driven through an RC delay from the buck-boost voltage directly rather than from the hot swap output.

For example, in some systems the

3.3V buck-boost rail must come up ﬁrst, followed by both buck rails in unison. This can be accomplished by driving the buck enables from the hot swap output, HSO, as shown in Figure 7. The bucks do not power up in the normal sequence since their enables are low to start. Once the buck-boost reaches regulation, the hot swap output is enabled, which in turn enables the two buck converters. Since the hot swap output is not powered until the buck-boost is in regulation, this conﬁguration ensures that the buck converters do not become active until after the buck-boost is in regulation, as shown by the waveforms in Figure 8.

If an additional delay is required

Conclusion

The LTC3101 is perfectly suited for the needs of the next generation of extended functionality compact portable electronic devices.

The job of the power system designer is simpliﬁed by its compact solution footprint and ability to generate six commonly required output voltage rails automatically from two independent wide input voltage range power sources. The LTC3101’s low quiescent current and a high efﬁciency, low loss PowerPath architecture maximize battery life. A wide range of output voltages, programmable duration µP reset generator, and independent enables offer ﬂexibility and easy customization.

before the bucks are enabled, this

Figure 7. Sequencing the buck enables using the hot swap output rail

Linear Technology Magazine • June 2009

Figure 8. Power-up sequencing, buck-boost followed by the buck outputs

L DESIGN FEATURES

ADC

0.1µF

12V

OUT

12V

1.5A

20k

10k

12V

330µF

OUT

GND

MON

LTC4217DHC-12

INTV

TIMER

FLT

AUTO

RETRY

TEMPERATURE (°C)

–50

0.9

0.8

0.7

0.6

0.5

0.4

0.3 –25 0 25 50 75 150125100

ISET

(V)

Improve Hot Swap Performance and Save Design Time with Hot Swap Controller that Integrates 2A MOSFET and Sense Resistor

Introduction

The LTC4217 Hot Swap controller turns a board’s supply voltage on and off in a controlled manner allowing the board to be safely inserted and removed from a live backplane. No surprise there, this is generally what Hot Swap controllers do, but the LTC4217 has a feature that gives it an advantage over other Hot Swap controllers. It simpliﬁes the design of Hot Swap systems by integrating the controller, MOSFET and sense resistor in a single IC. This saves signiﬁcant design time that would otherwise be spent choosing an optimum controller/MOSFET combination, setting current limits, and carefully designing a layout that protects the MOSFET from excessive power dissipation.

One signiﬁcant advantage of an integrated solution over discrete solutions is that the current limit accuracy is well known. In discrete solutions, the overall precision of the current limit is a function of adding the tolerances of contributing components, while in the LTC4217, it appears as a single 2A speciﬁcation.

The integrated solution also simpliﬁes layout issues by optimizing MOSFET and sense resistor connections. The inrush current, current limit threshold and timeout can be set to default values with no external

components or easily adjusted using resistors and capacitors to better suit a large range of applications. The part is able to cover a wide 2.9V to 26.5V voltage range and includes a temperature and current monitor. The MOSFET is kept in the safe-operating–area (SOA) by using a time-limited foldback current limit and overtemperature protection.

The LTC4217 can be easily applied in its basic conﬁguration, or, with a few additional external components, set up for applications with special requirements.

Monitoring the MOSFET

The LTC4217 features MOSFET current and temperature monitoring. The current monitor outputs a current proportional to the MOSFET current, while a voltage proportional to the MOSFET temperature is available. This allows external circuits to predict possible failure and shutdown the system.

The current in the MOSFET passes through a sense resistor, and the voltage on the sense resistor is converted to a current that is sourced out the I

pin. The gain is 50µA from I

MON

for 1A of MOSFET current. The output current can be converted to a voltage using an external resistor to drive a

Figure 2. 12V, 1.5A card resident application

by David Soo

Figure 1. V

comparator or ADC. The voltage compliance for the I (INTVCC – 0.7V).

The MOSFET temperature corresponds linearly to the voltage on the I

pin, with the temperature proﬁle

SET

shown in Figure 1. The open circuit voltage on this pin at room temperature is 0.63V. In addition, the overtemperature shutdown circuit turns off the MOSFET when the controller die temperature exceeds 145°C, and turns it on again when the temperature drops to 125°C.

12V Application

Figure 2 shows the LTC4217-12 in a

MON

12V Hot Swap application with default settings. The only external component required is the capacitor on the INTVCC pin. The current limit, inrush current control, and protection timer are internally set at levels that protect the integrated MOSFET. The input voltage monitors are preset for a 12V supply using internal resistive dividers from the V pins. The UV condition occurs when VDD falls below 9.23V; OV when VDD exceeds 15.05V.

ply voltage on and off in a controlled

supply to drive the UV and OV

The LTC4217 turns a board’s sup-

Linear Technology Magazine • June 2009

vs temperature

ISET

pin is from 0V to

MON

DESIGN FEATURES L

150k

20k

ADC

224k

0.1µF

20k

12V

10k

0.47µF

330µF

OUT

12V

0.8A

OUT

GND

MON

20k

SET

0.1µF

GATE

LTC4217FE

INTV

TIMER

FLT

140k

20k

FB VOLTAGE (V)

CURRENT LIMIT VALUE (A)

0.5

1.0

1.5

2.0

2.5

0.2 0.4 0.6 0.8 1.0 1.2

t1 t2

SLOPE = 0.3V/ms

GATE

OUT

VDD + 6.15

manner, allowing the board to be safely inserted and removed from a live backplane. Several conditions must be present before the internal MOSFET can be turned on. First the VDD supply exceeds its 2.73V undervoltage lockout level and the internally generated INTVCC crosses 2.65V. Next the UV and OV pins must indicate that the input power is within the acceptable range. These conditions must be satisﬁed for the duration of 100ms to ensure that any contact bounce during the insertion has ended.

The MOSFET is then turned on by a controlled 0.3V/ms gate ramp as shown in Figure 3. The voltage ramp of the output capacitor follows the slope of the gate ramp thereby setting the supply inrush current at:

= CL • (0.3V/ms)

INRUSH

To reduce inrush current further, use a shallower voltage ramp than the default 0.3V/ms by adding a ramp

Linear Technology Magazine • June 2009

Figure 5. 0.8A, 12V card resident applicaiton

Figure 3. Supply turn-on

capacitor (with a 1k series resistor) from gate to ground.

As OUT approaches the VDD supply, the powergood indicator (PG) becomes active. The deﬁnition of power good is the voltage on the FB pin exceeds

1.235V while the GATE pin is high. The FB pin monitors the output voltage via an internal resistive divider from the OUT pin. Once the OUT voltage crosses the 10.5V threshold and the GATE to OUT voltage exceeds 4.2V, the PG pin ceases to pull low and indicates that the power is good. Once OUT reaches the VDD supply, the GATE ramps until clamped at 6.15V above OUT.

The LTC4217 features an adjustable current limit with foldback that protects against short circuits or excessive load current. The default current limit is 2A and can be adjusted lower by placing a resistor between the I

pin and ground. To prevent exces-

SET

sive power dissipation in the switch during active current limit, the available current is reduced as a function

Figure 4. Current limit threshold foldback

of the output voltage sensed by the FB pin as shown in Figure 4.

An overcurrent fault occurs when the current limit circuitry has been engaged for longer than the delay set by the timer. Tying the TIMER pin to INTVCC conﬁgures the part to use a preset 2ms overcurrent time-out and a 100ms cool-down time. After the 100ms cool-down, the switch is allowed to turn on again if the overcurrent fault has been cleared. Bringing the UV pin below 0.6V and then high clears the fault. Tying the FLT pin to the UV pin allows the part to self-clear the fault and turn on again after the 100ms cool-down.

Programmable Features

The LTC4217 application shown in Figure 5 demonstrates the adjustable features.

The UV and OV resistive dividers set undervoltage and overvoltage turnoff thresholds while the FB divider determines the power good trip point. The R-C network on the GATE pin decreases the gate ramp to 0.24V/ms from the default 0.3V/ms to reduce the inrush current.

The 20k I sistive divider with an internal 20k resistor to reduce the current limit threshold (before foldback) to onehalf of the original threshold for a 1A current limit. The graph in Figure 6 shows the current limit threshold as the I

resistor varies.

SET

As in the previous application, the UV and FLT signals are tied together so that the part auto-retries turn-on following shutdown for an overcurrent fault.

resistor forms a re-

SET

continued on page 25

L DESIGN FEATURES

OUT

(AC)

50mV/DIV

20V/DIV

10A/DIV

LOAD

10A/DIV

5µs/DIV

LOAD STEP 0A TO 10A VIN = 12V V

OUT

= 1.2V MODE = 0V SW FREQ = 400kHz

OUT

(AC)

50mV/DIV

20V/DIV

10A/DIV

LOAD

10A/DIV

5µs/DIV

LOAD STEP 10A TO 0A VIN = 12V V

OUT

= 1.2V MODE = 0V SW FREQ = 400kHz

Compact No R

SENSE

Controllers Feature Fast Transient Response and Regulate to Low V from Wide Ranging V

Introduction

The trend in digital electronics is to lower voltages and increasing load currents. This puts pressure on DC/DC converters to produce low voltages from increasingly voltage-variable supplies, such as stacked batteries and unregulated intermediate power buses, so power converters must be optimized for low output voltages, low duty factors, and wide control bandwidths. To meet these requirements, the DC/DC controller IC must offer high voltage accuracy, good line and load regulation, and fast transient response. The constant on-time valley current mode architecture used in the LTC3878 and LTC3879 is ideally suited to low duty factor operation, offering a compact solution with excellent system performance.

The LTC3878 and LTC3879 are

a new generation of No R

SENSE

controllers that meet the demanding requirements of low voltage supplies for digital electronics. The LTC3878 is a pin compatible replacement for the LTC1778 in designs where EXTVCC is not required. The LTC3879 adds separate RUN and TRACK/SS pins for applications requiring voltage tracking. Both devices offer continuously programmable current limit, using the bottom MOSFET VDS voltage to sense current.

Valley Current Mode Control Simpliﬁes Loop Compensation…

There are two common implementations of current mode control. Peak current mode control regulates the high side MOSFET on-time, while valley current mode regulates the bottom side MOSFET off-time. The current mode loop bandwidth is in-

™

Figure 1. Transient response, positive load step

versely proportional to the on-time for a peak current controller and inversely proportional to the off-time for a valley mode controller. A peak current mode controller with an on-time of 50ns must have a closed current loop bandwidth exceeding 20MHz. For a valley current mode controller, the current loop bandwidth is determined by the typical off-time of 220ns, resulting in a closed current loop bandwidth requirement of only 4.5MHz. Consequently, valley current mode control has less stringent bandwidth requirements for the same system performance when compared to a peak current mode control in a similar application. This allows the LTC3878 and LTC3879 to offer high performance, low duty factor operation at reasonable current loop bandwidths.

The constant on-time valley current mode control of the LTC3878 and LTC3879 simplifies compensation design by eliminating the need for slope compensation. A ﬁxed frequency valley mode controller requires slope compensation when operating at less than 50% duty factor to prevent subcycle oscillation. Subcycle oscillation occurs because the PWM pulse width

OUT

by Terry J. Groom

Figure 2. Transient response, load release

is not uniquely determined by inductor current alone. This oscillation cannot exist in constant-on-time control because the PWM pulse width is uniquely determined by the internal open loop pulse generator. True current mode control and constant on-time combine to give the LTC3878 and LTC3879 performance advantages over other constant on-time regulators or ﬁxed frequency valley current mode control architectures.

…and Improves Transient Response Time

In a buck controller, transient response is largely determined by how quickly the inductor current responds to loop disturbances. The most demanding loop disturbances are load steps and load releases.

The inherent speed advantage of a constant on-time architecture lies in the fact that the regulator is pulse frequency modulated (PFM) insead of pulse width modulated (PWM). Although the switching frequency is ﬁxed in steady state operation, it can increase or decrease as required in response to an output load step or load release.

Linear Technology Magazine • June 2009

DESIGN FEATURES L

f Hz

MAX

ON OFF MIN

t t

(

)

( )

f g EA R

CGO m C

LIMIT

OUT

REF

OUT

= • •( )

.1 6

5A/DIV

OUT

0.5V/DIV

TRACK/SS

0.5V/DIV

20ms/DIV

VIN = 12V V

OUT

= 1.2V

SW FREQ = 400kHz

TRACK/SS

LTC3879

BOOST

0.22µF M1

RJK0305DPB

VCC

4.7µF

220pF

33pF

CMDSH-3

0.56µH

OUT1

330µF

2.5V s2

OUT2

47µF

6.3V s2

IN1

10µF

50V

IN2

100µF 50V

OUT

1.2V 15A

4.5V TO 28V

PGOOD

100k

80.6k

27k

FB1

10.0k

10.0k TG

152

RNG

143

MODE PGND

134

125

SGND INV

116

107

RUN

432k

FB2

10.0k

M2 RJK0330DPB

IN1

: UMK325BJ106MM s3

OUT1

: SANYO 2R5TPE330M9 s2

OUT2

: MURATA GRM31CR60J476M s2

L1: VISHAY IHLP4040DZ-11 0.56µH

0.1µF

LOAD CURRENT (A)

EFFICIENCY (%)

100

0.01 1 10 100

0.1

CONTINUOUS MODE

DISCONTINUOUS

MODE

VIN = 12V V

OUT

= 1.2V

SW FREQ = 400kHz

The maximum frequency in response to a load step is determined by the on-time plus the off-time:

In low duty factor applications the maximum frequency is typically much greater than the nominal operating frequency, producing excellent transient characteristics.

Figure 1 shows the load step response of a 12V-to-1.2V converter operating at 400kHz. In this case the on-time is equal to 250ns and the minimum off-time is 220ns. The maximum frequency available to respond to a load step is 2.12MHz, which is over ﬁve times the nominal switching frequency. Note the increase in switching frequency of the VSW waveform in response to the 10A load step. The increase in switching frequency causes the inductor current to ramp faster in constant on-time PFM controllers than is possible in a true ﬁxed frequency PWM.

In response to a load release (Figure 2), the minimum frequency is effectively zero, since the bottom gate is held high as long as needed to ramp the inductor current down to the internal regulation set point. In this example, the inductor current ramps from 11A to –8A in 13µs as the output recovers from the load step. For both load transient cases, variable frequency has an inherent speed advantage over ﬁxed frequency in transient recovery.

Start-Up Options

The LTC3878 offers the simplicity of current limited start-up through the combined RUN/SS pin. When RUN/SS is greater than 0.7V all internal bias is activated. Once RUN/SS exceeds 1.5V, switching begins. The current limit is gradually increased as the RUN/SS pin voltage ramps until reaching full

Figure 3. Start-up into a prebiased output

Transient settling requires both the large signal ramping of inductor current and the stable settling of the output to the desired regulation point. Excessive output overshoot or ringing indicates marginal system stability likely caused by inadequate compensation. A rough compensation check can be made by calculating the gain crossover frequency, given by the following equation (where V for the LTC3878 and V

REF

= 0.8V

REF

= 0.6V for

the LTC3879):

As a rule of thumb, the gain crossover frequency should be less than 20% of the switching frequency. With any analog system, transient response is determined by closed loop bandwidth. In order to optimize for transient performance, it is desirable to have a small inductor and a wide closed loop bandwidth. A small inductor is desired for quick output current response, while the closed loop bandwidth and phase margin determines how quickly the output settles after a load step.

output at approximately 3V.

separate RUN and TRACK/SS pins. All internal bias is activated when RUN exceeds 0.7V. Switching begins when RUN exceeds 1.5V. The TRACK/SS pin can also be used for input voltage tracking, where the LTC3879’s output tracks the voltage on the TRACK/SS pin until it exceeds 0.6V. Once TRACK/SS exceeds 0.6V the output regulates to the internal 0.6V reference. An internal 1µA pull-up current is available to create a soft-start voltage ramp when a small capacitor is connected to TRACK/SS. Together, RUN and TRACK/SS enable a number

Figure 4. Efﬁciency for application in Figure 5

The LTC3879 adds the ﬂexibility of

Linear Technology Magazine • June 2009

Figure 5. Wide input range to 1.2V at 15A, operating at 400kHz

L DESIGN FEATURES

TRACK/SS

MASTER

LTC3879

BOOST

0.22µF M1

RJK0305DPB

VCC

4.7µF

330pF

47pF

CMDSH-3

0.44µH

OUT1

330µF

2.5V s3

OUT2

100µF

6.3V s2

IN1

10µF

16V

IN2

180µF 16V

OUT

1.2V 20A

4.5V TO 14V

PGOOD

100k

57.6k

TR2

10.0k

15k

FB1

10.0k

TR1

10.0k

152

RNG

143

MODE PGND

134

125

SGND INV

116

107

RUN

576k

FB2

10.0k

470pF

0.1µF

1Ω

M2 RJK0330DPB

IN1

: TDK C3225X5R1C106MT s2

OUT1

: SANYO 2R5TPE330M9 s3

OUT2

: MURATA GRM31CR60J107ME39 s2

L1: PULSE PA0513.441NLT

5ms/DIV

VIN = 12V V

OUT

= 1.2V

SW FREQ = 400kHz

MASTER

0.5V/DIV

OUT

0.5V/DIV

Figure 6. Coincident tracking example produces 1.2V at 20A, operating at 300kHz

of start-up supply sequencing and tracking options.

Both the LTC3878 and LTC3879 have the ability to start up onto prebiased outputs. Because current limit is ramped in the LTC3878, prebiased output voltages are not an issue. The LTC3879 output tracks the input on the TRACK/SS pin. To accommodate prebiased outputs, the LTC3879 will not switch until the TRACK/SS pin exceeds the VFB voltage. Once TRACK/ SS exceeds VFB the output follows the TRACK/SS pin in continuous conduction mode until the output regulates to the internal reference.

In Figure 3 the LTC3879 output is prebiased to 0.5V. The TRACK/SS pin ramps from zero and crosses the prebiased output feedback point at approximately 28ms, when switching begins. Once switching begins the output enjoys a smooth soft-start ramp. The LTC3879 operates in continuous conduction mode during start-up, regardless of the mode setting, allowing regulation of the output voltage to the TRACK/SS input pin voltage during soft-start.

High Efﬁciency

The LTC3879 and LTC3879 offer excellent efﬁciency through the combination of strong gate drivers and short dead time. The top gate driver offers a 2.5Ω pull up resistance and a 1.2Ω pull down, while the bottom gate driver offers a 2.5Ω pull up and a 0.7Ω pull

down. Dead time has been measured as low as 12ns, minimizing switching loss. Efﬁciency has been measured at

91.8% in a 1.2V/20A application. The LTC3878 and LTC3879 offer

both discontinuous conduction mode (DCM) and continuous conduction mode (CCM) operation. Figure 4 shows peak efﬁciency over 90% for 12V and 15A in CCM. In CCM, either the top MOSFET or the bottom MOSFET is active and the output inductor is continuously conducting. In DCM, the top and bottom MOSFET can be off simultaneously in order to improve low current efﬁciency. In Figure 4, at 100mA, the efﬁciency is greater than 70% in DCM, compared to only 20% in CCM. Improvements in efﬁciency in DCM are seen when the load is less than the DC average of the steady state ripple current, causing the regulator to enter discontinuous conduction.

Application Example:

4.5V-to-28V In to 1.2V Out

with 90% Peak Efﬁciency

Figure 5 shows an application that converts a wide 4.5V-to-28V input voltage to a 1.2V ±5% output at 15A. The nominal ripple current is chosen to be 35% resulting in a 0.55µH inductor and ripple current of 5.1A. Because the top MOSFET is on for a short time, an RJF0305DPB (R nal), C

MILLER

= 150pF, V sufﬁcient. The stronger RJK0330DPB is chosen for the bottom MOSFET, with

= 10mΩ (nomi-

DS(ON)

MILLER

= 3V) is

a typical R

of 3.8mΩ. This results

DS(ON)

in 90% peak efﬁciency. Note that the efficiency, transient and start-up waveforms in Figures 1–4 were taken from this design example.

Tracking

Figure 6 shows a LTC3879 in a

1.2V/20A output, 300kHz application design with coincident rail tracking. In coincident tracking, two supplies ramp up in unison until the lower voltage supply reaches regulation, at which point the higher voltage supply continues to ramp to its regulated value. Coincident tracking is implemented by making the resistor divider from the master voltage to the TRACK/SS pin equal to the feedback divider from V to VFB. In Figure 6, the output is 1.2V, so the divider is equal to 0.6V/1.2V, or 0.5. This design tracks any master supply that is equal to or greater than

1.2V. The TRACK/SS pin should be greater than 0.65V in regulation to ensure that the LTC3879 has sufﬁcient

Figure 7. Coincident tracking waveforms for application in Figure 6

Linear Technology Magazine • June 2009

OUT

DESIGN FEATURES L

18V TO 72V

GND

5V, 2A

OUT

–V

OUT

PA1277NL

162k

6.81k

10pF

2200pF

1µF

330pF

22.1k

M1 Si4848DY

LT4430ES6

0.47µF

PS2801-1

UPS840

OUT

100µF ×2

BAT54C

BAS516

1µF

×2

4.7nF

10k

51.1Ω

BAV21W

BAS516

6.81k

0.04Ω

105k

4.7µF

0.1µF

INTVCC

GATE

SENSE

SYNC

GND

LT3758

SHDN/UVLO

FBX

8.66k 36.5k

100pF

OPTO

COMP

GND

margin to switch from tracking the TRACK/SS input voltage to regulating to the internal reference.

Figure 7 shows typical tracking waveforms of the application in Figure 6. V voltage, V

and the reference supply

OUT

MASTER

, are equal and track together during start-up until they reach 1.2V, at which point V lates to 1.2V while V

MASTER

regu-

OUT

continues

ramping to 1.8V.

Conclusion

The LTC3878 and the LTC3879 support a VIN range from 4V to 38V (40V abs max). The regulated output voltage is programmable from 90% VIN down to

LT3757/58, continued from page 4

An 18V–72V Input, 5V/2A Output Isolated Flyback Converter

The basic design shown in Figure 7 can be modiﬁed to provide DC isolation between the input and output with the addition of a reference, such as the LT4430, on the secondary side of the transformer and an optocoupler

0.8V (for the LTC 3878) and 0.6V (for the LTC3879). The output regulation accuracy is ±1% over the full –40°C to 85°C temperature range. The operating frequency is resistor programmable and is compensated for variations in VIN. Current limit is continuously programmable and is measured without a sense resistor by using the voltage drop across the external synchronous bottom MOSFET.

The valley current mode architecture is ideal for low duty factor operation and allows very low output voltages at reasonable current loop bandwidths. Compensation is easy to design and offers robust and stable

to provide feedback from the isolated secondary to the LT3758. Figure 8 shows an 18V–72V input, 5V/2A output isolated ﬂyback converter.

Conclusion

The LT3757 and LT3758 are versatile control ICs optimized for a wide variety of single-ended DC/DC converter to-

operation even with low ESR ceramic output capacitors. The LTC3878 offers current limited start-up, while the LTC3879 has separate run and output voltage tracking pins. The LTC3878 is available in the GN16 package, and the LTC3879 is available in thermally enhanced MSE16 and QFN (3mm × 3mm) packages. Excellent performance and compact size make the LTC3878 and LTC3879 well suited to small, tightly constrained applications such as distributed power supplies, embedded computing and point of load applications.

pologies. Both offer a particularly wide input voltage range. These ICs produce space saving, cost efﬁcient and high performance solutions in any of these topologies. The range of applications extends from single-cell, lithiumion powered systems to automotive, industrial and telecommunications power supplies. L

Linear Technology Magazine • June 2009

Figure 8. 18V–72V input, 5V/2A isolated ﬂyback converter

L DESIGN FEATURES

LOAD CURRENT (mA)

0.1

EFFICIENCY (%)

POWER LOSS (mW)

350

300

250

200

150

100

1 10 100

OUTP

OUTN

LT3582

SWP

CAPP

NEG

–12V

85mA

CAPP

SDA

SCL

GND

SWN

OUTN

I2C

INTERFACE

OUTP

SHDN

INPUT

4.5V TO 5.5V

POS

12V 80mA

6.8µH

4.7µF

C6 10nF

10nF

RAMPNRAMPP

4.7µF

C4 1µF

L2 6.8µH

OPTIONAL ON LT3582-12

 

REG0/OTP0 = B0h REG1/OTP1 = D8h REG2/OTP2 = 03h

D1-D2: DIODES INC. B0540WS-7 L1-L2: COILCRAFT XPL2010-682 C1: 4.7µF, 6.3V, X5R, 0805

C2: 4.7µF, 16V, X5R, 0805 C3: 1s 4.7µF OR 2s 4.7µF OR 10µF 16V, X5R, 0805

C4: 1µF, 16V, X5R, 0603 C5-C6: 10nF, 0603

Space-Saving, Dual Output DC/DC Converter Yields Plus/Minus Voltage Outputs with (Optional) I2C Programming

Introduction

There are many applications that require both positive and negative DC voltages generated from a single input supply. The LT3582 is a highly integrated dual switching regulator that produces positive and negative voltages for AMOLEDs, CCDs, op amps, and general ±5V and ±12V supplies. The LT3582 uses a novel control scheme resulting in low output voltage ripple and high conversion efﬁciency over a wide load current range. The total solution size is very small due to the tiny 3mm × 3mm 16-pin QFN package, integrated feedback resistors, integrated loop compensation networks and the single-inductor negative output topology (see Figure 1).

The LT3582-5 and LT3582-12 are factory conﬁgured for accurate ±5V and ±12V outputs respectively, making it easy to squeeze a high performance solution into a small space. For other voltage combinations, the LT3582 offers I2C digitally programmable out- puts of 3.2V to 12.775V and –1.2V to –13.95V that can be made permanent

with on-chip OTP (One-Time-Programming) memory. The input supply range is 2.55V to 5.5V and switch current limits are 350mA and 600mA for the boost and inverting switches, respectively. In addition, the LT3582 features power up sequencing with ramping from ground to regulation, power down discharging, positive output disconnect and soft-start.

Accurate Output Voltages without External Feedback Resistors

The LT3582 series uses integrated feedback resistors to select the output voltages. The LT3582-5 and LT358212 are pre-conﬁgured at the factory for ±5V and ±12V outputs with ±1.5% accuracy or better. The LT3582 allows other output voltages to be conﬁgured using the I2C interface. There are nine bits to conﬁgure the positive output voltage from 3.2V to 12.775V in 25mV steps and another eight bits to conﬁgure the negative output voltage from –1.2V to –13.95V in 50mV steps. Default settings can be stored

by Mathew Wich

Figure 1. Dual output supplies in a small board footprint

in One-Time-Programmable memory and, if left unlocked, the voltages can be subsequently changed on the ﬂy using the I2C interface.

Great Performance Includes Low Ripple and High Efﬁciency Across the Load Range

The LT3582 is among several novel parts from Linear Technology that modulate peak switch current and switch off time to reduce ripple and improve light load efﬁciency (also see the LT3494, LT3495, LT8410 and

Figure 2. ±12V supplies from a single 5V input

Linear Technology Magazine • June 2009

DESIGN FEATURES L

SWP

OUTP

CAPP

LT3582 SERIES

DISCONNECT

CONTROL

LOAD

VOUTP

10mV/DIV

AC COUPLED

SWP

5V/DIV

0.2A/DIV

5µs/DIV

CAPP

2V/DIV

0.2A/DIV

RAMPP

0.2V/DIV

OUTP

2V/DIV

1ms/DIV

VOUTP

5V/DIV

VOUTN

5V/DIV

RAMPP

0.5V/DIV V

RAMPN

0.5V/DIV

5ms/DIV

RAMPN

RAMPP

1V/DIV

RAMPN

1V/DIV

VOUTN

5V/DIV

VOUTP

5V/DIV

5ms/DIV

VOUTP

10mV/DIV

AC COUPLED

SWP

5V/DIV

0.2A/DIV

200ns/DIV

VOUTP

10mV/DIV

AC COUPLED

SWP

5V/DIV

0.2A/DIV

2µs/DIV

Figure 3. Switching waveforms at 1mA load for the boost application shown in Figure 2

Figure 6. Power-Up Sequencing (V

followed by V

OUTP

OUTN

)

LT8415). Under light load conditions, the LT3582 chooses an optimum combination of frequency and peak switch current to improve efﬁciency while moderating the output ripple. Figures 3–5 show how the frequency and peak inductor current vary from light to heavy loads. At very light loads (typically < 1mA), peak switching currents are dramatically reduced to further reduce ripple when frequencies are in the audio band.

Figure 8. V

Linear Technology Magazine • June 2009

soft-start ramping from ground

OUTP

Figure 4. Switching waveforms at 10mA load for the boost application shown in Figure 2

Figure 7. Output disconnect PMOS

Adjustable Power-Up Sequencing and Soft-Start Options

The LT3582 has digitally conﬁgurable power-up sequencing that forces the outputs to power up in one of four ways:

ramps up ﬁrst, followed by

OUTP

(shown in ﬁgure 6)

OUTN

ramps up ﬁrst, followed by

OUTN

OUTP

both outputs ramp up

simultaneously

both outputs are disabled

The LT3582-5 and LT3582-12 are factory conﬁgured for both outputs to ramp up simultaneously.

The power-up ramp rates of the output voltages are also adjustable. Slowly ramping the outputs (also known as soft-start) reduces what would otherwise be high peak switching currents during start-up. Without soft-start, high start-up current is inherent in switching regulators due to V far from its ﬁnal value. The regulator tries to charge the output capacitors

OUT

Figure 5. Switching waveforms at 100mA load for the boost application shown in Figure 2

as quickly as possible, which results in large peak currents.

The output voltage ramp rates are proportional to the ramp rates of the RAMPP and RAMPN pin voltages. Upon chip enable, a programmable current (1µA, 2µA, 4µA or 8µA) linearly charges capacitors (typically about 10nF) connected to the RAMPP and RAMPN pins. By varying the capacitor sizes or charging currents, a wide range of output voltage ramp rates can be accommodated.

being

Figure 9. Power-down discharge waveforms

L DESIGN FEATURES

+ –

SWN

FBN

VCN VCP

OTP

RAMPN

RAMPP

OUTN

CAPP

GND

SHDN

CHIP ENABLE

222k

SCL

SDA

OTP

SWP CAPP V

OUTP

–

OTP ADJUST

0.80V

0.75V

CAPP

OUTP

OUTN

FBP

FBN

50mV

OUTPUT SEQUENCING

BY OTP

PEAK TOFF

CONTROL

PEAK TOFF

CONTROL

Q S

Q R

VARIABLE DELAY VARIABLE DELAY

–

DISCONNECT

CONTROL

SERIAL INTERFACE,

LOGIC AND OTP

OUTPUT SEQUENCING

0.80V

FBP

Output Disconnect and Improved Efﬁciency

The LT3582 series has a PMOS output disconnect switch connected between CAPP and V normal operation the switch is closed

(see Figure 7). During

OUTP

and current is limited to about 155mA to help protect against output shorts. During shutdown, the PMOS switch is open providing up to 5V–5.5V of isolation between CAPP and V cases this allows V to ground.

In normal operation, the output disconnect switch represents ~1.4Ω of resistance in series with the output leading to a 1%–2% efﬁciency loss under heavy load conditions. The CAPP pin can be externally shorted to the V

pin to eliminate the power loss

OUTP

in the switch and improve efﬁciency.

OUTP

to discharge

OUTP

. In most

Figure 10. LT3582 block diagram

Unique Ability to Ramp Output Up From Ground

Smart control of the output disconnect PMOS also gives the LT3582 the unique ability to generate a smooth V

voltage ramp starting from

OUTP

ground and continuing all the way up to regulation (see Figures 6 and 8). This ability is not possible with typical boost converters because the current path from VIN through the inductor (L1) and Schottky diode (D1) to the output prevents it from starting at 0V (see Figure 7).

The disconnect control circuitry in the LT3582 allows V ground when disabled. Once enabled, the gate of the output disconnect PMOS is precisely controlled such that V

rises smoothly from ground up

OUTP

to regulation where the PMOS is fully turned on to reduce power losses.

to discharge to

OUTP

Power Down Discharge Assist

The power down discharge feature assists in discharging the outputs after shutdown (see Figure 9). This option is factory enabled on the LT3582-5 and LT3582-12 and can be enabled through the I2C interface in conjunction with the “both together” power-up setting on the LT3582.

Upon SHDN falling and when power -down discharge is enabled, internal transistors activate to assist in discharging the outputs toward ground. After both outputs are within ~0.5V to ~1.5V of ground, the chip powers down.

Digital Control and One-Time Programming

The LT3582 series supports the Standard Mode I2C interface. Although using this interface is not required

Linear Technology Magazine • June 2009

DESIGN FEATURES L

LOAD CURRENT (mA)

0.1

EFFICIENCY (%)

POWER LOSS (mW)

350

300

250

200

150

100

1 10 100

VIN = 3.3V

LT3582

SWP

CAPP

NEG

–5V

90mA

CAPP

SDA

SCL

GND

SWN

OUTN

I2C

INTERFACE

OUTP

SHDN

INPUT

2.7V TO 4.2V

POS

4.6V 100mA

1.5µH

C2 10µF

C6 10nF

10nF

RAMPNRAMPP

C3 10µF

C1 10µF

C4 10µF

L2 1.5µH

REG0/OTP0 = 1Ch REG1/OTP1 = 4Ch REG2/OTP2 = 07h

D1-D2: PANASONIC M21D3800L LOW VF SCHOTTKY L1-L2: TDK MLP3216S1R5L C1-C4: TAIYO YUDEN JMK212BJ106MK, 6.3V, X5R 0805 C5-C6: 0402 X5R

VDS (V)

0.1

0.01

(A)

0.1

1 10 100

TA = 25°C MULTIPLE PULSE DUTY CYCLE = 0.2

10s

100ms

10ms

1ms

SET

(Ω)

CURRENT LIMIT THRESHOLD VALUE (A)

0.5

1.0

1.5

2.0

2.5

10k 100k 1M 10M

for the LT3582-5 or LT3582-12, it does permit reading of the chip’s conﬁguration and the ability to disable the power switches through the interface.

Additional I2C functionality is available with the LT3582 including re-programmability of the output voltages, and setting the power up sequencing and power down discharge.

A default power-up conﬁguration can be made permanent in the LT3582 through the One-Time-Programmable memory. The chip will always use the default conﬁguration from OTP memory upon power -up. Unless locked by programming a speciﬁc OTP memory bit, the chip conﬁguration can be changed after power-up by writing new settings through the I2C interface.

Conclusion

The LT3582 is an easy-to-use compact solution for DC/DC converter applications where positive and negative outputs are required. It is accurate, efﬁcient and includes an outsized number of features for its diminutive 3mm × 3mm 16-pin QFN package. It is offered in ±5V (LT3582-5), ±12V (LT3582-12) and I2C-programmable (LT3582) output versions.

LTC4217, continued from page 17

This example places a 20k resistor

on the I

pin to set the gain of the

MON

current monitor output to 1V per amp of MOSFET current.

Instead of tying the TIMER pin to the INTVCC pin for a default 2ms overcurrent timeout, an external

0.47µF capacitor is used to set a

5.7ms timeout. During an overcurrent event the external timing capacitor is charged with a100µA pull-up current. If the voltage on the capacitor reaches the 1.2V threshold, the MOSFET turns

Figure 6. Current limit adjustment

Linear Technology Magazine • June 2009

Figure 11. Tiny AMOLED power supply is 0.8mm (max) thin

off. The equation for setting timing capacitor’s value is as follows:

CT = TCB • 0.083(µF/ms)

extending the circuit breaker timeout beyond 2ms. The SOA graph for the MOSFET used in LTC4217 is shown in Figure 7. The worse case power

While the MOSFET is cooling off, the LTC4217 discharges the timing capacitor. When the capacitor voltage reaches 0.2V an internal 100ms timer is started. Following this cool down period the fault is cleared (when using auto-retry) and the MOSFET is allowed to turn on again.

It is important to consider the safe

dissipation occurs when the voltage versus current proﬁle of the foldback current limit is at maximum. This occurs when the current is 1A and the voltage is one half of the 12V or 6V (see Figure 4, FB pin at 0.7V). In this case the power is 6W, which dictates a maximum time of 100ms (Figure 7, at 6V and 1A).

operating area of the MOSFET when

Conclusion

The primary role of the LTC4217 is to control hot insertion and provide the electronic circuit breaker function. Additionally the part includes

Figure 7. MOSFET SOA curve

protection of the MOSFET with focus on SOA compliance, thermal protection and precise 2A current limit. It is also adaptable over a large range of applications due to adjustable inrush current, overcurrent fault timer and current limit threshold. A high level of integration makes the LTC4217 easy to use yet versatile.

L DESIGN IDEAS

0.3A

0.47µFC50.47µF

0.47µF

D1 D2 D3

5.6k

0Ω

5.6k

68.1k

2k 2k

ISN1

M1 M2 M3

330mΩ 330mΩ

TG1

43V to 58V

33µH

L2 33µH

33µH

OVP1

OVP2

ISP1

0.3A

ISN2

TG2

ISP2

0.3A

ISN3

C1 1µF s3

330mΩ

TG3

ISP3

5.6k

68.1k

OVP3

C1: MURATA GRM31CR72A105KA01L C4–C6: MURATA GRM21BR72A474KA73 D1–D3: DIODES B1100

L1–L3: WE 7447789133 M1–M3: ZETEX ZXMP6A13F M4–M6: PHILIPS BC858B

1nF

SW1 SW2

LT3492

GND

SW3

FADJ

TG1-3

OVP1-3

C1-3

REF

CTRL1-3

ISP1-3 ISN1-3 V

PWM1-3

SHDN

PWM1-3

SHDN

1µF

6.3V

430k

100k

8 LEDs 8 LEDs 8 LEDs

(1MHz)

14 LEDs

1µF

100V

1µF

100V

1µF

100V

9V TO 40V

60mA

1020k

20k

OVP1

ISN1

TG1

1.65Ω

2.2nF

22µH

ISP1

D1 D2 D3

M1 M2 M3

SW1 SW2

LT3492

GND

SW3

TG1-3

OVP1-3

C1-3

REF

CTRL1-3

ISP1-3 ISN1-3 V

PWM1-3

SHDN

PWM1-3

SHDN

C1–C4: MURATA GRM31CR72A105KA01L D1–D3: DIODES B1100

L1–L3: COILCRAFT MSS6132 M1–M3: ZETEX ZXMP6A13F

1µF

6.3V

14 LEDs

60mA

1020k

20k

OVP2

ISN2

TG2

1.65Ω

ISP2

14 LEDs

60mA

1020k

20k

OVP3

215k

18.2k

49.9k

ISN3

TG3

1.65Ω

ISP3

1µF

FADJ

(1MHz)

Triple Output LED Driver Works with Inputs to 60V and Delivers 3000:1 PWM Dimming

Introduction

The LT3492 is a 60V triple output LED driver for high input and/or high output voltage backlighting or direct lighting applications. A single 4mm × 5mm IC can drive a large number of LEDs, reducing overall solution cost when compared to less capable drivers. A built-in gate driver for a disconnect PMOS in series with the LED string, along with other techniques, enables a 3000:1 PWM dimming ratio. When coupled with the part’s analog dimming functions, the overall dimming ratio can be as high as 30,000:1. The LT3492 can be conﬁgured into buckmode, boost-mode or buck-boost mode, depending on the available input voltage source and the number and conﬁguration of LEDs to be driven.

High Input Voltage Triple Buck Mode LED Driver

Many “regulated” supplies actually have fairly loose tolerances. For ex-

Figure 1. Triple buck mode LED driver with open LED protection

by Hua (Walker) Bai

DESIGN IDEAS Triple Output LED Driver Works with

Inputs to 60V and Delivers 3000:1

PWM Dimming ...................................26

Hua (Walker) Bai

Bidirectional Power Manager Provides Efﬁcient Charging and Automatic USB On-The-Go with a Single Inductor

.........................................................28

Sauparna Das

Supercapacitor Charger with Adjustable Output Voltage and Adjustable Charging Current Limit

.........................................................30

Jim Drew

Monolithic Triple Output Converter for

Li-Ion Powered Handheld Devices ......32

Chuen Ming Tan

Ultralow Power Boost Converters Require Only 8.5µA of Standby

Quiescent Current .............................34

Xiaohua Su

15VIN, 4MHz Monolithic Synchronous Buck Regulator Delivers 5A in

4mm × 4mm QFN ...............................35

Tom Gross

Linear Technology Magazine • June 2009

Figure 2. Triple boost mode 60mA × 14 LED driver

DESIGN IDEAS L

PWM

5V/DIV

LED

50mA/DIV

1µs/DIV

9V TO 40V

2.2nF

SW1 SW2

LT3492

GND

SW3

ISP1-3 ISN1-3 V

PWM1-3

SHDN

PWM

SHDN

1µF

6.3V

4 LEDs

150mA

1150k

20k

OVP3

18.2k

ISN3

TG3

0.68Ω

27µH

ISP3

ISN2

0.68Ω

ISP2

2.2µF 25V

ISN1

D1 D2 D3

0.68Ω

ISP1

1150k

20k

OVP2

1150k

20k

OVP1

2.2µF 50V

4 LEDs

150mA

TG2TG1

4 LEDs

150mA

M1 M2 M3

C1: MURATA GRM31CR71H225KA88L C3, C5, C7: MURATA GRM21BR71E225K

D1–D3: DIODES B1100 L1–L3: COILCRAFT MSS6132 M1–M3: ZETEX ZXMP6A13F

FADJ

TG1-3

OVP1-3

C1-3

REF

CTRL1-3

215k

49.9k

(1MHz)

350

300

250

200

150

100

10 20 30 40 50 60

OUTPUT VOLTAGE (V)

OUTPUT CURRENT(mA)

PVIN= 9V, 85% EFFICIENCY

ample, a 48V supply can range between 43V and 58V, well above most LED drivers’ safe operating voltage ratings. The LT3492’s 60V input voltage rating makes it an easy ﬁt in such volatile voltage environments.

Figure 1 shows a triple buck-mode

LED driver for high voltage inputs. Each channel can drive up to eight 300mA white LEDs in series, a limit set by assuming 4V maximum forward voltage and a 43V minimum input voltage. Red LEDs or infrared LEDs have much lower forward voltage, therefore each output can drive as many as 20 infrared LEDs. The VIN pin in Figure 1 is tied to a 5V supply, as opposed to PVIN, to improve circuit efﬁciency.

Triple Boost Mode Driver Supports 14 LEDs per Output from a 9V–40V Input

Figure 2 shows a triple boost mode LED driver that delivers 60mA to each LED string. Due to the LT3492’s 60V switch rating, each output can support up to 14 LEDs. The 9V-to-40V input range covers a diverse range of applications, including regulated 12V, 24V, 32V to 36V, etc. Unlike in a buck mode regulator, where the output current capability is determined by the switch current limit, the current driving capability of a boost regulator

Figure 3. Maximum output current capability of an LT3492 boost circuit

is a function of the ratio of output voltage to minimum input voltage. Figure 3 shows the maximum output current vs output voltage for a 9V minimum input (assuming 85% efﬁciency at 1MHz). For applications that require less than 40V output, the LT3496 should be considered instead.

Triple Buck-Boost Mode LED Driver Regulates During Load Dump Events

Buck-boost mode is used when the LED string voltage falls within the input voltage range. Figure 4 shows a buck-boost application that uses one inductor per driver. The LED string is returned to the input—returning all LED strings to the same potential allows easy heat sinking. To prevent body diode conduction, the drain of the disconnect PMOS is tied to the

anode of the LED string. The high input voltage of the circuits in Figure 2 and Figure 4 is a real beneﬁt in automotive applications, where the ability to ride through 40V load dump events while maintaining LED current regulation is required. Figure 5 shows the greater than 3000:1 PWM dimming ratio achievable with the LT3492. This high PWM dimming ratio helps improve the picture quality of an LCD display under various dynamic conditions.

Conclusion

The LT3492 is a high voltage triple output LED driver with 60V rated switches, allowing high input voltage and/or high output voltage operations with accurate LED current. It can run in buck mode, boost mode or buckboost mode with 3000:1 PWM dimming capability.

Figure 5. High dimming ratio (>3000:1) improves quality of an LCD display

Linear Technology Magazine • June 2009

Figure 4. Triple buck-boost mode 150mA × 4 LED driver

2727

L DESIGN IDEAS

BUS

CURRENT (mA)

BUS

(V)

4.5

5.0

5.5

4.0

3.5

3.0 200

400

600100

300

500

700

BUS

= 4.75V

VBUS

= 500mA

OUT

= BAT = 3.8V

USB 2.0 SPECIFICATIONS REQUIRE THAT HIGH POWER DEVICES NOT OPERATE IN THIS REGION

LOAD CURRENT (mA)

CURRENT (mA)

250

500

750

800

–250

–500

200

400

600

1000

BUS

CURRENT

BATTERY CURRENT (CHARGING)

BUS

= 5V BAT = 3.8V 5x MODE

BATTERY CURRENT

(DISCHARGING)

BUS

USB

ON-THE-GO

3.3µH

10µF

0.1µF

3.01k 1k

CLPROG PROG

LTC4160/

LTC4160-1

OUT

BAT

Li-Ion

OVGATE

OVSENS

OPTIONAL

OVERVOLTAGE

PROTECTION

SYSTEM

LOAD

6.2k

10µF

Bidirectional Power Manager Provides Efﬁcient Charging and Automatic USB On-The-Go with a Single Inductor

by Sauparna Das

Introduction

Imagine that your car won’t start—the battery is dead, the kids are getting fussy, you’re stranded in the middle of nowhere, and your cell phone won’t turn on because you forgot to charge it. What do you do now? Fortunately, you remember that your new camera is in the car, and it has a fully charged battery. Even better, this camera supports USB On-The-Go using a bidirectional power manager. You connect a USB micro-AB cable between the cell phone and camera and instantly start charging your phone. The phone powers up and you’re able to call for help.

The LTC4160 is a versatile, high efﬁciency power manager and battery charger that incorporates a bidirectional switching regulator, full featured battery charger, an ideal diode (with a controller for an optional external ideal diode), and an optional overvoltage protection circuit. The bidirectional switching regulator is able to power a portable system and charge its battery or provide a 5V output for USB OnThe-Go using a single inductor (Figure

1). This reduces component count and board space, key attributes for a power management IC in today’s feature rich portable devices. In shutdown, the part only draws 8µA of current, thus maximizing battery life.

Bidirectional Switching Power Path for USB On-The-Go

The LTC4160 contains a bidirectional switching regulator between V and V V

BUS

a step down converter and provides power to the application and battery charger (Figure 1). The switching regulator includes a precision average input current limit with multiple settings. Two of the settings correspond to the USB 100mA and 500mA limits.

The bidirectional switching regulator is able to power a portable system and charge

ly 300mV above the battery when the switcher is not in input current limit and the battery voltage is above 3.3V. This technique, known as Bat-Track output control, provides very efﬁcient charging, which minimizes loss and heat and eases thermal constraints. For battery voltages below 3.3V, V regulates to 3.6V when the switcher

. When power is applied to

OUT

,the switching regulator acts as

its battery or provide a 5V

output for USB On-The-Go

using a single inductor.

The voltage on V

is approximate-

OUT

is not in input current limit. This instant-on feature provides power to the system even when the battery is

BUS

completely discharged.

prioritized over charging the battery. If the combined system load and charge current exceed the current available at the input, the battery charger reduces its charge current to maintain power to the application. If the load alone exceeds the input current limit, then additional current is supplied by the battery via the ideal diode(s).

the bidirectional switching regulator steps up the voltage on V 5V on V ing regulator is capable of delivering at least 500mA. Power to V from the battery via the ideal diode(s). A precision output current limit circuit, similar to the one in step-down mode, prevents a load on V drawing more than 680 mA (Figure 1). The switching regulator also features true output disconnect which prevents body diode conduction of the PMOS switch. This allows V volts during a short circuit condition or while shut down, drawing zero current from the battery. When V

OUT

3.2V, the LTC4160 allows a portable

Power to the application is always

For USB On-The-Go applications,

to produce

. In this mode the switch-

BUS

OUT

BUS

to go to zero

BUS

OUT

comes

from

is ≥

Figure 1. The LTC4160 provides bidirectional power transfer. Left plot: V currents vs load current when input power is available.

voltage vs V

BUS

current in On-The-Go mode. Right plot: battery and V

BUS

Linear Technology Magazine • June 2009

BUS

DESIGN IDEAS L

TO µC

ENCHARGER

ENOTG

NTCBIAS

LIM0

LIM1

TO USB

TRANSCEIVER

MICRO-AB

BUS

USB

ON-THE-GO

3.3µH

C3 22µF 0805

0.1µF 0402

17 8

3.01k 1k

CLPROG

NTC PROG

LTC4160/LTC4160-1

VBUSGD

CHRG

FAULT

OUT

IDGATE

BAT

BUS

–

GND

LDO3V3

RTC

SYSTEM LOAD

GND

Li-ion

1µF

OVGATE

OVSENS

C1, C3: TAYIO YUDEN JMK212BJ226MG J1: HIROSE ZX62-AB-5PA L1: COILCRAFT LPS4018-332LM

M1: FAIRCHILD FDN372S M2: SILICONIX Si2333DS R1: 1/10 WATT RESISTOR

BUS

POWERS UP WHEN ID PIN HAS LESS THAN 10Ω TO GND (MICRO-A PLUG CONNECTED)

6.2k

10k

C1 22µF 0805

TO µC

product to meet the speciﬁcation for a high power USB device by maintaining V

above 4.75V for currents up

BUS

to 500mA.

Automatic USB On-The-Go

When two On-The-Go devices are connected, one is the A-device and the other is the B-device, depending on the orientation of the cable, which has a micro-A and a micro-B plug. The A-device provides power to the B-device and starts as the host. MicroA/micro-B cables include an ID pin in addition to the four standard pins (V

, D–, D+, and GND)—the micro-

BUS

A plug has its ID pin shorted to GND while on the micro-B plug the ID pin is ﬂoating. The impedance on the ID pin allows the USB power manager to determine whether it receives power from an external device or whether it should power up V to an external device.

Step-up mode can be enabled by either the ENOTG pin or the ID pin. The ENOTG pin can be connected to a microcontroller. The ID pin, on the other hand, is designed to be connected directly to the ID pin of a micro-AB receptacle. The pin is active low and contains an internal 2.5µA pull up current source. When the ID pin is ﬂoating or a micro-B plug is connected to the AB receptacle, the internal current source pulls ID up to the max of V

, V

BUS

and BAT. When a micro-A

OUT

to provide power

BUS

plug is connected to the receptacle, the short between ID and ground in the micro-A plug overrides the pull-up current source and pulls the ID pin on the LTC4160 down to ground. This activates the bidirectional switching regulator in step-up mode and powers up V

. A complete application

BUS

schematic is shown in Figure 2.

Other Features

The LTC4160 also includes a battery charger featuring programmable charge current (1.2A max), cell preconditioning with bad cell detection and termination, CC-CV charging, C/10 end of charge detection, safety timer termination, automatic recharge and a thermistor signal conditioner for temperature qualiﬁed charging. For the LTC4160, the nominal ﬂoat voltage is 4.2V. The LTC4160-1 provides a nominal ﬂoat voltage of 4.1V.

The overvoltage protection circuit can be used to protect the low voltage USB/Wall adapter input from the inadvertent application of high voltage or a failed wall adapter. This circuit controls the gate of an external high voltage N-channel MOSFET, and in conjunction with an external 6.2k resistor, can provide protection up to 68V.

The LTC4160 includes an integrated ideal diode and a controller for an optional external ideal diode. This provides a low loss power path from

the battery to V

when input power

OUT

is limited or unavailable. When input power is removed, the ideal diode(s) prevent V

from collapsing, with

OUT

only the output capacitor required for the switching regulator.

Conclusion

The LTC4160 is a feature rich power manager that is especially suited for USB On-The-Go applications, enabling bidirectional USB power transfer between portable devices. The part can directly detect the impedance on the ID pin of a micro-AB receptacle to automatically tell the internal bidirectional switching regulator to provide a 5V output on V The switching regulator can supply at least 500mA and comes with a current limit of 680mA. In addition, the LTC4160 can efﬁciently take power from 5V inputs (USB, Wall adapter, etc.) to power a portable application and charge its battery using a single inductor. Its unique switching architecture and Bat-Track output control provides fast and efﬁcient charging. Furthermore, an optional overvoltage protection circuit can provide protection against voltages of up to 68V on the V

pin. The combination of bi-

BUS

directional power transfer, automatic USB On-The-Go functionality and high voltage protection make the LTC4160 a must have for today’s high end portable devices.

for USB On-The-Go.

BUS

Linear Technology Magazine • June 2009

Figure 2. LTC4160 with automatic USB On-The-Go

2929

L DESIGN IDEAS

C N V V

N T

CHARGE

EFF FC UV

RECHARGE

( )

• –

•

GND

OUT

LT3663 BUCK

GND

TOP RAIL

TOP ENA

TOP RAIL

–

BOT RAIL

BOT ENA

GND

BIAS GND TOP CAP

TOP CAP

–

BOT CAP

–

GND

CAP

12V

GND

BOT

50F

TOP

50F

RUN

GND

OUT

LT3663 BUCK

CONTROL

CIRCUIT

(FIGURE 3)

GND

RUN

ENA

LIM

RUN

OUT

2.65V

1.2A

SENSE

BOOST

IR05H40CSPTR

3.3µH

L1: TDK VLCF5020T-3R3N1R7-1

LT3663

GND FB

4.7µF 50V 1206

40V

2.0A DFLS240

10k

ILIM

28.7k

0.1µF 16V

47µF 1206 16V

FB2

86.6k

FB1

200k

Supercapacitor Charger with Adjustable Output Voltage and Adjustable Charging Current Limit

Introduction

For applications using larger value supercapacitors (tens to hundreds of farads), a charger circuit with a relatively high charging current is needed to minimize the recharge time of the system. Supercapacitors are used as energy hold up devices in applications such as solid state RAID disks, where information stored in high speed volatile memory must be transferred to non-volatile ﬂash memory when power is lost. This transfer time may take minutes, requiring hundreds of farads to hold up the power supply until the transfer is complete. The requirement for the recharge time of these banks of supercapacitors is typically less than one hour. To accomplish this, a high charging current is required. This article describes a supercapacitor charging circuit using the LT3663 that meets these difﬁcult requirements.

The LT3663 is a 1.2A, 1.5MHz stepdown switching regulator with output current limit ideal for supercapacitor applications. The part has an input voltage range of 7.5V to 36V,has adjustable output voltage and adjustable output current limit. The output voltage is set with a resistor divider network in the feedback loop while the output current limit is set by a single resistor connected from the I

Figure 2. Capacitor charger circuit using the LT3663

Figure 1. Block diagram for charging two supercapacitors in series

pin to ground. With its internal compensation network and internal boost diode, the LT3663 requires a minimal number of external components.

Power Ride-Through Application

A procedure for selecting the size of the supercapacitor is outlined in the September 2008 edition of Linear Technology, in an article titled “Replace Batteries in Power Ride-Through Applications with Supercaps and 3mm × 3mm Capacitor Charger.” The procedure determines the effective supercapacitor (C

0.3Hz, based on the power level to

LIM

) capacitance at

EFF

by Jim Drew

be held up, the minimum operating voltage of the DC/DC converter supporting the load, the distributed circuit resistances including the ESR of the supercapacitors, and the required hold up time.

Once the size of the supercapacitor is known, the charging current can be determined to meet the recharge time requirements. The recharge time (T

RECHARGE

to recharge the supercapacitors from the minimum operating voltage (VUV) of the DC/DC converter to the full charge voltage (VFC) of the supercapacitors. The voltage on the individual supercapacitors at the start of the recharge cycle is the minimum operating voltage divided by the number (N) of supercapacitors in series. From here on, this article describes an application with two supercapacitors in series. The recharge current (I is determined by the capacitor charge control law:

This assumes that the voltage across the supercapacitor doesn’t discharge below the VUV/N value. This assumption is valid if the time period

Linear Technology Magazine • June 2009

) is the time required

)

CHARGE

DESIGN IDEAS L

C V

CHARGE

EFF FC

CHARGE

•

IN OUT

BIAS

(FIGURE 1

CONNECTIONS)

(FIGURE 1 CONNECTIONS)

GND

BYP

GND

–

SHDN

LT1761-3.3

3.3V

10µF

16V

0.01µF

10µF

TOP ENA

TOP CAP +

TOP CAP

–

150k

REF

LT1634-1.25

BOT_CHRG_L

100k 100k

OUT A V+

–IN A OUT B

–IN B

+IN B

+IN A

GND

0.1µF

–

LT1784

–

3.3V

0.01µF

BOT CAP

–

100k 100k

–

LT1784

–

3.3V

0.01µF

100k

10k

100k

R2 402k

R1 10k

10k

R4 402k

100k 100k

10k

–

LT1784

0.01µF

TOP RAIL

–

4N25

BOT ENA

2N7002

10k

GND

LT6702

while input power isn’t available is such that the supercapacitor’s leakage current hasn’t signiﬁcantly reduced the voltage across the capacitor. The voltage across the supercapacitor may actually rise slightly after the DC/DC converter shuts down due to the dielectric absorption effect. The initial charge time T

for a fully discharged

CHARGE

bank of supercapacitors is:

Figure 1 shows a block diagram of the components for this supercapacitor charger application.

Charging Circuit Using the LT3663

To set the charging current, a resistor R

is connected from the I

ILIM

the LT3663 to ground. Table 1 shows the nominal charging currents for various values of R

The full charge voltage is set by the resistor divider network in the

ILIM

pin of

LIM

Figure 3. Charger control circuit

feedback loop. Table 2 shows various full charge voltages versus the value of R ground) when resistor R tied between the V pin) is 200k. Figure 2 shows the charging circuit for each supercapacitor.

Control Circuit

(resistor from the FB pin to

FB2

pin and the FB

OUT

(resistor

FB1

for Charging Supercapacitors

The control circuit in Figure 3 is used to balance the voltages of the supercapacitors while they are charging. This is accomplished by prioritizing

Charging Current (A)

Table 1. Charging current vs R

0.4 140

0.6 75

Value (kΩ)

ILIM

0.8 48.7

1.0 36.5

1.2 28.7

ILIM

charge current to the lower voltage supercapacitor—speciﬁcally by enabling the charging circuit for the supercapacitor with the lower voltage while disabling the circuit for the other supercapacitor.

If the top charging circuit is enabled while the bottom charging circuit is disabled, the bottom supercapacitor is charged by the input return current from the top charger. This return current is a fraction of the charging current so the top supercapacitor charges faster. The control circuit

Full Charge Voltage (V)

Table 2. Full charge voltage vs R

2.65 86.6

2.5 93.1

2.4 100

2.2 115

2.0 133

continued on page 33

FB2

(kΩ)

FB2

Linear Technology Magazine • June 2009

3131

L DESIGN IDEAS

IN1

IN2

SW2

OUT1

FB2

FB1

SW3

LTC3521

SHDN1 SHDN2 SHDN3

102k

22pF

137k

68.1k

OUT3

10µF

OUT2

10µF

22pF

4.7µF

Li-ion

4.7µH

6.8µH

OUT2

1.8V 600mA

1.5M

332k

OUT1

22µF

OUT1

3.3V 1A

2.4V TO 4.2V

100k

PGOOD1 PGOOD2 PGOOD3

PWM

SW1A

SW1B

FB3

OUT1

: TDK C3216X5R0J226M

OUT2

, C

OUT3

: TDK C2012X5R0J106M L1: Sumida CDRH5D16NP-4R7NC L2, L3: Sumida CDRH2D18/LD-6R8NC

PGND2GNDPGND1

PWM

BURST

OFF

OUT3

1.2V

600mA

EFFICIENCY (%)

LOAD CURRENT (mA)

1k1

100

BUCK-BOOST MODE BUCK-BOOST MODE, BURST MODE OPERATION BUCK MODE BUCK MODE, BURST MODE OPERATION

VIN = 3.6V

OUT

= 3.3V

OUT

= 1.8V

Monolithic Triple Output Converter for Li-Ion Powered Handheld Devices

Introduction

Handheld devices often require several voltage rails to power microprocessors, communication I/O and other peripherals with a range of voltages from as low as 1V to as high as 3.3V. Producing these voltage rails from a single-cell Li-Ion battery requires multiple converters that are efﬁcient, can operate in a combination of buck and boost modes, and ﬁt into the already crowded board space of the handheld device. To meet these challenges, the LTC3521 triple-output converter combines a buck-boost converter and two synchronous buck converters in a 4mm × 4mm QFN package (Figure

1). The LTC3521 is a monolithic de-

vice internally compensated and with built-in soft-start capacitors. External components are limited to feedback resistors, output inductors and capacitors (Figure 2). The internal switching frequency of 1MHz makes it possible to select a wide range of tiny, low proﬁle capacitors and inductors. A complete 3-output converter occupies less than

0.5in2 (Figure 1), delivering up to 5W

of total output power with less than a 15°C temperature rise.

Figure 1. Buck-boost and buck converters occupy less than 0.5in2 of board space.

The LTC3521’s wide input voltage range allows its buck-boost converter to operate from 1.8V to 5.5V. Its proprietary buck-boost switching algorithm makes it possible to produce seamless transitions between buck and boost modes, using only a single inductor in a ﬁxed frequency operation. Smooth buck-boost transitions are especially useful in 3.3V applications where battery run time depends on using the entire 2.4V–4.2V operating range of a single cell Li-Ion battery. The

by Chuen Ming Tan

buck-boost converter can support up to 1A loads.

The LTC3521 buck converters feature internally compensated current mode control that ensures a rapid transient response over a wide range of output capacitor values. The buck converters can supply a load current of up to 600mA each over the entire input voltage range and its output can be set as low as 0.6V. The buck converter transitions smoothly to 100% duty cycle operation to extend battery life in low dropout operation. Other useful features of LTC3521 include short circuit protection, individual open-drain power good indicator, which allows for undervoltage fault detection, and sequenced start-up. Each converter can be independently enabled. With all converters disabled, the total supply current is reduced to under 1µA.

High Efﬁciency

Due to its high efﬁciency, the LTC3521 is able to operate in a tiny package, delivering 1A of output current on the buck-boost converter and 600mA each on the buck converters. As shown in Figure 3, all the converters can easily operate at efﬁciencies above 90% in PWM mode. Peak efﬁciency

Figure 2. A 3-output converter takes a single Li-Ion battery to

3.3V at 1A (buck-boost mode), 1.8V at 600mA and 1.2V at 600mA.

Figure 3. Efﬁciency vs load current for the circuit in Figure 2

Linear Technology Magazine • June 2009

occurs at the midpoint of the avail-

OUT1

100mV/DIV

OUT2

100mV/DIV

OUT3

100mV/DIV

100µs/DIV

OUT1

1V/DIV

OUT2

1V/DIV

OUT3

1V/DIV

500µs/DIV

ENABLE TOP

ENABLE BOTTOM

ENABLE PINS

5V/DIV

TOP

500mV/DIV

BOT

500mV/DIV

5s/DIV

TOP

= 50F

BOT

= 50F

INITIAL V

TOP = INITIAL VCBOT

ENABLE PINS

5V/DIV

TOP

500mV/DIV

BOT

500mV/DIV

5s/DIV

TOP

= 50F

BOT

= 100F

INITIAL V

TOP < INITIAL VCBOT

ENABLE TOP

ENABLE BOTTOM

able output current range—ensuring high efﬁciency under most operating conditions. When the application enters low power mode, the converters can be independently set to Burst Mode operation to further improve efﬁciency at light loads. In Burst Mode operation, the total quiescent current of the converters is reduced to 35µA. During noise critical phases, Burst Mode operation can be temporarily forced to low noise by dynamically driving the PWM pin high.

Supply Sequencing

Digital applications with multiple supplies typically specify sequenced start-up and shut-down of the supplies. Supply sequencing is important to prevent powering up I/O pins that are driven by unpowered core logic. Without deﬁned logic states, erratic ﬂuctuations may occur at the I/O pins. LTC3521 provides individual control of shutdown and PGOOD indicator pins, which can be used for supply sequencing. The three outputs of LTC3521 can be powered sequentially by tying the SHDN and PGOOD pins

Figure 4. Sequenced power-up waveforms

as shown in Figure 2. A low-to-high transition on SHDN3 pin powers up channel 3. When channel 3 is powered up, PGOOD3 pulls SHDN2 high to turn on channel 2. When channel 2 is powered up and PGOOD2 is high, SHDN1 is pulled high, ﬁnally turning on all three outputs (Figure 4).

Inter-Channel Performance

While in PWM mode, all three converters operate synchronously from a common 1MHz oscillator. This minimizes the interaction between the converters so that load steps on the output of one converter have minimum impact on the others. For example, Figure 5 shows the output voltages as two

DESIGN IDEAS L

Figure 5. Alternating light to full load step responses show little crosstalk between channels.

separate 20mA to 600mA load steps are applied to the buck channels and a 0A to 1A load step is applied to the buck-boost channel. In this case, even with small 10µF output capacitors on the buck converters and 22µF on the buck-boost converter, the interaction among channels is minimal.

Conclusion

The LTC3521 provides a highly integrated monolithic solution for applications requiring multiple voltage rails in a compact footprint. Its high efﬁciency and exceptional performance make the LTC3521 well suited for even the most demanding handheld applications.

LT3663, continued from page 31

consists of a 3.3V LDO (U6) and a precision 1.25V reference (U7). U1 and U2 are conﬁgured as difference ampliﬁers with a gain of one to measure the voltage across each supercapacitor while U3 is a level shifted difference ampliﬁer used to determine the voltage difference between the two supercapacitors. By level shifting the output of U3 to the reference voltage, the two comparators in U4 determine which supercapacitor needs charging.

An additional pair of level shifting resistors (R14 and R15, R16 and R17) are used to allow both supercapacitors to charge when they are within a 50mV window. When both supercapacitors are being charged, the bottom supercapacitor charges faster because it is being charged by its charging current plus the input return current of the top charger. This effect can be seen in Figure 4. The enable signal of the bottom charger is toggling as the bot-

Linear Technology Magazine • June 2009

tom supercapacitor is being charged faster than the top supercapacitor to maintain the 50mV difference between the two supercapacitors. Figure 5 shows the effect of a 2-to-1 mismatch in capacitance value where the top is a 50F supercapacitor and the bottom is a 100F. Here the voltage on the bottom supercapacitor rises more slowly and the top supercapacitor charger enable signal toggles to allow it to maintain

Conclusion

The LT3663 allows for a low component count supercapacitor charging circuit with adjustable full charge voltage and adjustable current limit ideal for larger value supercapacitors. A control circuit can monitor and balance the voltage across each supercapacitor, even if the supercapacitors are grossly mismatched in capacitance or initial voltage.

voltage balance.

Figure 4. Charging with equal value capacitors Figure 5. Charging with mismatched capacitors

3333

L DESIGN IDEAS

SW CAP

GND

CHIP ENABLE

FBP

LT8410

2.2µF

0.1µF

100µH

0.1µF

0.1µF*

OUT

= 16V

2.5V to 16V

CCVOUT

REF

SHDN

604K

412K

*HIGHER VALUE CAPACITOR IS REQUIRED WHEN THE VIN IS HIGHER THAN 5V

LOAD CURRENT (mA)

0.01

EFFICIENCY (%)

100

100100.1 1

VIN = 12V

VIN = 5V

VIN = 3.6V

LOAD CURRENT (mA)

0.01

OUT

PEAK-TO-PEAK RIPPLE (mV)

1010.1

VIN = 3.6V

Ultralow Power Boost Converters Require Only 8.5µA of Standby Quiescent Current

Introduction

Industrial remote monitoring systems and keep-alive circuits spend most of their time in standby mode. Many of these systems also depend on battery power, so power supply efﬁciency in standby state is very important to maximize battery life. The LT8410/-1 high efﬁciency boost converter is ideal for these systems, requiring only

8.5µA of quiescent current in standby mode. The device integrates high value (12.4M/0.4M) output feedback resistors, signiﬁcantly reducing input current when the output is in regulation with no load. Other features include an integrated 40V switch and Schottky diode, output disconnect with current limit, built in soft-start, overvoltage protection and a wide input range, all in a tiny 8-pin 2mm × 2mm DFN package.

by Xiaohua Su

Figure 1. 2.5V–16V To 16V boost converter

Application Example

Figure 1 details the LT8410 boost converter generating a 16V output from a 2.5V-to-16V input source. The LT8410/-1 controls power delivery by varying both the peak inductor current and switch off time. This control scheme results in low output voltage ripple as well as high efﬁciency over a wide load range. Figures 2 and 3 show efﬁciency and output peak-to-peak ripple for Figure 1’s circuit. Output ripple voltage is less than 10mV despite the circuit’s small (0.1µF) output capacitor.

The soft-start feature is implemented by connecting an external capacitor to the V is not needed, the capacitor can be removed. Output voltage is set by a resistor divider from the V ground with the center tap connected to the FBP pin, as shown in Figure 1. The FBP pin can also be biased directly by an external reference.

pin. If soft-start

REF

pin to

REF

Figure 2. Efﬁciency vs load current for Figure 1 converter

The SHDN pin of the LT8410/-1 can serve as an on/off switch or as an undervoltage lockout via a simple resistor divider from VCC to ground.

Ultralow Quiescent Current Boost Converter with Output Disconnect

Low quiescent current in standby mode and high value integrated feedback resistors allow the LT8410/-1 to regulate a 16V output at no load from a

3.6V input with about 30µA of average input current. Figures 4, 5 and 6 show typical quiescent and input currents in regulation with no load.

The device also integrates an output disconnect PMOS, which blocks the output load from the input during shutdown. The maximum current

Figure 3. Output peak-to-peak ripple vs load current for Figure 1 converter at 3.6V

through the PMOS is limited by circuitry inside the chip, allowing it to survive output shorts.

Compatible with High Impedance Batteries

A power source with high internal impedance, such as a coin cell battery, may show normal output on a voltmeter, but its voltage can collapse under heavy current demands. This makes it incompatible with high current DC/DC converters. With very low switch current limits (25mA for the LT8410 and 8mA for the LT8410-1), the LT8410/-1 can operate very efﬁciently from high impedance sources without causing inrush current problems. This feature also helps preserve battery life.

continued on page 36

Linear Technology Magazine • June 2009

DESIGN IDEAS L

RT TRACK MODE SGND PGND

PGOOD

SW SW SW SW SW SW

BOOST

0.1µF 25V

0.33µH

2.2µF

0.1µF

0.1µF 25V

71.5k

10.0k

20.0k

C3 47µF

6.3V

C4 47µF

6.3V

OUT

1.8V 5A

C2 22µF 16V

22µF

16V

12V

RUN

CLKIN CLKOUT

PHMODE

100k

10Ω

LTC3605

ITH INTV

PGND

∆V

100mV/DIV

2A/DIV

20µs/DIV

SW1

10V/DIV

SW2

10V/DIV

500ns/DIV

5A/DIV

15V

, 4MHz Monolithic

Synchronous Buck Regulator Delivers 5A in 4mm × 4mm QFN

Introduction

The LTC3605 is a high efﬁciency, monolithic synchronous step-down switching regulator that is capable of delivering 5A of continuous output current from input voltages of 4V to 15V. Its compact 4mm × 4mm QFN package has very low thermal impedance from the IC junction to the PCB, such that the regulator can deliver maximum power without the need of a heat sink. A single LTC3605 circuit can power a 1.2V microprocessor directly from a 12V rail—no need for an intermediate voltage rail.

The LTC3605 employs a unique controlled on-time/constant frequency current mode architecture, making it ideal for low duty cycle applications and high frequency operation. There are two phase-lock loops inside the LTC3605: one servos the regulator on-time to track the internal oscillator frequency, which is determined by an external timing resistor, and the other servos the internal oscillator to an external clock signal if the part is synchronized. Due to the controlled ontime design, the LTC3605 can achieve very fast load transient response while minimizing the number and value of external output capacitors.

The LTC3605’s switching frequency is programmable from 800kHz to 4MHz, or the regulator can be synchronized to an external clock for noise-sensitive applications.

Furthermore, multiple LTC3605s can be used in parallel to increase the available output current. The LTC3605 produces an out-of-phase clock signal so that parallel devices can be interleaved to reduce input and output current ripple. A multiphase, or PolyPhase®, design also generates lower high frequency EMI noise than a single-phase design, due to the lower switching currents of each phase. This conﬁguration also helps with the thermal design issues normally associated with a single high output current device.

1.8V

OUT

Buck Regulator

The LTC3605 is speciﬁcally designed for high efﬁciency at low duty cycles such as 12VIN-to-1.8VOUT at 5A, as

Figure 1. 12V to 1.8V at 5A buck converter operating at 2.25MHz

shown in Figure 1. High efﬁciency is achieved with a low R synchronous MOSFET switch (35mΩ) and a 70mΩ R

DS(ON)

MOSFET switch.

This circuit runs at 2.25MHz, which reduces the value and size of the output capacitors and inductor. Even with the high switching frequency, the efﬁciency of this circuit is about 80% at full load.

Figure 2 shows the fast load transient response of the application circuit shown in Figure 1. It takes only 10µs to recover from a 4A load step with less than 100mV of output

, 2.25MHz

voltage deviation and only two 47µF ceramic output capacitors. Note that compensation is internal, set up by tying the compensation pin (ITH) to the internal 3.3V regulator rail (INTVCC).

Tom Gross

bottom

DS(ON)

top synchronous

Linear Technology Magazine • June 2009

Figure 2. Load step response of the circuit in Figure 1

Figure 3. Multiphase operation waveforms of the circuit in Figure 4. The switch voltage and inductor ripple currents operate 180° out of phase with respect to each other.

3535

L DESIGN IDEAS

TRACK CLKOUT RT MODE SGND

BOOST

VON

FILT1

0.1µF

RUN PGND

ITH

CLKIN

FILT1

10Ω

PGOOD

100k

PHMODE INTV

LTC3605

RT MODE SGND PGND

PHMODE

INTV

BOOST

RUN PGOOD CLKOUT

ITH

TRACK CLKIN

LTC3605

162k

FB1

10.0k

162k

FB2

10.0k

ITH

FILT2

10Ω

FILT2

0.1µF

IN2

22µF

BST1

0.1µF

0.33µH

BST1

OUT1

47µF

OUT

1.2V 10A

INTVCC2

2.2µF

BST2

0.1µF

OUT2

47µF

INTVCC1

2.2µF

10pF

0.1µF

IN1

22µF

12V

ITH

390pF

BST2

VCC VOLTAGE (V)

QUIESCENT CURRENT (µA)

161284

OUTPUT VOLTAGE (V)

AVERAGE INPUT CURRENT (µA)

1000

100

4020 3010

VCC = 3.6V

TEMPERATURE (°C)

–40

QUIESCENT CURRENT (µA)

12080400

VCC = 3.6V

This connects an internal series RC to the compensation point of the loop, while introducing active voltage positioning to the output voltage: 1.5% at no load and –1.5% at full load. The hassle of using external components for compensation is eliminated. If one wants to further optimize the loop, and remove voltage positioning, an external RC ﬁlter can be applied to the ITH pin.

1.2V

Several LTC3605 circuits can run in parallel and out of phase to deliver high total output current with a minimal amount of input and output capacitance—useful for distributed power systems.

The 1.2V ulator shown in Figure 4 can support 10A of output current. Figure 3 shows the 180° out-of-phase operation of the two LTC3605s. The LTC3605 requires no external clock device to operate up to 12 devices synchronized out of phase—the CLKOUT and CLKIN pins of the devices are simply cascaded, where each slave’s CLKIN pin takes the CLKOUT signal of its respective master. To produce the required phase offsets, simply set the voltage level on

LT8410, continued from page 34

Conclusion

The LT8410/-1 is a smart choice for applications which require low standby quiescent current and/or require low input current, and is especially suited for power supplies

Figure 4. Quiescent current vs temperature (not switching)

, 10A, 2-Phase Supply

OUT

2-phase LTC3605 reg-

OUT

Figure 4. 12V to 1.2V at 10A 2-phase buck converter

the PHMODE pin of each device to INTVCC, SGND or INTVCC/2 for 180°, 120° or 90° out-of-phase signals, respectively, at the CLKOUT pin.

LTC3605s can run in parallel to produce 60A of output current. PolyPhase operation can also be used in multiple output applications to lower the amount of input ripple current,

Conclusion

The LTC3605 offers a compact, monolithic, regulator solution for high current applications. Due to its PolyP h ase capability, up to 12

with high impedance sources. The ultralow quiescent current and high value integrated feedback resistors keep average input current very low, signiﬁcantly extending battery oper-

Figure 5. Quiescent current vs VCC voltage (not switching)

reducing the necessary input capacitance. This feature, plus its ability to operate at input voltages as high as 15V, make the LTC3605 an ideal part for distributed power systems.

ating time. The LT8410/-1 is packed with features without compromising performance or ease of use and is available in a tiny 8-pin 2mm × 2mm package.

Figure 6. Average input current in regulation with no load

Linear Technology Magazine • June 2009

New Device Cameos

NEW DEVICE CAMEOS L

Ultralow Power, 14-Bit 150Msps ADC Reduces Digital Feedback in Data Conversion Systems

The LTC2262 is a low power 14-bit, 150Msps Analog-to-Digital Converter (ADC) that dissipates only 149mW, less than one-third the power of competitive solutions. This new benchmark enables portable applications limited by stringent power budgets to extend their performance capabilities, as well as providing higher operating efﬁciency and reduced recurring operating costs for 3G/4G LTE and WiMAX basestation equipment. In addition to offering considerably lower power, the LTC2262 integrates two unique features for reducing digital feedback in situations where even good layout practice may fail. These features in combination with low power ease the task of designing with high speed ADCs in a wide variety of applications, including portable medical imaging and ultrasound, portable test and instrumentation, non-destructive test equipment, software deﬁned radios and cellular basestations.

Digital feedback occurs when energy from ADC outputs couples back into the analog section, causing interaction that appears as odd shaping in the noise ﬂoor and spurs in the ADC output spectrum. The worst situation is at midscale, where all outputs are changing from ones to zeroes, or vice versa, generating large ground currents that couple back into the input.

To combat this effect, the LTC2262’s proprietary alternate bit polarity (ABP) mode inverts all of the odd bits before the output buffers to equalize the number of ones and zeroes switching. This method effectively cancels the large ground plane currents that contribute to digital feedback. In addition to the alternate bit polarity mode, an optional data output randomizer is also available for reducing interference from the digital outputs. The randomizer decorrelates the digital output to reduce the likelihood of repetitive

code patterns that couple back into the ADC input, causing unwanted tones in the output spectrum. Both digital feedback reduction techniques have proven to improve spurious free dynamic range (SFDR) performance by 10dB –15dB.

Operating from a low 1.8V analog supply, the LTC2262 achieves signiﬁcant power savings without sacriﬁcing AC performance. This ADC offers signal to noise ratio (SNR) performance of 72.8dB and spurious free dynamic range (SFDR) of 88dB at baseband. Ultralow jitter of 0.17ps undersampling of IF frequencies with excellent noise performance.

The LTC2262’s innovative digital outputs can be set to full rate CMOS, double data rate CMOS, or double data rate LVDS. Double data rate digital outputs allow data to be transmitted on both the rising edge and the falling edge of the clock, reducing the number of data lines needed by half. A separate output power supply allows the CMOS output swing to range from

1.2V to 1.8V.

Offered in a 6mm × 6mm QFN package, the LTC2262 includes a clock duty cycle stabilizer circuit to facilitate non-50% clock duty cycles, programmable digital output timing, programmable LVDS output current and optional LVDS output termination. These features combine to make the data transmission between the ADC and the digital receiver more ﬂexible.

allows

RMS

Quad 12-Bit/10-Bit/8-Bit DACs Include 10ppm/°C Reference

The LTC2634 quad 12-bit, 10-bit and 8-bit rail-to-rail digital-to-analog converters (DACs) integrate a precision reference in tiny 3mm × 3mm QFN and MSOP packages. The LTC2634 is the latest offering in Linear’s family of tiny 12-bit, 10-bit and 8-bit DACs with internal references. The LTC2634 joins the previously released LTC2636 octal and LTC2630/LTC2640 single channel DACs, offering a versatile

selection of the smallest DACs for numerous applications.

The LTC2634’s small size and internal reference is important for a variety of industrial, automotive and ATE applications. By integrating a 10ppm/°C reference, the LTC2636 offers further space reduction for compact circuit boards. The LTC2634 offers 12-bit performance of ±2.5LSB (max) INL error and less than 2.4nV•s crosstalk, ensuring that a voltage change on one DAC has minimal effect on the other DACs. Operating from a single 2.7V to

5.5V supply, supply current is a low 125µA per DAC.

The LTC2634 DACs are available in a number of ordering options to meet a wide range of applications. In addition to selecting one of three resolution options, designers can also choose between a 2.5V or 4.096V full-scale range. Ordering options provide the choice between powering up the DACs at zero-scale or mid-scale, offering ﬂexibility for designs that cannot be forced to ground when power is ﬁrst applied. Designers can choose between an MSOP-10 package or a 16-pin 3mm × 3mm QFN package that includes a hardware load-DAC (LDAC) pin, a clear pin that asynchronously forces the DAC outputs to their respective reset state, and a serial data output pin.

10MHz to 6GHz Low Power RMS Detector with 40dB Dynamic Range for Accurate RF Power Measurement

The LT5581 is a broadband 6GHz RMS detector, featuring 40dB dynamic range and a low operating supply cur rent of 1.4mA. The device is well suited for a wide range of power monitor and control applications in portable and battery-powered wireless systems, cellular basestations, picocells and femtocells, ﬁber optic transmitters and instrumentation. The LT5581 outputs a DC voltage that is linearly proportional to the log input power, providing an easy-to-use, mV/dB scaling with exceptional linearity of better than ±1dB across 40dB range.

Linear Technology Magazine • June 2009

3737

L NEW DEVICE CAMEOS

CURRENT (A)

0.5

0.6

0.7

0.8

0.9

1.0

0.3

0.4

0.1

0.2

0.5

0.6

0.7

0.8 0.9

1.0

0.3

0.4

0.1

0.2

INPUT

SUPPLY

CURRENT

VIN = 24V

CHARGER

INPUT

CURRENT

SYSTEM LOAD CURRENT (A)

VIN (V)

OUT(MAX)

(A)

1.0

1.2

1.4

1.6

1.8

2.0

0.6

0.8

0.2

0.4

0.25

0.30

0.35

0.40

0.45

0.50

0.15

0.20

0.05

0.10

16 181210 32 4014

24 26 3022

OUT(MAX)

INPUT

The LT5581’s RMS measurement capability provides accurate RF power readings to within ±0.2dB regardless of waveforms that have high crest-factor modulated content, multicarrier or multitone. Moreover, the LT5581 offers exceptional accuracy of ±1dB over its operating temperature range of –40°C to 85°C.

Operating over a wide supply voltage range of 2.7V to 5.25V, the LT5581’s low power consumption makes it ideal for battery-powered communication

LT3650, continued from page 7

monitored input supply become excessive. The CLP pin can be conﬁgured to implement an input current limit function for systems having multiple loads that share the LT3650 VIN supply. The LT3650 reduces maximum battery charge current if the voltage on the CLP pin exceeds the voltage on VIN by 50mV. Total load current on the input power supply can be monitored by connecting a sense resistor from the CLP pin to VIN, and connecting any external loads to the VIN pin. The LT3650 servos the charger maximum output current such that 50mV is maintained across the CLP sense resistor.

A Full Complement of Battery Charger Features

Figure 7 shows a battery charger that incorporates many of the LT3650’s unique features. This charger incor-

and multimedia devices. Yet, it has the accuracy performance to meet the performance required by basestations, picocells and femtocells, cable infrastructure and optical communication systems. Additionally, the LT5581’s wide frequency range extends to applications including WiMAX and wireless systems in the 5GHz ISM bands. The LT5581’s single-ended RF input does not require an external RF transformer, thus simplifying the application design while reducing costs.

porates top off charging with a 3-hour backup safety timer, and directly accepts input voltages from 12V to 40V (32V operating maximum). This charger uses a 9.1V Zener diode to level-shift the input supply, incorporating an undervoltage lockout function for VIN < 10V.

Battery pack temperature-sensing is enabled by connecting an NTC thermistor to the NTC pin. Charging is suspended if the battery temperature does not remain within a 0°C to 40°C range. The charger uses a resistor divider to modulate the voltage on RNG/SS, which reduces the maximum battery charge current if VIN is below 20V, useful for current-limited input sources such as wall adapters. A capacitor on the RNG/SS pin enables a soft-start function. A secondary system load is supported, with the input supply protected by an input

The LT5581 has a fast response time of 1µs rise time to a full power swing, suitable for time-division duplexing systems.

The LT5581 also incorporates a shutdown feature. When the LT5581’s Enable input pin is pulled low, the chip draws a typical shutdown current of

0.2µA, and a maximum of 6µA. The device is offered in a tiny 8-lead, 3mm × 2mm DFN surface mount package.

current limit feature, incorporated by connecting the input supply to the CLP pin via a 0.05Ω sense resistor. The maximum charge current is automatically reduced to keep the total input supply current from exceeding the 1A limit set by the sense resistor.

Conclusion

The LT3650 provides a versatile and easy-to-use platform for a wide variety of efﬁcient Li-Ion battery charger solutions. Low power dissipation makes continuous charging up to 2A practical, deriving power directly from input supplies up to 32V without the need for an intermediate DC/DC converter. The compact size of the IC coupled with modest external component requirements allows construction of space saving, cost-effective, and feature-rich Li-Ion battery chargers.

Figure 8. Charger maximum input current (IIN) and maximum output current (I

7. Charge current reduction for V supply current below 0.5A

OUT(MAX)

) vs VIN for the battery charger shown in Figure

< 20V keeps the charger input

Figure 9. Charger maximum input current, system load current, and total input supply current for the battery charger shown in Figure 7 for V 24V. Battery charger output current is reduced to maintain a maximum input supply current of 1A, which corresponds to 50mV across the 0.05Ω resistor that is connected between the CLP and V

Linear Technology Magazine • June 2009

pins of the LT3650.

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Linear Technology Corp. Ltd. Room 816, 8/F China Electronics Plaza B No. 3 Dan Ling Rd., Hai Dian District Beijing, 100080, PRC Tel: +86 (10) 6801-1080 Fax: +86 (10) 6805-4030

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Linear Technology GmbH

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Linear Technology GmbH

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Linear Technology GmbH Jesinger Strasse 65 D-73230 Kirchheim/Teck Germany Tel: +49 (0)7021 80770 Fax: +49 (0)7021 807720

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Linear Technology (UK) Ltd.

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Linear Technology Corporation

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Milpitas, CA 95035-7417

www.linear.com

Linear Technology Magazine • June 2009

LINEAR TECHNOLOGY LT3757 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual

Two New Controllers for Boost, Flyback, SEPIC and Inverting DC/DC Converters Accept Inputs up to 100V

by Wei Gu

Linear in the News…

DESIGN FEATURES L

Charge Li-Ion Batteries Directly from High Voltage Automotive and Industrial Supplies Using Standalone Charger in a 3mm × 3mm DFN

by Jay Celani

Power Management IC Combines USB On-The-Go and USB Charging in Compact Easy-to-Use Solution

by George H. Barbehenn and Sauparna Das

Power Management IC with Pushbutton Control Generates Six Voltage Rails from USB or 2 AA Cells Via Low Loss PowerPath Topology

Improve Hot Swap Performance and Save Design Time with Hot Swap Controller that Integrates 2A MOSFET and Sense Resistor

by David Soo

by Terry J. Groom

Space-Saving, Dual Output DC/DC Converter Yields Plus/Minus Voltage Outputs with (Optional) I2C Programming

by Mathew Wich

L DESIGN IDEAS

Triple Output LED Driver Works with Inputs to 60V and Delivers 3000:1 PWM Dimming

by Hua (Walker) Bai

by Sauparna Das

Supercapacitor Charger with Adjustable Output Voltage and Adjustable Charging Current Limit

by Jim Drew

Monolithic Triple Output Converter for Li-Ion Powered Handheld Devices

by Chuen Ming Tan

Ultralow Power Boost Converters Require Only 8.5µA of Standby Quiescent Current

by Xiaohua Su

Tom Gross

New Device Cameos

NEW DEVICE CAMEOS L