LINEAR TECHNOLOGY LT3693 Technical data

FEATURES

LT3693

36V, 3.5A, 2.4MHz

Step-Down Switching Regulator

U

DESCRIPTIO

■

Wide Input Range: 3.6V to 36V

■

3.5A Maximum Output Current

■

Adjustable Switching Frequency: 200kHz to 2.4MHz

■

Low Shutdown Current: IQ < 1μA

■

Integrated Boost Diode

■

Synchronizable Between 250kHz to 2MHz

■

Power Good Flag

■

Saturating Switch Design: 95m On-Resistance

■

0.790V Feedback Reference Voltage

■

Output Voltage: 0.79V to 30V

■

Thermal Protection

■

Soft-Start Capability

■

Small 10-Pin Thermally Enhanced MSOP and

(3mm × 3mm) DFN Packages

U

APPLICATIO S

■

Automotive Battery Regulation

■

Power for Portable Products

■

Distributed Supply Regulation

■

Industrial Supplies

■

Wall Transformer Regulation

The LT®3693 is an adjustable frequency (200kHz to

2.4MHz) monolithic buck switching regulator that accepts
input voltages up to 36V. A high effi ciency 95m switch
is included on the die along with a boost Schottky diode
and the necessary oscillator, control, and logic circuitry.
Current mode topology is used for fast transient response
and good loop stability. Shutdown reduces input supply
current to less than 1μA while a resistor and capacitor on
the RUN/SS pin provide a controlled output voltage ramp
(soft-start). A power good fl ag signals when V

reaches

OUT

91% of the programmed output voltage. The LT3693 is
available in 10-Pin MSOP and 3mm × 3mm DFN packages
with exposed pads for low thermal resistance.

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners.

TYPICAL APPLICATIO

5V Step-Down Converter

V

IN

6.3V TO 36V

V

OFF ON

15k

10mF

680pF

63.4k

RUN/SS BOOST

V

C

RT

PG

SYNC

IN

U

LT3693

GND

BD

SW

Effi ciency

V

OUT

5V

3.5A

0.47mF

4.7mH

100k

536k

47mF

3693 TA01a

FB

100

90

80

70

EFFICIENCY (%)

60

V

= 5V

OUT

L = 4.7μH
f = 600kHz

50

0 0.5

VIN = 12V

1.5

VIN = 34V

2

VIN = 24V

1

OUTPUT CURRENT (A)

2.5

3

3.5

3693 G01

3693f

1

LT3693

WW

W

ABSOLUTE AXI U RATI GS

U

(Note 1)

VIN, RUN/SS Voltage .................................................36V

BOOST Pin Voltage ...................................................56V

BOOST Pin Above SW Pin .........................................30V

FB, RT, V

Voltage .......................................................5V

C

PG, BD Voltage .........................................................30V

SYNC Voltage ............................................................20V

PIN CONFIGURATION

TOP VIEW

10

BD

1

BOOST

2

11

3

SW

4

V

IN

5

RUN/SS

10-LEAD (3mm × 3mm) PLASTIC DFN

EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB

DD PACKAGE

θJA = 45°C/W, θJC = 10°C/W

RT

9

V

C

FB

8

7

PG

6

SYNC

Operating Junction Temperature Range (Note 2)

LT3693E ............................................. –40°C to 125°C

LT3693I.............................................. –40°C to 125°C

Storage Temperature Range ................... –65°C to 150°C

Lead Temperature (Soldering, 10 sec)

(MSE Only) ....................................................... 300°C

TOP VIEW

10

1

BD

2

BOOST

SW
V

IN

RUN/SS

10-LEAD PLASTIC MSOP

EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB

θJA = 45°C/W, θJC = 10°C/W

3
4
5

MSE PACKAGE

11

RT

9

V

C

FB

8

PG

7

SYNC

6

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT3693EDD#PBF LT3693EDD#TRPBF LDGB 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C
LT3693IDD#PBF LT3693IDD#TRPBF LDGB 10-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C

LT3693EMSE#PBF LT3693EMSE#TRPBF LTDFZ 10-Lead Plastic MSOP –40°C to 125°C

LT3693IMSE#PBF LT3693IMSE#TRPBF LTDFZ 10-Lead Plastic MSOP –40°C to 125°C

Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based fi nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi cations, go to: http://www.linear.com/tapeandreel/

ELECTRICAL CHARACTERISTICS

The ● denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T

= 25°C. VIN = 10V, V

A

noted. (Note 2)

PARAMETER CONDITIONS MIN TYP MAX UNITS

Minimum Input Voltage

V

Quiescent Current from V

Quiescent Current from BD V

IN

= 0.2V 0.01 0.5 μA

RUN/SS

= 3V, Not Switching

V

BD

= 0, Not Switching 1.3 2.3 mA

V

BD

= 0.2V 0.01 0.5 μA

RUN/SS

= 3V, Not Switching

V

BD

RUN/SS

= 10V, V

= 15V, VBD = 3.3V unless otherwise

BOOST

●

●

●

3 3.6 V

0.45 1.2 mA

0.9 1.8 mA

3693f

2

LT3693

ELECTRICAL CHARACTERISTICS

The ● denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T
noted. (Note 2)

PARAMETER CONDITIONS MIN TYP MAX UNITS

Minimum Bias Voltage (BD Pin) 2.7 3 V

Feedback Voltage

FB Pin Bias Current (Note 3) V

FB Voltage Line Regulation 4V < V

Error Amp g

Error Amp Gain 2000

Source Current 60 μA

V

C

Sink Current 60 μA

V

C

Pin to Switch Current Gain 5.3 A/V

V

C

Clamp Voltage 2.0 V

V

C

Switching Frequency R

Minimum Switch Off-Time

Switch Current Limit Duty Cycle = 5% 4.6 5.4 6.0 A

Switch V

Boost Schottky Reverse Leakage V

Minimum Boost Voltage (Note 4)

BOOST Pin Current I

RUN/SS Pin Current V

RUN/SS Input Voltage High 2.5 V

RUN/SS Input Voltage Low 0.2 V

PG Threshold Offset from Feedback Voltage VFB Rising 65 mV

PG Hysteresis 10 mV

PG Leakage V

PG Sink Current V

SYNC Low Threshold 0.5 V

SYNC High Threshold 0.8 V

SYNC Pin Bias Current V

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.

Note 2: The LT3693E is guaranteed to meet performance specifi cations
from 0°C to 125°C. Specifi cations over the –40°C to 125°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LT3693I specifi cations are
guaranteed over the –40°C to 125°C temperature range.

m

CESAT

= 25°C. VIN = 10V, V

A

= 0, Not Switching 110 μA

V

BD

= 0.8V, VC = 0.4V

FB

< 36V 0.002 0.01 %/V

IN

= 8.66k

T

R

= 29.4k

T

R

= 187k

T

ISW = 3.5A 335 mV

= 10V, VBD = 0V 0.02 2 μA

SW

= 1A 35 60 mA

SW

= 2.5V 5 8 μA

RUN/SS

= 5V 0.1 1 μA

PG

= 0.4V

PG

= 0V 0.1 μA

SYNC

Note 3: Bias current fl ows out of the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed to

guarantee full saturation of the switch.

RUN/SS

= 10V, V

= 15V, VBD = 3.3V unless otherwise

BOOST

780
775

●

●

2.2

1.0

200

●

●

●

200 800 μA

790
790

10 40 nA

525 μMho

2.45

1.1

230

60 150 nS

1.5 2.0 V

800
805

2.7

1.25
260

mV
mV

MHz
MHz

kHz

3693f

3

LT3693

UW

TYPICAL PERFOR A CE CHARACTERISTICS

= 25°C unless otherwise noted.

T

A

Effi ciency

100

90

80

70

EFFICIENCY (%)

60

V

OUT

L = 4.7μH
f = 600kHz

50

0 0.5

VIN = 24V

= 5V

1

OUTPUT CURRENT (A)

Maximum Load Current

5.5

5.0

4.5

4.0

3.5

LOAD CURRENT (A)

V
T

3.0

L = 4.7μH
f = 600kHz

2.5
5

TYPICAL

MINIMUM

= 3.3V

OUT

= 25°C

A

10 20

INPUT VOLTAGE (V)

15

1.5

VIN = 12V

VIN = 34V

2

2.5

25 30

3

3693 G01

3693 G06

3.5

Effi ciency

100

90

80

70

EFFICIENCY (%)

60

V

OUT

L = 3.3μH
f = 600kHz

50

0 0.5

VIN = 12V

VIN = 24V

= 3.3V

1

OUTPUT CURRENT (A)

Maximum Load Current

5.5

5.0

4.5

4.0

LOAD CURRENT (A)

V

= 5V

OUT

3.5

= 25°C

T

A

L = 4.7μH
f = 600kHz

3.0
10 20

5

VIN = 34V

2

1.5

15

INPUT VOLTAGE (V)

2.5

TYPICAL

MINIMUM

3.5

3

3693 G02

25 30

3693 G07

Effi ciency

100

90

80

70

EFFICIENCY (%)

VIN = 12V

60

= 5V

V

OUT

L = 4.7μH
f = 600kHz

50

0 0.5

OUTPUT CURRENT (A)

Switch Current Limit

6.0

5.5

5.0

4.5

4.0

SWITCH CURRENT LIMIT(A)

3.5

3.0
0

2

1.5

1

20 60

40

DUTY CYCLE (%)

2.5

3.5

3

3693 G03

80 100

3.0

2.5

TOTAL POWER LOSS (W)

2.0

1.5

1.0

0.5

3693 G08

Switch Current Limit

6.5

6.0

5.5

5.0

4.5

4.0

3.5

3.0

SWITCH CURRENT LIMIT (A)

2.5

2.0
–50 25–25 0 50 75 100 150125

DUTY CYCLE = 10 %

DUTY CYCLE = 90 %

TEMPERATURE (°C)

4

3693 G09

Switch Voltage Drop

700

600

500

400

300

VOLTAGE DROP (mV)

200

100

0

0

12 45

SWITCH CURRENT (A)

Boost Pin Current

120

105

90

75

60

45

30

BOOST PIN CURRENT (mA)

15

3

3693 G10

0

0312 45

SWITCH CURRENT (A)

3693 G11

3693f

UW

0

TYPICAL PERFOR A CE CHARACTERISTICS

LT3693

TA = 25°C unless otherwise noted.

Feedback Voltage

840

820

800

780

FEEDBACK VOLTAGE (mV)

760

–50 25–25 0 50 75 100 150125

TEMPERATURE (°C)

Minimum Switch On-Time

140

120

100

80

60

40

MINIMUM SWITCH ON TIME (ns)

20

3693 G12

Switching Frequency

1.20
RT = 34.0k

1.15

1.10

1.05

1.00

0.95

FREQUENCY (MHz)

0.90

0.85

0.80

–50 25–25 0 50 75 100 150125

TEMPERATURE (oC)

Soft-Start

7

6

5

4

3

2

SWITCH CURRENT LIMIT (A)

1

3693 G13

Frequency Foldback

1200

RT = 34.0k

1000

800

600

400

SWITCHING FREQUENCY (kHz)

200

0

0

200 400

100 300

FB PIN VOLTAGE (mV)

RUN/SS Pin Current

12

10

8

6

4

RUN/SS PIN CURRENT (μA)

2

500

700 900

600

800

3693 G14

0

–50 25–25 0 50 75 100 150125

TEMPERATURE (°C)

Boost Diode

1.4

1.2

1.0

(V)

F

0.8

0.6

BOOST DIODE V

0.4

0.2

0

0

0.5 1.0 1.5

BOOST DIODE CURRENT (A)

3693 G15

3693 G18

2.0

0

0.5 1 2

0

1.5

RUN/SS PIN VOLTAGE (V)

Error Amp Output Current

50

40

30

20

10

0

–10

PIN CURRENT (μA)

C

–20

V

–30

–40

–50

–200

–100 100

FB PIN ERROR VOLTAGE (mV)

2.5 3 3.5

3693 G16

0 200

3693 G19

0

0

510

RUN/SS PIN VOLTAGE (V)

Minimum Input Voltage

5.0

4.5

4.0

3.5

3.0

INPUT VOLTAGE (V)

V

= 3.3V

OUT

2.5

= 25oC

T

A

L = 4.7MH
f = 600kHz

2.0
10 100 1000

1

LOAD CURRENT (mA)

20 30 35

15 25

3693 G17

1000

3693 G20

3693f

5

LT3693

0

UW

TYPICAL PERFOR A CE CHARACTERISTICS

TA = 25°C unless otherwise noted.

Minimum Input Voltage

6.5

6.0

5.5

5.0

INPUT VOLTAGE (V)

V

= 5V

OUT

4.5

= 25 oC

T

A

L = 4.7MH
f = 600kHz

4.0
1 1000

10 100 1000

LOAD CURRENT (mA)

Switching Waveforms;
Discontinuous Operation

V

SW

5V/DIV

I

L

0.2A/DIV

3693 G21

VC Voltages

2.50

2.00

1.50

VOLTAGE (V)

1.00

C

V

0.50

CURRENT LIMIT CLAMP

SWITCHING THRESHOLD

0

–50 25–25 0 50 75 100 150125

TEMPERATURE (°C)

3693 G22

Switching Waveforms;
Continuous Operation

V

SW

5V/DIV

I

L

0.5A/DIV

Power Good Threshold

95

90

85

80

THRESHOLD VOLTAGE (%)

75

–50 25–25 0 50 75 100 150125

TEMPERATURE (°C)

3693 G23

V

OUT

10mV/DIV

VIN = 12V
V

= 3.3V

OUT

I

= 110mA

LOAD

1μs/DIV

3693 G25

V

OUT

10mV/DIV

VIN = 12V
V

= 3.3V

OUT

I

= 1A

LOAD

1μs/DIV

3693 G26

3693f

6

UUU

PI FU CTIO S

LT3693

BD (Pin 1): This pin connects to the anode of the boost
Schottky diode. BD also supplies current to the internal
regulator.

BOOST (Pin 2): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.

SW (Pin 3): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.

(Pin 4): The VIN pin supplies current to the LT3693’s

V

IN

internal regulator and to the internal power switch. This
pin must be locally bypassed.

RUN/SS (Pin 5): The RUN/SS pin is used to put the
LT3693 in shutdown mode. Tie to ground to shut down
the LT3693. Tie to 2.5V or more for normal operation. If
the shutdown feature is not used, tie this pin to the V

IN

pin. RUN/SS also provides a soft-start function; see the
Applications Information section.

SYNC (Pin 6): This is the external clock synchronization
input. Ground this pin when not used. Tie to a clock source
for synchronization. Clock edges should have rise and
fall times faster than 1μs. Do not leave pin fl oating. See
synchronizing section in Applications Information.

PG (Pin 7): The PG pin is the open collector output of an
internal comparator. PG remains low until the FB pin is
within 9% of the fi nal regulation voltage. PG output is valid
when V

is above 3.6V and RUN/SS is high.

IN

FB (Pin 8): The LT3693 regulates the FB pin to 0.790V.
Connect the feedback resistor divider tap to this pin.

(Pin 9): The VC pin is the output of the internal error

V

C

amplifi er. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.

RT (Pin 10): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.

Exposed Pad (Pin 11): Ground. The Exposed Pad must
be soldered to PCB.

BLOCK DIAGRA

V

V

IN

IN

4

C1

RUN/SS

5

RT

10

R

T

SYNC

6

SOFT-START

PG

7

GND

11 8

W

INTERNAL 0.79V REF

ERROR AMP

+

0.7V

–

FB

R1

R2

–

BOOST

SW

BD

1

2

3

V

C

9

C3

L1

D1

C

C

C

F

R

C

C2

3693 BD

V

OUT

+

SLOPE COMP

5

OSCILLATOR

200kHzTO2.4MHz

+
–

SWITCH

LATCH

R

S

VC CLAMP

Q

3693f

7

LT3693

OPERATION

The LT3693 is a constant frequency, current mode step­down regulator. An oscillator, with frequency set by RT,
enables an RS fl ip-fl op, turning on the internal power
switch. An amplifi er and comparator monitor the current
fl owing between the V
off when this current reaches a level determined by the
voltage at V
voltage through an external resistor divider tied to the FB
pin and servos the V
increases, more current is delivered to the output; if it
decreases, less current is delivered. An active clamp on the

pin provides current limit. The VC pin is also clamped to

V

C

the voltage on the RUN/SS pin; soft-start is implemented
by generating a voltage ramp at the RUN/SS pin using an
external resistor and capacitor.

An internal regulator provides power to the control circuitry.
The bias regulator normally draws power from the V
but if the BD pin is connected to an external voltage higher
than 3V bias power will be drawn from the external source
(typically the regulated output voltage). This improves

. An error amplifi er measures the output

C

and SW pins, turning the switch

IN

pin. If the error amplifi er’s output

C

pin,

IN

effi ciency. The RUN/SS pin is used to place the LT3693
in shutdown, disconnecting the output and reducing the
input current to less than 0.5μA.

The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for effi cient opera­tion.

The oscillator reduces the LT3693’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup
and overload.

The LT3693 contains a power good comparator which trips
when the FB pin is at 91% of its regulated value. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3693 is
enabled and V

is above 3.6V.

IN

8

3693f

APPLICATIONS INFORMATION

LT3693

FB Resistor Network

The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resis­tors according to:

RR

12

⎛
⎜

⎝

079

OUT

.

⎞

1=

–

⎟

⎠

V

V

Reference designators refer to the Block Diagram.

Setting the Switching Frequency

The LT3693 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2.4MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.

SWITCHING FREQUENCY (MHz) RT VALUE (kΩ)

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.0

1.2

1.4

1.6

1.8

2.0

2.2

2.4

Figure 1. Switching Frequency vs. RT Value

215
140
100

78.7

63.4

53.6

45.3

39.2
34

26.7

22.1

18.2
15

12.7

10.7

9.09

Operating Frequency Tradeoffs

Selection of the operating frequency is a tradeoff between
effi ciency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower effi ciency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (f

SW(MAX)

) for a

given application can be calculated as follows:

VV

+

D OUT

–

+

()

()

DINSW

f

SW MAX

=

()

tVVV

ON MIN

where VIN is the typical input voltage, V
voltage, V

is the catch diode drop (~0.5V) and VSW is the

D

is the output

OUT

internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high V

IN/VOUT

ratio. Also, as shown
in the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
switching frequency is because the LT3693 switch has fi nite
minimum on and off times. The switch can turn on for a
minimum of ~150ns and turn off for a minimum of ~150ns.
Typical minimum on time at 25°C is 80ns. This means that
the minimum and maximum duty cycles are:

DC f t

DC f t

where fSW is the switching frequency, the t
minimum switch on time (~150ns), and the t

=

MIN SW

MAX SW

ON MIN

=

1–

()

OFF MIN

()

ON(MIN)

OFF(MIN)

is the

is
the minimum switch off time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.

A good choice of switching frequency should allow ad­equate input voltage range (see next section) and keep
the inductor and capacitor values small.

Input Voltage Range

The maximum input voltage for LT3693 applications
depends on switching frequency and Absolute Maxi­mum Ratings of the V

and BOOST pins (36V and 56V

IN

respectively).

While the output is in start-up, short-circuit, or other
overload conditions, the switching frequency should be
chosen according to the following equation:

VV

+

V

IN MAX

()

where V
V

IN(MAX)

is the output voltage, VD is the catch diode drop

OUT

(~0.5V), V
load), f
t

ON(MIN)

SW

is the minimum switch on time (~100ns). Note that

=

ft

SW

OUT D

ON MIN

()

VV

+–

DSW

is the maximum operating input voltage,

is the internal switch drop (~0.5V at max

SW

is the switching frequency (set by RT), and

a higher switching frequency will depress the maximum

3693f

9

LT3693

APPLICATIONS INFORMATION

operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve safe operation at
high input voltages.

If the output is in regulation and no short-circuit, start­up, or overload events are expected, then input voltage
transients of up to 36V are acceptable regardless of the
switching frequency. In this mode, the LT3693 may enter
pulse skipping operation where some switching pulses
are skipped to maintain output regulation. In this mode
the output voltage ripple and inductor current ripple will
be higher than in normal operation.

The minimum input voltage is determined by either the
LT3693’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:

VV

+

V

IN MIN

()

where V

IN(MIN)

=

1–

OUT D

ft

SW

OFF MIN

()

VV

+

–

DSW

is the minimum input voltage, and t

OFF(MIN)

is the minimum switch off time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.

Inductor Selection

For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔI

increases with higher VIN or V

L

OUT

and decreases with higher inductance and faster switch­ing frequency. A reasonable starting point for selecting
the ripple current is:

= 0.4(I

ΔI

L

where I

OUT(MAX)

OUT(MAX)

)

is the maximum output load current. To
guarantee suffi cient output current, peak inductor current
must be lower than the LT3693’s switch current limit (I

LIM

).

The peak inductor current is:

I

L(PEAK)

= I

OUT(MAX)

+ ΔIL/2

ripple current. The LT3693’s switch current limit (I

LIM

) is

5.5A at low duty cycles and decreases linearly to 4.5A at
DC = 0.8. The maximum output current is a function of
the inductor ripple current:

= I

I

OUT(MAX)

LIM

– ΔIL/2

Be sure to pick an inductor ripple current that provides
suffi cient maximum output current (I

OUT(MAX)

).

The largest inductor ripple current occurs at the highest

. To guarantee that the ripple current stays below the

V

IN

specifi ed maximum, the inductor value should be chosen
according to the following equation:

⎛

VV

OUT D

L

=

⎜

fI

Δ

⎝

SW L

⎛

⎞

+

VV

1–

⎜

⎟
⎠

⎝

OUT D

V

()

IN MAX

⎞

+

⎟⎟
⎠

where VD is the voltage drop of the catch diode (~0.4V),
V

IN(MAX)

voltage, f

is the maximum input voltage, V

is the switching frequency (set by RT), and

SW

is the output

OUT

L is in the inductor value.

The inductor’s RMS current rating must be greater than
the maximum load current and its saturation current
should be about 30% higher. For robust operation in fault
conditions (start-up or short circuit) and high input volt­age (>30V), the saturation current should be above 5A.
To keep the effi ciency high, the series resistance (DCR)
should be less than 0.05

, and the core material should
be intended for high frequency applications. Table 1 lists
several vendors and suitable types.

Table 1. Inductor Vendors

VENDOR URL PART SERIES TYPE

Murata www.murata.com LQH55D Open

TDK www.componenttdk.com SLF10145 Shielded

Toko www.toko.com D75C

D75F

Sumida www.sumida.com CDRH74

CR75
CDRH8D43

NEC www.nec.com MPLC073

MPBI0755

Shielded
Open

Shielded
Open
Shielded

Shielded
Shielded

where I

L(PEAK)

is the peak inductor current, I

the maximum output load current, and ΔI

10

OUT(MAX)

is the inductor

L

is

3693f

APPLICATIONS INFORMATION

LT3693

Of course, such a simple design guide will not always re­sult in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
load is lower than 3.5A, then you can decrease the value
of the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher effi ciency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details of
maximum output current and discontinuous mode opera­tion, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (V
is a minimum inductance required to avoid subharmonic
oscillations. See AN19.

Input Capacitor

Bypass the input of the LT3693 circuit with a ceramic
capacitor of X7R or X5R type. Y5V types have poor
performance over temperature and applied voltage, and
should not be used. A 10μF to 22μF ceramic capacitor is
adequate to bypass the LT3693 and will easily handle the
ripple current. Note that larger input capacitance is required
when a lower switching frequency is used. If the input
power source has high impedance, or there is signifi cant
inductance due to long wires or cables, additional bulk
capacitance may be necessary. This can be provided with
a lower performance electrolytic capacitor.

Step-down regulators draw current from the input sup­ply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3693 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 10μF capacitor is capable of this task, but only if it is
placed close to the LT3693 and the catch diode (see the
PCB Layout section). A second precaution regarding the

OUT/VIN

> 0.5), there

ceramic input capacitor concerns the maximum input
voltage rating of the LT3693. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT3693 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT3693’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safety section).

For space sensitive applications, a 4.7μF ceramic capaci­tor can be used for local bypassing of the LT3693 input.
However, the lower input capacitance will result in in­creased input current ripple and input voltage ripple, and
may couple noise into other circuitry. Also, the increased
voltage ripple will raise the minimum operating voltage
of the LT3693 to ~3.7V.

Output Capacitor and Output Ripple

The output capacitor has two essential functions. Along
with the inductor, it fi lters the square wave generated by the
LT3693 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3693’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:

C

OUT

where fSW is in MHz, and C
output capacitance in μF. Use X5R or X7R types. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with
a higher value capacitor if the compensation network is
also adjusted to maintain the loop bandwidth. A lower
value of output capacitor can be used to save space and
cost but transient performance will suffer. See the Fre­quency Compensation section to choose an appropriate
compensation network.

100

=

Vf

OUT SW

is the recommended

OUT

3693f

11

LT3693

APPLICATIONS INFORMATION

Table 2. Capacitor Vendors

VENDOR PHONE URL PART SERIES COMMANDS

Panasonic [phone] www.panasonic.com Ceramic,

Polymer,
Tantalum

Kemet [phone] www.kemet.com Ceramic,

Tantalum T494, T495

Sanyo [phone] www.sanyovideo.com Ceramic,

Polymer,
Tantalum

Murata [phone] www.murata.com Ceramic

AVX www.avxcorp.com Ceramic,

Tantalum TPS Series

Taiyo Yuden [phone] www.taiyo-yuden.com Ceramic

EEF Series

POSCAP

When choosing a capacitor, look carefully through the
data sheet to fi nd out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher volt­age rating, may be required. High performance tantalum
or electrolytic capacitors can be used for the output
capacitor. Low ESR is important, so choose one that is
intended for use in switching regulators. The ESR should
be specifi ed by the supplier, and should be 0.05

or less.
Such a capacitor will be larger than a ceramic capacitor
and will have a larger capacitance, because the capacitor
must be large to achieve low ESR. Table 2 lists several
capacitor vendors.

Catch Diode

The catch diode conducts current only during switch off
time. Average forward current in normal operation can be
calculated from:

I

D(AVG)

= I

(VIN – V

OUT

OUT

)/V

IN

where I

is the output load current. The only reason to

OUT

consider a diode with a larger current rating than necessary
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a schottky diode with a
reverse voltage rating greater than the input voltage. Table
3 lists several Schottky diodes and their manufacturers.

Table 3. Diode Vendors

V

PART NUMBER

On Semiconductor
MBRA340 40 3 500

Diodes Inc.
PDS340
B340A
B340LA

R

(V)

40
40
40

I

(A)

AVE

3
3
3

V

F

AT 3A

(mV)

500
500
450

12

3693f

APPLICATIONS INFORMATION

LT3693

Frequency Compensation

The LT3693 uses current mode control to regulate the
output. This simplifi es loop compensation. In particular, the
LT3693 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the

pin, as shown in Figure 2. Generally a capacitor (CC)

V

C

and a resistor (R

) in series to ground are used. In addi-

C

tion, there may be lower value capacitor in parallel. This
capacitor (C

) is not part of the loop compensation but

F

is used to fi lter noise at the switching frequency, and is
required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.

Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the com­pensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stabil­ity using a transient load. Figure 2 shows an equivalent
circuit for the LT3693 control loop. The error amplifi er is a
transconductance amplifi er with fi nite output impedance.
The power section, consisting of the modulator, power
switch and inductor, is modeled as a transconductance
amplifi er generating an output current proportional to
the voltage at the V
integrates this current, and that the capacitor on the V

) integrates the error amplifi er output current, resulting

(C

C

pin. Note that the output capacitor

C

pin

C

in two poles in the loop. In most cases a zero is required
and comes from either the output capacitor ESR or from
a resistor R

in series with CC. This simple model works

C

well as long as the value of the inductor is not too high
and the loop crossover frequency is much lower than the
switching frequency. A phase lead capacitor (C

) across

PL

the feedback divider may improve the transient response.
Figure 3 shows the transient response when the load cur­rent is stepped from 1A to 3A and back to 1A.

LT3693

CURRENT MODE

POWER STAGE

= 5.3mho

g

m

3M

V

C

R

C

C

F

C

C

ERROR

AMPLIFIER

gm =

525Mmho

Figure 2. Model for Loop Response

V

OUT

100mV/DIV

I

L

1A/DIV

Figure 3. Transient Load Response of the LT3693 Front Page
Application as the Load Current is Stepped from 1A to 3A.
V

= 5V

OUT

–

+

GND

SW

0.8V

10Ms/DIV

OUTPUT

C

R1

FB

R2

PL

ESR

C1

POLYMER

OR

TANTALUM

+

3693 F03

C1

CERAMIC

3693 F02

3693f

13

LT3693

)

APPLICATIONS INFORMATION

V

4.7MF

V

4.7MF

V

4.7MF

V

OUT

BD

BOOST

IN

V

LT3693

IN

SW

GND

C3

is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (see Figure 4b). For lower output voltages the
boost diode can be tied to the input (Figure 4c), or to
another supply greater than 2.8V. Tying BD to V

reduces

IN

the maximum input voltage to 28V. The circuit in Figure 4a

(4a) For V

OUT

> 2.8V

is more effi cient because the BOOST pin current and BD
pin quiescent current comes from a lower voltage source.

V

OUT

C3

< 2.8V

C3

D2

V

OUT

BD

BOOST

IN

IN

V

LT3693

IN

SW

GND

(4b) For 2.5V < V

BD

BOOST

V

LT3693

IN

SW

GND

OUT

You must also be sure that the maximum voltage ratings
of the BOOST and BD pins are not exceeded.

6.0

5.5

TO START
(WORST CASE)

5.0

4.5

4.0
TO RUN

3.5

INPUT VOLTAGE (V)

3.0

V

= 3.3V

OUT

= 25oC

T

A

2.5

L = 8.2MH
f = 600kHz

2.0

1

10 100 1000

LOAD CURRENT (mA)

10000

3693 FO5

(4c) For V

< 2.5V; V

OUT

IN(MAX

= 28V

Figure 4. Three Circuits For Generating The Boost Voltage

BOOST and BIAS Pin Considerations

Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost volt­age that is higher than the input voltage. In most cases
a 0.47μF capacitor will work well. Figure 2 shows three
ways to arrange the boost circuit. The BOOST pin must be
more than 2.3V above the SW pin for best effi ciency. For
outputs of 3V and above, the standard circuit (Figure 4a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it

8.0

TO START

7.0

(WORST CASE)

6.0

5.0
TO RUN

4.0

INPUT VOLTAGE (V)

V

= 5V

OUT

= 25oC

T

3.0

A

L = 8.2MH
f = 600kHz

2.0

1 1000010 100 1000

LOAD CURRENT (mA)

3693 F06

Figure 5. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit

3693f

14

APPLICATIONS INFORMATION

The minimum operating voltage of an LT3693 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
or the LT3693 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to
zero once the circuit has started. Figure 5 shows a plot
of minimum load to start and to run as a function of input
voltage. In many cases the discharged output capacitor
will present a load to the switcher, which will allow it to
start. The plots show the worst-case situation where V
is ramping very slowly. For lower start-up voltage, the
boost diode can be tied to V

; however, this restricts the

IN

input range to one-half of the absolute maximum rating
of the BOOST pin.

At light loads, the inductor current becomes discontinu­ous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above V

. At higher load currents, the inductor

OUT

current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3693, requiring a higher
input voltage to maintain regulation.

Soft-Start

The RUN/SS pin can be used to soft-start the LT3693,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC fi lter to
create a voltage ramp at this pin. Figure 6 shows the start­up and shut-down waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20μA when the RUN/SS
pin reaches 2.5V.

IN

LT3693

I

L

RUN

15k

RUN/SS

GND

2ms/DIV

Figure 6. To Soft-Start the LT3693, Add a Resisitor
and Capacitor to the RUN/SS Pin

Synchronization

Synchronizing the LT3693 oscillator to an external fre­quency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.3V
and peaks that are above 0.8V (up to 6V).

The LT3693 may be synchronized over a 250kHz to 2MHz
range. The R

resistor should be chosen to set the LT3693

T

switching frequency 20% below the lowest synchronization
input. For example, if the synchronization signal will be
250kHz and higher, the R

should be chosen for 200kHz.

T

To assure reliable and safe operation the LT3693 will only
synchronize when the output voltage is near regulation
as indicated by the PG fl ag. It is therefore necessary to
choose a large enough inductor value to supply the required
output current at the frequency set by the R
Inductor Selection section. It is also important to note that
slope compensation is set by the R

value: When the sync

T

frequency is much higher than the one set by R
compensation will be signifi cantly reduced which may
require a larger inductor value to prevent subharmonic
oscillation.

1A/DIV

V

RUN/SS

2V/DIV

V

OUT

2V/DIV

3693 F07

resistor. See

T

, the slope

T

3693f

15

LT3693

APPLICATIONS INFORMATION

Shorted and Reversed Input Protection

If the inductor is chosen so that it won’t saturate exces­sively, an LT3693 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3693 is absent. This may occur in battery charging ap­plications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT3693’s
output. If the V

pin is allowed to fl oat and the RUN/SS

IN

pin is held high (either by a logic signal or because it is
tied to V

), then the LT3693’s internal circuitry will pull

IN

its quiescent current through its SW pin. This is fi ne if
your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the V

pin is grounded while

IN

the output is held high, then parasitic diodes inside the
LT3693 can pull large currents from the output through
the SW pin and the V

pin. Figure 7 shows a circuit that

IN

will run only when the input voltage is present and that
protects against a shorted or reversed input.

PCB Layout

For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 8 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents fl ow in the LT3693’s V

and SW pins, the catch

IN

diode (D1) and the input capacitor (C1). The loop formed
by these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and V

nodes small so that the ground

C

traces will shield them from the SW and BOOST nodes.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT3693 to additional ground planes within the circuit
board and on the bottom side.

D4

MBRS140

V

IN

V

IN

RUN/SS

V

C

BOOST

LT3693

SW

GND FB

3693 F08

Figure 7. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output. It Also
Protects the Circuit from a Reversed Input. The LT3693
Runs Only When the Input is Present

V

OUT

BACKUP

L1

OUT

C1

VIAS TO SYNC

D1

VIAS TO LOCAL GROUND PLANE

VIAS TO V

V

OUT

C2

GND

VIAS TO RUN/SS

VIAS TO PG

C

R

RT

R

PG

C

R

C

R2

R1

VIAS TO V

IN

OUTLINE OF LOCAL
GROUND PLANE

3693 F09

Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation

16

3693f

APPLICATIONS INFORMATION

LT3693

Hot Plugging Safely

The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3693 circuits. However, these capaci­tors can cause problems if the LT3693 is plugged into a
live supply (see Linear Technology Application Note 88 for
a complete discussion). The low loss ceramic capacitor,
combined with stray inductance in series with the power
source, forms an under damped tank circuit, and the
voltage at the V

pin of the LT3693 can ring to twice the

IN

nominal input voltage, possibly exceeding the LT3693’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT3693 into an

CLOSING SWITCH

+

LOW
IMPEDANCE
ENERGIZED

24V SUPPLY

SIMULATES HOT PLUG

I

IN

STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR

V

IN

LT3693

4.7MF

(9a)

energized supply, the input network should be designed
to prevent this overshoot. Figure 9 shows the waveforms
that result when an LT3693 circuit is connected to a 24V
supply through six feet of 24-gauge twisted pair. The
fi rst plot is the response with a 4.7μF ceramic capacitor
at the input. The input voltage rings as high as 50V and
the input current peaks at 26A. A good solution is shown
in Figure 9b. A 0.7 resistor is added in series with the
input to eliminate the voltage overshoot (it also reduces
the peak input current). A 0.1μF capacitor improves high
frequency fi ltering. For high input voltages its impact on
effi ciency is minor, reducing effi ciency by 1.5 percent for
a 5V output at full load operating from 24V.

DANGER

RINGING V

MAY EXCEED

IN

ABSOLUTE MAXIMUM RATING

20Ms/DIV

20V/DIV

10A/DIV

V

IN

I

IN

0.77

+

+

22MF

35V

AI.EI.

+

LT3693

4.7MF0.1MF

LT3693

4.7MF

(9b)

(9c)

20V/DIV

10A/DIV

20V/DIV

10A/DIV

V

IN

I

IN

20Ms/DIV

V

IN

I

IN

20Ms/DIV

3693 F10

Figure 9. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT3693 is Connected to a Live Supply

3693f

17

LT3693

APPLICATIONS INFORMATION

High Temperature Considerations

The PCB must provide heat sinking to keep the LT3693
cool. The Exposed Pad on the bottom of the package must
be soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these lay­ers will spread the heat dissipated by the LT3693. Place
additional vias can reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to

= 35°C/W or less. With

JA

100 LFPM airfl ow, this resistance can fall by another 25%.
Further increases in airfl ow will lead to lower thermal re­sistance. Because of the large output current capability of
the LT3693, it is possible to dissipate enough heat to raise
the junction temperature beyond the absolute maximum of
125°C. When operating at high ambient temperatures, the
maximum load current should be derated as the ambient
temperature approaches 125°C.

Power dissipation within the LT3693 can be estimated by
calculating the total power loss from an effi ciency measure­ment and subtracting the catch diode loss and inductor
loss. The die temperature is calculated by multiplying the
LT3693 power dissipation by the thermal resistance from
junction to ambient.

Other Linear Technology Publications

Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.

TYPICAL APPLICATIONS

V

IN

6.5V TO 36V

10MF

680pF

D: ON SEMI MBRA340
L: NEC MPLC0730L4R7

15k

5V Step-Down Converter

V

IN

ON OFF

63.4k

RUN/SS BOOST

V

C

RT

PG

SYNC

f = 600kHz

LT3693

GND

BD

SW

V

OUT

5V

3.5A

0.47MF

FB

100k

L

4.7MH

D

536k

47MF

3693 TA02

18

3693f

TYPICAL APPLICATIONS

V

IN

4.6V TO 36V

4.7MF

680pF

D: ON SEMI MBRA340
L: NEC MPLC0730L3R3

19k

3.3V Step-Down Converter

V

IN

ON OFF

63.4k

RUN/SS BOOST

V

C

LT3693

RT

PG

SYNC

f = 600kHz

GND

BD

SW

LT3693

V

OUT

3.3V

3.5A

0.47MF

FB

100k

L

3.3MH

D

316k

47MF

3693 TA03

V

4V TO 36V

4.7mF

IN

15.4k

680pF

D1: ON SEMI MBRA340
D2: MBR0540
L: NEC MPLC0730L3R3

ON OFF

63.4k

2.5V Step-Down Converter

LT3693

GND

BD

SW

FB

V

IN

RUN/SS BOOST

V

C

RT

PG

SYNC

f = 600kHz

1mF

100k

D1

D2

215k

L

3.3mH

3693 TA04

V

OUT

2.5V

3.5A

47mF

3693f

19

LT3693

TYPICAL APPLICATIONS

5V, 2MHz Step-Down Converter

V

8.6V TO 22V

TRANSIENT TO 36V

4.7mF

V

15V TO 36V

10mF

IN

15k

680pF

D: ON SEMI MBRA340
L: NEC MPLC0730L2R2

IN

17.4k

680pF

D: ON SEMI MBRA340
L: NEC MBP107558R2P

ON OFF

12.7k

ON OFF

63.4k

LT3693

GND

BD

SW

FB

V

IN

RUN/SS BOOST

V

C

RT

PG

SYNC

f = 2MHz

12V Step-Down Converter

LT3693

GND

BD

SW

FB

V

IN

RUN/SS BOOST

V

C

RT

PG

SYNC

f = 600kHz

0.47mF

100k

0.47mF

50k

D

D

536k

715k

L

2.2mH

L

8.2mH

3693 TA05

3693 TA06

V

OUT

5V

2.5A

22mF

V
12V

3.5A

47mF

OUT

20

3693f

TYPICAL APPLICATIONS

V

IN

3.6V TO 27V

4.7MF

16.9k

680pF

D: ON SEMI MBRA340
L: NEC MPLC0730L3R3

1.8V Step-Down Converter

V

IN

ON OFF

78.7k

RUN/SS BOOST

V

C

LT3693

RT

PG

SYNC

f = 500kHz

GND

BD

SW

LT3693

V

OUT

1.8V

3.5A

0.47MF

FB

100k

L

3.3MH

D

127k

47MF

3693 TA08

3693f

21

LT3693

PACKAGE DESCRIPTION

DD Package

10-Lead Plastic DFN (3mm × 3mm)

(Reference LTC DWG # 05-08-1699)

0.675 ±0.05

3.50 ±0.05

1.65 ±0.05
(2 SIDES)2.15 ±0.05

PACKAGE
OUTLINE

0.25 ± 0.05

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

PIN 1

TOP MARK

(SEE NOTE 6)

0.200 REF

NOTE:

1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE

0.50
BSC

2.38 ±0.05

(2 SIDES)

3.00 ±0.10

(4 SIDES)

0.75 ±0.05

1.65 ± 0.10

(2 SIDES)

0.00 – 0.05

R = 0.115

TYP

2.38 ±0.10

(2 SIDES)

BOTTOM VIEW—EXPOSED PAD

106

15

0.50 BSC

0.38 ± 0.10

0.25 ± 0.05

(DD) DFN 1103

22

3693f

PACKAGE DESCRIPTION

2.794 ± 0.102
(.110 ± .004)

MSE Package

10-Lead Plastic MSOP, Exposed Die Pad

(Reference LTC DWG # 05-08-1664 Rev B)

BOTTOM VIEW OF

EXPOSED PAD OPTION

0.889 ± 0.127

(.035 ± .005)

1

LT3693

2.06 ± 0.102
(.081 ± .004)

1.83 ± 0.102
(.072 ± .004)

5.23

(.206)

MIN

0.305 ± 0.038

(.0120 ± .0015)

TYP

RECOMMENDED SOLDER PAD LAYOUT

0.254

(.010)

GAUGE PLANE

0.18

(.007)

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

DETAIL “A”

DETAIL “A”

2.083 ± 0.102
(.082 ± .004)

0.50

(.0197)

BSC

0° – 6° TYP

0.53 ± 0.152

(.021 ± .006)

3.20 – 3.45

(.126 – .136)

SEATING

PLANE

3.00 ± 0.102

(.118 ± .004)

(NOTE 3)

4.90 ± 0.152

(.193 ± .006)

1.10

(.043)

MAX

0.17 – 0.27

(.007 – .011)

TYP

10

12

0.50

(.0197)

BSC

8910

3

7

6

45

0.497 ± 0.076
(.0196 ± .003)

REF

3.00 ± 0.102
(.118 ± .004)

(NOTE 4)

0.86

(.034)

REF

0.1016 ± 0.0508
(.004 ± .002)

MSOP (MSE) 0307 REV B

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa­tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

3693f

23

LT3693

TYPICAL APPLICATIO

U

1.2V Step-Down Converter

V

IN

3.6V TO 27V

4.7mF

ON OFF

17k

78.7k

470pF

D: ON SEMI MBRA340
L: NEC MPLC0730L3R3

V

IN

RUN/SS BOOST

V

C

LT3693

RT

PG

SYNC

GND

f = 500kHz

BD

0.47mF

SW

D

FB

52.3k

100k

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PART NUMBER DESCRIPTION COMMENTS

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Linear Technology Corporation

24

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

), 200kHz, High Effi ciency Step-Down DC/DC Converter VIN: 5.5V to 60V, V

OUT

TSSOP16/E Package

), 500kHz Step-Down Switching Regulator in SOT-23 VIN: 3.6V to 36V, V

OUT

), 500kHz, High Effi ciency Step-Down DC/DC Converter VIN: 3.6V to 36V, V

OUT

ThinSOT

TM

3mm × 3mm DFN Package

), 2.8MHz, High Effi ciency Step-Down DC/DC Converter VIN: 3.6V to 25V, V

OUT

MS8E Package

), 1.1MHz, High Effi ciency Step-Down DC/DC

OUT

VIN: 3.6V to 25V, V
TSSOP16E Package

), 200kHz/500kHz, High Effi ciency Step-Down DC/DC

OUT

VIN: 3.3V to 60V, V
TSSOP16E Package

), 200kHz/500kHz, High Effi ciency Step-Down DC/DC

OUT

VIN: 3.3V to 60V, V
TSSOP16 Package

), Micropower Step-Down DC/DC Converter with

OUT

VIN: 3.3V to 60V, V
3mm × 3mm DFN10 and TSSOP16E Packages

), 2.4MHz, High Effi ciency

OUT

VIN: 3.6V to 38V, V
3mm × 3mm DFN10 and MSOP10E Packages

), 2.8MHz, High Effi ciency

OUT

VIN: 3.6V to 34V, V
3mm × 3mm DFN10 and MSOP10E Packages

), 750kHz High Effi ciency Step-Down

OUT

VIN: 3.6V to 36V, V
2mm x 3mm DFN6 Package

), 3MHz,

OUT

VIN: 3.6V to 34V, V
3mm × 3mm DFN8 and MSOP8E Packages

), 3MHz,

OUT

VIN: 3.7V to 37V, V
4mm × 4mm QFN24 and TSSOP16E Packages

: 3.6V to 34V, V

IN

3mm × 3mm DFN10 and MSOP10E Packages

), 2.8MHz,

OUT

VIN: 3.6V to 34V, V
3mm × 3mm DFN10 and MSOP10E Packages

), 2.4MHz,

OUT

VIN: 3.6V to 38V, V
3mm × 3mm DFN10 and MSOP10E Packages

●

www.linear.com

L

3.3mH

3693 TA09

Package

V

OUT

1.2V

3.5A

100mF

= 1.2V, IQ = 2.5mA, ISD = 25μA,

OUT(MIN)

= 1.2V, IQ = 1.6mA, ISD < 1μA,

OUT(MIN)

= 1.2V, IQ = 1.9mA, ISD < 1μA,

OUT(MIN)

= 0.8V, IQ = 0.8mA, ISD < 1μA,

OUT(MIN)

= 1.2V, IQ = 3.8mA, ISD < 30μA,

OUT(MIN)

= 1.2V, IQ = 100μA, ISD < 1μA,

OUT(MIN)

= 1.2V, IQ = 100μA, ISD < 1μA,

OUT(MIN)

= 1.25V, IQ = 100μA, ISD < 1μA,

OUT(MIN)

= 0.78V, IQ = 70μA, ISD < 1μA,

OUT(MIN)

= 1.26V, IQ = 50μA, ISD < 1μA,

OUT(MIN)

= 0.8V, IQ = 1.9mA, ISD < 1μA,

OUT(MIN)

= 0.78V, IQ = 2mA, ISD = 2μA,

OUT(MIN)

= 0.8V, IQ = 4.6mA, ISD = 1μA,

OUT(MIN)

= 0.78V, IQ = 75μA, ISD = 1μA,

OUT(MIN)

= 1.26V, IQ = 850μA, ISD < 1μA,

OUT(MIN)

= 0.78V, IQ = 70μA, ISD < 1μA,

OUT(MIN)

LT 0907 • PRINTED IN USA

© LINEAR TECHNOLOGY CORPORATION 2007

3693f