LINEAR TECHNOLOGY LT3508 Technical data

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LT3508

Dual Monolithic 1.4A

Step-Down Switching

Regulator

FEATURES

Wide Input Voltage Range: 3.7V to 36V

Two 1.4A Output Switching Regulators with Internal

Power Switches

Adjustable 250kHz to 2.5MHz Switching Frequency

Synchronizable over the Full Frequency Range

Anti-Phase Switching Reduces Ripple

Uses Small Inductors and Ceramic Capacitors

Accurate Programmable Undervoltage Lockout

Independent Tracking, Soft-Start and Power Good

Circuits Ease Supply Sequencing

Output Adjustable Down to 800mV

Small 4mm × 4mm 24-Pin QFN or 16-Pin Thermally

Enhanced TSSOP Surface Mount Packages

APPLICATIONS

Automotive

DSP Power Supplies

Wall Transformer Regulation

DSL and Cable Modems

PCI Express

DESCRIPTION

The LT®3508 is a dual current mode PWM step-down DC/DC converter with internal power switches capable of generating two 1.4A outputs. The wide input voltage range of 3.7V to 36V makes the LT3508 suitable for regulating power from a wide variety of sources, including automotive batteries, 24V industrial supplies and unregulated wall adapters. Both converters are synchronized to a single oscillator programmable up to 2.5MHz and run with opposite phases, reducing input ripple current. Its high operating frequency allows the use of small, low cost inductors and ceramic capacitors, resulting in low, predictable output ripple. Each regulator has independent tracking and softstart circuits and generates a power good signal when its output is in regulation, easing power supply sequencing and interfacing with microcontrollers and DSPs.

Cycle-by-cycle current limit, frequency foldback and thermal shutdown provide protection against shorted outputs, and soft-start eliminates input current surge during startup. The low current (<2A) shutdown mode enables easy power management in battery-powered systems.

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

TYPICAL APPLICATION

3.3V and 5V Dual Output Step-Down Converter with Output Sequencing

5.6V TO 36V

OUT1

3.3V

1.4A

35.7k

11.5k

22µF

6.8µH

4.7µF

0.22µF 0.22µF

51k

150pF

1nF

VINSHDN

BOOST1

SW1

LT3508

TRACK/SS1

GND

= 700kHz

BOOST2

SW2

RT/SYNC

FB2FB1

PG1 PG2TRACK/SS2

52.3k

ON OFF

43k

100pF

10µH

56.2k

10.7k

100k

3508 TA01a

OUT2 5V

1.4A

10µF

POWER GOOD

VIN = 12V

EFFICIENCY (%)

Efﬁ ciency

= 5V

OUT2

= 3.3V

OUT1

0.5 1

LOAD CURRENT (A)

1.5

3508 TA01b

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LT3508

ABSOLUTE MAXIMUM RATINGS

VIN Pin Voltage ............................................(–0.3V), 40V

BOOST Pin Voltage ...................................................60V

BOOST Above SW Voltage ........................................30V

SHDN, PG Voltage .....................................................40V

TRACK/SS, FB, R

/SYNC, VC Voltage ..........................6V

Operating Junction Temperature Range (Note 2)

LT3508E ............................................. –40°C to 125°C

LT3508I .............................................. –40°C to 125°C

LT3508H ............................................ –40°C to 150°C

PIN CONFIGURATION

TOP VIEW

TRACK/SS1

BOOST1

BOOST2

TRACK/SS2

EXPOSED PAD (PIN 17) IS GND AND MUST BE SOLDERED TO PCB

SW1

IN1

IN2

SW2

16-LEAD PLASTIC TSSOP

θJA = 40°C/W, θJC = 10°C/W

FE PACKAGE

FB1

PG1

/SYNC

SHDN

PG2

FB2

(Note 1)

Storage Temperature Range

QFN .................................................... –65°C to 150°C

TSSOP ............................................... –65°C to 150°C

Lead Temperature (Soldering, 10 sec)

TSSOP .............................................................. 300°C

TOP VIEW

/SYNC

VC1PG1

24 23 22 21 20 19

FB1

TRACK/SS1

EXPOSED PAD (PIN 25) IS GND AND MUST BE SOLDERED TO PCB

GND

24-LEAD (4mm s 4mm) PLASTIC QFN

7 8 9

BOOST1

θJA = 40°C/W, θJC = 10°C/W

10 11 12

IN1VIN2

SW1

UF PACKAGE

SHDN

PG2

SW2

BOOST2

FB2

TRACK/SS2

GND

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT3508EFE#PBF LT3508EFE#TRPBF 3508FE 16-Lead Plastic TSSOP –40°C to 125°C

LT3508IFE#PBF LT3508IFE#TRPBF 3508FE 16-Lead Plastic TSSOP –40°C to 125°C

LT3508HFE#PBF LT3508HFE#TRPBF 3508HFE 16-Lead Plastic TSSOP –40°C to 150°C

LT3508EUF#PBF LT3508EUF#TRPBF 3508

LT3508IUF#PBF LT3508IUF#TRPBF 3508

LT3508HUF#PBF LT3508HUF#TRPBF 3508H

24-Lead (4mm × 4mm) Plastic QFN 24-Lead (4mm × 4mm) Plastic QFN 24-Lead (4mm × 4mm) Plastic QFN

–40°C to 125°C

–40°C to 150°C

Consult LTC Marketing for parts speciﬁ ed with wider operating temperature ranges.*Temperature grades are identiﬁ ed by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based ﬁ nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel speciﬁ cations, go to: http://www.linear.com/tapeandreel/

3508fb

LT3508

ELECTRICAL CHARACTERISTICS

The l denotes the speciﬁ cations which apply over the full operating temperature range, otherwise speciﬁ cations are at T

= 25°C. VIN = 12V, V

PARAMETER CONDITIONS MIN TYP MAX UNITS

Minimum Operating Voltage, V

Quiescent Current Not Switching 4.3 5.2 mA

IN1

Quiescent Current Not Switching 320 500 µA

IN2

Shutdown Current (V

IN1

IN2

+ V

IN2

= 12V

IN1

= 0.3V 0.1 2 µA

SHDN

FB Voltage

FB Pin Bias Current (Note 3) V

= 0.800V, VC = 0.4V

FB Voltage Line Regulation 5V < VIN < 40V 0.01 %/V

Error Amp Transconductance 300 µS

Error Amp Voltage Gain 600 V/V

to Switch Current Gain 2.5 A/V

Switching Frequency R

Switching Phase R

Maximum Duty Cycle (Note 4) R

Foldback Frequency R

= 33.2k

= 33.2k 150 180 210 Deg

= 33.2k

= 7.50k

= 169k

= 33.2k, VFB = 0V 120 kHz

Switch Current Limit (Note 5) Duty Cycle = 15%

Switch V

CESAT

ISW = 1.5A 300 mV

Switch Leakage Current 0.01 1 µA

Minimum Boost Voltage 1.7 2.5 V

Boost Pin Current I

TRACK/SS Pin Current V

PG Threshold Offset V

PG Voltage Output Low V

PG Pin Leakage V

= 1.5A 35 50 mA

= 0V 0.8 1.2 2.2 µA

TRACK/SS

Rising 56 75 110 mV

= 0.6V, IPG = 250µA 0.13 0.4 V

= 2V 0.01 1 µA

SHDN Threshold Voltage 2.53 2.63 2.73 V SHDN Input Current (Note 6) V

= 60mV Above Threshold Voltage 6 8 10 µA

SHDN

SHDN Threshold Current Hysteresis 5.5 7.5 9.5 µA

SYNC Threshold Voltage 1 1.25 1.5 V

SYNC Input Frequency 0.25 2.5 MHz

= 17V unless otherwise noted. (Note 2)

BOOST

0.790

0.784

0.92 1 1.06 MHz

84 90

2.0 2.6 3.2 A

3.4 3.7 V

2.5 3.0 V

0.800 0.814

0.816

50 300 nA

80 98

% % %

V V

Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime.

Note 2: The LT3508E is guaranteed to meet performance speciﬁ cations from 0°C to 125°C junction temperature. Speciﬁ cations over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3508I is guaranteed over the full –40°C to 125°C operating junction temperature range. The LT3508H is guaranteed over the full –40°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C.

Note 3: Current ﬂ ows out of pin. Note 4: V

LT3508 when V

=12V. Circuitry increases the maximum duty cycle of the

BOOST

> VIN + 2.5V. See “Minimum Operating Voltage” in

BOOST

the Applications Information section for details. Note 5: Current limit is guaranteed by design and/or correlation to static

test. Slope compensation reduces current limit at higher duty cycles.

Note 6: Current ﬂ ows into pin. Note 7: This IC includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature range when overtemperature protection is active. Continuous operation above the speciﬁ ed maximum operating junction temperature may impair device reliability.

3508fb

LT3508

TYPICAL PERFORMANCE CHARACTERISTICS

Efﬁ ciency, V

TA = 25°C f = 700kHz

EFFICIENCY (%)

Feedback Voltage

0.810

0.805

0.800

= 5V

OUT

VIN = 12V

VIN = 24V

VIN = 32V

0.5 1

LOAD CURRENT (A)

3508 G01

1.5

Efﬁ ciency, V

TA = 25°C f = 700kHz

EFFICIENCY (%)

0.5 1

LOAD CURRENT (A)

Switch Current Limit vs Temperature

3.0

2.5

2.0

1.5

= 3.3V Efﬁ ciency, V

OUT

VIN = 12V

VIN = 24V

VIN = 32V

1.5

3508 G02

TA = 25°C f = 1MHz

EFFICIENCY (%)

Switch Current Limit vs Duty Cycle

3.0 TA = 25°C

2.5

2.0

1.5

= 1.8V

OUT

VIN = 3.3V

VIN = 5V

VIN = 12V

0.5 1

LOAD CURRENT (A)

TYPICAL

MINIMUM

1.5

3508 G03

FEEDBACK VOLTAGE (V)

0.795

0.790 –50

–25 0 25 50

TEMPERATURE (°C)

Switching Frequency vs R

1000

TA = 25°C

100

(k)

0.1 FREQUENCY (MHz)

1.0

CURRENT LIMIT (A)

0.5

75 100 125 150

3508 G04

–50 –25 0 25 50 75 100 125 150

Switching Frequency

110

3508 G07

vs Temperature

1.2 RT = 33.2k

1.0

0.8

0.6

0.4

SWITCHING FREQUENCY (MHz)

0.2

–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

3508 G05

3508 G08

1.0

CURRENT LIMIT (A)

0.5

20 40 60 80

DUTY CYCLE (%)

Switching Frequency Foldback

3.0 TA = 25°C

2.5

2.0

1.5

1.0

SWITCHING FREQUENCY (MHz)

0.5

200 400 600 800

100 300 500 700

FEEDBACK VOLTAGE (mV)

RT = 7.50k

RT = 33.2k

100

3508 G06

RT = 169k

3508 G09

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TYPICAL PERFORMANCE CHARACTERISTICS

LT3508

Quiescent Current VC Voltages

5.0 TA = 25°C

4.5

4.0

3.5

3.0

2.5

2.0

1.5

INPUT CURRENT (mA)

1.0

0.5

515

IN1

IN2

INPUT VOLTAGE (V)

3508 G10

2.5

2.0

1.5

VOLTAGE (V)

1.0

0.5

–50 –25 0 25 50 75 100 125 150

Switch Voltage Drop

350

TA = 25°C

300

250

200

150

CLAMP VOLTAGE

TO SWITCH

TEMPERATURE (°C)

3508 G11

Boost Pin Current

TA = 25°C

Error Amp Output Current

OUTPUT CURRENT (µA)

–50 –25 0 25 50 75 100 125 150

SINKING

SOURCING

TEMPERATURE (°C)

3508 G12

100

SWITCH VOLTAGE (mV)

SHDN Pin Current

120

100

SHDN PIN CURRENT (µA)

515

0.5

SWITCH CURRENT (A)

TA = –45°C

TA = 25°C

10 20

SHDN PIN VOLTAGE (V)

1 1.5

3508 G13

TA = 125°C

3508 G15

BOOST PIN CURRENT (mA)

0.5

SWITCH CURRENT (A)

1 1.5

3508 G14

Undervoltage Lockout

4.0

3.5

3.0

2.5

2.0

1.5

INPUT VOLTAGE (V)

1.0

0.5

–50 –25 0 25 50 75 100 125 150

IN1

IN2

TEMPERATURE (°C)

3508 G16

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LT3508

PIN FUNCTIONS

BOOST1, BOOST2: The BOOST pins are used to provide drive voltages, higher than the input voltage, to the internal NPN power switches. Tie through a diode to a 2.8V or higher supply, such as V

OUT

or VIN.

Exposed Pad: The Exposed Pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation.

FB1, FB2: The LT3508 regulates each feedback pin to

0.800V. Connect the feedback resistor divider taps to these pins.

GND: Tie the GND pins directly to the Exposed Pad and ground plane.

PG1, PG2: The power good pins are the open-collector outputs of an internal comparator. PG remains low until the FB pin is within 10% of the ﬁ nal regulation voltage. As well as indicating output regulation, the PG pins can be used to sequence the two switching regulators. These pins can be left unconnected. The PG outputs are valid when V

is greater than 2.4V and SHDN is high. The PG

IN1

comparators are disabled in shutdown.

/SYNC: The RT/SYNC pin is used to set the internal

oscillator frequency. Tie a 33.2k resistor from R

/SYNC

to GND for a 1MHz switching frequency. To synchronize the part to an external frequency, drive the R

/SYNC pin

with a logic-level signal with positive and negative pulse widths of at least 80ns.

SHDN: The shutdown pin is used to put the LT3508 in shutdown mode. Pull the pin below 0.3V to shut down the LT3508. The 2.63V threshold can function as an accurate undervoltage lockout (UVLO), preventing the regulator

from operating until the input voltage has reached the programmed level. Do not drive SHDN more than 6V above V

IN1

SW1, SW2: The SW pins are the outputs of the internal power switches. Connect these pins to the inductors, catch diodes and boost capacitors.

TRACK/SS1, TRACK/SS2: The TRACK/SS pins are used to soft-start the two channels, to allow one channel to track the other output, or to allow both channels to track another output. For tracking, tie a resistor divider to this pin from the tracked output. For soft-start, tie a capacitor to this pin. An internal 1.2µA soft-start current charges the capacitor to create a voltage ramp at the pin. Leave these pins disconnected if unused.

, VC2: The VC pins are the outputs of the internal error

amps. The voltages on these pins control the peak switch currents. These pins are normally used to compensate the control loops, but can also be used to override the loops. Pull these pins to ground with an open drain to shut down each switching regulator.

IN1

: The V

pin supplies current to the LT3508 internal

IN1

circuitry and to the internal power switch connected to SW1 and must be locally bypassed. V

must be greater

IN1

than 3.7V for channel 1 or channel 2 to operate.

IN2

: The V

pin supplies current to the internal power

IN2

switch connected to SW2 and must be locally bypassed. Connect this pin directly to V nel 2 is coming from a different source. V greater than 3V and V

must be greater than 3.7V for

IN1

unless power for chan-

IN1

must be

IN2

channel 2 to operate.

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BLOCK DIAGRAM

SHDN

LT3508

IN1

RT/SYNC

TRACK/SS

1.2µA

0.75V

INT REG

AND REF

MASTER

OSC

CLK1

CLK2

SLOPE

SLAVE

CLK

OSC

0.625V

–

LIMIT

CLAMP

ERROR

AMP

– +

TRACK/SS

75mV

0.80V

BOOST

OUT

GND

–

3508 F01

Figure 1. Block Diagram of the LT3508 with Associated External Components (One of Two Switching Regulators Shown)

3508fb

LT3508

OPERATION

The LT3508 is a dual constant frequency, current mode regulator with internal power switches. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3508 is shut down and draws minimal current from the input source tied to the

pins. If the SHDN pin exceeds 1V, the internal bias

circuits turn on, including the internal regulator, reference and oscillator. The switching regulators will only begin to operate when the SHDN pin exceeds 2.63V.

The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the RS ﬂ ip-ﬂ op and turns on the internal NPN power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at V

, current comparator C1 resets the

ﬂ ip-ﬂ op, turning off the switch. The current in the inductor ﬂ ows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the V

pin controls

the current through the inductor to the output. The internal error ampliﬁ er regulates the output current by continually adjusting the V on the V

pin voltage. The threshold for switching

pin is 0.8V, and an active clamp of 1.75V limits

the output current.

The switching frequency is set either by the resistance to GND at the R signal driving the R

/SYNC pin or the frequency of the logic-level

/SYNC pin. A detection circuit monitors

for the presence of a SYNC signal on the pin and switches

between the two modes. Unique circuitry generates the appropriate slope compensation ramps and generates the 180° out-of-phase clocks for the two channels.

The switching regulator performs frequency foldback during overload conditions. An ampliﬁ er senses when

is less than 0.625V and begins decreasing the oscil-

lator frequency down from full frequency to 12% of the nominal frequency when V

= 0V. The FB pin is less than

0.8V during start-up, short-circuit and overload conditions. Frequency foldback helps limit switch current under these conditions.

The switch driver operates either from V

or from the

BOOST pin. An external capacitor and Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efﬁ cient operation.

The TRACK/SS pin serves as an alternative input to the error ampliﬁ er. The ampliﬁ er will use the lowest voltage of either the reference of 0.8V or the voltage on the TRACK/SS pin as the positive input of error ampliﬁ er. Since the TRACK/SS pin is driven by a constant current source, a single capacitor on the pin will generate a linear ramp on the output voltage. Tying the TRACK/SS pin to a resistor divider from the output of one of the switching regulators allows one output to track another.

The PG output is an open-collector transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when the LT3508 is enabled (SHDN is high) and V

is greater

IN1

than ~2.4V.

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APPLICATIONS INFORMATION

LT3508

Setting the Output Voltage

The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to:

⎛ ⎜

⎝

OUT

⎞

–

⎟

⎠

R2 should be 20k or less to avoid bias current errors. Reference designators refer to the Block Diagram.

Minimum Operating Voltage

The minimum operating voltage is determined either by the LT3508’s undervoltage lockout or by its maximum duty cycle. If V

IN1

and V

are tied together, the undervoltage

IN2

lockout is at 3.7V or below. If the two inputs are used separately, then V or below and V below. Because the internal supply runs off V nel 2 will not operate unless V

has an undervoltage lockout of 3.7V

IN1

has an undervoltage lockout of 3V or

IN2

, chan-

IN1

> 3.7V. The duty cycle

IN1

is the fraction of time that the internal switch is on and is determined by the input and output voltages:

OUT F

VV V

IN SW F

+–

where VF is the forward voltage drop of the catch diode (~0.4V) and V

is the voltage drop of the internal switch

(~0.4V at maximum load).

Example: I

= 1.5A/50mA = 30, DC

IN MIN()

= 1.5A and I

33 04

–. . .=

= 50mA, V

BOOST

= 1/(1+1/30) = 96%:

MAX

04 04 38

VVV

OUT

= 3.3V,

Maximum Operating Voltage

The maximum operating voltage is determined by the Absolute Maximum Ratings of the V

and BOOST pins,

and by the minimum duty cycle:

MIN

where t

= t

ON(MIN)

• f

is equal to 130ns (for TJ > 125°C t

ON(MIN)

is equal to 150ns) and f is the switching frequency. Running at a lower switching frequency allows a lower minimum duty cycle. The maximum input voltage before pulse skipping occurs depends on the output voltage and the minimum duty cycle:

IN PS

OUT F

MIN

–=

FSW()

Unlike many ﬁ xed frequency regulators, the LT3508 can extend its duty cycle by turning on for multiple cycles. The LT3508 will not switch off at the end of each clock cycle if there is sufﬁ cient voltage across the boost capacitor (C3 in Figure 1). Eventually, the voltage on the boost capacitor falls and requires refreshing. Circuitry detects this condition and forces the switch to turn off, allowing the inductor current to charge up the boost capacitor. This places a limitation on the maximum duty cycle as follows:

MAX

+11

where βSW is equal to the SW pin current divided by the BOOST pin current as shown in the Typical Performance Characteristics section. This leads to a minimum input voltage of:

IN MIN

OUT F

MAX

–=

FSW()

Example: f = 790kHz, V

= 3.3V, DC

OUT

= 130ns • 790kHz

MIN

= 0.103:

33 04

IN PS()

0 103

–. .=

04 04 36

VVV

The LT3508 will regulate the output current at input voltages greater than V

. For example, an application with an

IN(PS)

output voltage of 1.8V and switching frequency of 1.5MHz has a V

of 11.3V, as shown in Figure 2. Figure 3 shows

IN(PS)

operation at 18V. Output ripple and peak inductor current have signiﬁ cantly increased. Exceeding V

IN(PS)

is safe if the output is in regulation, if the external components have adequate ratings to handle the peak conditions and if the peak inductor current does not exceed 3.2A. A saturating inductor may further reduce performance. Do not exceed V greater than 5V, use V

during start-up or overload conditions (for outputs

IN(PS)

= 5V to calculate V

OUT

IN(PS)

). For operation above 20V in pulse skipping mode, program the switching frequency to 1.1MHz or less.

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LT3508

APPLICATIONS INFORMATION

OUT

100mV/DIV

(AC)

500mA/DIV

3508 F02

= 1.8V and

OUT

Figure 2. Operation Below V

2µs/DIV

. VIN = 10V, V

IN(PS)

fSW = 1.5MHz

OUT

100mV/DIV

(AC)

500mA/DIV

2µs/DIV

Figure 3. Operation Above V

. VIN = 18V, V

IN(PS)

3508 F03

OUT

= 1.8V and fSW = 1.5MHz. Output Ripple and Peak Inductor Current Increase

Table 1. Programming the Switching Frequency

SWITCHING FREQUENCY (MHz)

2.5 7.50

2.2 9.76

2 11.5

1.8 14

1.6 16.9

1.4 20.5

1.2 26.1

1 33.2

0.9 38.3

0.8 44.2

0.7 52.3

0.6 61.9

0.5 76.8

0.45 88.7

0.4 100

0.35 115

0.3 140

0.25 169

(kΩ)

Setting the Switching Frequency

The switching frequency is programmed either by driving

/SYNC pin with a logic level SYNC signal or by tying

the R

a resistor from the R selecting the value of R

/SYNC pin to ground. A graph for

for a given operating frequency

is shown in the Typical Application section. Suggested programming resistors for various switching frequencies are shown in Table 1.

Choosing a high switching frequency will allow the smallest overall solution size. However, at high input voltages the efﬁ ciency can drop signiﬁ cantly with increasing switching frequency. The choice of switching frequency will also impact the input voltage range, inductor and capacitor selection, and compensation. See the related sections for details.

Inductor Selection and Maximum Output Current

A good ﬁ rst choice for the inductor value is:

MHz

LV V

()

OUT F

.12

• f

where VF is the voltage drop of the catch diode (~0.4V) and L is in H. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efﬁ ciency, the series resistance (DCR) should be less than 0.1Ω. Table 2 lists several vendors and types that are suitable.

Table 2. Inductor Vendors

VENDOR URL PART SERIES TYPE

Coilcraft www.coilcraft MSS7341 Shielded

Murata www.murata.com LQH55D Open

TDK www.component.tdk.com SLF7045

SLF10145

Toko www.toko.com DC62CB

D63CB

D75C D75F

Sumida www.sumida.com CR54

CDRH74

CDRH6D38

CR75

Shielded Shielded

Shielded Shielded Shielded

Open

Open Shielded Shielded

Open

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APPLICATIONS INFORMATION

LT3508

The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efﬁ ciency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (V

OUT/VIN

> 0.5), a minimum inductance

is required to avoid sub-harmonic oscillations:

kHz

LVV

()

MIN OUT F

•

800

The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3508 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3508 will deliver depends on the switch current limit, the inductor value, and the input and output voltages.

When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor:

1–

DC V V

()

ΔI

()

OUT F

•

where f is the switching frequency of the LT3508 and L is the value of the inductor. The peak inductor and switch current is:

III

SW PK L PK OUT

==+

() ()

To maintain output regulation, this peak current must be

. I

less than the LT3508’s switch current limit I

LIM

at least 2A for at low duty cycles and decreases linearly

to 1.55A at DC = 90%. The maximum output current is a function of the chosen inductor value:

ΔΔ

IIIADC

OUT MAX LIM

()

–•–.•–==

21025

()

Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit.

One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3508 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when I

is less than ΔIL/2.

OUT

Input Capacitor Selection

Bypass the V

pins of the LT3508 circuit with a ceramic

capacitor of X7R or X5R type. For switching frequencies above 500kHz, use a 4.7µF capacitor or greater. For switching frequencies below 500kHz, use a 10µF or higher capacitor. If the V is necessary. If the V

pins are tied together only a single capacitor

pins are separated, each pin will

need its own bypass. The following paragraphs describe the input capacitor considerations in more detail.

Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3508 input and to force this switching current into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. With two switchers operating at the same frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple. However, a conservative value is the RMS input current for the channel that is delivering most power (V

VVV

()

IN RMS OUT

()

and is largest when VIN = 2V

OUT IN OUT

• V

OUT

times I

OUT

–

(50% duty cycle). As

OUT

the second, lower power channel draws input current,

3508fb

LT3508

APPLICATIONS INFORMATION

the input capacitor’s RMS current actually decreases as the out-of-phase current cancels the current drawn by the higher power channel. Considering that the maximum load current from a single channel is ~1.4A, RMS ripple current will always be less than 0.7A.

The high frequency of the LT3508 reduces the energy storage requirements of the input capacitor. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types.

An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10µF in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an oversurge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor.

A ﬁ nal caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3508. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details see Application Note 88.

Output Capacitor Selection

The output capacitor has two essential functions. Along with the inductor, it ﬁ lters the square wave generated by the LT3508 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the

LT3508’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good value is:

VVMHz

OUT

where C

50 1

OUT

•

OUT

is in µF. Use X5R or X7R types. This choice will provide low output ripple and good transient response. Transient performance can be improved with a high value capacitor if the compensation network is also adjusted to maintain the loop bandwidth. A lower value of output capacitor can be used, but transient performance will suffer. With an external compensation network, the loop gain can be lowered to compensate for the lower capacitor value. Look carefully at the capacitor’s data sheet to ﬁ nd out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor, or one with a higher voltage rating, may be required. High performance electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be speciﬁ ed by the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 3 lists several capacitor vendors.

Table 3. Capacitor Vendors

VENDOR PART SERIES COMMENTS

Panasonic Ceramic

Kemet Ceramic

Sanyo Ceramic

Murata Ceramic

AVX Ceramic

Taiyo Yuden Ceramic

TDK Ceramic

Polymer Tantalum

Tantalum T494, T495

Polymer Tantalum

Tantalum TPS Series

EEF Series

POSCAP

3508fb

APPLICATIONS INFORMATION

LT3508

Diode Selection

The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from:

–

()

D AVG

()

IVV

OUT IN OUT

The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current.

Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 4 lists several Schottky diodes and their manufacturers. If operating at high ambient temperatures, consider using a Schottky with low reverse leakage.

Table 4. Schottky Diodes

at 1A

PART NUMBER

On Semiconductor

MBR0520L 20 0.5

MBR0540 40 0.5 620

MBRM120E 20 1 530

MBRM140 40 1 550

Diodes Inc.

B0530W 30 0.5

B120 20 1 500

B130 30 1 500

B140HB 40 1

DFLS140 40 1.1 510

B240 40 2 500

(V)

AVE

(A)

(mV)

VF at 2A

(mV)

BOOST Pin Considerations

The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22µF capacitor and fast switching diode (such as the CMDSH-3 or MMSD914LT1) will work well. For applications 1MHz or faster, a 0.1µF capacitor is sufﬁ cient. Use a 0.47µF capacitor or greater for applicaitons running below 500kHz. Figure 4 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efﬁ ciency. For outputs of 3.3V

and higher, the standard circuit (Figure 4a) is best. For outputs between 2.8V and 3.3V, use a small Schottky diode (such as the BAT-54). For lower output voltages, the boost diode can be tied to the input (Figure 4b). The circuit in Figure 4a is more efﬁ cient because the boost pin current comes from a lower voltage source. Finally, the anode of the boost diode can be tied to another source that is at least 3V (Figure 4c). For example, if you are generating a 3.3V output, and the 3.3V output is on whenever the particular channel is on, the anode of the BOOST diode can be connected to the 3.3V output. In any case, be sure that the maximum voltage at the BOOST pin is both less than 60V and the voltage difference between the BOOST and SW pins is less than 30V.

BOOST

MAX V

BOOST

MAX V

> 3V

IN2

BOOST

MAX V MINIMUM VALUE FOR V

LT3508

GND

– VSW V

VIN + V

BOOST

LT3508

GND

– VSW V

BOOST

LT3508

GND

– VSW V

BOOST

IN2

OUT

IN IN

IN2

+ V

OUT

(4a)

(4b)

IN2

(4c)

Figure 4. Generating the Boost Voltage

OUT

3508 F04

3508fb

LT3508

APPLICATIONS INFORMATION

The minimum operating voltage of an LT3508 application is limited by the undervoltage lockout (≈3.7V) and by the maximum duty cycle. The boost circuit also limits the minimum input voltage for proper start-up. If the input voltage ramps slowly, or the LT3508 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load current generally goes to zero once the circuit has started. Figure 5 shows a plot of minimum load to start and to run as a function of input voltage. Even without an output load current, in many

Minimum Input Voltage, V

6.5 TA = 25°C

= 3.3V

OUT

6.0

5.5

5.0

4.5

INPUT VOLTAGE (V)

4.0

3.5

3.0

INPUT VOLTAGE (V)

Figure 5. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit

TO RUN

10 100 10000

Minimum Input Voltage, V

TA = 25°C

= 5V

OUT

TO RUN

TO START

LOAD CURRENT (mA)

TO START

100

LOAD CURRENT (mA)

OUT

1000

= 3.3V

OUT

3508 F05a

= 5V

10000

3508 G05b

cases the discharged output capacitor will present a load to the switcher that will allow it to start. The plots show the worst case, where V

is ramping very slowly.

Frequency Compensation

The LT3508 uses current mode control to regulate the output. This simpliﬁ es loop compensation. In particular, the LT3508 does not require the ESR of the output capacitor for stability, so you are free to use ceramic capacitors to achieve low output ripple and small circuit size.

Frequency compensation is provided by the components tied to the V

) and a resistor (RC) in series to ground are used. In

pin, as shown in Figure 1. Generally a capacitor

addition, there may be a lower value capacitor in parallel. This capacitor (C

) is not part of the loop compensation

but is used to ﬁ lter noise at the switching frequency, and is required only if a phase-lead capacitor is used or if the output capacitor has high ESR.

Loop compensation determines the stability and transient performance. Designing the compensation network is a bit complicated and the best values depend on the application and in particular the type of output capacitor. A practical approach is to start with one of the circuits in this data sheet that is similar to your application and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load.

Figure 6 shows an equivalent circuit for the LT3508 control loop. The error ampliﬁ er is a transconductance ampliﬁ er with ﬁ nite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance ampliﬁ er generating an output current proportional to the voltage at the V

pin. Note that

the output capacitor integrates this current, and that the capacitor on the V

pin (CC) integrates the error ampliﬁ er

output current, resulting in two poles in the loop. In most cases a zero is required and comes from either the output capacitor ESR or from a resistor R

in series with CC.

This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency

3508fb

APPLICATIONS INFORMATION

LT3508

is much lower than the switching frequency. A phase-lead capacitor (C

) across the feedback divider may improve

the transient response.

LT3508

CURRENT MODE

POWER STAGE

= 2.5S

GND

AMPLIFIER

300µS

Figure 6. Model for Loop Response

ERROR

–

0.8V

POLYMER

TANTALUM

OUTPUT

ESR

CERAMIC

3508 F06

Shutdown and Undervoltage Lockout

Figure 7 shows how to add undervoltage lockout (UVLO) to the LT3508. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where the problems might occur.

An internal comparator will force the part into shutdown below the minimum V

of 3.7V. This feature can be

IN1

used to prevent excessive discharge of battery-operated systems.

If an adjustable UVLO threshold is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.63V. Current hysteresis is added above the SHDN threshold. This can be used to set voltage hysteresis of the UVLO using the following:

–

μA

263

–.

263

μA

–

Example: switching should not start until the input is above

4.75V and is to stop if the input falls below 4V.

VVVV

475 40

., .

475 4

.–

μA

663

475 263

.–.

100

–

μA

200

Keep the connection from the resistor to the SHDN pin short and make sure the interplane or surface capacitance to switching nodes is minimized. If high resistor values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node.

Soft-Start

LT3508

SHDN

R4C1

2.6V

–

TRACK/SS

The output of the LT3508 regulates to the lowest voltage present at either the TRACK/SS pin or an internal 0.8V reference. A capacitor from the TRACK/SS pin to ground is charged by an internal 1.2A current source resulting in a linear output ramp from 0V to the regulated output whose duration is given by:

8µA

7.5µA

Figure 7. Undervoltage Lockout

3508 F07

RAMP

At power up, internal open-collector ouputs discharge both TRACK/SS pins. The pins clamp at 1.3V.

•..08

μA

3508fb

LT3508

APPLICATIONS INFORMATION

Output Tracking and Sequencing

Complex output tracking and sequencing between channels can be implemented using the LT3508’s TRACK/SS and PG pins. Figure 8 shows several conﬁ gurations for output tracking and sequencing of 5V and 3.3V applications.

Independent Start-Up Ratiometric Start-Up

OUT1

1V/DIV

0.1µF

0.047µF

20ms/DIV

TRACK/SS1

LT3508

TRACK/SS2

(8a)

OUT1

OUT2

3.3V

1V/DIV

20ms/DIV

TRACK/SS1

0.22µF

TRACK/SS2

Independent soft-start for each channel is shown in Figure 8a. The output ramp time for each channel is set by the soft-start capacitor as described in the soft-start section.

Coincident Start-Up

LT3508

(8b)

OUT1

OUT2

OUT1

OUT2

3.3V

1V/DIV

0.1µF

TRACK/SS1

TRACK/SS2

10.0k

20ms/DIV

LT3508

28.7k

OUT1

OUT2

(8c)

OUT1

OUT2

3.3V

1V/DIV

0.1µF

0.047µF

Output Sequencing

20ms/DIV

TRACK/SS1

TRACK/SS2

LT3508

OUT1

PG1

OUT2

(8d)

OUT1

OUT2

3.3V

Figure 8

1V/DIV

EXTERNAL

Controlled Power Up and Down

OUT1

OUT2

EXTERNAL SOURCE

20ms/DIV

SOURCE

+ –

TRACK/SS1

TRACK/SS2

10.0k

LT3508

28.7k

OUT1

OUT2

(8e)

3.3V

3508fb

APPLICATIONS INFORMATION

LT3508

Ratiometric tracking is achieved in Figure 8b by connecting both the TRACK/SS pins together. In this conﬁ guration the TRACK/SS pin source current is doubled (2.4A) which must be taken into account when calculating the output rise time.

By connecting a feedback network from V TRACK/SS2 pin with the same ratio that set the V

OUT1

to the

OUT2

voltage, absolute tracking shown in Figure 8c is implemented. A small V

voltage offset will be present due

OUT2

to the TRACK/SS2 1.2A source current. This offset can be corrected for by slightly reducing the value of R2. Use a resistor divider such that when V

is in regulation,

OUT1

TRACK/SS2 is pulled up to 1V or greater. If TRACK/SS is below 1V, the output may regulate FB to a voltage lower than the 800mV reference voltage.

Figure 8d illustrates output sequencing. When V

OUT1

is within 10% of its regulated voltage, PG1 releases the TRACK/SS2 soft-start pin allowing V

OUT2

to softstart. In this case PG1 will be pulled up to 1.3V by the TRACK/SS pin.

If precise output ramp up and down is required, drive the TRACK/SS pins as shown in Figure 8e.

Multiple Inputs

For applications requiring large inductors due to high V to V

ratios, a 2-stage step down approach may reduce

OUT

inductor size by allowing an increase in frequency. A dual step-down application (Figure 9) steps down the input voltage (V voltage to power the second output (V

) to the highest output voltage, then uses that

IN1

IN2

). V

OUT1

must be able to provide enough current for its output plus the input current at V

when V

IN2

is at its maximum load.

OUT2

For applications with multiple input voltages, the LT3508 can accommodate input voltages as low as 3V on V

IN2

. This can be useful in applications regulating outputs from a PCI Express bus, where the 12V input is power limited and the 3.3V input has power available to drive other outputs. In this case, tie the 12V input to V

3.3V input to V

. See the Typical Application section for

IN2

and the

IN1

an example circuit.

Do not tie TRACK/SS1 and TRACK/SS2 together if using multiple inputs. If V

is below 3V, TRACK/SS2 pulls low

IN2

and would hold TRACK/SS1 low as well if the two pins are tied together, which would prevent channel 1 from operating.

Shorted and Reverse Input Protection

If the inductor is chosen so that it won’t saturate excessively, an LT3508 step-down regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT3508 is absent. This may occur in battery charging

OUT1

0.9A

5.7V TO 36V

L1 6.8µH

56.2k

10.7k

C4 10µF

R5 39k

4.7µF

ON OFF

0.1µF

D3 D4

C6 100pF

1nF

IN1VIN2

SHDN

BOOST1

SW1

TRACK/SS1

GND

3.3nF

LT3508

RT/SYNC

= 1MHz

BOOST2

SW2

FB2FB1

PG1 PG2TRACK/SS2

33.2k

330pF

47k

0.1µF

OUT1

3.3µH

18.7k

15.0k

Figure 9. 1MHz, Wide Input Range 5V and 1.8V Outputs

OUT2

1.8V 1A

C5 47µF

R7 100k

POWER

3508 F09

GOOD

3508fb

LT3508

APPLICATIONS INFORMATION

applications or in battery back-up systems where a battery or some other supply is diode OR-ed with the LT3508’s output. If the V

pin is allowed to ﬂ oat and the SHDN pin

is held high (either by a logic signal or because it is tied

), then the LT3508’s internal circuitry will pull its

to V

quiescent current through its SW pin. This is ﬁ ne if your system can tolerate a few mA in this state. If you ground the SHDN pin, the SW pin current will drop to essentially zero. However, if the V

pin is grounded while the output

is held high, then parasitic diodes inside the LT3508 can pull large currents from the output through the SW pin and the V

pin. Figure 10 shows a circuit that will run

only when the input voltage is present and that protects against a shorted or reversed input.

PARASITIC DIODE

LT3508

3508 F10

OUT

PCB Layout

For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 11 shows the recommended PCB layout with trace and via locations. Note that large, switched currents ﬂ ow in the LT3508’s V SW pins, the catch diode (D1) and the input capacitor (C

and

The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and V

nodes

small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT3508 to additional ground planes within the circuit board and on the bottom side.

Figure 10. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output

(11a) Example Layout for FE16 Package (11b) Example Layout for QFN Package

Figure 11. A Good PCB Layout Ensures Proper Low EMI Operation

3508fb

APPLICATIONS INFORMATION

LT3508

High Temperature Considerations

The die temperature of the LT3508 must be lower than the maximum rating of 125°C (150°C for the H grade). This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT3508. The maximum load current should be derated as the ambient temperature approaches 125°C (150°C for the H grade). The die temperature is calculated by multiplying the LT3508 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT3508 can be estimated by calculating the total power loss from an efﬁ ciency measurement and subtracting the catch diode loss. Thermal resistance depends on the layout of the circuit board, but values from 30°C/W to 60°C/W are typical. Die temperature rise was measured on a 4-layer, 6.5cm × 7.5cm circuit board in still air at a load current of 1.4A (f

= 700kHz). For a 12V input to

3.3V output the die temperature elevation above ambient

was 13°C; for 24VIN to 3.3V 12V

to 5V

the rise was 14°C and for 24VIN to 5V

OUT

the rise was 18°C; for

OUT

the rise was 19°C.

Outputs Greater Than 6V

For outputs greater than 6V, add a resistor of 1k to 2.5k across the inductor to damp the discontinuous ringing of the SW node, preventing unintended SW current. The 12V output circuit in the Typical Applications section shows the location of this resistor.

Other Linear Technology Publications

Application Notes 19, 35 and 44 contain more detailed descriptions and design information for step-down regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 318 shows how to generate a dual polarity output supply using a step-down regulator.

TYPICAL APPLICATIONS

3.9V TO 16V

OUT2

OUT1

1.8V

1.4A

L1 3.3µH

18.7k

15.0k

C4 47µF

C1 TO C5: X5R OR X7R D1, D2: MMSD4148 D3: DIODES INC. B140 D4: DIODES INC. B240A

1MHz, 3.3V and 1.8V Outputs with Sequencing

FB2FB1

PG1 PG2TRACK/SS2

33.2k

ON OFF

0.1µF

39k

150pF

4.7µF

0.1µF

D3 D4

R5 47k

C6 330pF

1nF

IN1

BOOST1

SW1

LT3508

TRACK/SS1

GND

= 1MHz

SHDN

IN2

BOOST2

RT/SYNC

SW2

L2 4.7µH

35.7k

11.5k

R7 100k

3508 TA02

OUT2

3.3V

1.4A

C5 10µF

POWER GOOD

3508fb

LT3508

TYPICAL APPLICATIONS

3.3V and 5V Dual Output Step-Down Converter with Output Sequencing

5.7V TO 36V

OUT1

3.3V

1.4A

L1 6.8µH

35.7k

11.5k

C4 22µF

C1 TO C5: X5R OR X7R D1, D2: MMSD4148 D3: DIODES INC. B140 D4: DIODES INC. B240A

FB2FB1

PG1 PG2TRACK/SS2

52.3k

ON OFF

0.22µF

43k

100pF

L2 10µH

56.2k

10.7k

R7 100k

3508 TA03

OUT2 5V

1.4A

C5 10µF

POWER GOOD

4.7µF

0.22µF

D3 D4

R5 51k

C6 150pF

1nF

IN1

BOOST1

SW1

LT3508

TRACK/SS1

GND

= 700kHz

SHDN

IN2

BOOST2

RT/SYNC

SW2

OUT1

0.9A

5.7V TO 36V

L1 6.8µH

56.2k

10.7k

C4 10µF

C1 TO C5: X5R OR X7R D1, D2: MMSD4148 D3: DIODES INC. B240A D4: DIODES INC. B120

1MHz, Wide Input Range 5V and 1.8V Outputs

4.7µF

ON OFF

0.1µF

D3 D4

R5 39k

C6 100pF

1nF

IN1VIN2

SHDN

BOOST1

SW1

TRACK/SS1

GND

3.3nF

LT3508

RT/SYNC

= 1MHz

BOOST2

SW2

FB2FB1

PG1 PG2TRACK/SS2

33.2k

330pF

OUT1

0.1µF

18.7k

47k

3.3µH

15.0k

R7 100k

3508 TA04

OUT2

1.8V 1A

C5 47µF

POWER GOOD

3508fb

TYPICAL APPLICATIONS

14V TO 36V

OUT2

OUT1

12V

1.4A*

L1 15µH

R2 1k

R1 154k

11.0k

4.7µF

C1 TO C5: X5R OR X7R D1, D2: MMSD4148 D3: DIODES INC. B240A D4: DIODES INC. B140 R2: USE 0.25W RESISTOR. FOR CONTINUOUS OPERATION ABOVE 30V, USE TWO 2k, 0.25W RESISTORS IN PARALLEL

*DERATE OUTPUT CURRENT AT HIGHER AMBIENT TEMPERATURES AND INPUT VOLTAGES TO MAINTAIN JUNCTION TEMPERATURE BELOW THE ABSOLUTE MAXIMUM

LT3508

1MHz, 5V and 12V Outputs

FB2FB1

PG1 PG2TRACK/SS2

33.2k

ON OFF

39k

100pF

0.1µF

L2 6.8µH

56.2k

10.7k

R8 100k

3508 TA06

OUT2 5V

1.4A*

C5 10µF

POWER GOOD

4.7µF

0.1µF

R5 43k

C6 100pF

1nF

IN1

BOOST1

SW1

LT3508

TRACK/SS1

GND

= 1MHz

SHDN

IN2

BOOST2

RT/SYNC

SW2

3508fb

LT3508

PACKAGE DESCRIPTION

2.74

(.108)

FE Package

16-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663)

Exposed Pad Variation BA

16 1514 13 12 11

4.90 – 5.10* (.193 – .201)

2.74

(.108)

10 9

6.60 ±0.10

4.50 ±0.10

RECOMMENDED SOLDER PAD LAYOUT

0.09 – 0.20

(.0035 – .0079)

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

SEE NOTE 4

0.65 BSC

4.30 – 4.50* (.169 – .177)

0.50 – 0.75

(.020 – .030)

MILLIMETERS

(INCHES)

(.108)

0.45 ±0.05

2.74

1.05 ±0.10

1345678

0.25 REF

0° – 8°

0.65

(.0256)

BSC

0.195 – 0.30

(.0077 – .0118)

TYP

4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT

*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE

2.74

(.108)

1.10

(.0433)

MAX

0.05 – 0.15

(.002 – .006)

FE16 (BA) TSSOP 0204

6.40

(.252)

BSC

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PACKAGE DESCRIPTION

LT3508

UF Package

24-Lead Plastic QFN (4mm × 4mm)

(Reference LTC DWG # 05-08-1697)

0.70 ±0.05

4.50 ± 0.05

3.10 ± 0.05

2.45 ± 0.05

(4 SIDES)

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

4.00 ± 0.10

(4 SIDES)

PIN 1 TOP MARK (NOTE 6)

NOTE:

1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE

0.25 ±0.05

0.50 BSC

PACKAGE OUTLINE

0.75 ± 0.05

2.45 ± 0.10

(4-SIDES)

0.200 REF

0.00 – 0.05

BOTTOM VIEW—EXPOSED PAD

R = 0.115

TYP

2423

PIN 1 NOTCH R = 0.20 TYP OR

0.35 × 45° CHAMFER

0.40 ± 0.10

(UF24) QFN 0105

0.25 ± 0.05

0.50 BSC

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

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LT3508

TYPICAL APPLICATION

5V, 1.8V Output from PCI Express

12V

40.2k

R10

14.7k

OUT1

0.9A

RELATED PARTS

L1 6.8µH

52.3k

10k

C6 10µF

C1 TO C6: X5R OR X7R D1, D2: MMSD4148 D3: DIODES INC. B140 D4: DIODES INC. B120

R5 43k

C9 100pF

4.7µF

0.1µF

D3 D4

0.047µF

SHDN

BOOST1

SW1

TRACK/SS1

C10

0.047µF

IN1VIN2

LT3508

GND

= 1MHz

BOOST2

SW2

RT/SYNC

FB2FB1

PG1 PG2TRACK/SS2

33.2k

47k

330pF

0.1µF

3.3µH

18.7k

15.0k

IN2

R7 100k

3508 TA05

3.3V

OUT2

1.8V

1.4A

C5 47µF

POWER GOOD

4.7µF

PART NUMBER DESCRIPTION COMMENTS

LT1765 25V, 2.75A (I

), 1.25MHz, High Efﬁ ciency Step-Down

OUT

VIN: 3V to 25V, V

DC/DC Converter

LT1766 60V, 1.2A (I

DC/DC Converter

LT1767 25V, 1.2A (I

), 200kHz, High Efﬁ ciency Step-Down

OUT

), 1.25MHz, High Efﬁ ciency Step-Down

OUT

VIN: 5.5V to 60V, V Packages

VIN: 3V to 25V, V

DC/DC Converter

LT1940/LT1940L Dual Monolithic 1.4A, 1.1MHz Step-Down Switching

VIN: 3.6V to 25V, V

Regulators

LTC3407 Dual 600mA, 1.5MHz, Synchronous Step-Down

VIN: 2.5V to 5.5V, V

Regulator

LT3493 1.2A, 750kHz Step-Down Switching Regulator in

2mm × 3mm DFN

LT3501/LT3510 Dual 3A/2A, 1.5MHz High Efﬁ ciency Step-Down

Switching Regulators

LT3506/LT3506A Dual Monolithic 1.6A, 1.1MHz Step-Down Switching

Regulators

LTC3701 Two Phase, Dual, 500kHz, Constant Frequency, Current

: 3.6V to 36V, V

: 3.6V to 25V, V

TSSOP20E Package

VIN: 3.6V to 25V, V TSSOPE Packages

VIN: 2.5V to 10V, V

Mode, High Efﬁ ciency Step-Down DC/DC Controller

LTC3736 Dual Two Phase, No R

with Output Tracking

LTC3737 Dual Two Phase, No R

Output Tracking

No R

is a trademark of Linear Technology Corporation.

SENSE

™, Synchronous Controller

SENSE

DC/DC Controller with

SENSE

: 2.75V to 9.8V, V

SSOP-24 Packages

: 2.75V to 9.8V, V

SSOP-24 Packages

= 1.2V, IQ = 1mA, S8, TSSOP16E Packages

OUT(MIN)

= 1.2V, IQ = 2.5mA, TSSOP16/TSSOP16E

OUT(MIN)

= 1.2V, IQ = 1mA, MS8, MS8E Packages

OUT(MIN)

= 1.25V, IQ = 3.8mA, TSSOP16E Packages

OUT(MIN)

= 0.6V, IQ = 40µA, MSE Package

OUT(MIN)

= 0.78V, IQ = 1.9mA, 2mm × 3mm DFN Package

OUT(MIN)

= 0.8V, IQ = 3.7mA, ISD < 10µA,

OUT(MIN)

= 0.8V, IQ = 3.8mA, 16-Lead DFN and 16-Lead

OUT(MIN)

= 0.8V, IQ = 460µA, SSOP-16 Package

OUT(MIN)

= 0.6V, IQ = 300µA, 4mm × 4mm QFN or

OUT(MIN)

= 0.6V, IQ = 220µA, 4mm × 4mm QFN or

OUT(MIN)

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507

●

www.linear.com

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LT 0208 REV B • PRINTED IN USA

LINEAR TECHNOLOGY LT3508 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual