LINEAR TECHNOLOGY LT3475, LT3475-1 Technical data

FEATURES

LT3475/LT3475-1

Dual Step-Down

1.5A LED Driver

U

DESCRIPTIO

■

True Color PWMTM Delivers Constant Color with

3000:1 Dimming Range

■

Wide Input Range: 4V to 36V Operating, 40V

Maximum

■

Accurate and Adjustable Control of LED Current

from 50mA to 1.5A

■

High Side Current Sense Allows Grounded Cathode

LED Operation

■

Open LED (LT3475) and Short Circuit Protection

■

LT3475-1 Drives LED Strings Up to 25V

■

Accurate and Adjustable 200kHz to 2MHz

Switching Frequency

■

Anti-Phase Switching Reduces Ripple

■

Uses Small Inductors and Ceramic Capacitors

■

Available in the Compact 20-Lead TSSOP Thermally

Enhanced Surface Mount Package

U

APPLICATIO S

■

Automotive and Avionic Lighting

■

Architectural Detail Lighting

■

Display Backlighting

■

Constant-Current Sources

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners. Patents Pending.

The LT®3475/LT3475-1 are dual step-down DC/DC
converters designed to operate as a constant-current
source. An internal sense resistor monitors the output
current allowing accurate current regulation ideal for
driving high current LEDs. The high side current sense al­lows grounded cathode LED operation. High output current
accuracy is maintained over a wide current range, from
50mA to 1.5A, allowing a wide dimming range. Unique
PWM circuitry allows a dimming range of 3000:1, avoid­ing the color shift normally associated with LED current
dimming.

The high switching frequency offers several advantages,
permitting the use of small inductors and ceramic capaci­tors. Small inductors combined with the 20 lead TSSOP
surface mount package save space and cost versus
alternative solutions. The constant switching frequency
combined with low-impedance ceramic capacitors result
in low, predictable output ripple.

With its wide input range of 4V to 36V, the LT3475/LT3475-1
regulate a broad array of power sources. A current mode
PWM architecture provides fast transient response and
cycle-by-cycle current limiting. Frequency foldback and
thermal shutdown provide additional protection.

TYPICAL APPLICATIO

Dual Step-Down 1.5A LED Driver

V

IN

2.2μF

5V TO 36V

10μH 10μH

0.1μF

1.5A LED

CURRENT

0.22μF

*DIMMING

CONTROL

4.7μF

BOOST1 BOOST2

SW1 SW2

OUT1 OUT2
LED1 LED2

PWM1 PWM2
V

C1

REF R

V

ADJ1

V

IN

LT3475

GND

SHDN

U

V

V

ADJ2

Effi ciency

95

VIN = 12V

0.22μF

DIMMING*
CONTROL

C2

T

fSW = 600kHz*SEE APPLICATIONS SECTION FOR DETAILS

24.3k

0.1μF

1.5A LED
CURRENT

2.2μF

3475 TA01

90

85

80

75

70

EFFICIENCY (%)

65

60

55

0

TWO SERIES CONNECTED
WHITE 1.5A LEDS

SINGLE WHITE 1.5A LED

0.5 1
LED CURRENT (A)

3475 TA01b

1.5

3475fb

1

LT3475/LT3475-1

WW

W

U

ABSOLUTE AXI U RATI GS

(Note 1)

VIN Pin .........................................................(-0.3V), 40V

BOOST Pin Voltage ...................................................60V

BOOST Above SW Pin ...............................................30V

OUT, LED, Pins (LT3475) ...........................................15V

OUT, LED Pins (LT3475-1) .........................................25V

PWM Pin ...................................................................15V

Pin ......................................................................6V

V

ADJ

, RT, REF Pins ..........................................................3V

V

C

SHDN Pin ...................................................................V

Maximum Junction Temperature (Note 2)............. 125°C

Operating Temperature Range (Note 3)

LT3475E/LT3475E-1 ............................. –40°C to 85°C

LT3475I/LT3475I-1 ............................. –40°C to 125°C

Storage Temperature Range ................... –65°C to 150°C

Lead Temperature Range (Soldering, 10 sec) .......300°C

IN

PIN CONFIGURATION

TOP VIEW

1

OUT1

2

LED1

3

BOOST1

4

SW1

5

V

IN

V

IN

SW2

BOOST2

LED2

OUT2

20-LEAD PLASTIC TSSOP

T

= 125°C, θJA = 30°C/W, θJC = 8°C/W

JMAX

EXPOSED PAD (PIN 21) IS GROUND AND MUST

BE ELECTRICALLY CONNECTED TO THE PCB.

6

7

8

9

10

21

FE PACKAGE

PWM1

20

V

19

ADJ1

V

18

C1

REF

17

SHDN

16

GND

15

R

14

T

V

13

C2

V

12

ADJ2

PWM2

11

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT3475EFE#PBF LT3475EFE#TRPBF LT3475EFE 20-Lead Plastic TSSOP –40°C to 85°C

LT3475IFE#PBF LT3475IFE#TRPBF LT3475IFE 20-Lead Plastic TSSOP –40°C to 125°C

LT3475EFE-1#PBF LT3475EFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 85°C

LT3475IFE-1#PBF LT3475IFE-1#TRPBF LT3475FE-1 20-Lead Plastic TSSOP –40°C to 125°C

Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based fi nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi cations, go to: http://www.linear.com/tapeandreel/

The

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifi cations are at TA = 25°C. VIN = 12V, V

PARAMETER CONDITIONS MIN TYP MAX UNITS

Minimum Input Voltage

Input Quiescent Current Not Switching 6 8 mA

Shutdown Current

SHDN = 0.3V, V

● denotes the specifi cations which apply over the full operating

BOOST

= V

OUT

BOOST

= 0V

= 16V, V

= 4V unless otherwise noted (Note 3)

OUT

●

3.7 4 V

0.01 2 μA

2

3475fb

LT3475/LT3475-1

ELECTRICAL CHARACTERISTICS

The ● denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T

= 25°C. VIN = 12V, V

A

PARAMETER CONDITIONS MIN TYP MAX UNITS

LED Pin Current V

ADJ

V

ADJ

Tied to V

Tied to V

REF

REF

• 2/3

• 7/30

LT3475E/LT3475E-1 0°C to 85°C

REF Voltage

Reference Voltage Line Regulation 4V < V

Reference Voltage Load Regulation 0 < I

Pin Bias Current (Note 4)

V

ADJ

Switching Frequency R

Maximum Duty Cycle R

Switching Phase R

Foldback Frequency R

< 40V 0.05 %/V

IN

< 500μA 0.0002 %/μA

REF

= 24.3k

T

= 24.3k

T

R

= 4.32k

T

R

= 100k

T

= 24.3k 150 180 210 Deg

T

= 24.3k, V

T

= 0V 80 kHz

OUT

SHDN Threshold (to Switch) 2.5 2.6 2.74 V

SHDN Pin Current (Note 5) V

SHDN

=

2.6V

PWM Threshold 0.3 0.8 1.2 V

Switching Threshold 0.8 V

V

C

Source Current VC = 1V 50 μA

V

C

Sink Current VC = 1V 50 μA

V

C

LED to V

LED to V

V

V

V

Transresistance 500 V/A

C

Current Gain 1 mA/μA

C

to Switch Current Gain 2.6 A/V

C

Clamp Voltage 1.8 V

C

Pin Current in PWM Mode VC = 1V, V

C

PWM

= 0.3V

OUT Pin Clamp Voltage (LT3475) 13.5 14 14.5 V

OUT Pin Current in PWM Mode V

OUT

= 4V, V

PWM

= 0.3V

Switch Current Limit (Note 6) 2.3 2.7 3.2 A

Switch V

CESAT

BOOST Pin Current I

ISW =1.5A 350 500 mV

=1.5A 25 40 mA

SW

Switch Leakage Current 0.1 10 μA

Minimum Boost Voltage Above SW 1.8 2.5 V

BOOST

= 16V, V

= 4V unless otherwise noted (Note 3)

OUT

0.97

●

0.94

0.336

0.325

●

0.31

●

1.22 1.25 1.27 V

●

●

530 600 640 kHz

●

90 95

1.00

1.03

1.04

0.350

0.364

0.375

0.385

40 400 nA

80
98

7911 μA

●

●

10 400 nA

25 50 μA

A
A
A
A
A

%
%
%

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.

Note 2: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specifi ed maximum operating junction
temperature may impair device reliability.

Note 3: The LT3475E and LT3475E-1 are guaranteed to meet performance
specifi cations from 0°C to 85°C. Specifi cations over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3475I and LT3475I-1
are guaranteed to meet performance specifi cations over the –40°C to
125°C operating temperature range.

Note 4: Current fl ows out of pin.
Note 5: Current fl ows into pin.
Note 6: Current limit is guaranteed by design and/or correlation to static

test. Slope compensation reduces current limit at higher duty cycles.

3475fb

3

LT3475/LT3475-1

UW

TYPICAL PERFOR A CE CHARACTERISTICS

LED Current vs V

1.50

TA = 25°C

1.25

1.00

0.75

0.50

LED CURRENT (A)

0.25

0

0

0.25 0.5 0.75 1

Switch Current Limit
vs Duty Cycle

3.0

2.5

2.0

1.5

1.0

CURRENT LIMIT (A)

0.5

0

0

20 40 60 80

ADJ

V

(V)

ADJ

TYPICAL

MINIMUM

DUTY CYCLE (%)

3475 G01

TA = 25°C

3475 G04

1.25

100

LED Current vs Temperature Switch On Voltage

1.2

1.0

0.8

0.6

0.4

LED CURRENT (A)

0.2

0

–50

–25 0

V

= V

• 2/3

ADJ

REF

V

= V

ADJ

TEMPERATURE (˚C)

• 7/30

REF

50 100 125

25 75

3475 G02

600

TA = 25°C

500

400

300

200

SWITCH ON VOLTAGE (mV)

100

0

0

Switch Current Limit vs
Temperature

3.5

3.0

2.5

2.0

1.5

CURRENT LIMIT (A)

1.0

0.5

0

–50

–25 0

50 100 125

25 75

TEMPERATURE (°C)

3475 G05

Current Limit vs Output Voltage

3.0

TA = 25°C

2.5

2.0

1.5

1.0

CURRENT LIMIT (A)

0.5

0

0

0.5 1.0
SWITCH CURRENT (A)

0.5 2.5 3.0 3.52.01.0

1.5

1.5

V

(V)

OUT

2.0

3475 G03

4.0

3475 G06

Oscillator Frequency
vs Temperature

700

RT = 24.3kΩ

650

600

550

500

OSCILLATOR FREQUENCY (kHz)

450

400

–50

–25 0

TEMPERATURE (˚C)

4

50 100 125

25 75

3475 G07

Oscillator Frequency Foldback Oscillator Frequency vs R

700

TA = 25°C

= 24.3kΩ

R

T

600

500

400

300

200

OSCILLATOR FREQUENCY (kHz)

100

0

0.5 1.0 1.5 2.5

0

V

OUT

2.0

(V)

3475 G08

TA = 25°C

1000

OSCILLATOR FREQUENCY (kHz)

10

1

RT (kΩ)

T

10 100

3475 G09

3475fb

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Boost Pin Current

35

TA = 25°C

30

25

20

15

10

BOOST PIN CURRENT (mA)

5

0

0.5 1.0 2.0

0

SWITCH CURRENT (A)

1.5

3475 G10

Quiescent Current

7

TA = 25°C

6

5

4

3

2

INPUT CURRENT (mA)

1

0

10 20 40

0

30

VIN (V)

3475 G11

LT3475/LT3475-1

Open-Circuit Output Voltage and
Input Current

50

TA = 25°C

45

40

35

30

25

20

15

OUTPUT VOLTAGE (V)

10

5

0

0

INPUT CURRENT

LT3475-1

10

V

LT3475-1

LT3475

OUTPUT VOLTAGE

LT3475

20

(V)

IN

14

12

INPUT CURRENT (mA)

10

8

6

4

2

0

30

40

3475 G12

Minimum Input Voltage, Single

Reference Voltage

1.28

1.27

1.26

(V)

1.25

REF

V

1.24

1.23

1.22
–50

–25 0

50 100 125

25 75

TEMPERATURE (˚C)

3475 G13

1.5A White LED

6

TA = 25°C

5

4

TO RUN

(V)

3

IN

V

2

1

0

0

0.5 1
LED CURRENT (A)

UUU

PI FU CTIO S

OUT1, OUT2 (Pins 1, 10): The OUT pin is the input to the
current sense resistor. Connect this pin to the inductor
and the output capacitor.

LED1, LED2 (Pins 2, 9): The LED pin is the output of
the current sense resistor. Connect the anode of the LED
here.

(Pins 5, 6): The VIN pins supply current to the internal

V

IN

circuitry and to the internal power switches and must be
locally bypassed.

SW1, SW2 (Pins 4, 7): The SW pin is the output of the
internal power switch. Connect this pin to the inductor,
switching diode and boost capacitor.

Minimum Input Voltage, Two Series
Connected 1.5A White LEDs

10

TA = 25°C

TO START

LED VOLTAGE

1.5

3475 G14

(V)

IN

V

9

8

7

6

5

TO RUN

0

TO START

LED VOLTAGE

0.5 1
LED CURRENT (A)

3475 G15

BOOST1, BOOST2 (Pins 3, 8): The BOOST pin is used to
provide a drive voltage, higher than the input voltage, to
the internal bipolar NPN power switch.

GND (Pins 15, Exposed Pad Pin 21): Ground. Tie the GND
pin and the exposed pad directly to the ground plane. The
exposed pad metal of the package provides both electrical
contact to ground and good thermal contact to the printed
circuit board. The exposed pad must be soldered to the
circuit board for proper operation. Use a large ground plane
and thermal vias to optimize thermal performance.

1.5

3475fb

5

LT3475/LT3475-1

UUU

PI FU CTIO S

RT (Pin 14): The RT pin is used to set the internal
oscillator frequency. Tie a 24.3k resistor from R

to GND

T

for a 600kHz switching frequency.

SHDN

(Pin 16): The

SHDN

pin is used to shut down the

switching regulator and the internal bias circuits. The

2.6V switching threshold can function as an accurate
undervoltage lockout. Pull below 0.3V to shut down the
LT3475/LT3475-1. Pull above 2.6V to enable the LT3475/
LT3475-1. Tie to V

IN

SHDN

function is unused.

if the

REF (Pin 17): The REF pin is the buffered output of the
internal reference. Either tie the REF pin to the V

ADJ

pin
for a 1.5A output current, or use a resistor divider to
generate a lower voltage at the V

pin. Leave this pin

ADJ

unconnected if unused.

BLOCK DIAGRAM

VC1, VC2 (Pins 18, 13): The VC pin is the output of the
internal error amp. The voltage on this pin controls the
peak switch current. Use this pin to compensate the
control loop.

, V

V

ADJ1

the internal voltage-to-current amplifi er. Connect the V

(Pins 19, 12): The V

ADJ2

pin is the input to

ADJ

ADJ

pin to the REF pin for a 1.5A output current. For lower
output currents, program the V
formula: I

= 1.5A • V

LED

ADJ

/1.25V.

pin using the following

ADJ

PWM1, PWM2 (Pins 20, 11): The PWM pin controls the
connection of the V
the PWM pin is low, the V

pin to the internal circuitry. When

C

pin is disconnected from the

C

internal circuitry and draws minimal current. If the PWM
feature is unused, leave this pin unconnected.

V

IN

C

IN

V

SHDN

IN

INT REG

AND

D1 D2

BOOST1

C1 C2

Q1 Q2

SW1 SW2

L1 L2

D3 D4

C

OUT1

D

LED1

C

C1

DRIVER

OUT1

0.067Ω 100Ω 0.067Ω100Ω

LED1 LED2

gm1 gm2

PWM 1

V

C1

Q3 Q4

1.25k

C1 C2

QR

QS

∑

UVLO

SLOPE COMP SLOPE COMP

MOSC 1 MOSC 2

SLAVE

OSC

FREQUENCY

FOLDBACK

–
+

2V 2V

V

ADJ1

R

T

V

R

IN

T

MASTER

OSC

∑

SLAVE

OSC

FREQUENCY
FOLDBACK

–
+

1.25V

REF

V

ADJ2

EXPOSED

PAD

QR

QS

DRIVER

1.25k

GND

BOOST2

OUT2

PWM2

V

C2

D

LED 2

C

C

OUT2

C2

3475 BD

6

3475fb

OPERATION

LT3475/LT3475-1

The LT3475 is a dual constant frequency, current mode
regulator with internal power switches capable of gen­erating constant 1.5A outputs. Operation can be best
understood by referring to the Block Diagram.

If the SHDN pin is tied to ground, the LT3475 is shut
down and draws minimal current from the input source
tied to V

. If the SHDN pin exceeds 1V, the internal bias

IN

circuits turn on, including the internal regulator, reference
and oscillator. The switching regulators will only begin to
operate when the SHDN pin exceeds 2.6V.

The switcher is a current mode regulator. Instead of directly
modulating the duty cycle of the power switch, the feedback
loop controls the peak current in the switch during each
cycle. Compared to voltage mode control, current mode
control improves loop dynamics and provides cycle-by­cycle current limit.

A pulse from the oscillator sets the RS fl ip-fl op and turns
on the internal NPN bipolar power switch. Current in the
switch and the external inductor begins to increase. When
this current exceeds a level determined by the voltage at
VC, current comparator C1 resets the fl ip-fl op, turning
off the switch. The current in the inductor fl ows through
the external Schottky diode and begins to decrease. The
cycle begins again at the next pulse from the oscillator.
In this way, the voltage on the VC pin controls the current
through the inductor to the output. The internal error
amplifi er regulates the output current by continually
adjusting the VC pin voltage. The threshold for switching
on the VC pin is 0.8V, and an active clamp of 1.8V limits
the output current.

The voltage on the V

pin sets the current through the

ADJ

LED pin. The NPN, Q3, pulls a current proportional to the
voltage on the V

pin through the 100Ω resistor. The gm

ADJ

amplifi er servos the VC pin to set the current through the

0.067Ω resistor and the LED pin. When the voltage drop
across the 0.067Ω resistor is equal to the voltage drop
across the 100Ω resistor, the servo loop is balanced.

Tying the REF pin to the V

pin sets the LED pin current

ADJ

to 1.5A. Tying a resistor divider to the REF pin allows the

programming of LED pin currents of less than 1.5A. LED
pin current can also be programmed by tying the V

ADJ

pin

directly to a voltage source.

An LED can be dimmed with pulse width modulation
using the PWM pin and an external NFET. If the PWM
pin is unconnected or is pulled high, the part operates
nominally. If the PWM pin is pulled low, the V

pin is dis-

C

connected from the internal circuitry and draws minimal
current from the compensation capacitor. Circuitry draw­ing current from the OUT pin is also disabled. This way,

pin and the output capacitor store the state of

the V

C

the LED pin current until the PWM is pulled high again.
This leads to a highly linear relationship between pulse
width and output light, allowing for a large and accurate
dimming range.

pin allows programming of the switching frequency.

The R

T

For applications requiring the smallest external components
possible, a fast switching frequency can be used. If low
dropout or very high input voltages are required, a slower
switching frequency can be programmed.

During startup V

will be at a low voltage. The NPN,

OUT

Q3, can only operate correctly with suffi cient voltage
of ≈1.7V at V
the V

pin high until V

C

, A comparator senses V

OUT

rises above 2V, and Q3 is op-

OUT

and forces

OUT

erating correctly.

The switching regulator performs frequency foldback
during overload conditions. An amplifi er senses when
V

is less than 2V and begins decreasing the oscillator

OUT

frequency down from full frequency to 15% of the nominal
frequency when V

= 0V. The OUT pin is less than 2V

OUT

during startup, short circuit, and overload conditions.
Frequency foldback helps limit switch current under these
conditions.

The switch driver operates either from V

or from the

IN

BOOST pin. An external capacitor and Schottky diode
are used to generate a voltage at the BOOST pin that
is higher than the input supply. This allows the driver
to saturate the internal bipolar NPN power switch for
effi cient operation.

3475fb

7

LT3475/LT3475-1

APPLICATIONS INFORMATION

Open Circuit Protection

The LT3475 has internal open-circuit protection. If the LED
is absent or is open circuit, the LT3475 clamps the voltage
on the LED pin at 14V. The switching regulator then oper­ates at a very low frequency to limit the input current. The
LT3475-1 has no internal open circuit protection. With the
LT3475-1, be careful not to violate the ABSMAX voltage of
th BOOST pin; if V

> 25V, external open circuit protection

IN

circuitry (as shown in Figure 1) may be necessary.The
output voltage during an open LED condition is shown in
the Typical Performance Characteristics section.

Undervoltage Lockout

Undervoltage lockout (UVLO) is typically used in situations
where the input supply is current limited, or has high source
resistance. A switching regulator draws constant power
from the source, so the source current increases as the
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.

An internal comparator will force the part into shut­down when V

falls below 3.7V. If an adjustable UVLO

IN

threshold is required, the SHDN pin can be used. The
threshold voltage of the SHDN pin comparator is 2.6V. An
internal resistor pulls 9μA to ground from the SHDN pin
at the UVLO threshold.

Choose resistors according to the following formula:

TH

2.6V

–2.6V

R1

–9μA

R2 =

V

VTH = UVLO Threshold

Example: Switching should not start until the input is
above 8V.

= 8V

V

TH

R1=100k

R2 =

2.6V

8V – 2.6V

= 57.6k

–9μA

100k

OUT

10k

GND

3475 F01

V

C

LT3475

V

C

3475 F02

22V

100k

Figure 1. External Overvoltage Protection
Circuitry for the LT3475-1

V

IN

C1

V

IN

R1

SHDN

R2

Figure 2. Undervoltage Lockout

2.6V

9μA

Keep the connections from the resistors to the SHDN pin
short and make sure the coupling to the SW and BOOST
pins is minimized. If high resistance values are used, the
SHDN pin should be bypassed with a 1nF capacitor to
prevent coupling problems from switching nodes.

Setting the Switching Frequency

The LT3475 uses a constant frequency architecture that
can be programmed over a 200kHz to 2MHz range with a
single external timing resistor from the R
A graph for selecting the value of R

T

pin to ground.

T

for a given operating

frequency is shown in the Typical Applications section.

Table 1. Switching Frequencies

SWITCHING FREQUENCY (MHz) RT (kΩ)

2 4.32

1.5 6.81

1.2 9.09

1 11.8

0.8 16.9

0.6 24.3

0.4 40.2

0.3 57.6

0.2 100

3475fb

8

APPLICATIONS INFORMATION

LT3475/LT3475-1

Table 1 shows suggested RT selections for a variety of
switching frequencies.

Operating Frequency Selection

The choice of operating frequency is determined by
several factors. There is a tradeoff between effi ciency and
component size. A higher switching frequency allows the
use of smaller inductors at the cost of increased switching
losses and decreased effi ciency.

Another consideration is the maximum duty cycle. In certain
applications, the converter needs to operate at a high duty
cycle in order to work at the lowest input voltage possible.
The LT3475 has a fi xed oscillator off time and a variable
on time. As a result, the maximum duty cycle increases
as the switching frequency is decreased.

Input Voltage Range

The minimum operating voltage is determined either by the
LT3475’s undervoltage lockout of 4V, or by its maximum
duty cycle. The duty cycle is the fraction of time that the
internal switch is on and is determined by the input and
output voltages:

V

OUT

+ V

F

F

DC =

VIN–VSW+ V

()

()

where VF is the forward voltage drop of the catch diode
(~0.4V) and V

is the voltage drop of the internal switch

SW

(~0.4V at maximum load). This leads to a minimum input
voltage of:

+ V

V

IN MIN

()

V

OUT

=

DC

MAX

F

–VF+ V

SW

The maximum operating voltage is determined by the
absolute maximum ratings of the V

and BOOST pins,

IN

and by the minimum duty cycle.

+ V

MIN

V

=

= t

OUT

DC

ON(MIN)

MIN

F

–VF+ V

• f

SW

is equal to 140ns and f is the switching

V

IN MAX

()

with DC

where t

ON(MIN)

frequency.

Example: f = 750kHz, V

DC

V

= 140ns • 750kHz = 0.105

MIN

3.4V + 0.4V

IN MAX

=

()

0.105

= 3.4V

OUT

– 0.4V + 0.4V = 36V

The minimum duty cycle depends on the switching fre­quency. Running at a lower switching frequency might
allow a higher maximum operating voltage. Note that
this is a restriction on the operating input voltage; the
circuit will tolerate transient inputs up to the Absolute
Maximum Ratings of the V
voltage should be limited to the V

and BOOST pins. The input

IN

operating range (36V)

IN

during overload conditions (short circuit or start up).

Minimum On Time

The LT3475 will regulate the output current at input volt­ages greater than V

IN(MAX)

. For example, an application

with an output voltage of 3V and switching frequency of

1.2MHz has a V

IN(MAX)

of 20V, as shown in Figure 3. Figure

4 shows operation at 35V. Output ripple and peak inductor

with DC

where t

= 1–t

MAX

0FF(MIN)

OFF(MIN)

is equal to 167ns and f is the switching

frequency.

Example: f = 600kHz, V

DC

V

= 1− 167ns • 600kHz = 0.90

MAX

4V + 0.4V

IN MIN

=

()

0.9

• f

= 4V

OUT

– 0.4V + 0.4V = 4.9V

V

OUT

500mV/DIV

(AC COUPLED)

1A/DIV

V

20V/DIV

I

L

SW

Figure 3. Operation at V
V

= 3V and fSW = 1.2MHHz

OUT

IN(MAX)

3475 F03

= 20V.

3475fb

9

LT3475/LT3475-1

APPLICATIONS INFORMATION

current have signifi cantly increased. Exceeding V

IN(MAX)

is safe if the external components have adequate ratings
to handle the peak conditions and if the peak inductor
current does not exceed 3.2A. A saturating inductor may
further reduce performance.

V

OUT

500mV/DIV

(AC COUPLED)

I

L

1A/DIV

V

SW

20V/DIV

3475 F04

Figure 4. Operation above V
Ripple and Peak Inductor Current Increases

IN(MAX)

. Output

Inductor Selection and Maximum Output Current

A good fi rst choice for the inductor value is:

L = (V

OUT

+ VF)•

1.2MHz
f

where VF is the voltage drop of the catch diode (~0.4V),
f is the switching frequency and L is in μH. With this value
the maximum load current will be above 1.6A at all duty
cycles. The inductor’s RMS current rating must be greater
than the maximum load current and its saturation current
should be at least 30% higher. For highest effi ciency,
the series resistance (DCR) should be less than 0.15Ω.
Table 2 lists several vendors and types that are suitable.
For robust operation at full load and high input voltages

> 30V), use an inductor with a saturation current

(V

IN

higher than 3.2A.

Table 2. Inductors

PART NUMBER

Sumida

CR43-3R3 3.3 1.44 0.086 3.5

CR43-4R7 4.7 1.15 0.109 3.5

CDRH4D16-3R3 3.3 1.10 0.063 1.8

CDRH4D28-3R3 3.3 1.57 0.049 3.0

CDRH4D28-4R7 4.7 1.32 0.072 3.0

CDRH6D26-5R0 5.0 2.20 0.032 2.8

CDRH6D26-5R6 5.6 2.0 0.036 2.8

CDRH5D28-100 10 1.30 0.048 3.0

CDRH5D28-150 15 1.10 0.076 3.0

CDRH73-100 10 1.68 0.072 3.4

CDRH73-150 15 1.33 0.130 3.4

CDRH104R-150 15 3.1 0.050 4.0

Coilcraft

DO1606T-332 3.3 1.30 0.100 2.0

DO1606T-472 4.7 1.10 0.120 2.0

DO1608C-332 3.3 2.00 0.080 2.9

DO1608C-472 4.7 1.50 0.090 2.9

MOS6020-332 3.3 1.80 0.046 2.0

MOS6020-472 10 1.50 0.050 2.0

DO3316P-103 10 3.9 0.038 5.2

DO3316P-153 15 3.1 0.046 5.2

VALUE

(μH)

I

RMS

(A)

DCR

()

HEIGHT

(mm)

The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current, and
reduces the output voltage ripple. If your load is lower than
the maximum load current, then you can relax the value of the
inductor and operate with higher ripple current. This allows
you to use a physically smaller inductor, or one with a lower
DCR resulting in higher effi ciency. In addition, low inductance
may result in discontinuous mode operation, which further
reduces maximum load current. For details of maximum
output current and discontinuous mode operation, see Linear
Technology’s Application Note 44. Finally, for duty cycles
greater than 50% (V

OUT/VIN

> 0.5), a minimum inductance

is required to avoid sub-harmonic oscillations:

10

L

= (V

MIN

OUT

+ VF)•

800kHz

f

3475fb

APPLICATIONS INFORMATION

LT3475/LT3475-1

The current in the inductor is a triangle wave with an average
value equal to the load current. The peak switch current
is equal to the output current plus half the peak-to-peak
inductor ripple current. The LT3475 limits its switch cur­rent in order to protect itself and the system from overload
faults. Therefore, the maximum output current that the
LT3475 will deliver depends on the switch current limit,
the inductor value, and the input and output voltages.

When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor

ΔIL=

1–DC

()

V

OUT

+ V

F

()

L•f

()

where f is the switching frequency of the LT3475 and L
is the value of the inductor. The peak inductor and switch
current is

ΔI

L

+

2

I

SW PK

= I

()

LPK

= I

()

OUT

To maintain output regulation, this peak current must be
less than the LT3475’s switch current limit I

LIM

. I

LIM

is at
least 2.3A at low duty cycles and decreases linearly to 1.8A
at DC = 0.9. The maximum output current is a function of
the chosen inductor value:

Input Capacitor Selection

Bypass the input of the LT3475 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type will work if there is
additional bypassing provided by bulk electrolytic capaci­tors or if the input source impedance is low. The following
paragraphs describe the input capacitor considerations in
more detail.

Step-down regulators draw current from the input supply
in pulses with very fast rise and fall times. The input ca­pacitor is required to reduce the resulting voltage ripple at
the LT3475 input and to force this switching current into a
tight local loop, minimizing EMI. The input capacitor must
have low impedance at the switching frequency to do this
effectively, and it must have an adequate ripple current rat­ing. With two switchers operating at the same frequency
but with different phases and duty cycles, calculating the
input capacitor RMS current is not simple. However, a
conservative value is the RMS input current for the channel

• I

that is delivering most power (V

OUT

OUT

):

C

INRMS

= I

OUT

V

OUT(VIN–VOUT

•

V

IN

)

I

OUT

<

2

I

OUT MAX

= 2.3A• 1–0.25•DC

= I

()

LIM

ΔI

L

–

2

ΔI

L

()

–

2

Choosing an inductor value so that the ripple current is
small will allow a maximum output current near the switch
current limit.

One approach to choosing the inductor is to start with the
simple rule given above, look at the available inductors,
and choose one to meet cost or space goals. Then use
these equations to check that the LT3475 will be able to
deliver the required output current. Note again that these
equations assume that the inductor current is continu­ous. Discontinuous operation occurs when I
than ΔI

L

/2.

OUT

is less

and is largest when VIN = 2V

(50% duty cycle). As the

OUT

second, lower power channel draws input current, the
input capacitor’s RMS current actually decreases as the
out-of-phase current cancels the current drawn by the
higher power channel. Considering that the maximum
load current from a single channel is ~1.5A, RMS ripple
current will always be less than 0.75A.

The high frequency of the LT3475 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 10μF. The combination
of small size and low impedance (low equivalent series
resistance or ESR) of ceramic capacitors makes them the
preferred choice. The low ESR results in very low voltage
ripple. Ceramic capacitors can handle larger magnitudes
of ripple current than other capacitor types of the same
value. Use X5R and X7R types.

3475fb

11

LT3475/LT3475-1

APPLICATIONS INFORMATION

An alternative to a high value ceramic capacitor is a
lower value ceramic along with a larger electrolytic
capacitor. The electrolytic capacitor likely needs to be greater
than 10μF in order to meet the ESR and ripple current
requirements. The input capacitor is likely to see high
surge currents when the input source is applied. Tanta­lum capacitors can fail due to an over-surge of current.
Only use tantalum capacitors with the appropriate surge
current rating. The manufacturer may also recommend
operation below the rated voltage of the capacitor.

A fi nal caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plug­ging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3475. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.

Output Capacitor Selection

RMS current rating of the output capacitor is usually not
of concern. It can be estimated with the formula:

I

C(RM S)

= ΔIL/12

The low ESR and small size of ceramic capacitors make
them the preferred type for LT3475 applications. Not all
ceramic capacitors are the same, however. Many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coeffi cients. In particular Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and at temperature extremes.
Because loop stability and transient response depend on
the value of C

, this loss may be unacceptable. Use X7R

OUT

and X5R types. Table 3 lists several capacitor vendors.

Table 3. Low ESR Surface Mount Capacitors.

VENDOR TYPE SERIES

Taiyo-Yuden Ceramic X5R, X7R

AVX Ceramic X5R, X7R

TDK Ceramic X5R, X7R

For most LEDs, a 2.2μF, 6.3V ceramic capacitor (X5R or
X7R) at the output results in very low output voltage ripple
and good transient response. Other types and values will
also work. The following discusses tradeoffs in output
ripple and transient performance.

The output capacitor fi lters the inductor current to
generate an output with low voltage ripple. It also stores
energy in order to satisfy transient loads and stabilizes the
LT3475’s control loop. Because the LT3475 operates at a
high frequency, minimal output capacitance is necessary.
In addition, the control loop operates well with or without
the presence of output capacitor series resistance (ESR).
Ceramic capacitors, which achieve very low output ripple
and small circuit size, are therefore an option.

You can estimate output ripple with the following
equation:

V

RIPPLE

where ΔI

= ΔIL / (8 • f • C

is the peak-to-peak ripple current in the

L

) for ceramic capacitors

OUT

inductor. The RMS content of this ripple is very low so the

Diode Selection

The catch diode (D3 from the Block Diagram) conducts
current only during switch off time. Average forward cur­rent in normal operation can be calculated from:

I

D(AVG)

= I

OUT

(VIN – V

OUT

)/V

IN

The only reason to consider a diode with a larger current
rating than necessary for nominal operation is for the
worst-case condition of shorted output. The diode cur­rent will then increase to one half the typical peak switch
current limit.

Peak reverse voltage is equal to the regulator input
voltage. Use a diode with a reverse voltage rating greater
than the input voltage. Table 4 lists several Schottky
diodes and their manufacturers.

Diode reverse leakage can discharge the output capacitor
during LED off times while PWM dimming. If operating at
high ambient temperatures, use a low leakage Schottky
for the widest PWM dimming range.

12

3475fb

APPLICATIONS INFORMATION

LT3475/LT3475-1

Table 4. Schottky Diodes

V

V

R

(V)

(A)

I

AVE

(A)

at 1A

F

(mV)

V

at 2A

F

(mV)

On Semiconductor

MBR0540 40 0.5 620

MBRM120E 20 1 530

MBRM140 40 1 550

Diodes Inc

B120 20 1 500

B130 30 1 500

B140HB 40 1 530

DFLS140 40 1.1 510

B240 40 2 500

International Rectifi er

10BQ030 30 1 420

BOOST Pin Considerations

The capacitor and diode tied to the BOOST pin gener­ate a voltage that is higher than the input voltage. In
most cases, a 0.22μF capacitor and fast switching diode
(such as the CMDSH-3 or MMSD914LT1) will work well.
Figure 5 shows three ways to arrange the boost circuit.
The BOOST pin must be more than 2.5V above the SW

pin for full effi ciency. For outputs of 3.3V and higher, the
standard circuit (Figure 5a) is best. For outputs between

2.8V and 3.3V, use a small Schottky diode (such as the
BAT-54). For lower output voltages, the boost diode can be
tied to the input (Figure 5b). The circuit in Figure 5a is more
effi cient because the BOOST pin current comes from a
lower voltage source. The anode of the boost diode can
be tied to another source that is at least 3V. For example, if
you are generating a 3.3V output, and the 3.3V output is on
whenever the LED is on, the BOOST pin can be
connected to the 3.3V output. For LT3475-1 applications
with higher output voltages, an additional Zener diode
may be necessary (Figure 5d) to maintain pin voltage
below the absolute maximum. In any case, be sure that
the maximum voltage at the BOOST pin is both less than
60V and the voltage difference between the BOOST and
SW pins is less than 30V.

The minimum operating voltage of an LT3475 application
is limited by the undervoltage lockout (~3.7V) and by the
maximum duty cycle. The boost circuit also limits the
minimum input voltage for proper start up. If the input
voltage ramps slowly, or the LT3475 turns on when the
output is already in regulation, the boost capacitor may
not be fully charged. Because the boost capacitor charges

V

BOOST

MAX V

V

BOOST

MAX V

D2

BOOST

LT3475

V

IN

– VSW≅ V

BOOST

BOOST

LT3475

V

IN

– VSW – V

BOOST

GND

IN

≅ 2V

IN

GND

Z

≅ VIN + V

(5d)

SW

SW

OUT

(5b)

– V

C3

V

OUT

D2

C3

V

OUT

3475 F05

Z

D2

BOOST

BOOST

LT3475

V

IN

– VSW≅ V

BOOST

GND

OUT

≅ VIN + V

SW

OUT

V

IN

V
MAX V

C3

V

OUT

V

IN

(5a)

V

> 3V

IN2

V

IN

D2

BOOST

LT3475

V

IN

GND

V

– VSW≅ V

BOOST

MAX V

BOOST

MINIMUM VALUE FOR V

≅ V

IN2

+ V

IN2

SW

C3

V

OUT

3475 F05

IN

=

3V

IN2

V

IN

(5c)

Figure 5. Generating the Boost Voltage

3475fb

13

LT3475/LT3475-1

APPLICATIONS INFORMATION

with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load current generally goes to zero
once the circuit has started. The typical performance char­acteristics section shows a plot of minimum load to start
and to run as a function of input voltage. Even without an
output load current, in many cases the discharged output
capacitor will present a load to the switcher that will allow
it to start. The plots show the worst case, where V

IN

is

ramping very slowly.

Programming LED Current

The LED current can be set by adjusting the voltage on the

pin. For a 1.5A LED current, either tie V

V

ADJ

to REF or

ADJ

to a 1.25V source. For lower output currents, program the

using the following formula: I

V

ADJ

= 1.5A • V

LED

ADJ

/1.25V.
Voltages less than 1.25V can be generated with a voltage
divider from the REF pin, as shown in Figure 6. In order
to have accurate LED current, precision resistors are
preferred (1% or better is recommended). Note that the

pin sources a small amount of bias current, so use

V

ADJ

the following formula to choose resistors:

the voltage on the V

pin by tying a low on resistance

ADJ

FET to the resistor divider string. This allows the se­lection of two different LED currents. For reliable op­eration program an LED current of no less than 50mA.
The maximum current dimming ratio (I
calculated from the maximum LED current (I
minimum LED current (I

I

MAX/IMIN

= I

RATIO

) as follows:

MIN

RATIO

MAX

) can be

) and the

Another dimming control circuit (Figure 8) uses the PWM
pin and an external NFET tied to the cathode of the LED.
An external PWM signal is applied to the PWM pin and the
gate of the NFET (For PWM dimming ratios of 20 to 1 or
less, the NFET can be omitted). The average LED current is
proportional to the duty cycle of the PWM signal. When the
PWM signal goes low, the NFET turns off, turning off the
LED and leaving the output capacitor charged. The PWM
pin is pulled low as well, which disconnects the V

pin,

C

storing the voltage in the capacitor tied there. Use the C-RC
string shown in Figure 8 and Figure 9 tied to the V

pin for

C

proper operation during startup. When the PWM pin goes
high again, the LED current returns rapidly to its previous
on state since the compensation and output capacitors are
at the correct voltage. This fast settling time allows the

R2 =

1.25V – V

To minimize the error from variations in V

R1

V

ADJ

ADJ

+ 50nA

pin current,

ADJ

use resistors with a parallel resistance of less than 4k. Use
resistor strings with a high enough series resistance so as not
to exceed the 500μA current compliance of the REF pin.

Dimming Control

There are several different types of dimming control
circuits. One dimming control circuit (Figure 7) changes

REF

R1

R2

Figure 6. Setting V

LT3475

V

ADJ

GND

3475 F06

with a Resistor Divider

ADJ

REF

R1

R2

DIM

Figure 7. Dimming with a MOSFET and Resistor Divider

PWM

100Hz TO

10kHz

Figure 8. Dimming Using PWM Signal

PWM

LED

V

ADJ

LT3475

GND

LT3475

GND

V

C

3475 F08

3475 F07

3.3nF

10k

0.1μF

3475fb

14

F

APPLICATIONS INFORMATION

LT3475/LT3475-1

LT3475 to maintain diode current regulation with PWM
pulse widths as short as 7.5 switching cycles (12.5μs for

= 600kHz). Maximum PWM period is determined by

f

SW

the system and is unlikely to be longer than 12ms. Using
PWM periods shorter than 100μs is not recommended.
The maximum PWM dimming ratio (PWM
calculated from the maximum PWM period (t
minimum PWM pulse width (t

t

MAX/tMIN

Total dimming ratio (DIM

= PWM

RATIO

RATIO

) as follows:

MIN

) is the product of the PWM

RATIO

) can be

) and

MAX

dimming ratio and the current dimming ratio.

Example:

I
t
I
PWM
DIM

= 1A, I

MAX

= 3.3μs (fSW = 1.4MHz)

MIN

= 1A/0.1A =10:1

RATIO

RATIO

RATIO

MIN

= 0.1A, t

MAX

= 9.9ms

= 9.9ms/3.3μs = 3000:1

= 10 • 3000 = 30000:1

To achieve the maximum PWM dimming ratio, use the
circuit shown in Figure 9. This allows PWM pulse widths
as short as 4.5 switching cycles (7.5μs for f

= 600kHz).

SW

Note that if you use the circuit in Figure 9, the rising edge
of the two PWM signals must align within 100ns.

220pF

R

T

1M

PWM1

Figure 9. Extending the PWM Dimming Range

R

T

LT3475

GND

V

3475 F09

C

3.3nF

10k

0.1μ

Layout Hints

As with all switching regulators, careful attention must
be paid to the PCB layout and component placement. To
maximize effi ciency, switch rise and fall times are made
as short as possible. To prevent electromagnetic interfer­ence (EMI) problems, proper layout of the high frequency
switching path is essential. The voltage signal of the SW
and BOOST pins have sharp rise and fall edges. Minimize
the area of all traces connected to the BOOST and SW
pins and always use a ground plane under the switching
regulator to minimize interplane coupling. In addition, the
ground connection for frequency setting resistor R
capacitors at V

, VC2 pins (refer to the Block Diagram)

C1

and

T

should be tied directly to the GND pin and not shared
with the power ground path, ensuring a clean, noise-free
connection.

PWM2SHDNPWM1

201918171615141312

123456789

V

IN

VIA TO LOCAL GND PLANE

Figure 10. Recommended Component Placement

11

10

3475 F10

3475fb

15

LT3475/LT3475-1

TYPICAL APPLICATIONS

V

5V TO 36V

D3

L2

μH

10

C5

2.2μF

6.3V

LED 1 LED 2

0.1

C1 TO C5: X5R OR X7R
D1, D2: DFLS140
D3, D4: MBR0540
LED CURRENT = 1A

Dual Step-Down 1A LED Driver

IN

C1

μF

4.7
50V

C4

0.22μF

6.3V

D1

C6

μF

R2
1k

R3

2k

V

IN

BOOST1 BOOST2

SW1

OUT1 OUT2

LED1 LED2

V

C1

REF R

V

ADJ1

LT3475

GND

SHDN

V

SW2

V

ADJ2

C2

T

C3

0.22

μF

6.3V

0.1μF

R1

24.3k

fSW = 600kHz

D4

L1

10μH

D2

C2

μF

2.2

6.3V

C7

3475 TA02

CURRENT

Dual Step-Down 1.5A LED Driver with 1200 : 1 True Color PWM Dimming

V

IN

6V TO 36V

D3

L2

10

C4

μF

2.2

6.3V

LED 11.5A LED

R3
10k

M1 M2

D1, D2: B260
D3, D4: MBR0540
C1 TO C5: X5R OR X7R
M1, M2: Si2302ADS
M3: 2n7002L

C8

0.1

μF

PWM1 PWM2

C1

μF

4.7
50V

C2

0.22μF

C6

3.3nF

6.3V

D1

μH

V

IN

BOOST1 BOOST2

LT3475

SW1

OUT1 OUT2

LED1 LED2

PWM1 PWM2

V

C1

REF

V

ADJ1

GND

C8
220p

M3

SHDN

1M
R2

V

SW2

V

ADJ2

D4

C3

0.22

μF

6.3V

D2

C2

R

T

10μH

C7

3.3nF

R1

24.3k

fSW = 600kHz

L1

2.2μF

6.3V

C5

LED 2

R4
10k

C9

0.1

μF

3475 TA03

1.5A LED
CURRENT

3475fb

16

TYPICAL APPLICATIONS

V

IN

5V TO 36V

D3

L2

μH

10

C4

μF

2.2

6.3V

C6

μF

0.1

C1

4.7
50V

C2

0.22μF

6.3V

D1

Step-Down 3A LED Driver

μF

V

IN

BOOST1 BOOST2

LT3475

SW1

OUT1 OUT2

V

C1

REF

V

ADJ1

GND

SHDN

SW2

LED1

LED2

V

ADJ2

LT3475/LT3475-1

D4

C3

0.22

μF

6.3V

D2

V

C2

R

T

R1

24.3k

L1

10μH

2.2

C7

0.1μF

LED 1

6.3V

C5

μF

3A LED
CURRENT

D1, D2: B240A
D3, D4: MBR0540
C1 TO C5: X5R OR X7R

10V TO 36V

C4

μF

2.2
10V

1.5A LED

CURRENT

fSW = 600kHz

3475 TA04

Dual Step-Down LED Driver with Series Connected LEDs

V

D3

IN

L2

μH

15

LED 1

C6

0.1

C1

μF

4.7
50V

C2

0.22μF
10V

D1

μF

V

IN

BOOST1 BOOST2

SW1

OUT1

LED1

V

C1

REF

V

ADJ1

LT3475

GND

SHDN

SW2

OUT2

LED2

V

V

ADJ2

C3

0.22

μF

10V

D2

C2

R

T

15μH

R1

24.3k

L1

LED 2

C7

0.1μF

LED 4LED 3

D4

C5

2.2μF
10V

1.5A LED
CURRENT

D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R

fSW = 600kHz

3475 TA05

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17

LT3475/LT3475-1

TYPICAL APPLICATIONS

V

IN

5V TO 28V

C1

4.7
35V

L2

μH

10

Dual Step-Down 1.5A Red LED Driver

μF

D3

C2

0.22μF
35V

V

IN

BOOST1 BOOST2

SW1

SHDN

LT3475

SW2

0.22
35V

D4

C3

μF

L1

10μH

C4

μF

2.2

6.3V

1.5A LED

CURRENT

D1, D2: B240A
D3, D4: MMSD4148T1
C1 TO C5: X5R OR X7R

0.1

D1

OUT1

LED1

V

C1

C6

μF

REF

V

ADJ1

GND

OUT2

LED2

V

V

ADJ2

D2

C5

2.2μF

6.3V

C2

R

T

R1

24.3k

C7

0.1μF

LED 2LED 1

fSW = 600kHz

3475 TA06

1.5A LED
CURRENT

18

3475fb

PACKAGE DESCRIPTION

LT3475/LT3475-1

FE Package

20-Lead Plastic TSSOP (4.4mm)

(Reference LTC DWG # 05-08-1663)

Exposed Pad Variation CB

3.86

(.152)

6.60 ±0.10

4.50 ±0.10

RECOMMENDED SOLDER PAD LAYOUT

0.09 – 0.20

(.0035 – .0079)

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE

SEE NOTE 4

0.65 BSC

4.30 – 4.50*
(.169 – .177)

0.50 – 0.75

(.020 – .030)

MILLIMETERS

(INCHES)

2.74

(.108)

0.45 ±0.05

1.05 ±0.10

0.25
REF

6.40 – 6.60*
(.252 – .260)

3.86

(.152)

20 19 18 17 16 15

1345678910

2

0° – 8°

0.65

(.0256)

BSC

0.195 – 0.30

(.0077 – .0118)

TYP

4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT

*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE

111214 13

2.74

(.108)

1.20

(.047)

MAX

0.05 – 0.15

(.002 – .006)

FE20 (CB) TSSOP 0204

6.40

(.252)

BSC

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
no responsibility is assumed for its use. Linear Technology Corporation makes no representation that
the interconnection of its circuits as described herein will not infringe on existing patent rights.

However,

3475fb

19

LT3475/LT3475-1

TYPICAL APPLICATION

Dual Step-Down 1.5A LED Driver with Four Series Connected LED Output

V

21V TO 36V

L1

33μH

R1

1k

R4
10k

D7

C4

2.2μF
25V

R6

Q1

100k

D1, D4: 7.5V ZENER DIODE
D2, D3: MMSD4148
D5, D6: B240A
D7, D8: 22V ZENER DIODE
R1, R2: USE 0.5W RESISTOR OF TWO 2k 0.25W RESISTORS IN PARALLEL
Q1, Q2: MMBT3904
C1 TO C5: X5R or X7R
*DERATE LED CURRENT AT ELEVATED AMBIENT TEMPERATURES TO MAINTAIN LT3475-1 JUNCTION TEMPERATURE BELOW 125 °C.

12V TO 18V LED VOLTAGE 12V TO 18V LED VOLTAGE

0.1μF

1.5A LED

CURRENT*

IN

C1

4.7μF

D2D1

50V

C2

0.22μF
16V

D5

C6

V

IN

BOOST1 BOOST2

SW1 SW2

OUT1 OUT2
LED1

V

C1

REF R

V

ADJ1

LT3475-1

GND

SHDN

LED2

V

ADJ2

V

C2

T

C3

0.22μF
16V

R3

24.3k

fSW = 600kHz

D3D6D4

L2

33μH

R2

1k

C7

0.1μF

1.5A LED

CURRENT*

C5

2.2μF
25V

100k

R5
10k

D8

R7

Q2

3475 TA08

RELATED PARTS

PART NUMBER DESCRIPTION COMMENTS

LT1618 Constant-Current, 1.4MHz, 1.5A Boost

Converter

LT3466 Dual Full Function Step-Up LED Driver Drivers Up to 20 LEDs, V

LT3474 36V, 1A (I

), 2MHz Step-Down

LED

LED Driver

LT3477 42V, 3A, 3.5MHz Boost, Buck-Boost,

Buck LED Driver

LT3479 3A, Full-Featured DC/DC Converter with

Soft-Start and Inrush Current Protection

LT3846 Dual 1.3A, 2MHz, LED Driver V

Linear Technology Corporation

20

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear.com

= 1.6V, V

V

IN(MIN)

IN(MAX)

MS10 Package

V

IN(MIN)

= 4V, V

= 36V, 400:1 True Color PWM, ISD < 1μA,

IN(MAX)

TSSOP16E Package

V

= 2.5V, V

IN(MIN)

IN(MAX)

QFN, TSSOP20E Packages

= 2.5V, V

V

IN(MIN)

IN(MAX)

DFN, TSSOP Packages

: 2.5V to 24V, V

IN

OUT(MAX)

DFN, TSSOP16E Packages

= 18V, V

: 2.7V to 24V, V

IN

= 25V, V

= 24V, V

= 35V, Analog/PWM, ISD < 1μA,

OUT(MAX)

= 40V, DFN, TSSOP16E Packages

OUT(MAX)

= 40V, Analog/PWM, ISD < 1μA,

OUT(MAX)

= 40V, Analog/PWM, ISD < 1μA,

OUT(MAX)

= 36V, 1000:1 True Color PWMTM Dimmin,

© LINEAR TECHNOLOGY CORPORATION 2006

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LT 1007 REV B • PRINTED IN USA