Linear Technology Analog Devices LT8301 Datasheet

42VIN Micropower No-Opto

2.7V TO 36V

+

= 12V)
= 24V)
= 36V)

Isolated Flyback Converter

with 65V/1.2A Switch

FEATURES DESCRIPTION

LT8301

n

AEC-Q100 Qualified for Automotive Applications

n

2.7V to 42V Input Voltage Range

n

1.2A, 65V Internal DMOS Power Switch

n

Low Quiescent Current:

100µA in Sleep Mode
350µA in Active Mode

n

Boundary Mode Operation at Heavy Load

n

Low-Ripple Burst Mode® Operation at Light Load

n

Minimum Load <0.5% (Typ) of Full Output

n

V

Set with a Single External Resistor

OUT

n

No Transformer Third Winding or Opto-Isolator

Required for Regulation

n

Accurate EN/UVLO Threshold and Hysteresis

n

Internal Compensation and Soft-Start

n

Output Short-Circuit Protection

n

5-Lead TSOT-23 Package

APPLICATIONS

n

Isolated Telecom, Automotive, Industrial, Medical

Power Supplies

n

Isolated Auxiliary/Housekeeping Power Supplies

The LT®8301 is a micropower isolated flyback converter.
By sampling the isolated output voltage directly from the
primary-side flyback waveform, the part requires no third
winding or opto-isolator for regulation. The output voltage
is programmed with a single external resistor. Internal
compensation and soft-start further reduce external com­ponent count. Boundary mode operation provides a small
magnetic solution with excellent load regulation. Low
ripple Burst Mode operation maintains high efficiency at
light load while minimizing the output voltage ripple. A

1.2A, 65V DMOS power switch is integrated along with
all high voltage circuitry and control logic into a 5-lead
ThinSOT™ package.

The LT8301 operates from an input voltage range of 2.7V
to 42V and can deliver up to 6W of isolated output power.
The high level of integration and the use of boundary
and low ripple burst modes result in a simple to use, low
component count, and high efficiency application solution
for isolated power delivery.

All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. Patents, including 5438499, 7463497, and 7471522.

TYPICAL APPLICATION

2.7V to 36VIN/5V

V

IN

10µF

EN/UVLO

Micropower Isolated Flyback Converter

OUT

V

IN

SW

LT8301

R

FB

GND

3:1

•

40µH 4.4µH

•

154k

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100µF

8301 TA01a

V

OUT

5V
6mA TO 0.40A (V
6mA TO 0.70A (V
6mA TO 1.00A (V
6mA TO 1.15A (V

–

V

OUT

IN
IN
IN
IN

= 5V)

Efficiency vs Load Current

90

85

80

75

EFFICIENCY (%)

70

65

60

0

0.2

0.4 0.6 0.8

LOAD CURRENT (A)

VIN = 5V

= 12V

V

IN

= 24V

V

IN

= 36V

V

IN

1.0 1.2

8301 TA01b

Rev. A

1

LT8301

EN/UVLO 1

TOP VIEW

IN

4 SW

PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS

(Note 1)

SW (Note 2) ............................................................. 65V

........................................................................... 42V

V

IN

EN/UVLO ................................................................... V

RFB ...................................................... VIN – 0.5V to V

IN
IN

Current into RFB ................................................... 200µA

Operating Junction Temperature Range (Notes 3, 4)

LT8301E, LT8301I .............................. –40°C to 125°C

GND 2

R

3

FB

S5 PACKAGE

5-LEAD PLASTIC TSOT-23

= 150°C/W

θ

JA

5 V

LT8301H ............................................ –40°C to 150°C

LT8301MP

......................................... –55°C to 150°C

Storage Temperature Range .................. –65°C to 150°C

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LT8301ES5#TRMPBF LT8301ES5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8301IS5#TRMPBF LT8301IS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8301HS5#TRMPBF LT8301HS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 150°C
LT8301MPS5#TRMPBF LT8301MPS5#TRPBF LTGMF 5-Lead Plastic TSOT-23 –55°C to 150°C

AUTOMOTIVE PRODUCTS**

LT8301ES5#WTRMPBF LT8301ES5#WTRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8301IS5#WTRMPBF LT8301IS5#WTRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 125°C
LT8301HS5#WTRMPBF LT8301HS5#WTRPBF LTGMF 5-Lead Plastic TSOT-23 –40°C to 150°C
Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.

Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix.

**Versions of this part are available with controlled manufacturing to support the quality and reliability requirements of automotive applications. These
models are designated with a #W suffix. Only the automotive grade products shown are available for use in automotive applications. Contact your local
Analog Devices account representative for specific product ordering information and to obtain the specific Automotive Reliability reports for thesemodels.

2

Rev. A

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LT8301

ELECTRICAL CHARACTERISTICS

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 5V, V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT

V

IN

I

Q

I

HYS

f

MIN

t

ON(MIN)

t

OFF(MAX)

I

SW(MAX)

I

SW(MIN)

R

DS(ON)

I

LKG

I

RFB

Input Voltage Range

UVLO Threshold Rising

V

IN

Falling

VIN Quiescent Current V

V

EN/UVLO
EN/UVLO

= 0.2V

= 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)

EN/UVLO Shutdown Threshold For Lowest Off I
EN/UVLO Enable Threshold Falling

Hysteresis

EN/UVLO Hysteresis Current V

EN/UVLO

V

EN/UVLO

V

EN/UVLO

= 0.2V

= 1.1V

= 1.3V

Minimum Switching Frequency 9.4 10 10.6 kHz
Minimum Switch-On Time 170 ns
Maximum Switch-Off Time Backup Timer 190 µs
Maximum SW Current Limit
Minimum SW Current Limit
Switch On-Resistance ISW = 500mA 0.4 Ω
Switch Leakage Current VIN = 42V, VSW = 65V 0.1 0.5 µA
RFB Regulation Current

Regulation Current Line Regulation 2.7V ≤ VIN ≤ 42V 0.02 0.1 %/V

R

FB

= VIN unless otherwise noted.

EN/UVLO

l

Q

l

l

l

l

2.7 42 V

2.5

2.65 V

2.3

0.8
215
100
350

2 µA

µA
µA
µA

0.2 0.55 V

1.204 1.228

1.248 V

0.014

–0.1

2.2

–0.1

0

2.5

0

0.1

2.8

0.1

µA
µA
µA

1.200 1.375 1.550 A

0.22 0.29 0.36 A

97.5 100 102.5 µA

V

V

Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.

Note 2: The SW pin is rated to 65V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 65V as shown
in Figure5.

Note 3: The LT8301E is guaranteed to meet performance specifications
from 0°C to 125°C operating junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.

The LT8301I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8301H is guaranteed over the full –40°C to
150°C operating junction temperature range. The LT8301MP is guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.

Note 4: The LT8301 includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 150°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.

For more information www.analog.com

Rev. A

3

LT8301

350

8301 G03

1.2

SWITCHING FREQUENCY (kHz)

I

(µA)

8301 G07

8301 G02

TYPICAL PERFORMANCE CHARACTERISTICS

Output Load and Line Regulation Output Short-Circuit Protection

5.20
FRONT PAGE APPLICATION

5.15

5.10

5.05

5.00

4.95

OUTPUT VOLTAGE (V)

4.90

4.85

4.80

0.4 0.8

0.2

0

0.6 1.0

LOAD CURRENT (A)

VIN = 5V

= 12V

V

IN

= 24V

V

IN

= 36V

V

IN

1.2

8301 G01

Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Waveforms

6

FRONT PAGE APPLICATION

5

4

3

2

OUTPUT VOLTAGE (V)

1

0

0.2 0.6

0

VIN = 5V

= 12V

V

IN

= 24V

V

IN

= 36V

V

IN

0.4 0.8
LOAD CURRENT (A)

1.0

1.2

= 25°C, unless otherwise noted.

T

A

Switching Frequency
vs Load Current

FRONT PAGE APPLICATION

300

250

200

150

100

50

1.4

1.6

0

0

0.2 0.4
LOAD CURRENT (A)

VIN = 5V
V

IN

V

IN

V

IN

0.8

0.6 1.0

= 12V
= 24V
= 36V

V

OUT

50mV/DIV

V

SW

20V/DIV

FRONT PAGE APPLICATION
VIN = 12V

= 600mA

I

LOAD

VIN Shutdown Current

10

8

6

Q

4

2

0

0

TJ = 150°C
T
T

5 15

= 25°C

J

= –55°C

J

10

5µs/DIV

25 45

30

20

VIN (V)

V

OUT

50mV/DIV

V

SW

20V/DIV

8301 G04

FRONT PAGE APPLICATION

= 12V

V

IN

= 200mA

I

LOAD

5µs/DIV

8301 G05

VIN Quiescent Current,
Sleep Mode

140

130

120

110

(µA)

100

Q

I

90

80

70

60

5 45

35

40

0

20

10

15

V

(V)

IN

TJ = 150°C

= 25°C

T

J

= –55°C

T

J

25

40

30

35

V

OUT

50mV/DIV

V

SW

20V/DIV

FRONT PAGE APPLICATION
V

= 12V

IN

= 6mA

I

LOAD

VIN Quiescent Current,
Active Mode

400

380

360

(µA)

340

Q

I

320

300

280

5 45

0

10

TJ = 150°C

= 25°C

T

J

= –55°C

T

J

40

35

8301 G06

20µs/DIV

20

25

15

30

V

(V)

IN

4

Rev. A

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TYPICAL PERFORMANCE CHARACTERISTICS

V

(V)

1.245

8301 G10

150

I

(µA)

5

8301 G11

150

I

(µA)

105

8301 G12

150

RESISTANCE (mΩ)

1000

8301 G13

150

I

(A)

150

8301 G14

1.6

FREQUENCY (kHz)

150

8301 G15

600

FREQUENCY (kHz)

20

150

TIME (ns)

500

8301 G17

150

TIME (ns)

500

8301 G18

150

= 25°C, unless otherwise noted.

T

A

LT8301

EN/UVLO Enable Threshold EN/UVLO Hysteresis Current R

1.240

1.235

1.230

1.225

EN/UVLO

1.220

1.215

1.210

1.205

800

600

400

200

0

R

DS(ON)

TEMPERATURE (°C)

7550 125100250–25–50

7550 125100250–25–50

TEMPERATURE (°C)

4

3

HYS

2

1

0

TEMPERATURE (°C)

7550 125100250–25–50

Switch Current Limit Maximum Switching Frequency

MAXIMUM CURRENT LIMIT

MINIMUM CURRENT LIMIT

–25

0

50

25

TEMPERATURE (°C)

75

100

125

SW

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0

–50

RFB

Regulation Current

FB

104
103
102
101
100

99
98
97
96
95

500

400

300

200

100

0

–25 25

–50

TEMPERATURE (°C)

0 50

TEMPERATURE (°C)

7550 125100250–25–50

100

75

125

Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time

15

10

5

0

–50

–25 0 25 50

TEMPERATURE (°C)

75 100 125

400

300

200

100

0

TEMPERATURE (°C)

7550 125100250–25–50

For more information www.analog.com

400

300

200

100

0

TEMPERATURE (°C)

7550 125100250–25–50

Rev. A

5

LT8301

8301 BD

V

T1

+

–

PIN FUNCTIONS

EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8301. Pull the pin
below 0.2V to shut down the LT8301. This pin has an
accurate 1.228V threshold and can be used to program a
VIN undervoltage lockout (UVLO) threshold using a resis­tor divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of VIN UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.

GND (Pin 2): Ground. Tie this pin directly to local ground
plane.

RFB (Pin 3): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer primary

BLOCK DIAGRAM

IN

C

IN

R

FB

SW pin. The ratio of the RFB resistor to an internal 10k
resistor, times a trimmed 1.0V reference voltage, deter­mines the output voltage (plus the effect of any non-unity
transformer turns ratio). Minimize trace area at this pin.

SW (Pin 4): Drain of the 65V Internal DMOS Power
Switch. Minimize trace area at this pin to reduce EMI and
voltage spikes.

VIN (Pin 5): Input Supply. The VIN pin supplies current
to internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the R

pin. Locally

FB

bypass this pin to ground with a capacitor.

D

OUT

:1

N

PS

•

PRI

L

SEC

L

•

V

OUT

C

OUT

V

OUT

3 45

R

IN

1:4

FB

BOUNDARY

DETECTOR

M2M3

OSCILLATOR

SWV

–

25µA

R

REF

10kΩ

1.0V

g

m

+

–

S

A3

R Q

DRIVER

M1

+

R1

EN/UVLO

1

R2

2.5µA

1.228V

M4

–

A1

+

V

IN

REFERENCE

REGULATORS

+

A2

R

SENSE

–

GND

2

6

Rev. A

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OPERATION

LT8301

The LT8301 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to commu­nicate the output voltage information from the isolated
secondary side of the transformer to the primary side
for regulation. Historically, opto-isolators or extra trans­former windings communicate this information across
the isolation boundary. Opto-isolator circuits waste output
power, and the extra components increase the cost and
physical size of the power supply. Opto-isolators can also
cause system issues due to limited dynamic response,
nonlinearity, unit-to-unit variation and aging over life­time. Circuits employing extra transformer windings also
exhibit deficiencies, as using an extra winding adds to
the transformer’s physical size and cost, and dynamic
response is often mediocre.

The LT8301 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this man­ner, neither opto-isolator nor extra transformer winding
is required for regulation. Since the LT8301 operates
in either boundary conduction mode or discontinuous
conduction mode, the output voltage is always sampled
on the SW pin when the secondary current is zero. This
method improves load regulation without the need of
external load compensation components.

The LT8301 is a simple to use micropower isolated fly­back converter housed in a 5-lead TSOT-23 package. The
output voltage is programmed with a single external resis­tor. By integrating the loop compensation and soft-start
inside, the part further reduces the number of external
components. As shown in the Block Diagram, many of
the blocks are similar to those found in traditional switch­ing regulators including reference, regulators, oscillator,
logic, current amplifier, current comparator, driver, and
power switch. The novel sections include a flyback pulse
sense circuit, a sample-and-hold error amplifier, and a
boundary mode detector, as well as the additional logic for
boundary conduction mode, discontinuous conduction
mode, and low ripple Burst Mode operation.

Boundary Conduction Mode Operation

The LT8301 features boundary conduction mode opera­tion at heavy load, where the chip turns on the primary
power switch when the secondary current is zero.
Boundary conduction mode is a variable frequency, vari­able peak-current switching scheme. The power switch

on and the transformer primary current increases

turns
until an internally controlled peak current limit. After the
power switch turns off, the voltage on the SW pin rises to
the output voltage multiplied by the primary-to-secondary
transformer turns ratio plus the input voltage. When the
secondary current through the output diode falls to zero,
the SW pin voltage collapses and rings around VIN. A
boundary mode detector senses this event and turns the
power switch back on.

Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc­tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit sub-harmonic oscillation.

Discontinuous Conduction Mode Operation

As the load gets lighter, boundary conduction mode
increases the switching frequency and decreases the
switch peak current at the same ratio. Running at a higher
switching frequency up to several MHz increases switch­ing and gate charge losses. To avoid this scenario, the
LT8301 has an additional internal oscillator, which clamps
the maximum switching frequency to be less than 430kHz
(typ). Once the switching frequency hits the internal fre­quency clamp, the part starts to delay the switch turn-on
and operates in discontinuous conduction mode.

Low Ripple Burst Mode

Unlike traditional flyback converters, the LT8301 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling

Operation

For more information www.analog.com

Rev. A

7

LT8301

OPERATION

of the output voltage. The inherent minimum switch cur­rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.

As the load gets very light, the LT8301 starts to fold
back the switching frequency while keeping the mini­mum switch current limit. So the load current is able to
decrease while still allowing minimum switch-off time for

APPLICATIONS INFORMATION

Output Voltage

The R
only external resistor used to program the output voltage.
The LT8301 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.

Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the V
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:

V
VF = Output diode forward voltage

I
ESR = Total impedance of secondary circuit
N

The flyback voltage is then converted to a current I
the flyback pulse sense circuit (M2 and M3). This current
I

RFB

generate a ground-referred voltage. The resulting volt-

age feeds to the inverting input of the sample-and-hold

error amplifier. Since the sample-and-hold error ampli-

fier samples the voltage when the secondary current is

zero, the (I

assumed to be zero.

resistor as depicted in the Block Diagram is the

FB

supply. The

IN

= (V

FLBK

= Transformer secondary current

SEC

= Transformer effective primary-to-secondary

PS

OUT

+ VF + I

• ESR) • N

SEC

PS

turns ratio

RFB

also flows through the internal 10k R

• ESR) term in the V

SEC

FLBK

resistor to

REF

equation can be

by

the sample-and-hold error amplifier. Meanwhile, the part
switches between sleep mode and active mode, thereby
reducing the effective quiescent current to improve light
load efficiency. In this condition, the LT8301 operates in
low ripple Burst Mode. The 10kHz (typ) minimum switch­ing frequency determines how often the output voltage is
sampled and also the minimum load requirement.

An internal trimmed reference voltage,V

1.0V, feeds

IREF

to the non-inverting input of the sample-and-hold error
amplifier. The relatively high gain in the overall loop
causes the voltage across R
to V
V

V

I

. The resulting relationship between V

IREF

can be expressed as:

IREF

⎛
⎜

⎝

or

V

IREF

RFB

⎞

V

FLBK

R

FB

FLBK

⎟
⎠

⎛

=

⎜
⎝

•R

V
R

REF

IREF

REF

= V

⎞
⎟

⎠

•R

IREF

FB

= Internal trimmed reference voltage

= RFB regulation current = 100µA

Combination with the previous V

equation for V

, in terms of the RFB resistor, trans-

OUT

resistor to be nearly equal

REF

= I

RFB•RFB

equation yields an

FLBK

FLBK

and

former turns ratio, and diode forward voltage:

⎛

⎞

R

V

= 100µA •

OUT

FB

− V

⎜

N

⎝

PS

F

⎟
⎠

Output Temperature Coefficient

The first term in the V

equation does not have tem-

OUT

perature dependence, but the output diode forward volt­age V

has a significant negative temperature coefficient

F

(–1mV/°C to –2mV/°C). Such a negative temperature coef­ficient produces approximately 200mV to 300mV voltage
variation on the output voltage across temperature.

Rev. A

8

For more information www.analog.com

APPLICATIONS INFORMATION



LT8301

For higher voltage outputs, such as 12V and 24V, the

output diode temperature coefficient has a negligible

effect on the output voltage regulation. For lower voltage

outputs, such as 3.3V and 5V, however, the output diode

temperature coefficient does count for an extra 2% to 5%

output voltage regulation. For customers requiring tight

output voltage regulation across temperature, please refer

to other ADI parts with integrated temperature compensa-

tion features.

Selecting Actual R

Resistor Value

FB

The LT8301 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,

the scheme contains repeatable delays and error sources,

which will affect the output voltage and force a re-evalua-

tion of the RFB resistor value. Therefore, a simple two-step

process is required to choose feedback resistor RFB.
Rearrangement of the expression for V

Voltage section yields the starting value for R

• V

N

( )

RFB=

V
V

N

PS

PS

100µA

= Output voltage

OUT

= Output diode forward voltage = ~0.3V

F

= Transformer effective primary-to-secondary

OUT

+ V

F

in the Output

OUT

:

FB

turns ratio

Power up the application with the starting R

value and

FB

other components connected, and measure the regulated

output voltage, V

OUT(MEAS)

. The final RFB value can be

adjusted to:

V

=

V

OUT(MEAS)

OUT

•R

FB

R

FB(FINAL)

Once the final RFB value is selected, the regulation accu-

racy from board to board for a given application will be

very consistent, typically under ±5% when including

device variation of all the components in the system

(assuming resistor tolerances and transformer windings

matching within ±1%). However, if the transformer or

the output diode is changed, or the layout is dramatically

altered, there may be some change in V

OUT

.

Output Power

A flyback converter has a complicated relationship
between the input and output currents compared to a
buck or a boost converter. A boost converter has a rela­tively constant maximum input current regardless of input
voltage and a buck converter has a relatively constant
maximum output current regardless of input voltage. This
is due to the continuous non-switching behavior of the
two currents. A flyback converter has both discontinu­ous input and output currents which make it similar to
a non-isolated buck-boost converter. The duty cycle will
affect the input and output currents, making it hard to
predict output power. In addition, the winding ratio can
be changed to multiply the output current at the expense
of a higher switch voltage.

The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 50V dur

­ing the switch-off time. 15V of margin is left for leakage
inductance voltage spike. T

o achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 50V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.

One design example would be a 5V output converter with
a minimum input voltage of 8V and a maximum input volt­age of 32V. A three-to-one winding ratio fits this design
example perfectly and outputs equal to 5.42W at 32V but
lowers to 2.71W at 8V.

The following equations calculate output power:

P

= η • VIN•D •I

OUT

η = Efficiency =  85%

D = Duty Cycle =

I

SW(MAX)

= Maximum switch current limit = 1.2A (min)

SW(MAX)

V

( )

V

( )

OUT

OUT

+ V

• 0.5

+ V

F

•NPS+ V

F

•N

PS

IN

Rev. A

For more information www.analog.com

9

LT8301

7

8301 F01

40

OUTPUT POWER (W)

7

8301 F02

40

OUTPUT POWER (W)

7

8301 F03

40

OUTPUT POWER (W)

7

8301 F04

40

OUTPUT POWER (W)

APPLICATIONS INFORMATION

MAXIMUM

6

OUTPUT

CURRENT

5

4

3

2

1

0

0

10 20

INPUT VOLTAGE (V)

N = 5:1

N = 3:1

N = 2:1

N = 1:1

30

Figure1. Output Power for 3.3V Output

MAXIMUM

6

OUTPUT

CURRENT

5

4

3

2

N = 3:2

N = 1:1

N = 2:3

N = 1:3

MAXIMUM

6

OUTPUT

CURRENT

5

4

3

2

1

0

0

10 20

INPUT VOLTAGE (V)

N = 4:1

N = 3:1

N = 2:1

N = 1:1

30

Figure2. Output Power for 5V Output

MAXIMUM

6

OUTPUT

CURRENT

5

4

3

2

N = 4:5
N = 1:2

N = 1:3

N = 1:5

Primary Inductance Requirement

The LT8301 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction of
secondary current reflects the output voltage on the pri­mary SW pin. The sample-and-hold error amplifier needs
a minimum
put voltage. In order to ensure proper sampling, the sec­ondary winding needs to conduct current for a minimum
of 450ns. The following equation gives the minimum
value for primary-side magnetizing inductance:

t
I

L

≥

PRI

OFF(MIN)

SW(MIN)

1

0

0

10 20

INPUT VOLTAGE (V)

30

1

0

0

10 20

INPUT VOLTAGE (V)

30

Figure3. Output Power for 12V Output Figure4. Output Power for 24V Output

In addition to the primary inductance requirement for
the minimum switch-off time, the LT8301 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 170ns.
This minimum switch-on time is mainly for leading-edge

450ns to settle and sample the reflected out-

blanking the initial switch turn-on current spike. If the
inductor current exceeds the desired current limit during
that time, oscillation may occur at the output as the cur­rent control loop will lose its ability to regulate. Therefore,
the following equation relating to maximum input voltage
must also be followed in selecting primary-side magnetiz-

t

OFF(MIN)•NPS

I

SW(MIN)

• V

+ V

( )

OUT

F

= Minimum switch-off time = 450ns

= Minimum switch current limit = 290mA (typ)

ing inductance:

t

PRI

≥

ON(MIN)

L

t

ON(MIN)

= Minimum switch-on time = 170ns

• V

I

SW(MIN)

IN(MAX)

Rev. A

10

For more information www.analog.com

APPLICATIONS INFORMATION

LT8301

In general, choose a transformer with its primary mag­netizing inductance about 30% larger than the minimum
values calculated above.

A transformer with much larger

inductance will have a bigger physical size and may cause

Analog Devices has worked with several leading magnetic
component manufacturers to produce pre-designed fly­back transformers for use with the LT8301. Table1 shows
the details of these transformers.

instability at light load.

Turns Ratio

Selecting a Transformer

Transformer specification and design is perhaps the most
critical part of successfully applying the LT8301. In addi­tion to the usual list of guidelines dealing with high fre­quency isolated power supply transformer design, the
following information should be carefully considered.

Table1. Predesigned Transformers—Typical Specifications

TRANSFORMER
PART NUMBER

750313973 15.24 × 13.34 × 11.43 40 1 4:1 80 40 Würth Electronik 8 to 36 3.3 0.80
750370047 13.35 × 10.8 × 9.14 30 1 3:1:1 60 12.5 Würth Electronik 8 to 32 5 0.55
750313974 15.24 × 13.34 × 11.43 40 1 3:1 80 50 Würth Electronik 8 to 36 5 0.55
750313970 15.24 × 13.34 × 11.43 40 1 2:1 80 70 Würth Electronik 18 to 42 3.3 0.75
750310799 9.14 × 9.78 × 10.54 25 0.125 1:1:0.33 60 74 Würth Electronik 8 to 30 12 0.22
750313972 15.24 × 13.34 × 11.43 40 1 1:1 80 185 Würth Electronik 18 to 42 5 0.42
750313975 15.24 × 13.34 × 11.43 40 1 1:2 110 865 Würth Electronik 8 to 36 24 0.12
750313976 15.24 × 13.34 × 11.43 40 1 1:4 110 2300 Würth Electronik 8 to 32 48 0.05
12387-T036 15.5 × 12.5 × 11.5 40 2 4:1 160 25 Sumida 8 to 36 3.3 0.80
12387-T037 15.5 × 12.5 × 11.5 40 2 3:1 210 30 Sumida 8 to 36 5 0.55
12387-T040 15.5 × 12.5 × 11.5 40 1.5 2:1 210 50 Sumida 18 to 42 3.3 0.75
12387-T041 15.5 × 12.5 × 11.5 40 1.5 1:1 210 200 Sumida 18 to 42 5 0.42
12387-T038 15.5 × 12.5 × 11.5 40 2 1:2 220 460 Sumida 8 to 36 24 0.12
12387-T039 15.5 × 12.5 × 11.5 40 2 1:4 220 2200 Sumida 8 to 32 48 0.05
PA3948.003NL 15.24 × 13.08 × 11.45 40 1.45 4:1 210 26 Pulse Engineering 8 to 36 3.3 0.80
PA3948.004NL 15.24 × 13.08 × 11.45 40 1.95 3:1 220 29 Pulse Engineering 8 to 36 5 0.55
PA3948.001NL 15.24 × 13.08 × 11.45 40 1.45 2:1 410 70 Pulse Engineering 18 to 42 3.3 0.75
PA3948.002NL 15.24 × 13.08 × 11.45 40 1.45 1:1 405 235 Pulse Engineering 18 to 42 5 0.42
PA3948.005NL 15.24 × 13.08 × 11.45 40 1.60 1:2 220 1275 Pulse Engineering 8 to 36 24 0.12
PA3948.006NL 15.24 × 13.08 × 11.45 40 1.65 1:4 220 3350 Pulse Engineering 8 to 32 48 0.05

DIMENSIONS

(W × L × H) (mm)

L

PRI

(µH)

L

LKG

(µH) NP

:NS

Note that when choosing the R
voltage, the user has relative freedom in selecting a trans­former turns ratio to suit a given application. In contrast,
the use of simple ratios of small integers, e.g., 3:1, 2:1,
1:1, provides more freedom in settling total turns and
mutual inductance.

R

R

PRI

(mΩ)

SEC

(mΩ) VENDOR

resistor to set output

FB

TARGET APPLICATIONS

V

(V) V

IN

OUT

(V) I

OUT

(A)

For more information www.analog.com

Rev. A

11

LT8301

OUT

APPLICATIONS INFORMATION

Typically, choose the transformer turns ratio to maximize

available output power. For low output voltages (3.3V or

5V), a larger N:1 turns ratio can be used with multiple pri-

mary windings relative to the secondary to maximize the

transformer’s current gain (and output power). However,

remember that the SW pin sees a voltage that is equal

to the maximum input supply voltage plus the output

voltage multiplied by the turns ratio. In addition, leakage

inductance will cause a voltage spike (V

LEAKAGE

) on top of
this reflected voltage. This total quantity needs to remain
below the 65V absolute maximum rating of the SW pin to
prevent breakdown of the internal power switch. Together
these conditions place an upper limit on the turns ratio,
NPS, for a given application. Choose a turns ratio low
enough to ensure:

NPS<

65V − V

IN(MAX)

V

− V

+ V

LEAKAGE

F

For lower output power levels, choose a smaller N:1 turns
ratio to alleviate the SW pin voltage stress. Although a
1:N turns ratio makes it possible to have very high output
voltages without exceeding the breakdown voltage of the
internal power switch, the multiplied parasitic capacitance
through turns ratio may cause the switch turn-on cur­rent spike ringing beyond 170ns leading-edge blanking,
thereby producing light load instability in certain applica­tions. So any 1:N turns ratio should be fully evaluated
before its use with the LT8301.

The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci­fies turns ratio accuracy within ±1%.

Saturation Current

The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core
is saturated will not be transferred to the secondary and
will instead be dissipated in the core. When designing
custom transformers to be used with the LT8301, the
saturation current should always be specified by the
transformer manufacturers.

Winding Resistance

Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output volt

­age regulation will be maintained independent of winding
resistance due to the boundary/discontinuous conduction
mode operation of the LT8301.

Leakage Inductance and Snubbers

Transformer leakage inductance on either the primary
or secondary causes a voltage spike to appear on the
primary after the power switch turns off. This spike is
increasingly prominent at higher load currents where
more stored energy must be dissipated. It is very impor­tant to minimize transformer leakage inductance.

When designing an application, adequate margin should
be kept for the worst-case leakage voltage spikes even
under overload conditions. In most cases shown in
Figure5, the reflected output voltage on the primary plus

should be kept below 50V. This leaves at least 15V

V

IN

margin for the leakage spike across line and load condi­tions. A larger voltage margin will be required for poorly
wound transformers or for excessive leakage inductance.

In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig­gering the boundary mode detector, the LT8301 internally
blanks the boundary mode detector for approximately
350ns. Any remaining voltage ringing after 350ns may
turn the power switch back on again before the second­ary current falls to zero. So the leakage inductance spike
ringing should be limited to less than

350ns.

Rev. A

12

For more information www.analog.com

APPLICATIONS INFORMATION

DZ Snubber RC Snubber

LT8301

<65V

<50V

<65V

<50V

V

SW

V

LEAKAGE

t

> 450ns

OFF

t

< 350ns

SP

TIME

<65V

<50V

V

SW

V

LEAKAGE

t

> 450ns

OFF

t

< 350ns

SP

TIME

8301 F05

V

SW

V

LEAKAGE

t

> 450ns

OFF

t

< 350ns

SP

TIME

No Snubber with DZ Snubber with RC Snubber

Figure5. Maximum Voltages for SW Pin Flyback Waveform

L

ℓ

Z

D

•

•

L

ℓ

C

R

•

•

Figure6. Snubber Circuits

A snubber circuit is recommended for most applications.
Two types of snubber circuits shown in Figure6 that can
protect the internal power switch include the DZ (diode­Zener) snubber and the RC (resistor-capacitor) snubber.
The DZ snubber ensures well defined and consistent
clamping voltage and has slightly higher power efficiency,
while the RC snubber quickly damps the voltage spike
ringing and provides better load regulation and EMI per­formance. Figure5 shows the flyback waveforms with the
DZ and RC snubbers.

For the DZ snubber, proper care must be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leakage
inductance spike. Choose a diode that has a reverse-volt­age rating higher than the maximum SW pin voltage. The

8301 F06b8300 F06a

diode breakdown voltage should be chosen to bal

Zener

-

ance power loss and switch voltage protection. The best
compromise

is to choose the largest voltage breakdown.

Use the following equation to make the proper choice:
V

ZENER(MAX)

≤ 65V – V

IN(MAX)

For an application with a maximum input voltage of 32V,
choose a 20V Zener diode, the V

ZENER(MAX)

of which is

around 21V and below the 33V maximum.

The power loss in the clamp will determine the power
rating of the Zener diode. Power loss in the clamp is
highest at maximum load and minimum input voltage.
The switch current is highest at this point along with the
energy stored in the leakage inductance. A 0.25W Zener
will satisfy most applications when the highest V

ZENER

ischosen.

For more information www.analog.com

Rev. A

13

LT8301

(OPTIONAL)

V

8301 F07

R2

APPLICATIONS INFORMATION

Tables 2 and 3 show some recommended diodes and
Zener diodes.

Table2. Recommended Zener Diodes

PART

CMDZ5248B 18 0.25 SOD-323 Central Semiconductor
CMDZ5250B 20 0.25 SOD-323

Table3. Recommended Diodes

PART

CMHD4448 0.25 100 SOD-123 Central Semiconductor
DFLS1100 1 100 PowerDI-123 Diodes Inc.
DFLS1150 1 150 PowerDI-123 Diodes Inc.

V

I

MAX

(A)

POWER

ZENER

(V)

(W) CASE VENDOR

V

REVERSE

(V) CASE VENDOR

The recommended approach for designing an RC snub­ber is to measure the period of the ringing on the SW
pin when the power switch turns off without the snub­ber and then add capacitance (starting with 100pF) until
the period of the ringing is 1.5 to 2 times longer. The
change in period will determine the value of the parasitic
capacitance, from which the parasitic inductance can be
determined from the initial period, as well. Once the value
of the SW node capacitance and inductance is known, a
series resistor can be added to the snubber capacitance
to dissipate power and critically dampen the ringing. The
equation for deriving the optimal series resistance using
the observed periods ( t
snubber capacitance (C

and t

PERIOD
SNUBBER

PERIOD(SNUBBED)

) is:

) and

Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
may need to be sized for thermal dissipation.

Undervoltage Lockout (UVLO)

A resistive divider from VIN to the EN/UVLO pin imple­ments undervoltage lockout (UVLO). The EN/UVLO pin
falling threshold is set at 1.228V with 14mV hysteresis.
In addition, the EN/UVLO pin sinks 2.5µA when the volt­age at the pin is below 1.228V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:

V

IN(UVLO+)

V

IN(UVLO−)

1.242V •(R1+ R2)

=

R2

1.228V •(R1+ R2)

=

+ 2.5µA •R1

Figure7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8301 in shutdown with quiescent current less than 2µA.

IN

R1

EN/UVLO

LT8301

R2

RUN/STOP
CONTROL

C

C

=

PAR

L

PAR

R

SNUBBER

=

t

⎛
⎜

⎝

t

PERIOD

C

PAR

14

SNUBBER

PERIOD(SNUBBED)

t

PERIOD

2

2

• 4π

L

PAR

=

C

PAR

2

⎞

− 1

⎟
⎠

For more information www.analog.com

Figure7. Undervoltage Lockout (UVLO)

GND

Rev. A

APPLICATIONS INFORMATION

5V +0.3V

LT8301

Minimum Load Requirement

The LT8301 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8301 has to turn on and off at
least for a minimum amount of time and with a minimum
frequency. The LT8301 delivers a minimum amount of
energy even during light load conditions to ensure accu­rate output voltage information. The minimum energy
delivery creates a minimum load requirement, which can
be approximately estimated as:

OUT

2

• f

MIN

I

LOAD(MIN)

L

PRI

I

SW(MIN)

L

•I

PRI

=

SW(MIN)

2 • V

= Transformer primary inductance

= Minimum switch current limit = 360mA

(max)
f

= Minimum switching frequency = 10.6kHz (max)

MIN

The LT8301 typically needs less than 0.5% of its full out-

put power as minimum load. Alternatively, a Zener diode
with its breakdown of 20% higher than the output volt­age can serve as a minimum load if pre-loading is not
acceptable. For a 5V output, use a 6V Zener with cathode
connected to the output.

worst-case scenario where the output is directly shorted
to ground through a long wire and the huge ring after fold­ing back still falsely triggers the boundary mode detec­tor, a secondary overcurrent protection ensures that the
LT8301

can still function properly. Once the switch cur­rent hits 2.2A overcurrent limit, a soft-start cycle initiates
and throttles back both switch current limit and switching
frequency very hard. This output short protection pre­vents the switch current from running away and limits
the average output diode current.

Design Example

Use the following design example as a guide to design
applications for the LT8301. The design example involves
designing a 5V output with a 500mA load current and an
input range from 8V to 32V.

V
V

IN(MIN)

= 5V, I

OUT

= 8V, V

OUT

IN(NOM)

= 500mA

= 12V, V

IN(MAX)

= 32V,

Step 1: Select the Transformer Turns Ratio.

V

NPS<

LEAKAGE

65V − V

IN(MAX)

V

= Margin for transformer leakage spike = 15V

OUT

+ V

− V

F

LEAKAGE

VF = Output diode forward voltage = ~0.3V

Example:

Output Short-Circuit Protection

When the output is heavily overloaded or shorted, the
reflected SW pin waveform rings longer than the internal
blanking time. If no protection scheme is applied, after
the 450ns minimum switch-off time, the excessive ring
might falsely trigger the boundary mode detector and turn

The choice of transformer turns ratio is critical in deter­mining output current capability of the converter. Table4
shows the switch voltage stress and output current capa­bility at different transformer turns ratio.

65V − 32V −15V

NPS<

= 3.4

the power switch back on again before the secondary cur­rent falls to zero. The part then runs into continuous con­duction mode at maximum switching frequency, and the
switch current may run away

. To prevent the switch cur­rent from running away under this condition, the LT8301
gradually folds back both maximum switch current limit
and switching frequency as the output voltage drops

Table4. Switch Voltage Stress and Output Current Capability
vs Turns Ratio

V

N

PS

1:1 37.3 330 14-40
2:1 42.6 470 25-57
3:1 47.9 540 33-67

V

SW(MAX)
IN(MAX)

at

(V)

I

OUT(MAX)

V

IN(MIN)

at

(mA) DUTY CYCLE (%)

from regulation. As a result, the switch current remains
below 1.375A (typ) maximum switch current limit. In the

For more information www.analog.com

Since only NPS = 3 can meet the 500mA output current
requirement, N

= 3 is chosen in this example.

PS

Rev. A

15

LT8301

290mA

PS

3

APPLICATIONS INFORMATION

Step 2: Determine the Primary Inductance.

Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:

t
t

I

L

≥

PRI

L

≥

PRI

OFF(MIN)

ON(MIN)

SW(MIN)

t

OFF(MIN)•NPS

I

SW(MIN)

t

ON(MIN)

• V

I

SW(MIN)

IN(MAX)

= 450ns

= 170ns

= 290mA (typ)

• V

+ V

( )

OUT

F

Example:

L

L

450ns • 3• (5V + 0.3V)

≥

PRI

PRI

≥

170ns • 32V

290mA

= 25µH

= 19µH

Most transformers specify primary inductance with a tol­erance of ±20%. With other component tolerance consid­ered, choose a transformer with its primary inductance
30% larger than the minimum values calculated above.

= 40µH is then chosen in this example.

L

PRI

Once the primary inductance has been determined, the
maximum load switching frequency can be calculated as:

fSW=

ISW=

1

t

+ t

ON

V

OUT•IOUT

η • V

OFF

IN

•D

=

• 2

•I

L

PRI

SW

V

IN

1

+

NPS•(V

L

•I

PRI

SW

+ VF)

OUT

Example:

D =

I

f

(5V + 0.3V)• 3

(5V + 0.3V)• 3 + 12V

5V •0.5A • 2

=

SW

0.85 •12V • 0.57

= 199kHz

SW

= 0.57

= 0.86A

The transformer also needs to be rated for the correct
saturation current level across line and load conditions.
A saturation current rating larger than 2A is necessary
to work with the LT8301. The 750313974 from Würth is
chosen as the flyback transformer.

Step 3: Choose the Output Diode.

Two main criteria for choosing the output diode include
forward current rating and reverse voltage rating. The
maximum load requirement is a good first-order guess
as the average current requirement for the output diode.
A conservative metric is the maximum switch current limit
multiplied by the turns ratio,

I

DIODE(MAX)

= I

SW(MAX)

• N

PS

Example:
I

DIODE(MAX)

= 4.125A

Next calculate reverse voltage requirement using maxi­mum VIN:

V

+

IN(MAX)

N

V

REVERSE

= V

OUT

Example:

V

REVERSE

= 5V +

32V

= 15.6V

The CMSH5-20 (5A, 20V diode) from Central
Semiconductor is chosen.

Rev. A

16

For more information www.analog.com

OUT

OUT

2 • 5V • 0.05V

APPLICATIONS INFORMATION

3 •(5V+0.3V)

LT8301

Step 4: Choose the Output Capacitor.

The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the equation below to
calculate the output capacitance:

•I

• ΔV

2

SW

L

=

PRI

2 • V

C

OUT

Example:
Design for output voltage ripple less than 1% of V

OUT

,

i.e., 50mV.

2

= 60µF

C

40µH • (0.86A)

=

OUT

Remember ceramic capacitors lose capacitance with
applied voltage. The capacitance can drop to 40% of
quoted capacitance at the maximum voltage rating. So a
100µF, 10V rating ceramic capacitor is chosen.

Step 5: Design Snubber Circuit.

A 20V Zener with a maximum of 21V will provide optimal
protection and minimize power loss. So a 20V, 0.25W
Zener from Central Semiconductor (CMDZ5250B) is
chosen.

Choose a diode that is fast and has sufficient reverse volt­age breakdown:

V
V

REVERSE

SW(MAX)

> V

= V

SW(MAX)

IN(MAX)

+ V

ZENER(MAX)

Example:
V

REVERSE

> 53V

A 100V, 0.25A diode from Central Semiconductor
(CMHD4448) is chosen.

Step 6: Select the R

Resistor.

FB

Use the following equation to calculate the starting value

FB

RFB=

:

N

PS

•(V

OUT

100µA

+ VF)

for R

The snubber circuit protects the power switch from leak

­age inductance voltage spike. A DZ snubber is recom­mended for this application because of lower leakage
inductance and larger voltage margin. The Zener and the
diode need to be selected.

The maximum Zener breakdown voltage is set according
to the maximum V

V

ZENER(MAX)

:

IN

≤ 65V – V

IN(MAX)

Example:
V

ZENER(MAX)

≤ 65V – 32V = 33V

Example:

RFB=

100µA

= 159k

Depending on the tolerance of standard resistor values,
the precise resistor value may not exist. For 1% standard
values, a 158k resistor should be close enough. As dis

­cussed in the Application Information section, the final
RFB value should be adjusted on the measured output
voltage.

For more information www.analog.com

Rev. A

17

LT8301

R2

2 • 5V

APPLICATIONS INFORMATION

Step 7: Select the EN/UVLO Resistors.

Determine the amount of hysteresis required and calcu­late R1 resistor value:

V

IN(HYS)

= 2.5µA • R1
Example:
Choose 2V of hysteresis,
R1 = 806k
Determine the UVLO thresholds and calculate R2 resistor

value:

V

IN(UVLO+)

1.242V •(R1+ R2)

=

+ 2.5µA •R1

Example:
Set V

UVLO rising threshold to 7.5V,

IN

R2 = 232k
V
V

IN(UVLO+)

IN(UVLO–)

= 7.5V
= 5.5V

Step 8: Ensure minimum load.

The theoretical minimum load can be approximately esti­mated as:

2

•10.6kHz
= 5.5mA

I

LOAD(MIN)

40µH • (360mA)

=

Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con­verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 6mA. In this example, a 820Ω resistor is selected
as the minimum load.

Rev. A

18

For more information www.analog.com

TYPICAL APPLICATIONS

8301 TA02b

LT8301

V

2.7V TO 36V

2.7V to 36VIN/15V

IN

10µF

EN/UVLO

Micropower Isolated Flyback Converter

OUT

D2

T1

1:1

V

IN

LT8301

GND

SW

R

Z1

D1

150k

FB

40µH

•

40µH

•

D1: CENTRAL CMHD4448
D2: CENTRAL CMMR1U-02
T1: SUMIDA 12387-T041
Z1: CENTRAL CMDZ5248B

V
15V
2mA TO 130mA (V
2mA TO 230mA (V

10µF

2mA TO 320mA (V
2mA TO 370mA (V

V

8301 TA02a

OUT

OUT

+

= 5V)

IN

= 12V)

IN

= 24V)

IN

= 36V)

IN

–

Efficiency vs Load Curent

95

90

85

80

EFFICIENCY (%)

75

VIN = 5V

= 12V

V

70

65

0

100 200 300 400

LOAD CURRENT (mA)

IN

= 24V

V

IN

= 36V

V

IN

V

8V TO 36V

8V to 36VIN/3.3V

IN

4.7µF

806k

EN/UVLO

232k

Micropower Isolated Flyback Converter

OUT

D2

T1
4:1

V

IN

LT8301

GND

SW

R

Z1

D1

137k

FB

40µH

•

2.5µH

•

D1: CENTRAL CMHD4448
D2: NXP PMEG2020EH
T1: SUMIDA 12387-T036
Z1: CENTRAL CMDZ5250B

For more information www.analog.com

V

OUT

3.3V

8.5mA TO 0.95A (V

8.5mA TO 1.30A (V

47µF

8.5mA TO 1.50A (V

V

OUT

8301 TA03

+

= 12V)

IN

= 24V)

IN

= 36V)

IN

–

Rev. A

19

LT8301

8301 TA04b

TYPICAL APPLICATIONS

V

8V TO 36V

8V to 36VIN/24V

IN

4.7µF

806k

232k

V

EN/UVLO

LT8301

GND

Micropower Isolated Flyback Converter

OUT

D2

T1

40µH

121k

1:2

•

160µH

•

D1: CENTRAL CMHD4448
D2: ST STPS1150A
T1: WÜRTH 750313975
Z1: CENTRAL CMDZ5248B

Z1

IN

SW

R

FB

D1

V

OUT

24V

1.2mA TO 130mA (V

1.2mA TO 180mA (V

1.2mA TO 200mA (V

4.7µF

V

OUT

8301 TA04a

+

= 12V)

IN

= 24V)

IN

= 36V)

IN

–

Efficiency vs Load Curent

95

90

85

80

EFFICIENCY (%)

75

70

65

0

50 100 150 200

LOAD CURRENT (mA)

VIN = 12V

= 24V

V

IN

= 36V

V

IN

20

V

8V TO 36V

8V to 36VIN/48V

IN

4.7µF

806k

EN/UVLO

232k

Micropower Isolated Flyback Converter

OUT

D2

T1

1:4

V

IN

LT8301

GND

SW

R

Z1

D1

118k

FB

40µH

•

640µH

•

D1: CENTRAL CMHD4448
D2: DIODES BAV21W-7-F
T1: WÜRTH 750313976
Z1: CENTRAL CMDZ5252B

V

OUT

48V

0.6mA TO 70mA (V

0.6mA TO 90mA (V

1µF

0.6mA TO 100mA (V

V

OUT

8301 TA03

+

= 12V)

IN

= 24V)

IN

= 36V)

IN

–

Rev. A

For more information www.analog.com

TYPICAL APPLICATIONS

LT8301

2.7V TO 42V

V

12V TO 24V

V

IN

IN

VIN to (V

10µF

12V to 24V

4.7µF

+ 10V)/(VIN – 10V) Micropower Converter

IN

T1

1:1

V

IN

LT8301

GND

/Four 15V

IN

806k

EN/UVLO

232k

D1: CENTRAL CMHD4448
D2-D5: CENTRAL CMMR1U-02
T1: SUMIDA EPH2815-ADBN-A0349
Z1: CENTRAL CMDZ5248B

Micropower Isolated Flyback Converter

OUT

V

IN

LT8301

GND

SWEN/UVLO

R

FB

SW

R

FB

•

•

102k

Z1

30µH 30µH

D1

150k

D1

4.7µF

40µH40µH

D1, D2: DIODES INC. DFLS160
T1: SUMIDA 12387-T041
Z1: CENTRAL CMDZ12L

1:1:1:1:1

4.7µF

D2

T1

D2

•

2.2µF

•

D3

•

30µH

2.2µF

D4

•

30µH

2.2µF

D5

•

30µH

2.2µF

Z1

Z2

8301 TA06

V

IN

150mA

V

IN

150mA

V

IN

7.5k

7.5k

7.5k

7.5k

8301 TA07

+ 10V

– 10V

V
15V
60mA

V
V

15V
60mA

V
V

15V
60mA

V
V

15V
60mA

V

OUT1

OUT1
OUT2

OUT2
OUT3

OUT3
OUT4

OUT4

+

–
+

–
+

–
+

–

For more information www.analog.com

Rev. A

21

LT8301

5 PLCS (NOTE 3)

0.20 BSC

6. JEDEC PACKAGE REFERENCE IS MO-193

0.62

0.95

PACKAGE DESCRIPTION

S5 Package

5-Lead Plastic TSOT-23

(Reference LTC DWG # 05-08-1635 Rev B)

MAX

3.85 MAX

DATUM ‘A’

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DRAWING NOT TO SCALE

3. DIMENSIONS ARE INCLUSIVE OF PLATING

4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

5. MOLD FLASH SHALL NOT EXCEED 0.254mm

2.62 REF

RECOMMENDED SOLDER PAD LAYOUT

PER IPC CALCULATOR

0.30 – 0.50 REF

REF

1.22 REF

1.4 MIN

0.09 – 0.20
(NOTE 3)

2.80 BSC

1.50 – 1.75
(NOTE 4)

0.80 – 0.90

1.00 MAX

PIN ONE

0.95 BSC

2.90 BSC
(NOTE 4)

0.30 – 0.45 TYP

0.01 – 0.10

1.90 BSC

S5 TSOT-23 0302 REV B

22

Rev. A

For more information www.analog.com

LT8301

REVISION HISTORY

REV DATE DESCRIPTION PAGE NUMBER

A 04/19 Add AEC-Q100 Qualified Front Page Feature Bullet.

Add Automotive (W) Flow Parts to Order Information Section.

1
2

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

For more information www.analog.com

Rev. A

23

LT8301

8301 TA08c

TYPICAL APPLICATION

8V TO 36V

Efficiency vs Load Current Output Load and Line Regulation

95

90

85

80

EFFICIENCY (%)

75

70

65

0

8V to 36VIN/12V

V

IN

4.7µF

806k

EN/UVLO

232k

VIN = 12V

= 24V

V

IN

= 36V

V

IN

100 200 300 400

LOAD CURRENT (mA)

8301 TA08b

V

IN

LT8301

GND

Micropower Isolated Flyback Converter

OUT

D2

T1

40µH

118k

1:1

•

40µH

•

D1: CENTRAL CMHD4448
D2: DIODE INC. DFLS160
T1: WÜRTH 750313972
Z1: CENTRAL CMDZ5250B

12.4

12.3

12.2

12.1

12.0

11.9

OUTPUT VOLTAGE (V)

11.8

11.7

11.6
0

10µF

8301 TA08a

SW

R

Z1

D1

FB

+

V

OUT

12V

2.5mA TO 270mA (V

2.5mA TO 360mA (V

2.5mA TO 400mA (V

–

V

OUT

100

200

LOAD CURRENT (mA)

= 12V)

IN

= 24V)

IN

= 36V)

IN

VIN = 12V
V
V

300

= 24V

IN

= 36V

IN

400

RELATED PARTS

PART NUMBER DESCRIPTION COMMENTS

LT8300 100V

Micropower Isolated Flyback Converter with

IN

150V/260mA Switch

LT8302 42V

Micropower Isolated Flyback Converter with

IN

65V/3.6A Switch

LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V ≤ V
LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch,

LT3748 100V Isolated Flyback Controller 5V ≤ V
LT3798 Off-Line Isolated No Opto-Coupler Flyback Controller

with Active PFC

LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch
LT3757A/LT3759/

40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive

LT3758
LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch

®

3803/LTC3803-3/

LTC

200kHz/300kHz Flyback Controllers in SOT-23 VIN and V

LTC3803-5
LTC3805/LT C3805-5 Adjustable Frequency Flyback Controllers V

24

For more information www.analog.com

Low IQ Monolithic No-Opto Flybacks, 5-Lead TSOT-23

Low IQ Monolithic No-Opto Flybacks, 8-Lead SO-8E

≤ 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23

CC

MSOP-16(12)

≤ 100V, No Opto Flyback , MSOP-16 with High Voltage Spacing

IN

V

and V

IN

and V

IN

Limited Only by External Components

OUT

Limited by External Components

OUT

Limited by External Components

OUT

 ANALOG DEVICES, INC. 2014-2019

Rev. A

04/19

www.analog.com