intersil ISL6323A DATA SHEET

ISL6323A

Data Sheet

Monolithic Dual PWM Hybrid Controller Powering AMD SVI Split-Plane and PVI Uniplane Processors

The ISL6323A dual PWM controller delivers high efficiency and tight regulation from two synchronous buck DC/DC converters. The ISL6323A supports hybrid power control of AMD processors which operate from either a 6-bit parallel VID interface (PVI) or a serial VID interface (SVI). The dual output ISL6323A features a multi-phase controller to support uniplane VDD core voltage and a single phase controller to power the Northbridge (VDDNB) in SVI mode. Only the multi-phase controller is active in PVI mode to support uniplane VDD only processors.

A precision uniplane core voltage regulation system is provided by a two-to-four-phase PWM voltage regulator (VR) controller. The integration of two power MOSFET drivers, adding flexibility in layout, reduce the number of external components in the multi-phase section. A single phase PWM controller with integrated driver provides a second precision voltage regulation system for the North Bridge portion of the processor. This monolithic, dual controller with integrated driver solution provides a cost and space saving power management solution.

For applications which benefit from load line programming to reduce bulk output capacitors, the ISL6323A features output voltage droop . The multi-phase portion also includes advanced control loop features for optimal transient response to load apply and removal. One of these features is highly accurate, fully differential, continuous DCR current sensing for load line programming and channel current balance. Dual edge modulation is another unique feature, allowing for quicker initial response to high di/dt load transients.

The ISL6323A supports Power Savings Mode by dropping the number of phases to one when the PSI_L bit is set.

Ordering Information

PART NUMBER

(Note)

ISL6323ACRZ* ISL6323A CRZ 0 to +70 48 Ld 7x7 QFN L48.7x7 ISL6323AIRZ* ISL6323A IRZ -40 to +85 48 Ld 7x7 QFN L48.7x7 *Add “-T” suffix for tape and reel. Please refer to TB347 for details on reel

specifications. NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.

PART

MARKING

TEMP.

(°C)

PACKAGE

(Pb-free)

PKG.

DWG . #

March 23, 2009

FN6878.0

Features

• Processor Core Voltage Via Integrated Multi-Phase Power Conversion

• Configuration Flexibility

- 2-Phase Operation with Internal Drivers

- 3- or 4-Phase Operation with External PWM Drivers

• PSI_L Support with Phase Shedding for Improved Efficiency at Light Load

• Serial VID Interface Inputs

- Two Wire, Clock and Data, Bus

- Conforms to AMD SVI Specifications

• Parallel VID Interface Inputs

- 6-bit VID input

- 0.775V to 1.55V in 25mV Steps

- 0.375V to 0.7625V in 12.5mV Steps

• Precision Core Voltage Regulation

- Differential Remote Voltage Sensing

- ±0.6% System Accuracy Over-Temperature

- Adjustable Reference-Voltage Offset

• Optimal Processor Core Voltage Transient Response

- Adaptive Phase Alignment (APA)

- Active Pulse Positioning Modulation

• Fully Differential, Continuous DCR Current Sensing

- Accurate Load Line Programming

- Precision Channel Current Balancing

• Variable Gate Drive Bias: 5V to 12V

• Overcurrent Protection

• Multi-tiered Overvoltage Protection

• Selectable Switching Frequency up to 1MHz

• Simultaneous Digital Soft-Start of Both Outputs

• Processor NorthBridge Voltage Via Single Phase Power Conversion

• Precision Voltage Regulation

- Differential Remote Voltage Sensing

- ±0.6% System Accuracy Over Temperature

• Serial VID Interface Inputs

- Two Wire, Clock and Data, Bus

- Conforms to AMD SVI Specifications

• Overcurrent Protection

• Continuous DCR Current Sensing

• Variable Gate Drive Bias: 5V to 12V

• Simultaneous Digital Soft-Start of Both Outputs

• Selectable Switching Frequency up to 1MHz

• Pb-Free Plus Anneal Available (RoHS Compliant)

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

1-888-INTERSIL or 1-888-468-3774

| Intersil (and design) is a registered trademark of Intersil Americas Inc.

All other trademarks mentioned are the property of their respective owners.

ISL6323AISL6323A

Pinout

FB_NB

ISEN_NB+

RGND_NB

VID0/VFIXEN

VID1/SEL

VID2/SVD

VID3/SVC

VID4

VID5

VCC

RGND

ISL6323A

(48 LD QFN)

TOP VIEW

ISEN_NB-

ISEN4+

COMP_NB

48 47 46 45 44 43 42 41 40 39 38 37

13 14 15 16 17 18 19 20 21 22 23 24

OFS

VSEN

ISEN4-

DVC

RSET

ISEN3+

ISEN3-

GND

COMP

PVCC_NB

LGATE_NB

BOOT_NB

APA

ISEN1+

ISEN1-

UGATE_NB

ISEN2+

PHASE_NB

ISEN2-

VDDPWRGD

PWM4

PWM3

PWROK

PHASE1

UGATE1

BOOT1

LGATE1

PVCC1_2

LGATE2

BOOT2

UGATE2

PHASE2

Integrated Driver Block Diagram

PWM

SOFT-START

AND

FAULT LOGIC

GATE

CONTROL

LOGIC

SHOOT-

THROUGH

PROTECTION

PVCC

BOOT

UGATE

20KΩ

PHASE

10KΩ

LGATE

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Controller Block Diagram

ISL6323AISL6323A

ISEN_NB+

ISEN_NB-

VDDPWRGD

APA

COMP

OFS

DVC

RGND

PWROK

VID0/VFIXEN

VID1/SEL VID2/SVD VID3/SVC

VID4 VID5

VSEN

RSET

ISEN1+

ISEN1-

ISEN2+

ISEN2-

ISEN3+

ISEN3-

ISEN4+

ISEN4-

OFFSET

∑

SVI

SLAVE

BUS AND

PVI

DAC

NB_REF

LOGIC

RESISTOR MATCHING

CH1

CURRENT

SENSE

CH2

CURRENT

SENSE

CH3

CURRENT

SENSE

CH4

CURRENT

SENSE

CURRENT

SENSE

APA

ISEN3-

ISEN4-

NB_REF

E/A

RGND_NB

LOGIC

I_TRIP

∑

I_AVG

LOGIC

TRIANGLE WAVE

∑

CHANNEL CURRENT BALANCE

NB FAULT LOGIC

SOFT-START

AND

FAULT LOGIC

LOAD APPL Y

TRANSIENT

ENHANCEMENT

CLOCK AND GENERATOR

∑

I_AVG

∑

FB_NB

1 N

E/A

DROOP

CONTROL

PWM1

PWM2

COMP_NB

RAMP

EN_12V

ENABLE

LOGIC

PWM3

PWM4

POWER-ON

RESET

CHANNEL

DETECT

PWM3

SIGNAL

LOGIC

SIGNAL

LOGIC

MOSFET

PWM4

MOSFET

DRIVER

MOSFET

DRIVER

PH3/PH4

POR

EN_12V

ISEN3ISEN4-

BOOT_NB

UGATE_NB

PHASE_NB

LGATE_NB

PVCC_NB

VCC

PVCC1_2

BOOT1

UGATE1

PHASE1

LGATE1

BOOT2

UGATE2

PHASE2

LGATE2

PWM3

PWM4

GND

FN6878.0

Typical Application - SVI Mode

ISL6323AISL6323A

+5V

NC NC

FB COMP ISEN3+ ISEN3PWM3

APA

DVC

VCC

OFS FS

RSET

VFIXEN SEL SVD SVC VID4 VID5 PWROK VDDPWRGD

GND

VSEN

BOOT1

UGATE1

PHASE1

LGATE1

ISEN1-

ISEN1+

PVCC1_2

BOOT2

UGATE2

PHASE2

LGATE2

ISEN2-

ISEN2+

RGND

ISEN4+

ISEN4-

+12V

VDD

CPU

LOAD

+12V

BOOT1

UGATE1

PHASE1

LGATE1

PGND

ISL6614

BOOT2

UGATE2

PHASE2

LGATE2

PWM1

VCC PVCC

GND

PWM2

+12V

OFF

+12V

ISL6323A

COMP_NB

FB_NB

PWM4

PVCC_NB

BOOT_NB

UGATE_NB

PHASE_NB

LGATE_NB

ISEN_NB-

ISEN_NB+

+12V

LOAD

VDDNB

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Typical Application - PVI Mode

ISL6323AISL6323A

+5V

FB COMP ISEN3+ ISEN3PWM3

APA

DVC

VCC

OFS FS

RSET

VID0 VID1/SEL VID2 VID3

VID4 VID5 PWROK VDDPWRGD

GND

VSEN

BOOT1

UGATE1

PHASE1

LGATE1

ISEN1-

ISEN1+

PVCC1_2

BOOT2

UGATE2

PHASE2

LGATE2

ISEN2-

ISEN2+

RGND

ISEN4+

ISEN4-

+12V

VDD

CPU

LOAD

+12V

BOOT1

UGATE1

PHASE1

LGATE1 PGND

ISL6614

BOOT2

UGATE2

PHASE2

LGATE2

PWM1

VCC PVCC

GND

PWM2

+12V

OFF ON

+12V

ISL6323A

COMP_NB

FB_NB

PWM4

PVCC_NB

BOOT_NB

UGATE_NB

PHASE_NB

LGATE_NB

ISEN_NB-

ISEN_NB+

+12V

VDDNB

LOAD

NORTH BRIDGE REGULATOR DISABLED IN PVI MODE

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ISL6323AISL6323A

Absolute Maximum Ratings Thermal Information

Supply Voltage (VCC) . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6.2V

Supply Voltage (PVCC) . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to +15V

Absolute Boot Voltage (V Phase Voltage (V

GND - 8V (<400ns, 20µJ) to 24V (<200ns, V

PHASE

Upper Gate Voltage (V

- 3.5V (<100ns Pulse Width, 2µJ) to V

Lower Gate Voltage (V

PHASE

GND - 5V (<100ns Pulse Width, 2µJ) to PVCC+ 0.3V

). . . . . . . .GND - 0.3V to GND + 36V

BOOT

). . . . . . . . GND - 0.3V to 15V (PVCC = 12)

BOOT BOOT

= 12V) + 0.3V + 0.3V

). . . .V

UGATE

LGATE

PHASE

). . . . . . . GND - 0.3V to PVCC + 0.3V

BOOT-PHASE

- 0.3V to V

Input, Output, or I/O Voltage . . . . . . . . . GND - 0.3V to VCC + 0.3V

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.

NOTES:

is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See

1. θ

Tech Brief TB379.

2. For θ

, the “case temp” location is the center of the exposed metal pad on the package underside.

3. Limits established by characterization and are not production tested.

Electrical Specifications Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are

100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested.

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

BIAS SUPPLIES

Input Bias Supply Current I Gate Drive Bias Current - PVCC1_2 Pin I Gate Drive Bias Current - PVCC_NB Pin I VCC POR (Power-On Reset) Threshold VCC Rising 4.20 4.35 4.50 V

PVCC POR (Power-On Reset) Threshold PVCC Rising 4.20 4.35 4.50 V

PWM MODULATOR

Oscillator Frequency Accuracy, F

Typical Adjustment Range of Switching Frequency (Note 3) 0.08 1.0 MHz Oscillator Ramp Amplitude, V

P-P

Maximum Duty Cycle (Note 3) 99.5 %

CONTROL THRESHOLDS

EN Rising Threshold 0.80 0.88 0.92 V EN Hysteresis 70 130 190 mV PWROK Input HIGH Threshold 1.1 V PWROK Input LOW Threshold 0.95 V

; EN = high 15 22 25 mA

VCC PVCC1_2 PVCC_NB

; EN = high 1 1.8 3 mA

; EN = high 0.3 0.9 2 mA

VCC Falling 3.70 3.85 4.05 V

PVCC Falling 3.70 3.85 4.05 V

RT = 100kΩ (±0.1%) to Ground, (Droop Enabled)

= 100kΩ (±0.1%) to VCC,

(Droop Disabled), 0°C to +70°C R

= 100kΩ (±0.1%) to VCC,

(Droop Disabled), -40°C to +85°C

(Note 3) 1.50 V

Thermal Resistance θ

(°C/W) θJC (°C/W)

QFN Package (Notes 1, 2). . . . . . . . . . 27 2

Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . .+150°C

Maximum Storage Temperature Range. . . . . . . . . .-65°C to +150°C

Pb-free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . .see link below

http://www.intersil.com/pbfree/Pb-FreeReflow.asp

Recommended Operating Conditions

VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+5V ±5%

PVCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . .+5V to 12V ±5%

Ambient Temperature

ISL6323ACRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C

ISL6323AIRZ . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +85°C

225 250 265 kHz

245 275 310 kHz

240 275 310 kHz

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ISL6323AISL6323A

Electrical Specifications Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are

100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued)

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

VDDPWRGD Sink Current Open drain, V_VDDPWRGD = 400mV 4 mA PWM Channel Disable Threshold V

PIN_ADJUSTABLE OFFSET

OFS Source Current Accuracy (Positive Offset) R OFS Sink Current Accuracy (Negative Offset) R

REFERENCE AND DAC

System Accuracy (VDAC > 1.000V) -0.6 0.6 % System Accuracy (0.600V < VDAC < 1.000V) -1.0 1.0 % System Accuracy (VDAC < 0.600V) -2.0 2.0 % DVC Voltage Gain VDAC = 1V 2.0 V APA Current Tolerance V

ERROR AMPLIFIER

DC Gain R Gain-Bandwidth Product (Note 3) C Slew Rate (Note 3) C Maximum Output Voltage Load = 1mA 3.80 4.20 V Minimum Output Voltage Load = -1mA 1.3 1.6 V

SOFT-START RAMP

Soft-Start Ramp Rate 2.2 3.0 4.0 mV/µs

PWM OUTPUTS

PWM Output Voltage LOW Threshold I PWM Output Voltage HIGH Threshold I

CURRENT SENSING - CORE CONTROLLER

Sensed Current Tolerance V

CURRENT SENSING - NB CONTROLLER

Sensed Current Tolerance V

DROOP CURRENT

Tolerance V

OVERCURRENT PROTECTION

Overcurrent Trip Level - Average Channel Normal Operation, R

Overcurrent Limiting - Individual Channel Normal Operation, R

POWER GOOD

Core Overvoltage Threshold VSEN Rising VDAC

, V

ISEN3-

OFS OFS

APA

L L L

LOAD LOAD

ISENn-

4 Phases, T

ISEN_NB-

SET

ISENn-

4 Phases, T

ISEN4-, VISEN2-

= 10kΩ (± 0.1%) from OFS to GND 27.5 31 34.5 µA = 30kΩ (± 0.1%) from OFS to VCC 50.5 53.5 56.5 µA

= 1V 90 100 108 µA

= 10k to ground, (Note 3) 96 dB = 100pF, RL = 10k to ground, (Note 3) 20 MHz = 100pF, Load = ±400µA, (Note 3) 8 V/µs

= ±500µA 0.5 V = ±500µA 4.5 V

- V

ISENn+

= +25°C

- V

ISEN_NB+

= 23.2mV, R

= 23.2mV,

SET

= 37.6kΩ,

= 37.6kΩ, 4 Phases, TA = +25°C

- V

ISENn+

= +25°C

= 23.2mV, R

= 28.2kΩ 87 100 120 µA

SET

= 37.6kΩ,

4.4 V

68 88 µA

68 89 µA

68 88 µA

Dynamic VID Change (Note 3) 130 µA

= 28.2kΩ 142 µA

SET

Dynamic VID Change (Note 3) 190 µA

+225mV

VDAC +

250mV

VDAC

+275mV

FN6878.0

ISL6323AISL6323A

Electrical Specifications Recommended Operating Conditions, Unless Otherwise Specified. Parameters with MIN and/or MAX limits are

100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization and are not production tested. (Continued)

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Core Undervoltage Threshold VSEN Falling VDAC

-325mV

NB Undervoltage Threshold ISEN_NB+ Falling VDAC

-310mV

Power Good Hysteresis 50 mV

OVERVOLTAGE PROTECTION

OVP Trip Level 1.73 1.80 1.84 V OVP Lower Gate Release Threshold 350 400 mV

SWITCHING TIME (Note 3) [See “Timing Diagram” on page 9]

UGATE Rise Time t

LGATE Rise Time t

UGATE Fall Time t

RUGATE; VPVCC

90%

RLGATE; VPVCC

90%

FUGATE; VPVCC

= 12V, 3nF Load, 10% to

= 12V, 3nF Load, 90% to

10% LGATE Fall Time t UGATE Turn-On Non-overlap t

LGATE Turn-On Non-overlap t

FLGATE; VPVCC PDHUGATE

Adaptive

PDHLGATE

= 12V , 3nF Load, 90% to 10% 12 ns

; V

= 12V, 3nF Load,

PVCC

; V

= 12V , 3nF Load, Adaptive 10 ns

PVCC

GATE DRIVE RESISTANCE (Note 3)

Upper Drive Source Resistance V Upper Drive Sink Resistance V Lower Drive Source Resistance V Lower Drive Sink Resistance V

= 12V, 15mA Source Current 2.0 Ω

PVCC

= 12V, 15mA Sink Current 1.65 Ω

PVCC

= 12V, 15mA Source Current 1.25 Ω

PVCC

= 12V, 15mA Sink Current 0.80 Ω

PVCC

MODE SELECTION

VID1/SEL Input Low EN taken from LO to HI, VDDIO = 1.5V 0.6 V VID1/SEL Input High EN taken from LO to HI, VDDIO = 1.5V 1.00 V

PVI INTERFACE

VIDx Pull-down VDDIO = 1.5V 30 40 µA VIDx Input Low VDDIO = 1.5V 0.6 V VIDx Input High VDDIO = 1.5V 1.00 V

SVI INTERFACE

SVC, SVD Input HIGH (VIH) 0.95 V SVC, SVD Input LOW (VIL) 0.4 V Schmitt Trigger Input Hysteresis 0.14 0.35 0.45 V SVD Low Level Output Voltage 3mA Sink Current 0.285 V Maximum SVC, SVD Leakage (Note 3) ±5 µA

VDAC

-300mV VDAC

-275mV

VDAC

-270mV VDAC

-235mV

26 ns

18 ns

10 ns

FN6878.0

Timing Diagram

UGATE

LGATE

PDHUGATE

RUGATE

ISL6323AISL6323A

FUGATE

FLGATE

Functional Pin Description

VID1/SEL

This pin selects SVI or PVI mode operation based on the state of the pin prior to enabling the ISL6323A. If the pin is LO prior to enable, the ISL6323A is in SVI mode and the dual purpose pins [VID0/VFIXEN, VID2/SVC, VID3/SVD] use their SVI mode related functions. If the pin held HI prior to enable, the ISL6323A is in PVI mode and dual purpose pins use their VIDx related functions to decode the correct DAC code.

VID0/VFIXEN

If VID1 is LO prior to enable [SVI Mode], the pin is functions as the VFIXEN selection input from the AMD processor for determining SVI mode versus VFIX mode of operation. If VID1 is HI prior to enable [PVI Mode], the pin is used as DAC input VID0. This pin has an internal 30µA pull-down current applied to it at all times.

VID2/SVD

If VID1 is LO prior to enable [SVI Mode], this pin is the serial VID data bi-directional signal to and from the master device on AMD processor. If VID1 is HI prior to enable [PVI Mode], this pin is used to decode the programmed DAC code for the processor. In PVI mode, this pin has an internal 30µA pull-down current applied to it. There is no pulldown current in SVI mode.

VID3/SVC

If VID1 is LO prior to enable [SVI Mode], this pin is the serial VID clock input from the AMD processor. If VID1 is HI prior to enable [PVI Mode], the ISL6323A is in PVI mode and this pin is used to decode the programmed DAC code for the processor. In PVI mode, this pin has an internal 30µA pull-down current applied to it. There is no pulldown current in SVI mode.

VID4

This pin is active only when the ISL6323A is in PVI mode. When VID1 is HI prior to enable, the ISL6323A decodes the programmed DAC voltage required by the AMD processor. This pin has an internal 30µA pull-down current applied to it at all times.

RLGATE

PDHLGATE

VID5

VCC

VCC is the bias supply for the ICs small-signal circuitry. Connect this pin to a +5V supply and decouple using a quality 0.1µF ceramic capacitor.

PVCC1_2

The power supply pin for the multi-phase internal MOSFET drivers. Connect this pin to any voltage from +5V to +12V depending on the desired MOSFET gate-drive level. Decouple this pin with a quality 1.0µF ceramic capacitor.

PVCC_NB

The power supply pin for the internal MOSFET driver for the Northbridge controller. Connect this pin to any voltage from +5V to +12V depending on the desired MOSFET gate-drive level. Decouple this pin with a quality 1.0µF ceramic capacitor.

GND

GND is the bias and reference ground for the IC. The GND connection for the ISL6323A is through the thermal pad on the bottom of the package.

This pin is a threshold-sensitive (approximately 0.85V) system enable input for the controller. Held low , this pin disab les both CORE and NB controller operation. Pulled high, the pin enables both controllers for operation.

When the EN pin is pulled high, the ISL6323A will be placed in either SVI or PVI mode. The mode is determined by the latched value of VID1 on the rising edge of the EN signal.

A third function of this pin is to provide driver bias monitor for external drivers. A resistor divider with the center tap connected to this pin from the drive bias supply prevents enabling the controller before insufficient bias is provided to external driver. The resistors should be selected such th at

FN6878.0

ISL6323AISL6323A

when the POR-trip point of the external driver is reached, the voltage at this pin meets the above mentioned threshold level.

A resistor, placed from FS to Ground or from FS to VCC, sets the switching frequency of both controllers. Refer to Equation 1 for proper resistor calculation.

RT10

With the resistor tied from FS to Ground, Droop is enabled. With the resistor tied from FS to VCC, Droop is disabled.

10.61 1.035 fs()log–[]

(EQ. 1)

VSEN and RGND

VSEN and RGND are inputs to the core voltage re gulator (VR) controller precision differential remote-sense amplifier and should be connected to the sense pins of the remote processor core(s), VDDFB[H,L].

FB and COMP

These pins are the internal error amplifier inverting input and output respectively of the core VR controller. FB, VSEN and COMP are tied together through external R-C networks to compensate the regulator.

APA

Adaptive Phase Alignment (APA) pin for setting trip level and adjusting time constant. A 100µA current flows into the APA pin and by tying a resistor from this pin to COMP the trip level for the Adaptive Phase Alignment circuitry can be set.

OFS

The OFS pin provides a means to program a dc current for generating an offset voltage across the resistor between FB and VSEN The offset current is generated via an external resistor and precision internal voltage references. The polarity of the offset is selected by connecting the resistor to GND or VCC. For no offset, the OFS pin should be left unconnected.

ISEN1-, ISEN1+, ISEN2-, ISEN2+, ISEN3-, ISEN3+, ISEN4- and ISEN4+

These pins are used for differentially sensing the corresponding channel output currents. The sensed currents are used for channel balancing, protection, and core load line regulation.

Connect ISEN1-, ISEN2-, ISEN3-, and ISEN4- to the node between the RC sense elements surrounding the inductor of their respective channel. Tie the ISEN+ pins to the VCORE side of their corresponding channel’s sense capacitor.

UGATE1 and UGATE2

Connect these pins to the corresponding upper MOSFET gates. These pins are used to control the upper MOSFETs and are monitored for shoot-through prevention purposes. Maximum individual channel duty cycle is limited to 93.3%.

BOOT1 and BOOT2

These pins provide the bias voltage for the corresponding upper MOSFET drives. Connect these pins to appropriately-chosen external bootstrap capacitors. Internal bootstrap diodes connected to the PVCC1_2 pin provide the necessary bootstrap charge.

PHASE1 and PHASE2

Connect these pins to the sources of the corresponding upper MOSFETs. These pins are the return path for the upper MOSFET drives.

LGA TE1 and LGATE2

These pins are used to control the lower MOSFET s. Connect these pins to the corresponding lower MOSFETs’ gates.

PWM3 and PWM4

Pulse-width modulation outputs. Connect these pins to the PWM input pins of an Intersil driver IC if 3- or 4-phase operation is desired. Connect the ISEN- pins of the channels not desired to +5V to disable them and configure the core VR controller for 2- or 3-phase operation.

PWROK

System wide Power Good signal. If this pin is low, the two SVI bits are decoded to determine the “metal VID”. When pin is high, the SVI is actively running its protocol.

RSET

Connect this pin to VCC through a resistor to set the effective value of the internal RISEN current sense resistors. An external PTC thermistor network can also be used to thermally compensate the current sense resistors to account for changes in inductor DCR over-temperature.

VDDPWRGD

During normal operation this pin indicates whether both output voltages are within specified overvoltage a nd undervoltage limits. If either output voltage exceeds these limits or a reset event occurs (such as an overcurrent event), the pin is pulled low. This pin is always low pri or to the end of sof t-st art.

RGND_NB

This pin is an input to the NB VR controller precision differential remote-sense amplifier and should be connected to the sense pin of the North Bridge, VDDNBFBL.

DVC

The DVC pin is a buffered version of the reference to the error amplifier. A series resistor and capaci tor between the DVC pin and FB pin smooth the voltage transition d uring VID-on-the-fly operations.

FB_NB and COMP_NB

These pins are the internal error amplifier inverting input and output respectively of the NB VR controller. FB_NB, VDIFF_NB, and COMP_NB are tied together through external R-C networks to compensate the re gu l at o r.

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ISL6323AISL6323A

ISEN_NB-, ISEN_NB+

These pins are used for differentially sensing the North Bridge output current. The sensed current is used for protection and load line regulation if droop is enabled.

Connect ISEN_NB- to the node between the RC sense element surrounding the inductor. Tie the ISEN_NB+ pin to the VNB side of the sense capacitor.

UGATE_NB

Connect this pin to the corresponding upper MOSFET gate. This pin provides the PWM-controlled gate drive for the upper MOSFET and is monitored for shoot-through prevention purposes.

BOOT_NB

This pin provides the bias voltage for the corresponding upper MOSFET drive. Connect this pin to appropriately-chosen external bootstrap capacitor. The internal bootstrap diode connected to the PVCC_NB pin provides the necessary bootstrap charge.

PHASE_NB

Connect this pin to the source of the corresponding upper MOSFET. This pin is the return path for the upper MOSFET drive. This pin is used to monitor the voltage drop across the upper MOSFET for overcurrent protection.

channels. In a 3-phase converter, each channel switches 1/3 cycle after the previous channel and 1/3 cycle before the following channel. As a result, the three-phase converter has a combined ripple frequency three times greater than the ripple frequency of any one phase. In addition, the peak-topeak amplitude of the combined inductor currents is reduced in proportion to the number of phases (Equations 2 and 3). Increased ripple frequency and lower ripple amplitude mean that the designer can use less per-channel inductance and lower total output capacitance for any performance specification.

Figure 1 illustrates the multiplicative effect on output ripple frequency. The three channel currents (IL1, IL2, and IL3) combine to form the AC ripple current and the DC load current. The ripple component has three times the ripple frequency of each individual channel current. Each PWM pulse is terminated 1/3 of a cycle after the PWM pulse of the previous phase. The peak-to-peak current for each phase is about 7A, and the DC components of the inductor currents combine to feed the load.

IL1 + IL2 + IL3, 7A/DIV

IL3, 7A/DIV

LGATE_NB

Connect this pin to the corresponding MOSFET’s gate. This pin provides the PWM-controlled gate drive for the lower MOSFET. This pin is also monitored by the adaptive shootthrough protection circuitry to determine when the lower MOSFET has turned off.

Operation

The ISL6323A utilizes a multi-phase architecture to provide a low cost, space saving power conversion solution for the processor core voltage. The controller also implements a simple single phase architecture to provide the Northbridge voltage on the same chip.

Multi-phase Power Conversion

Microprocessor load current profiles have changed to the point that the advantages of multi-phase power conversion are impossible to ignore. The technical challenges associated with producing a single-phase converter that is both cost-effective and thermally viable have forced a change to the cost-saving approach of multi-phase. The ISL6323A controller helps simplify implementation by integrating vital functions and requiring minimal external components. The “Controller Block Diagram” on page 3 provides a top level view of the multi-phase power conversion using the ISL6323A controller.

Interleaving

The switching of each channel in a multi-phase converter is timed to be symmetrically out of phase with each of the other

PWM3, 5V/DIV

IL2, 7A/DIV

PWM2, 5V/DIV

IL1, 7A/DIV

PWM1, 5V/DIV

1μs/DIV

FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS

FOR 3-PHASE CONVERTER

T o understand the reduction of ripple current amplitude in the multi-phase circuit, examine Equation 2, which represents an individual channel peak-to-peak inductor current.

VINV

–()V

OUT

------------------------------------------------------=

LfSV

In Equation 2, V

and V

OUT

are the input and output

OUT

(EQ. 2)

voltages respectively, L is the single-channel inductor value, and f

is the switching frequency.

The output capacitors conduct the ripple component of the inductor current. In the case of multi-phase converters, the capacitor current is the sum of the ripple currents from each of the individual channels. Compare Equation 2 to the expression for the peak-to-peak current after the summation of N symmetrically phase-shifted inductor currents in Equation 3. Peak-to-peak ripple current decreases by an amount proportional to the number of channels. Outputvoltage ripple is a function of capacitance, capacitor equivalent series resistance (ESR), and inductor ripple

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current. Reducing the inductor ripple current allows the designer to use fewer or less costly output capacitors.

VINNV

–()V

OUT

------------------------------------------------------------=

CPP,

LfSV

OUT

(EQ. 3)

Another benefit of interleaving is to reduce input ripple current. Input capacitance is determined in part by the maximum input ripple current. Multi-phase topologies can improve overall system cost and size by lowering input ripple current and allowing the designer to reduce the cost of input capacitance. The example in Figure 2 illustrates input currents from a three-phase converter combining to reduce the total input ripple current.

The converter depicted in Figure 2 delivers 1.5V to a 36A load from a 12V input. The RMS input capacitor current is 5.9A. Compare this to a single-phase converter also stepping down 12V to 1.5V at 36A. The single-phase converter has

11.9A

input capacitor current. The single-phase converter

RMS

must use an input capacitor bank with twice the RMS current capacity as the equivalent three-phase converter.

INPUT-CAPACITOR CURRENT, 10A/DIV

CHANNEL 3 INPUT CURRENT 10A/DIV

CHANNEL 2 INPUT CURRENT 10A/DIV

CHANNEL 1 INPUT CURRENT 10A/DIV

1µs/DIV

FIGURE 2. CHANNEL INPUT CURRENTS AND INPUT-

CAPACITOR RMS CURRENT FOR 3-PHASE CONVERTER

Figures 25, 26 and 27 in the section entitled “Input Capacitor Selection” on page 32 can be used to determine the input-capacitor RMS current based on load current, duty cycle, and the number of channels. They are provided as aids in determining the optimal input capacitor solution.

Active Pulse Positioning Modulated PWM Operation

The ISL6323A uses a proprietary Active Pulse Positioning (APP) modulation scheme to control the internal PWM signals that command each channel’s driver to turn their upper and lower MOSFETs on and off. The time interval in which a PWM signal can occur is generated by an internal clock, whose cycle time is the inverse of the switching frequency set by the resistor between the FS pin and ground. The advantage of Intersil’s proprietary Active Pulse Positioning (APP) modulator is that the PWM signal has the ability to turn on at any point during this PWM time interval,

and turn off immediately after the PWM signal has transitioned high. This is important because it allows the controller to quickly respond to output voltage drops associated with current load spikes, while avoiding the ring back affects associated with other modulation schemes.

The PWM output state is driven by the position of the error amplifier output signal, V

, minus the current correction

COMP

signal relative to the proprietary modulator ramp waveform as illustrated in Figure 3. At the beginning of each PWM time interval, this modified V

signal is compared to the

COMP

internal modulator waveform. As long as the modified V

voltage is lower then the modulator waveform

COMP

voltage, the PWM signal is commanded low. The internal MOSFET driver detects the low state of the PWM signal and turns off the upper MOSFET and turns on the lower synchronous MOSFET. When the modified V

COMP

voltage crosses the modulator ramp, the PWM output transitions high, turning off the synchronous MOSFET and turning on the upper MOSFET. The PWM signal will remain high until the modified V

voltage crosses the modulator ramp

COMP

again. When this occurs the PWM signal will transition low again.

During each PWM time interval the PWM signal can only transition high once. Once PWM transitions high it can not transition high again until the beginning of the next PWM time interval. This prevents the occurrence of double PWM pulses occurring during a single period.

To further improve the transient response, ISL6323A also implements Intersil’s proprietary Adaptive Phase Alignment (APA) technique, which turns on all phases together under transient events with large step current. With both APP and APA control, ISL6323A can achieve excellent transient performance and reduce the deman d on the output capacitors.

Adaptive Phase Alignment (APA)

To further improve the transient response, the ISL6323A also implements Intersil’s proprietary Adaptive Phase Alignment (APA) technique, which turns on all of the channels together at the same time during large current step transient events. As Figure 3 shows, the APA circuitry works by monitoring the voltage on the APA pin and comparing it to a filtered copy of the voltage on the COMP pin. The voltage on the APA pin is a copy of the COMP pin voltage that has been negatively offset. If the APA pin exceeds the filtered COMP pin voltage an APA event occurs and all of the channels are forced on.

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EXTERNAL CIRCUIT

APA

FIGURE 3. ADAPTIVE PHASE ALIGNMENT DETECTION

APA

APA,TRIP

COMP

ISL6323A INTERNAL CIRCUIT

100µA

APA

LOW PASS

FILTER

ERROR

AMPLIFIER

CIRCUITRY

TO APA

The APA trip level is the amount of DC offset between the COMP pin and the APA pin. This is the voltage excursion that the APA and COMP pins must have during a transient event to activate the Adaptive Phase Alig nment circuitry. This APA trip level is set through a resistor, R

APA

, that connects from the APA pin to the COMP pin. A 100µA current flows across R

into the APA pin to set the APA

APA

trip level as described in Equation 4. An APA trip level of 500mV is recommended for most applications. A 0.1µF capacitor , C

, should also be placed across the R

APA

resistor to help with noise immunity.

APA TRIP,

APA

100 106–×⋅=

(EQ. 4)

PWM Operation

The timing of each core channel is set by the number of active channels. Channel detection on the ISEN3- and ISEN4- pins selects 2-Channel to 4-Channel operation for the ISL6323A. The switching cycle is defined as the time between PWM pulse termination signals of each channel. The cycle time of the pulse signal is the inverse of the switching frequency set by the resistor between the FS pin and ground. The PWM signals command the MOSFET driver to turn on/off the channel MOSFETs.

For 4-channel operation, the channel firing order is 1-2-3-4: PWM3 pulse happens 1/4 of a cycle after PWM4, PWM2 output follows another 1/4 of a cycle after PWM3, and PWM1 delays another 1/4 of a cycle after PWM2. For 3-channel operation, the channel firing order is 1-2-3.

Connecting ISEN4- to VCC selects three channel operation and the pulse times are spaced in 1/3 cycle increments. If ISEN3- is connected to VCC, 2-Channel operation is selected and the PWM2 pulse happens 1/2 of a cycle after PWM1 pulse.

Continuous Current Sampling

In order to realize proper current-balance, the currents in each channel are sampled continuously every switching cycle. During this time, the current-sense amplifier uses the ISEN inputs to reproduce a signal proportional to the

inductor current, I

. This sensed current, I

, is simply a

SEN

scaled version of the inductor current.

PWM

SWITCHING PERIOD

SEN

TIME

FIGURE 4. CONTINUOUS CURRENT SAMPLING

The ISL6323A supports Inductor DCR current sensing to continuously sample each channel’s current for channel-current balance. The internal circuitry, shown in Figure 3 represents Channel N of an N-Channel converter. This circuitry is repeated for each channel in the converter, but may not be active depending on how many channels are operating.

Inductor windings have a characteristic distributed resistance or DCR (Direct Current Resistance). For simplicity, the inductor DCR is considered as a separate lumped quantity, as shown in Figure 5. The channel current I

, flowing through the inductor, passes through the DCR.

Equation 5 shows the S-domain equivalent voltage, V

across the inductor.

VLs() I

sL DCR+⋅()⋅=

A simple R-C network across the inductor (R

, R2 and C)

(EQ. 5)

extracts the DCR voltage, as shown in Figure 5. The voltage across the sense capacitor, V proportional to the channel current I

sL⋅

⎛⎞

-------------

DCR

⋅()

1R2

C⋅⋅1+

⎝⎠

--------------------------------------------------------

s()

⎛⎞

------------------------

⎜⎟

⎝⎠

, can be shown to be

KDCRI

, shown in Equation 6.

⋅⋅⋅=

(EQ. 6)

Where:

---------------------

R2R1+

(EQ. 7)

If the R-C network components are selected such that the RC time constant matches the inductor L/DCR time constant (see Equation 8), then V

is equal to the voltage drop across

the DCR multiplied by the ratio of the resistor divider, K. If a resistor divider is not being used, the value for K is 1.

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INDUCTOR

VL(s)

ISENn-

ISENn+

VCC

RSET

VC(s)

DCR

OUT

SET

UGATE(n)

MOSFET

DRIVER

ISL6323A INTERNAL CIRCUIT

SAMPLE

LGATE(n)

SEN

VC(s)

ISEN

FIGURE 5. INDUCTOR DCR CURRENT SENSING

CONFIGURATION

⋅

-------------

DCR

1R2

---------------------

1R2

C⋅=

(EQ. 8)

The capacitor voltage VC, is then replicated across the effective internal sense resistor, R current through R current. This current, I

which is proportional to the inductor

ISEN

, is continuously sensed and is

SEN

. This develops a

ISEN

then used by the controller for load-line regulation, channel-current balancing, and overcurrent detection and limiting. Equation 9 shows that the proportion between the channel current, I by the value of the effective sense resistance, R

, and the sensed current, I

SEN

ISEN

, is driven

, and

the DCR of the inductor.

DCR

----------------- -

SENIL

⋅=

ISEN

The effective internal R

resistance is important to the

ISEN

(EQ. 9)

current sensing process because it sets the gain of the load line regulation loop when droop is enabled as well as the gain of the channel-current balance loop and the overcurrent trip level. The effective internal R

resistance is user

ISEN

programmable and is set through use of the RSET pin. Placing a single resistor, R VCC pin programs the effective internal R

, from the RSET pin to the

SET

resistance

ISEN

according to Equation 10.

--------- -

ISEN

400

⋅=

SET

(EQ. 10)

The North Bridge regulator samples the load current in the same manner as the Core regulator does. The R will program all the effective internal R

resistors to the

ISEN

SET

resistor

same value.

Channel-Current Balance

One important benefit of multi-phase operation is the thermal advantage gained by distributing the dissipated heat over multiple devices and greater area. By doing this the designer avoids the complexity of driving parallel MOSFETs and the expense of using expensive heat sinks and exotic magnetic materials.

In order to realize the thermal advantage, it is important that each channel in a multi-phase converter be controlled to carry about the same amount of current at any load level. To achieve this, the currents through each channel must be sampled every switching cycle. The sampled currents, I

from each active channel are summed together and divided by the number of active channels. The resulting cycle average current, I

, provides a measure of the total

AVG

load-current demand on the converter during each switching cycle. Channel-current balance is achieved by comparing the sampled current of each channel to the cycle average current, and making the proper adjustment to each channel pulse width based on the error. Intersil’s patented current-balance method is illustrated in Figure 6, with error correction for Channel 1 represented. In the figure, the cycle average current, I sample, I

, to create an error signal IER. The filtered error

signal modifies the pulse width commanded by V correct any unbalance and force I

, is compared with the Channel 1

AVG

toward zero. The same

COMP

method for error signal correction is applied to each active channel.

FILTER

AVG

MODULATOR

RAMP

WAVEFORM

÷ N

f(s)

COMP

NOTE: Channel 3 and 4 are optional.

FIGURE 6. CHANNEL-1 PWM FUNCTION AND CURRENT-

BALANCE ADJUSTMENT

PWM1

TO GATE

CONTROL

LOGIC

VID Interface

The ISL6323A supports hybrid power control of AMD processors which operate from either a 6-bit parallel VID interface (PVI) or a serial VID interface (SVI). The VID1/SEL pin is used to command the ISL6323A into either the PVI mode or the SVI mode. Whenever the EN pin is held LOW, both the

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multi-phase Core and single-phase North Bridge Regulators are disabled and the ISL6323A is continuously sampling voltage on the VID1/SEL pin. When the EN pin is toggled HIGH, the status of the VID1/SEL pin will latch the ISL6323A into either PVI or SVI mode. This latching occurs on the rising edge of the EN signal.If the VID1/SEL pin is held LOW during the latch, the ISL6323A will be placed into SVI mode. If the VID1/SEL pin is held HIGH during the latch, the ISL6323A will be placed into PVI mode. For the ISL6323A to properly enter into either mode, the level on the VID1/SEL pin must be stable no less than 1µs prior to the EN signal transitioning from low to high.

6-Bit Parallel VID Interface (PVI)

With the ISL6323A in PVI mode, the single-phase North Bridge regulator is disabled. Only the multi-phase controller is active in PVI mode to support uniplane VDD only processors. Table 1 shows the 6-bit parallel VID codes and the corresponding reference voltage.

TABLE 1. 6-BIT PARALLEL VID CODES

VID5 VID4 VID3 VID2 VID1 VID0 VREF

0 0 0 0 0 0 1.5500

0 0 0 0 0 1 1.5250

0000101.5000

0000111.4750

0001001.4500

0001011.4250

0001101.4000

0001111.3750

0010001.3500

0010011.3250

0010101.3000

0010111.2750

0011001.2500

0011011.2250

0011101.2000

0011111.1750

0100001.1500

0100011.1250

0100101.1000

0100111.0750

0101001.0500

0101011.0250

0101101.0000

0101110.9750

0110000.9500

TABLE 1. 6-BIT PARALLEL VID CODES (Continued)

VID5 VID4 VID3 VID2 VID1 VID0 VREF

0110010.9250

0110100.9000

0110110.8750

0111000.8500

0111010.8250

0111100.8000

0111110.7750

1000000.7625

1000010.7500

1000100.7375

1000110.7250

1001000.7125

1001010.7000

1001100.6875

1001110.6750

1010000.6625

1010010.6500

1010100.6375

1010110.6250

1011000.6125

1011010.6000

1011100.5875

1011110.5750

1100000.5625

1100010.5500

1100100.5375

1100110.5250

1101000.5125

1101010.5000

1101100.4875

1101110.4750

1110000.4625

1110010.4500

1110100.4375

1110110.4250

1111000.4125

1111010.4000

1111100.3875

1111110.3750

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Serial VID Interface (SVI)

The on-board Serial VID interface (SVI) circuitry allows the processor to directly drive the core voltage and Northbridge voltage reference level within the ISL6323A. The SVC and SVD states are decoded with direction from the PWROK and VFIXEN inputs as described in the sections that follow. The ISL6323A uses a digital to analog converter (DAC) to generate a reference voltage based on the decoded SVI value. See Figure 7 for a simple SVI interface timing diagram.

PRE-PWROK METAL VID

Typical motherboard start-up occurs with the VFIXEN input low. The controller decodes the SVC and SVD inputs to determine the Pre-PWROK metal VID setting. Once the POR circuitry is satisfied, the ISL6323A begins decoding the inputs per Table 2. Once the EN input exceeds the rising enable threshold, the ISL6323A saves the Pre-PWROK metal VID value in an on-board holding register and passes this target to the internal DAC circuitry.

TABLE 2. PRE-PWROK METAL VID CODES

SVC SVD

00 1.1 01 1.0 10 0.9 11 0.8

OUTPUT VOLTAGE

(V)

The Pre-PWROK metal VID code is decoded and latched on the rising edge of the enable signal. Once enabled, the ISL6323A passes the Pre-PWROK metal VID code on to internal DAC circuitry. The internal DAC circuitry begins to ramp both the VDD and VDDNB planes to the decoded PrePWROK metal VID output level. The digital soft-start circuitry actually stair steps the internal reference to the target gradually over a fix interval. The controlled ramp of both output voltage planes reduces in-rush current during the soft-start interval. At the end of the soft-start interval, the VDDPWRGD output transitions high indicating both output planes are within regulation limits

If the EN input falls below the enable falling threshold, the ISL6323A ramps the internal reference voltage down to near zero. The VDDPWRGD deasserts with the loss of enable. The VDD and VDDNB planes will linearly decrease to near zero.

TABLE 3. VFIXEN VID CODES

OUTPUT VOLTAGE

SVC SVD

00 1.4 01 1.2 10 1.0 11 0.8

(V)

VFIX MODE

In VFIX Mode, the SVC, SVD and VFIXEN inputs are fixed external to the controller through jumpers to either GND or VDDIO. These inputs are not expected to change, but the

VCC

SVC

SVD

ENABLE

PWROK

VDD AND VDDNB

VDDPWRGD

VFIXEN

1 3 42 5 6 7 8 9 10 11 12

METAL_VID METAL_VID

FIGURE 7. SVI INTERFACE TIMING DIAGRAM: TYPICAL PRE-PWROK METAL VID STAR-TUP

V_SVI V_SVI

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ISL6323A is designed to support the potential change of state of these inputs. If VFIXEN is high, the IC decodes the SVC and SVD states per Table 3.

Once enabled, the ISL6323A begins to soft-start both VDD and VDDNB planes to the programmed VFIX level. The internal soft-start circuitry slowly stair steps the reference up to the target value and this results in a controlled ramp of the power planes. Once soft-start has ended and both output planes are within regulation limits, the VDDPWRGD pin transitions high. If the EN input falls below the enable falling threshold, then the controller ramps both VDD and VDDNB down to near zero.

SVI MODE

Once the controller has successfully soft-started and VDDPWRGD transitions high, the Northbridge SVI interface can assert PWROK to signal the ISL6323A to prepare for SVI commands. The controller actively monitors the SVI interface for set VID commands to move the plane voltages to start-up VID values. Details of the SVI Bus protocol are

Once the set VID command is received, the ISL6323A decodes the information to determine which plane and the VID target required (see Table 4). The internal DAC circuitry steps the required output plane voltage to the new VID level. During this time one or both of the planes could be targeted. In the event the core voltage plane, VDD, is commanded to power off by serial VID commands, the VDDPWRGD signal remains asserted. The Northbridge voltage plane must remain active during this time.

If the PWROK input is deasserted, then the controller steps both VDD and VDDNB planes back to the stored PrePWROK metal VID level in the holding register from initial soft-start. No attempt is made to read the SVC and SVD inputs during this time. If PWROK is reasserted, then the on-board SVI interface waits for a set VID command.

If VDDPWRGD deasserts during normal operation, both voltage planes are powered down in a controlled fashion. The internal DAC circuitry stair steps both outputs down to near zero.

provided in the AMD Design Guide for Voltage Regulator Controllers Accepting Serial VID Codes specification.

TABLE 4. SERIAL VID CODES

SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V)

000_0000b 1.5500 010_0000b 1.1500 100_0000b 0.7500 110_0000b 0.3500 000_0001b 1.5375 010_0001b 1.1375 100_0001b 0.7375 110_0001b 0.3375* 000_0010b 1.5250 010_0010b 1.1250 100_0010b 0.7250 110_0010b 0.3250*

000_0011b 1.5125 010_0011b 1.1125 100_0011b 0.7125 110_0011b 0.3125* 000_0100b 1.5000 010_0100b 1.1000 100_0100b 0.7000 110_0100b 0.3000* 000_0101b 1.4875 010_0101b 1.0875 100_0101b 0.6875 110_0101b 0.2875*

000_0110b 1.4750 010_0110b 1.0750 100_0110b 0.6750 110_0110b 0.2750*

000_0111b 1.4625 010_0111b 1.0625 100_0111b 0.6625 110_0111b 0.2625* 000_1000b 1.4500 010_1000b 1.0500 100_1000b 0.6500 110_1000b 0.2500* 000_1001b 1.4375 010_1001b 1.0375 100_1001b 0.6375 110_1001b 0.2375* 000_1010b 1.4250 010_1010b 1.0250 100_1010b 0.6250 110_1010b 0.2250*

000_1011b 1.4125 010_1011b 1.0125 100_1011b 0.6125 110_1011b 0.2125*

000_1100b 1.4000 010_1100b 1.0000 100_1100b 0.6000 110_1100b 0.2000*

000_1101b 1.3875 010_1101b 0.9875 100_1101b 0.5875 110_1101b 0.1875*

000_1110b 1.3750 010_1110b 0.9750 100_1110b 0.5750 110_1110b 0.1750*

000_1111b 1.3625 010_1111b 0.9625 100_1111b 0.5625 110_1111b 0.1625* 001_0000b 1.3500 011_0000b 0.9500 101_0000b 0.5500 111_0000b 0.1500* 001_0001b 1.3375 011_0001b 0.9375 101_0001b 0.5375 111_0001b 0.1375* 001_0010b 1.3250 011_0010b 0.9250 101_0010b 0.5250 111_0010b 0.1250*

001_0011b 1.3125 011_0011b 0.9125 101_0011b 0.5125 111_0011b 0.1125* 001_0100b 1.3000 011_0100b 0.9000 101_0100b 0.5000 111_0100b 0.1000* 001_0101b 1.2875 011_0101b 0.8875 101_0101b 0.4875* 111_0101b 0.0875*

001_0110b 1.2750 011_0110b 0.8750 101_0110b 0.4750* 111_0110b 0.0750*

001_0111b 1.2625 011_0111b 0.8625 101_0111b 0.4625* 111_0111b 0.0625* 001_1000b 1.2500 011_1000b 0.8500 101_1000b 0.4500* 111_1000b 0.0500* 001_1001b 1.2375 011_1001b 0.8375 101_1001b 0.4375* 111_1001b 0.0375*

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TABLE 4. SERIAL VID CODES (Continued)

SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V) SVID[6:0] VOLTAGE (V)

001_1010b 1.2250 011_1010b 0.8250 101_1010b 0.4250* 111_1010b 0.0250*

001_1011b 1.2125 011_1011b 0.8125 101_1011b 0.4125* 111_1011b 0.0125*

001_1100b 1.2000 011_1100b 0.8000 101_1100b 0.4000* 111_1100b OFF

001_1101b 1.1875 011_1101b 0.7875 101_1101b 0.3875

001_1110b 1.1750 011_1110b 0.7750 101_1110b 0.3750

001_1111b 1.1625 011_1111b 0.7625 101_1111b 0.3625

NOTE:

*Indicates a VID not required for AMD Family 10h processors.

* 111_1101b OFF * 111_1110b OFF * 111_1111b OFF

POWER SAVINGS MODE: PSI_L

Bit 7 of the Serial VID codes transmitted as part of th e 8-bit data phase over the SVI bus is allocated for the PSI_L. If Bit 7 is 0, then the processor is at an optimal load for the regulator to enter power savings mode. If Bit 7 is 1, then the regulator should not be in power savings mode.

With the ISL6323A, Power Savings mode is realized through phase shedding. Once a Serial VID command with Bit 7 set to 0 is received, the ISL6323A will shed all phases in a sequential manner until only Channel 1 is switching. If active, Channel 4 will be shed first, followed by Channel 3 with Channel 2 being shed last. When a phase is shed, that phase will not go into a tri-state mode until that phase would have had its PWM go HIGH.

When leaving Power Savings Mode, through the reception of a Serial VID command with Bit 7 set to 1, the ISL6323A will sequentially turn on phases starting with Phase 2. When a phase is being reactivated, it will not leave a tri-state until the PWM of that phase goes HIGH.

If, while in Power Savings Mode, a Serial VID command is received that forces a VID level change while maintaining Bit 7 at 0, the ISL6323A will first exit the Power Savings Mode state as previously described. The output voltage will then be stepped up or down to the appropriate VID level. Finally, the ISL6323A will then re-enter Power Savings Mode.

Voltage Regulation

The integrating compensation network shown in Figure 8 insures that the steady-state error in the output voltage is limited only to the error in the reference voltage and offset errors in the OFS current source, remote-sense and error amplifiers. Intersil specifies the guaranteed tolerance of the ISL6323A to include the combined tolerances of each of these elements.

The output of the error amplifier, V modulator to generate the PWM signals. The PWM signals control the timing of the Internal MOSFET drivers and regulate the converter output so that the voltage at FB is equal to the voltage at REF. This will regulate the output voltage to be equal to Equation 11. The internal and external circuitry that controls voltage regulation is illustrated in Figure 8.

, is used by the

COMP

OUTVREFVOFS

– V

–=

DROOP

(EQ. 11)

The ISL6323A incorporates differential remote-sense amplification in the feedback path. The differential sensing removes the voltage error encountered when measuring the output voltage relative to the controller ground reference point resulting in a more accurate means of sensing output voltage.

EXTERNAL CIRCUIT ISL6323A INTERNAL CIRCUIT

COMP

OUT

FIGURE 8. OUTPUT VOLT AGE AND LOAD-LINE

+ V

DROOP

REGULATION WITH OFFSET ADJUSTMENT

OFS

VSEN

RGND

)

DROOP

CONTROL

∑

OSCILLATOR

8 I

AVG

OFS

ERROR

AMPLIFIER

VID

DAC

COMP

Load-Line (Droop) Regulation

By adding a well controlled output impedance, the output voltage can effectively be level shifted in a direction which works to achieve a cost-effective solution can help to reduce the output-voltage spike that results from fast load-current demand changes.

The magnitude of the spike is dictated by the ESR and ESL of the output capacitors selected. By positioning the no-load voltage level near the upper specification limit, a larger negative spike can be sustained without crossing the lower limit. By adding a well controlled output impedance, the

FN6878.0

ISL6323A

output voltage under load can effectively be level shifted down so that a larger positive spike can be sustained without crossing the upper specification limit.

As shown in Figure 8, with the FS resistor tied ground, a current eight times the average current of all active channels, 8*I regulation resistor R R

is proportional to the output current, effectively creating

, flows from FB through a load-line

AVG

. The resulting voltage drop across

an output voltage droop with a steady-state value defined as in Equation 12:

DROOPIAVGRFB

⋅=

(EQ. 12)

The regulated output voltage is reduced by the droop voltage V

. The output voltage as a function of load current is

DROOP

shown in Equation 13.

⎛⎞

OUT

OUTVREFVOFS

In Equation 13, V

–

REF

programmed offset voltage, I of the converter, K R

resistor connected to the RSET pin (KI is defined in

SET

-------------

–=

⋅⋅ ⋅⋅

⎜⎟

⎝⎠

is the reference voltage, V

is an internal gain determined by the

OUT

400

--------- -

---------------

⋅

SET

OFS

(EQ. 13)

is the

⎛⎞

DCR

⎝⎠

is the total output current

Equation 10), K is the DC gain of the RC filter across the inductor (K is defined in Equation 7), N is the number of active channels, and DCR is the Inductor DCR value.

Output-Voltage Offset Programming

The ISL6323A allows the designer to accurately adjust the offset voltage by connecting a resistor, R pin to VCC or GND. When R

is connected between OFS

OFS

and VCC, the voltage across it is regulated to 1.6V. This causes a proportional current (I and out of the OFS pin. If R

OFS

) to flow into the FB pin

OFS

is connected to ground, the voltage across it is regulated to 0.3V, and I OFS pin and out of the FB pin. The offset current flowing through the resistor between VDIFF and FB will generate the desired offset voltage which is equal to the product (I

OFSxRFB

). These functions are shown in Figures 9 and

10. Once the desired output offset voltage has been determined,

use the formulas in Equations 14 and 15 to set R For Positive Offset (connect R

0.3 RFB×

--------------------------

OFS

OFFSET

For Negative Offset (connect R

1.6 RFB×

--------------------------

OFS

OFFSET

OFS

to GND):

to VCC):

, from the OFS

OFS

flows into the

OFS

(EQ. 14)

(EQ. 15)

VDIFF

OFS

VCC

OFS

FIGURE 9. NEGATIVE OFFSET OUTPUT VOL TAGE

PROGRAMMING

OUT

OFS

FIGURE 10. POSITIVE OFFSET OUTPUT VOLT AGE

OFS

GND

PROGRAMMING

ISL6323A

VREF

E/A

GND

E/A

GND

1.6V

0.3V

VCC

1.6V

0.3V

VCC

Dynamic VID

The AMD processor does not step the output voltage commands up or down to the target voltage, but instead passes only the target voltage to the ISL6323A through either the PVI or SVI interface. The ISL6323A manages the resulting VID-on-the-Fly transition in a controlled manner, supervising a safe output voltage transition without discontinuity or disruption. The ISL6323A begins slewing the DAC at 3.25mV/µs until the DAC and target voltage are

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ISL6323A

equal. Thus, the total time required for a dynamic VID transition is dependent only on the size of the DAC change.

To further improve dynamic VID performance, ISL6323A also implements a proprietary DAC smoothing feature. The external series RC components connected between DVC and FB limit any stair-stepping of the output voltage during a VID-on-the-Fly transition.

Compensating Dynamic VID Transitions

During a VID transition, the resulting change in voltage on the FB pin and the COMP pin causes an AC current to flow through the error amplifier compensation components from the FB to the COMP pin. This current then flows through the feedback resistor, R overshoot or undershoot at the end of the VID transition. In order to ensure the smooth transition of the output voltage during a VID change, a VID-on-the-fly compensation network is required. This network is composed of a resistor and capacitor in series, R and the FB pin.

VSEN

DVC

VDAC+RGND

FIGURE 11. DYNAMIC VID COMPENSATION NETWORK

This VID-on-the-fly compensation network works by sourcing AC current into the FB node to offset the effects of the AC current flowing from the FB to the COMP pin during a VID transition. To create this compensation current the ISL6323A sets the voltage on the DVC pin to be 2x the voltage on the REF pin. Since the error amplifier forces the voltage on the FB pin and the REF pin to be equal, the resulting voltage across the series RC between DVC and FB is equal to the REF pin voltage. The RC compensation components, R create the desired amount of compensation current.

The amount of compensation current required is dependant on the modulator gain of the system, K1, and the error amplifier R-C components, R between the FB and COMP pins. Use Equations 17, 18 and 19 to calculate the RC component values, R for the VID-on-the-fly compensation network. For these equations: V

is the input voltage for the power train; V

is the oscillator ramp amplitude (1.5V); and R

, and can cause the output voltage to

DVC

and C

DVC

, between the DVC

DVC

= I

ISL6323A INTERNAL CIRCUIT

DVC

and C

, can then be selected to

DVC

and CC, that are in series

ERROR

AMPLIFIER

and C

DVC

and CC are

COMP

DVC

P-P

the error amplifier R-C components between the FB and COMP pins.

RCOMP

-----------

------- -

---------------- -

K1 1–

×=

(EQ. 16)

(EQ. 17)

(EQ. 18)

Advanced Adaptive Zero Shoot-Through Deadtime Control (Patent Pending)

The integrated drivers incorporate a unique adaptive deadtime control technique to minimize deadtime, resulting in high efficiency from the reduced freewheeling time of the lower MOSFET body-diode conduction, and to prevent the upper and lower MOSFETs from conducting simultaneously . This is accomplished by ensuring either rising gate turns on its MOSFET with minimum and sufficient delay after the other has turned off.

During turn-off of the lower MOSFET, the PHASE voltage is monitored until it reaches a -0.3V/+0.8V (forward/reverse inductor current). At this time the UGATE is released to rise. An auto-zero comparator is used to correct the r

DS(ON)

drop in the phase voltage preventing false detection of the -0.3V phase level during r

DS(ON)

conduction period. In the case of zero current, the UGA TE is released after 35ns delay of the LGATE dropping below 0.5V. When LGATE first begins to transition low, this quick transition can disturb the PHASE node and cause a false trip, so there is 20ns of blanking time once LGAT E falls until PHASE is monitored.

Once the PHASE is high, the advanced adaptive shoot-through circuitry monitors the PHASE and UGATE voltages during a PWM falling edge and the subsequent UGATE turn-off. If either the UGATE falls to less than 1.75V above the PHASE or the PHASE falls to less than +0.8V , the LGATE is released to turn-on.

Initialization

Prior to initialization, proper conditions must exist on the EN, VCC, PVCC1_2, PVCC_NB, ISEN3-, and ISEN4- pins. When the conditions are met, the controller begins soft-start. Once the output voltage is within the proper window of operation, the controller asserts PGOOD.

Power-On Reset

The ISL6323A requires VCC, PVCC1_2, and PVCC_NB inputs to exceed their rising POR thresholds before the ISL6323A has sufficient bias to guarantee proper operation.

The bias voltage applied to VCC must reach the internal power-on reset (POR) rising threshold. Once this threshold

is reached, the ISL6323A has enough bias to begin checking the driver POR inputs, EN, and channel detect portions of the initialization cycle. Hysteresis between the rising and falling thresholds assure the ISL6323A will not advertently

FN6878.0

ISL6323A

ISL6323A INTERNAL CIRCUIT

POR

CIRCUIT

SOFT-START

AND

FAULT LOGIC

FIGURE 12. POWER SEQUENCING USING THRESHOLD-

SENSITIVE ENABLE (EN) FUNCTION

ENABLE COMPARATOR

0.86V

CHANNEL

DETECT

EXTERNAL CIRCUIT

VCC

PVCC1_2

PVCC_NB

+12V

10.7kΩ

1.00kΩ

ISEN3-

ISEN4-

turn off unless the bias voltage drops substantially (see Electrical Specifications on page 6).

The bias voltage applied to the PVCC1_2 and PVCC_NB pins power the internal MOSFET drivers of each output channel. In order for the ISL6323A to begin operation, both PVCC inputs must exceed their POR rising threshold to guarantee proper operation of the internal drivers. Hysteresis between the rising and falling thresholds assure that once enabled, the ISL6323A will not inadvertently turn off unless the PVCC bias voltage drops substantially (see “Electrical Specifications” on page 6). Depending on the number of active CORE channels determined by the Phase Detect block, the external driver POR checking is supported by the Enable Comparator.

Enable Comparator

The ISL6323A features a dual function enable input (EN) for enabling the controller and power sequencing between the controller and external drivers or another voltage rail. The enable comparator holds the ISL6323A in shutdown until the voltage at EN rises above 0.86V . The enable comparator has about 110mV of hysteresis to prevent bounce. It is important that the driver ICs reach their rising POR level before the ISL6323A becomes enabled. The schematic in Figure 12 demonstrates sequencing the ISL6323A with the ISL66xx family of Intersil MOSFET drivers, which require 12V bias.

When selecting the value of the resistor divider the driver maximum rising POR threshold should be used for calculating the proper resistor values. This will prevent improper sequencing events from creating false trips during soft-start.

If the controller is configured for 2-phase CORE operation, then the resistor divider can be used for sequencing the controller with another voltage rail. The resistor divider to EN should be selected using a similar approach as the previous driver discussion.

The EN pin is also used to force the ISL6323A into either PVI or SVI mode. The mode is set upon the rising edge of the EN signal. When the voltage on the EN pin rises above

0.86V , the mode will be set depending upon the status of the VID1/SEL pin.

Phase Detection

The ISEN3- and ISEN4- pins are monitored prior to soft-start to determine the number of active CORE channel phases.

If ISEN4- is tied to VCC, the controller will configure the channel firing order and timing for 3-phase operation. If ISEN3- and ISEN4- are tied to VCC, the controller will set the channel firing order and timing for 2-phase operation (see “PWM Operation” on page 13 for details).

Soft-Start Output Voltage Targets

Once the POR and Phase Detect blocks and enable comparator are satisfied, the controller will begin the softstart sequence and will ramp the CORE and NB output voltages up to the SVI interface designated target level if the controller is set SVI mode. If set to PVI mode, the North Bridge regulator is disabled and the core is soft started to the level designated by the parallel VID code.

SVI Mode

Prior to soft-starting both CORE and NB outputs, the ISL6323A must check the state of the SVI interface inputs to determine the correct target voltages for both outputs. When the controller is enabled, the state of the VFIXEN, SVD and SVC inputs are checked and the target output voltages set for both CORE and NB outputs are set by the DAC (see “Serial VID Interface (SVI)” on page 16). These targets will only change if the EN signal is pulled low or after a POR reset of VCC.

Soft-Start

The soft-start sequence is composed of three periods, as shown in Figure 13. At the beginning of soft-start, the DAC immediately obtains the output voltage targets for both outputs by decoding the state of the SVI or PVI inputs. A 100µs fixed delay time, TDA, proceeds the output voltage rise. After this delay period the ISL6323A will begin ramping both CORE and NB output voltages to the programmed DAC level at a fixed rate of 3.25mV/µs. The amount of time required to ramp the output voltage to the final DAC voltage is referred to as TDB, and can be calculated as shown in Equation 19.

DAC

------------------------------

TDB

3.25 103–×

(EQ. 19)

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ISL6323A

After the DAC voltage reaches the final VID setting, PGOOD will be set to high.

400mV/DIV

TDA

5V/DIV

FIGURE 13. SOFT-START WAVEFORMS

TDB

VDDPWRGD

5V/DIV

100µs/DIV

CORE

400mV/DIV

Pre-Biased Soft-Start

The ISL6323A also has the ability to start up into a pre-charged output, without causing any unnecessary disturbance. The FB pin is monitored during soft-start, and should it be higher than the equival e nt internal ramping reference voltage, the output drives hold both MOSFETs off.

Once the internal ramping reference exceeds the FB pin potential, the output drives are enabled, allowing the output to ramp from the pre-charged level to the final level dictated by the DAC setting. Should the output be pre-charged to a level exceeding the DAC setting, the output drives are enabled at the end of the soft-start period, leading to an abrupt correction in the output voltage down to the DAC-set level. Both CORE and NB output support start-up into a pre-charged output.

OUTPUT PRECHARGED

ABOVE DAC LEVEL

OUTPUT PRECHARGED

BELOW DAC LEVEL

CORE

400mV/DIV

5V/DIV

Fault Monitoring and Protection

The ISL6323A actively monitors both CORE and NB output voltages and currents to detect fault conditions. Fault monitors trigger protective measures to prevent damage to either load. One common power good indicator is provided for linking to external system monitors. The schematic in Figure 15 outlines the interaction between the fault monitors and the power good signal.

Power Good Signal

The power good pin (VDDPWRGD) is an open-drain logic output that signals whether or not the ISL6323A is regulating both NB and CORE output voltages within the proper levels, and whether any fault conditions exist. This pin should be tied to a +5V source through a resistor.

During shutdown and soft-start, VDDPWRGD pulls low and releases high after a successful soft-start and both output voltages are operating between the undervoltage and overvoltage limits. PGOOD transitions low when an undervoltage, overvoltage, or overcurrent condition is detected on either output or when the controller is disabled by a POR reset or EN. In the event of an overvoltage or overcurrent condition, the controller latches off and PGOOD will not return high. Pending a POR reset of the ISL6323A and successful soft-start, the PGOOD will return high.

Overvoltage Protection

The ISL6323A constantly monitors the sensed output voltage on the VSEN pin to detect if an overvoltage event occurs. When the output voltage rises above the OVP trip level and exceeds the PGOOD OV limit actions are taken by the ISL6323A to protect the microprocessor load.

At the inception of an overvoltage event, both on-board lower gate pins are commanded low as are the active PWM outputs to the external drivers, the PGOOD signal is driven low, and the ISL6323A latches off normal PWM action. This turns on the all of the lower MOSFETs and pulls the output voltage below a level that might cause damage to the load. The lower MOSFETs remain driven ON until VDIFF falls below 400mV . The ISL6323A will continue to protect the load in this fashion as long as the overvoltage condition recurs. Once an overvoltage condition ends the ISL6323A latches off, and must be reset by toggling POR, before a soft-start can be re-initiated.

100µs/DIV

FIGURE 14. SOFT-START WAVEFORMS FOR ISL6323A-

BASED MULTIPHASE CONVERTER

FN6878.0

ISL6323A

142µA

OCL

100µA

NB ONLY

OCP

SOFT-START, FAULT

AND CONTROL LOGIC

ISEN_NB+

DAC - 300mV

CORE ONL Y

1.8V

DAC + 250mV

VSEN

DAC - 300mV

FIGURE 15. POWER GOOD AND PROTECTION CIRCUITRY

Pre-POR Overvoltage Protection

Prior to PVCC and VCC exceeding their POR levels, the ISL6323A is designed to protect either load from any overvoltage events that may occur. This is accomplished by means of an internal 10kΩ resistor tied from PHASE to LGATE, which turns on the lower MOSFET to control the output voltage until the overvoltage event ceases or the input power supply cuts off. For complete protection, the low side MOSFET should have a gate threshold well below the maximum voltage rating of the load/microprocessor.

In the event that during normal operation the PVCC or VCC voltage falls back below the POR threshold, the pre-POR overvoltage protection circuitry reactivates to protect from any more pre-POR overvoltage events.

Undervoltage Detection

The undervoltage threshold is set at VDAC - 300mV typical. When the output voltage (VSEN-RGND) is below the undervoltage threshold, PGOOD gets pulled low. No other action is taken by the controller. PGOOD will return high if the output voltage rises above VDAC - 250mV typical.

REPEAT FOR EACH

CORE CHANNEL

100µA

OCP

AVG

CORE ONLY

OVP1.8V

OVP

ISL6323A INTERNAL CIRCUITRY

VDDPWRGD

Open Sense Line Protection

In the case that either of the remote sense lines, VSEN or GND, become open, the ISL6323A is designed to detect this and shut down the controller. This event is detected by monitoring small currents that are fed out the VSEN and RGND pins. In the event of an open sense line fault, the controller will continue to remain off until the fault goes away, at which point the controller will re-initiate a soft-start sequence.

Overcurrent Protection

The ISL6323A takes advantage of the proportionality between the load current and the average current, I detect an overcurrent condition. See “Continuous Current Sampling” on page 13 and “Channel-Current Balance” on page 14 for more detail on how the average current is measured. Once the average current exceeds 100µA, a comparator triggers the converter to begin overcurrent protection procedures. The Core regulator and the North Bridge regulator have the same type of overcurrent protection.

The overcurrent trip threshold is dictated by the DCR of the inductors, the number of active channels, the DC gain of the inductor RC filter and the R

resistor . The ov ercurren t trip

SET

threshold is shown in Equation 20.

N– V

⋅

OCP

100μ A

-------------

DCR

⎛⎞

--- -

--------- -

⋅

⎝⎠

400

SET

----------------------------------------

⋅⋅

2Lf

Where:

---------------------

R1R2+

fSW = Switching Frequency

See “Continuous Current Sampling” on page 13.

Equation 20 is valid for both the Core regulator and the North Bridge regulator. This equation includes the DC load current as well as the total ripple current contributed by all the phases. For the North Bridge regulator, N is 1.

During soft-start, the overcurrent trip point is boosted by a factor of 1.4. Instead of comparing the average measured current to 100µA, the average current is compared to 140µA. Immediately after soft-start is over, the comparison level changes to 100µA. This is done to allow for start-up into an active load while still supplying output capacitor in-rush current.

CORE REGULATOR OVERCURRENT

At the beginning of overcurrent shutdown, the controller sets all of the UGATE and LGATE signals low, puts PWM3 and PWM4 (if active) in a high-impedance state, and forces VDDPWRGD low. This turns off all of the upper and lower MOSFETs. The system remains in this state for fixed period of 12ms. If the controller is still enabled at the end of this wait

OUT

, to

AVG

OUT

--------------- -

⋅–⋅⋅⋅=

(EQ. 20)

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ISL6323A

period, it will attempt a soft-start, as shown in Figure 16. If the fault remains, the trip-retry cycles will continue until either the fault is cleared or for a total of seven attempts. If the fault is not cleared on the final attempt, the controller disables UGATE and LGA TE signals for both Core and North Bridge and latches off requiring a POR of VCC to reset the ISL6323A.

It is important to note that during soft start, the overcurrent trip point is increased by a factor of 1.4. If the fault draws enough current to trip overcurrent during normal run mode, it may not draw enough current during the soft start ramp period to trip overcurrent while the output is ramping up. If a fault of this type is affecting the output, then the regulator will complete soft start and the trip-retry counter will be reset to zero. Once the regulator has completed soft start, the overcurrent trip point will return to it’s nominal setting and an overcurrent shutdown will be initiated. This will result in a continuous hiccup mode.

Note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there is no thermal hazard.

OUTPUT CURRENT, 50A/DIV

OUTPUT VOLTAGE, 500mV/DIV

OVERCURRENT PROTECTION IN POWER SAVINGS MODE

While in Power Savings Mode, the OCP trip point will be lower than when running in Normal Mode. Equation 20, with N = 1, will yield the OCP trip point for the Core regulator while in Power Savings mode.

If an overcurrent event should occur while the system is in Power Savings Mode, the ISL6323A will restart in the Normal state with the PSI_L bit set to 1.

Individual Channel Overcurrent Limiting

The ISL6323A has the ability to limit the current in each individual channel of the Core regulator without shutting down the entire regulator. This is accomplished by continuously comparing the sensed currents of each channel with a constant 140µA OCL reference current. If a channel’s individual sensed current exceeds this OCL limit, the UGATE signal of that channel is immediately forced low, and the LGATE signal is forced high. This turns off the upper MOSFET(s), turns on the lower MOSFET(s), and stops the rise of current in that channel, forcing the current in the channel to decrease. That channel’s UGATE signal will not be able to return high until the sensed channel current falls back below the 140µA reference.

General Design Guide

This design guide is intended to provide a high-level explanation of the steps necessary to create a multiphase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced below. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts for all common microprocessor applications.

FIGURE 16. OVERCURRENT BEHAVIOR IN HICCUP MODE

3ms/DIV

NORTH BRIDGE REGULATOR OVERCURRENT

The overcurrent shutdown sequence for the North Bridge regulator is identical to the Core regulator with the exception that it is a single phase regulator and will only disable the MOSFET drivers for the North Bridge. Once 7 retry attempts have been executed unsuccessfully, the controller will disable UGATE and LGATE signals for both Core and North Bridge and will latch off requiring a POR of VCC to reset the ISL6323A.

Note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there is no thermal hazard.

Power Stages

The first step in designing a multi-phase converter is to determine the number of phases. This determination depends heavily on the cost analysis which in turn depends on system constraints that differ from one design to the next. Principally, the designer will be concerned with whether components can be mounted on both sides of the circuit board, whether through-hole components are permitted, the total board space available for power-supply circuitry, and the maximum amount of load current. Generally speaking, the most economical solutions are those in which each phase handles between 25A and 30A. All surface-mount designs will tend toward the lower end of this current range. If through-hole MOSFET s and inductors can be used, higher per-phase currents are possible. In cases where board space is the limiting constraint, current can be pushed as high as 40A per phase, but these designs require heat sinks and forced air to cool the MOSFETs, inductors and heat-dissipating surfaces.

FN6878.0

ISL6323A

MOSFETS

The choice of MOSFETs depends on the current each MOSFET will be required to conduct, the switching frequency , the capability of the MOSFETs to dissipate heat, and the availability and nature of heat sinking and air flow .

LOWER MOSFET POWER CALCULATION

The calculation for power loss in the lower MOSFET is simple, since virtually all of the loss in the lower MOSFET is due to current conducted through the channel resistance (r output current, I d is the duty cycle (V

). In Equation 21, IM is the maximum continuous

DS(ON)

LOW 1,

is the peak-to-peak inductor current and

P-P

DS ON()

OUT/VIN

⎛⎞ ⎜⎟ ⎝⎠

----- -

LP-P()

----------------------------------------+⋅=

1d–()⋅

(EQ. 21)

An additional term can be added to the lower-MOSFET loss equation to account for additional loss accrued during the dead time when inductor current is flowing through the lower-MOSFET body diode. This term is dependent on the diode forward voltage at I frequency, f

, and the length of dead times, td1 and td2, at

, V

, the switching

D(ON)

the beginning and the end of the lower-MOSFET conduction interval respectively.

LOW 2,

DON()fS

⎛⎞

⋅⋅=

⎜⎟

------

⎝⎠

-----------+

P-P

⎛⎞

⎜⎟

td1⋅

------

–

⎜⎟

⎝⎠

P-P

----------2

td2⋅+

(EQ. 22)

The total maximum power dissipated in each lower MOSFET is approximated by the summation of P

LOW,1

and P

LOW,2

UPPER MOSFET POWER CALCULATION

In addition to r

losses, a large portion of the

DS(ON)

upper-MOSFET losses are due to currents conducted across the input voltage (V

) during switching. Since a

substantially higher portion of the upper-MOSFET losses are dependent on switching frequency, the power calculation is more complex. Upper MOSFET losses can be divided into separate components involving the upper-MOSFET switching times, the lower-MOSFET body-diode reverse-recovery charge, Q r

DS(ON)

conduction loss.

, and the upper MOSFET

When the upper MOSFET turns off, the lower MOSFET does not conduct any portion of the inductor current until the voltage at the phase node falls below ground. Once the lower MOSFET begins conducting, the current in the upper MOSFET falls to zero as the current in the lower MOSFET ramps up to assume the full inductor current. In Equation 23, the required time for this commutation is t approximated associated power loss is P

UP 1,VIN

⎛⎞

⋅⋅⋅≈

----- -

----------+

⎝⎠

P-P

⎛⎞

----

⎜⎟

⎝⎠

and the

UP,1

(EQ. 23)

At turn on, the upper MOSFET begins to conduct and this transition occurs over a time t approximate power loss is P

⎛⎞

UP 2,VIN

⋅⋅⋅≈

-------- -

⎜⎟

–

----- 2

⎝⎠

. In Equation 24, the

UP,2

⎛⎞

----

⎜⎟

⎝⎠

(EQ. 24)

A third component involves the lower MOSFET reverserecovery charge, Qrr. Since the inductor current has fully commutated to the upper MOSFET before the lower-MOSFET body diode can recover all of Q

, it is

conducted through the upper MOSFET across VIN. The power dissipated as a result is P

VINQrrf

UP 3,

⋅⋅=

UP,3

(EQ. 25)

Finally, the resistive part of the upper MOSFET is given in Equation 26 as P

UP 4,rDS ON()

UP,4

⎛⎞

⎜⎟

----- N

⎝⎠

+⋅≈

d⋅

---------12

(EQ. 26)

The total power dissipated by the upper MOSFET at full load can now be approximated as the summation of the results from Equations 23, 24, 25 and 26. Since the power equations depend on MOSFET parameters, choosing the correct MOSFETs can be an iterative process involving repetitive solutions to the loss equations for different MOSFETs and different switching frequencies.

Internal Bootstrap Device

All three integrated drivers feature an internal bootstrap schottky diode. Simply adding an external capacitor across the BOOT and PHASE pins completes the bootstrap circuit. The bootstrap function is also designed to prevent the bootstrap capacitor from overcharging due to the large negative swing at the PHASE node. This reduces voltage stress on the boot to phase pins.

The bootstrap capacitor must have a maximum voltage rating above PVCC + 4V and its capacitance value can be chosen from Equation 27:

GATE

BOOT_CAP

GATE

where Q at V

GS1

control MOSFETs. The ΔV allowable droop in the rail of the upper gate drive.

--------------------------------------

≥

ΔV

BOOT_CAP

QG1PVCC•

----------------------------------- -

GS1

is the amount of gate charge per upper MOSFET

•=

(EQ. 27)

gate-source voltage and NQ1 is the number of

BOOT_CAP

term is defined as the

FN6878.0

ISL6323A

1.6

1.4

1.2

(µF)

0.8

0.6

BOOT_CAP

0.4

0.2 20nC

0.0

FIGURE 17. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE

Gate Drive Voltage Versatility

The ISL6323A provides the user flexibility in choosing the gate drive voltage for efficiency optimization. The controller ties the upper and lower drive rails together. Simply applying a voltage from 5V up to 12V on PVCC sets both gate drive rail voltages simultaneously.

Package Power Dissipation

When choosing MOSFETs it is important to consider the amount of power being dissipated in the integrated drivers located in the controller. Since there are a total of three drivers in the controller package, the total power dissipated by all three drivers must be less than the maximum allowable power dissipation for the QFN package.

Calculating the power dissipation in the drivers for a desired application is critical to ensure safe operation. Exceeding the maximum allowable power dissipation level will push the IC beyond the maximum recommended operating junction temperature of +125°C. The maximum allowable IC power dissipation for the 7x7 QFN package is approximately 3.5W at room temperature. See “Layout Considerations” on page 32 for thermal transfer improvement suggestions.

50nC

VOLTAGE

= 100nC

GATE

0.30.0 0.1 0.2 0.4 0.5 0.6 0.90.7 0.8 1.0 ΔV

BOOT_CAP

(V)

Qg_TOTPQg_Q1PQg_Q2IQ

Qg_Q1

Qg_Q2QG2

-- -

⎛⎞

-- -

⋅ QG2NQ2⋅+

⎝⎠

In Equations 28 and 29, P power loss and P loss; the gate charge (Q

PVCC FSWNQ1N

⋅⋅ ⋅⋅⋅=

PVCC FSWNQ2N

⋅⋅⋅⋅=

N⋅

Qg_Q1

is the total lower gate drive power

Qg_Q2

VCC⋅++=

PHASE

PHASEFSWIQ

+⋅⋅=

is the total upper gate drive

and QG2) is defined at the

(EQ. 28)

(EQ. 29)

particular gate to source drive voltage PVCC in the corresponding MOSFET data sheet; I quiescent current with no load at both drive outputs; N and N phase, respectively; N phases. The I

are the number of upper and lower MOSFETs per

VCC product is the quiescent power of the

is the number of active

PHASE

is the driver total

controller without capacitive load and is typically 75mW at 300kHz.

PVCC

FIGURE 18. TYPICAL UPPER-GATE DRIVE TURN-ON PATH

PVCC

BOOT

PHASE

HI2

LO2

HI1

LO1

UGATE

LGATE

GI1R

GI2

When designing the ISL6323A into an application, it is recommended that the following calculation is used to ensure safe operation at the desired frequency for the selected MOSFETs. The total gate drive power losses, P

Qg_TOT

, due to the gate charge of MOSFETs and the integrated driver’s internal circuitry and their corresponding average driver current can be estimated with Equations 28 and 29, respectively.

FIGURE 19. TYPICAL LOWER-GATE DRIVE TURN-ON PATH

The total gate drive power losses are dissipated among the resistive components along the transition path and in the bootstrap diode. The portion of the total power dissipated in the controller itself is the power dissipated in the upper drive path resistance (P (P

), and in the boot strap diode (P

DR_UP

), the lower drive path resistance

DR_UP

BOOT

). The rest of the power will be dissipated by the external gate resistors (R

and RG2) and the internal gate resistors (R

) of the MOSFETs. Figures 18 and 19 show the typical

GI2

GI1

and

upper and lower gate drives turn-on transition path. The total

FN6878.0

ISL6323A

power dissipation in the controller itself, PDR, can be roughly estimated as Equation 30:

DRPDR_UPPDR_LOW PBOOTIQ

Qg_Q1

BOOT

DR_UP

DR_LOW

EXT1RG1

---------------------

⎛⎞

HI1

--------------------------------------

⎜⎟

⎝⎠

HI1REXT1

⎛⎞

HI2

--------------------------------------

⎜⎟

⎝⎠

HI2REXT2

GI1

-------------

LO1

----------------------------------------

LO1REXT1

LO2

----------------------------------------

LO2REXT2

EXT2RG2

VCC⋅()+++=

Qg_Q1

---------------------

⋅=

Qg_Q2

---------------------

⋅=

GI2

-------------

(EQ. 30)

Inductor DCR Current Sensing Component Selection and R

With the single R effective internal sense resistors for both the North Bridge and Core regulators, it is important to set the R and the inductor RC filter gain, K, properly. See “Continuous Current Sampling” on page 13 and “Channel-Current Balance” on page 14 for more details on the application of the R

There are 3 separate cases to consider when calculating these component values. If the system under design will never utilize the North Bridge regulator and the ISL6323 will always be in parallel mode, then follow the instructions for Case 3 and only calculate values for Core regulator components.

For all three cases, use the expected VID voltage that would be used at TDC for Core and North Bridge for the V and V

CASE 1

In Case 1, the DC voltage across the North Bridge inductor at full load is less than the DC voltage across a single phase of the Core regulator while at full load. Here, the DC voltage across the Core inductors must be scaled down to match the DC voltage across the North Bridge inductor, which will be impressed across the ISEN_NB pins without any gain. So, the R unpopulated and K = 1.

1. Choose a capacitor value for the North Bridge RC filter. A

2. Calculate the value for resistor R

resistor and the RC filter gain.

SET

variables, respectively.

DCR

⋅

MAX

resistor for the North Bridge inductor RC filter is left

0.1µF capacitor is a recommended starting point.

--------------------------------------

DCRNBCNB⋅

Value Calculation

SET

resistor setting the value of the

SET

Core

MAX

--------------------------

⋅<

DCR

Core

using Equation 32:

SET

value

CORE

(EQ. 31)

(EQ. 32)

3. Calculate the value for the R

resistor using

SET

Equation 33: (Derived from Equation 20).

SET

Where:

400

--------- -

K = 1

DCR

------------------------------

⋅⋅=

100μ A

K⋅

⎛⎞

⎜⎟

OCP

⎝⎠

VINVNB–

----------------------------------

⋅⋅

NBfSW

⋅+

-----------

(EQ. 33)

4. Using Equation 34 (also derived from Equation 20), calculate the value of K for the Core regulator.

--------- -

400

⋅⋅ ⋅=

SET

------------------------------

DCR

CORE

-----------------------------------------------------------------------------------------------------------

OCP

CORE

100μ A

VINN– V

⋅

---------------------------------------------

⋅⋅

CORE

COREfSW

(EQ. 34)

--------------------

⋅+

5. Choose a capacitor value for the Core RC filters. A 0.1µF capacitor is a recommended starting point.

6. Calculate the values for R

and R2 for Core.

Equations 35 and 36 will allow for their computation.

----------------------------------------------

Core

--------------------------

DCR

Core

----------------------------------------------

Core

⋅

Core

⋅=

Core

(EQ. 35)

(EQ. 36)

CASE 2

Core

MAX

DCR

⋅

MAX

--------------------------

⋅>

DCR

Core

(EQ. 37)

In Case 2, the DC voltage across the North Bridge inductor at full load is greater than the DC voltage across a single phase of the Core regulator while at full load. Here, the DC voltage across the North Bridge inductor must be scaled down to match the DC voltage across the Core inductors, which will be impressed across the ISEN pins without any gain. So, the R

resistor for the Core inductor RC filters is

left unpopulated and K = 1.

1. Choose a capacitor value for the Core RC filter. A 0.1µF capacitor is a recommended starting point.

2. Calculate the value for resistor R

Core

------------------------------------------------

DCR

⋅

CoreCCore

3. Calculate the value for the R

resistor using Equation 39

SET

(EQ. 38)

(Derived from Equation 20).:

SET

Where:

DCR

400

-------------------------------------- -

--------- -

⋅⋅=

K = 1

CORE

100μ A

K⋅

⎛⎞

⎜⎟

OCP

CORE

⎝⎠

VINV

–

------------------------------------------ -

CORE

⋅⋅

COREfSW

(EQ. 39)

⋅+

4. Using Equation 40 (also derived from Equation 20), calculate the value of K for the North bridge regulator.

100μ A

VINVNB–

----------------------------------

⋅⋅

NBfSW

⋅+

-----------

(EQ. 40)

--------- -

400

⋅⋅ ⋅=

SET

---------------------

DCR

-------------------------------------------------------------------------------

OCP

CORE

--------------------

FN6878.0

ISL6323A

5. Choose a capacitor value for the North Bridge RC filter. A

0.1µF capacitor is a recommended starting point.

6. Calculate the values for R

and R2 for North Bridge.

Equations 41 and 42 will allow for their computation.

------------------------------------ -

---------------------

DCR

------------------------------------ -

⋅

⋅=

(EQ. 41)

(EQ. 42)

CASE 3

Core

--------------------------

⋅=

DCR

Core

MAX

⋅

DCR

MAX

(EQ. 43)

In Case 3, the DC voltage across the North Bridge inductor at full load is equal to the DC voltage across a single phase of the Core regulator while at full load. Here, the full scale DC inductor voltages for both North Bridge and Core will be impressed across the ISEN pins without any gain. So, the R resistors for the Core and North Bridge inductor RC filters are left unpopulated and K = 1 for both regulators.

For this Case, it is recommended that the overcurrent trip point for the North Bridge regulator be equal to the overcurrent trip point for the Core regulator divided by the number of core phases.

1. Choose a capacitor value for the North Bridge RC filter. A

0.1µF capacitor is a recommended starting point.

2. Calculate the value for the North Bridge resistor R

--------------------------------------

DCRNBCNB⋅

(EQ. 44)

3. Choose a capacitor value for the Core RC filter. A 0.1µF capacitor is a recommended starting point.

5. Calculate the value for the Core resist or R

Core

------------------------------------------------

Core

SET

DCR

400

--------- -

K = 1

⋅

CoreCCore

DCR

CORE

-------------------------------------- -

⋅⋅=

100μ A

resistor using Equation 46:

SET

K⋅

⎛⎞

⎜⎟

OCP

CORE

⎝⎠

6. Calculate the value for the R

Where:

(EQ. 45)

VINV

–

------------------------------------------ -

CORE

⋅⋅

COREfSW

(EQ. 46)

--------------------

⋅+

7. Calculate the OCP trip point for the North Bridge regulator using Equation 47. If the OCP trip point is higher than desired, then the component values must be recalculated utilizing Case 1. If the OCP trip point is lower than desired, then the component values must be recalculated utilized Case 2.

OCP

100μ A

---------------------

DCR

–

⎛⎞

--------- -

⋅

SET

⎝⎠

400

INVNB

----------------------------------

⋅⋅

NBfSW

-----------

⋅+⋅⋅=

(EQ. 47)

CORE

NOTE: The values of R

must be greater than 20kΩ and

SET

less than 80kΩ. For all of the 3 cases, if the calculated value of R

is less than 20kΩ, then either the OCP trip point

SET

needs to be increased or the inductor must be changed to an inductor with higher DCR. If the R 80kΩ, then a value of R

that is less than 80kΩ must be

SET

resistor is greater than

SET

chosen and a resistor divider across both North Bridge and Core inductors must be set up with proper gain. This gain will represent the variable “K” in all equations. It is also very important that the R

resistor be tied between the RSET

SET

pin and the VCC pin of the ISL6323.

Inductor DCR Current Sensing Component Fine Tuning

UGATE(n)

MOSFET

DRIVER

ISL6323A INTERNAL CIRCUIT

SAMPLE

LGATE(n)

40kΩ

-----------------

SEN

SET

VC(s)

ISEN

2.4kΩ

FIGURE 20. DCR SENSING CONFIGURATION

Due to errors in the inductance and/or DCR it may be necessary to adjust the value of R constants correctly. The effects of time constant mismatch can be seen in the form of droop overshoot or undershoot during the initial load transient spike, as shown in Figure 21. Follow the steps below to ensure the R-C and inductor L/DCR time constants are matched accurately.

1. If the regulator is not utilizing droop, modify the circuit by placing the frequency set resistor between FS and Ground for the duration of this procedure.

DCR

INDUCTOR

and R2 to match the time

VL(s)

ISENn-

ISENn+

RSET

VC(s)

SET

OUT

VCC

FN6878.0

ISL6323A

2. Capture a transient event with the oscilloscope set to about L/DCR/2 (sec/div). For example, with L = 1µH and DCR = 1mΩ, set the oscilloscope to 500µs/div.

3. Record ΔV1 and ΔV2 as shown in Figure 21.

ΔV

2,NEW

OUT

TRAN

ΔI

, for the time

1,OLD

(EQ. 48)

ΔV

FIGURE 21. TIME CONSTANT MISMATCH BEHAVIOR

4. Select new values, R

1,NEW

and R constant resistors based on the original values, R and R

, using Equations 48 and 49.

2,OLD

V1Δ

1NEW,

1OLD,

---------- -

⋅=

V2Δ

function, the gain of the current signal, and the value of the compensation components, R

C2 (OPTIONAL)

FIGURE 22. COMPENSATION CONFIGURATION FOR

LOAD-LINE REGULATED ISL6323A CIRCUIT

and CC.

COMP

VSEN

ISL6323A

Since the system poles and zero are affected by the values of the components that are meant to compensate them, the solution to the system equation becomes fairly complicated. Fortunately, there is a simple approximation that comes very close to an optimal solution. Treating the system as though it were a voltage-mode regulator, by compensating the L-C poles and the ESR zero of the voltage mode approximation, yields a solution that is always stable with very close to ideal transient performance.

2NEW,

5. Replace R

V1Δ

2OLD,

and R2 with the new values and check to see

---------- -

⋅=

V2Δ

(EQ. 49)

that the error is corrected. Repeat the procedure if necessary.

Loadline Regulation Resistor

The loadline regulation resistor, labeled RFB in Figure 8, sets the desired loadline required for the application. Equation 50 can be used to calculate R

DROOP

MAX

DCR

---------------

K⋅⋅⋅

SET

Where R

--------------------------------------------------------------------- -

OUT

400

------------------------- -

--------- -

is the 2.4kΩ internal current sense resistor, KI

ISEN

(EQ. 50)

is defined in Equation 10 and K is defined in Equation 7. If no loadline regulation is required, FS resistor should be

tied between the FS pin and VCC. To choose the value for R

in this situation, please refer to “Compensation Without

Loadline Regulation” on page 30.

Compensation With Loadline Regulation

The load-line regulated converter behaves in a similar manner to a peak current mode controller because the two poles at the output filter L-C resonant frequency split with the introduction of current information into the control loop. The final location of these poles is determined by the system

Select a target bandwidth for the compensated system, f

The target bandwidth must be large enough to assure adequate transient performance, but smaller than 1/3 of the per-channel switching frequency. The values of the compensation components depend on the relationships of f to the L-C pole frequency and the ESR zero frequency. For each of the following three, there is a separate set of equations for the compensation components.

In Equation 51, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent series resistance of the bulk output filter capacitance; and V

P-P

is the peak-to-peak sawtooth signal amplitude as described in the Electrical Specifications on page 6.

Once selected, the compensation values in Equation 51 assure a stable converter with reasonable transient performance. In most cases, transient performance can be improved by making adjustments to R value of R

while observing the transient performance on an

. Slowly increase the

oscilloscope until no further improvement is noted. Normally, C

will not need adjustment. Keep the value of CC from

Equation 51 unless some performance issue is noted. The optional capacitor C

, is sometimes needed to bypass

noise away from the PWM comparator (see Figure 22). Keep a position available for C

, and be prepared to install a high-

frequency capacitor of between 22pF and 150pF in case any leading edge jitter problem is noted.

FN6878.0

ISL6323A

Case 1:

Case 2:

Case 3:

------------------------------- -f 2 π LC⋅⋅⋅

CRFB

------------------------------- 2 π LC⋅⋅⋅

CRFB

2 π f0V

----------------------------------------------------------

⋅=

0.66 V

0.66 VIN⋅

----------------------------------------------------=

⋅⋅ ⋅ ⋅

2 π V

PPRFBf0

-------------------------------------<≤

2 π C ESR⋅⋅ ⋅

2 π⋅()⋅

---------------------------------------------------------------- -

⋅=

0.66 V

-------------------------------------------------------------------------------------= 2 π⋅()

-------------------------------------> 2 π C ESR⋅⋅ ⋅

0.66 VIN⋅

VPPR

2 π f0V

----------------------------------------------

⋅=

0.66 V

0.66 VINESR C⋅⋅ ⋅

----------------------------------------------------------------= 2 π V

PPRFBf0

LC⋅⋅⋅ ⋅ ⋅

P-P

⋅

LC⋅⋅⋅

⋅

LC⋅⋅⋅ ⋅ ⋅

L⋅⋅ ⋅ ⋅

P-P

ESR⋅⋅

(EQ. 51)

L⋅⋅ ⋅ ⋅ ⋅

Compensation Without Loadline Regulation

The non load-line regulated converter is accurately modeled as a voltage-mode regulator with two poles at the L-C resonant frequency and a zero at the ESR frequency. A type III controller, as shown in Figure 23, provides the necessary compensation.

FIGURE 23. COMPENSATION CIRCUIT WITHOUT LOAD-LINE

REGULATION

COMP

ISL6323A

VSEN

CESR⋅

--------------------------------------------

R1R

⋅=

LC⋅ C ESR⋅–

--------------------------------------------= R

-----------------------------------------------------------------------------------------------------=

2 π⋅()

0.75 VIN⋅

f0f

⋅⋅ ⋅ ⋅ ⋅

LC⋅()RFBV

P-P

(EQ. 52)

⎛⎞

f0fHFLCR

P-P

------------------------------------------------------------------------------------------=

0.75 V

0.75 VIN2 π f

-----------------------------------------------------------------------------------------------------=

2 π⋅()

⋅⋅ ⋅⋅⋅

2π

⋅

⎝⎠

2 π f

f0f

⋅⋅ ⋅ ⋅ ⋅

LC⋅ 1–⋅⋅ ⋅()⋅⋅

LC⋅()RFBV

P-P

In the solutions to the compensation equations, there is a single degree of freedom. For the solutions presented in Equation 53, R

is selected arbitrarily. The remaining

compensation components are then selected according to Equation 53.

In Equation 53, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent-series resistance of the bulk output-filter capacitance; and V

is the peak-to-

peak sawtooth signal amplitude as described in Electrical Specifications on page 6.

Case 1:

Case 2:

------------------------------- -f 2 π LC⋅⋅⋅

CRFB

------------------------------- 2 π LC⋅⋅⋅

CRFB

2 π f0V

----------------------------------------------------------

⋅=

0.66 V

0.66 VIN⋅

----------------------------------------------------= 2 π V

⋅⋅ ⋅ ⋅

PPRFBf0

-------------------------------------<≤

2 π C ESR⋅⋅ ⋅

VPP2 π⋅()⋅

---------------------------------------------------------------- -

⋅=

0.66 V

-------------------------------------------------------------------------------------- -= 2 π⋅()

0.66 VIN⋅

P-PRFB

LC⋅⋅⋅ ⋅ ⋅

P-P

⋅

LC⋅⋅⋅

⋅

(EQ. 53)

LC⋅⋅⋅ ⋅ ⋅

The first step is to choose the desired bandwidth, f

, of the

compensated system. Choose a frequency high enough to assure adequate transient performance but not higher than 1/3 of the switching frequency. The type-III compensator has an extra high-frequency pole, f

. This pole can be used for

added noise rejection or to assure adequate attenuation at the error-amplifier high-order pole and zero frequencies. A good general rule is to choose f desired. Choosing f

=10f0, but it can be higher if

to be lower than 10f0 can cause problems with too much phase shift b elow the system bandwidth,

Case 3:

CRFB

-------------------------------------> 2 π C ESR⋅⋅ ⋅

0.66 VINESR C⋅⋅ ⋅

----------------------------------------------------------------- -= 2 π V

2 π f0V

----------------------------------------------

⋅=

0.66 V

P-PRFBf0

P-P

ESR⋅⋅

L⋅⋅ ⋅ ⋅

L⋅⋅ ⋅ ⋅ ⋅

FN6878.0

ISL6323A

Output Filter Design

The output inductors and the output capacitor bank together to form a low-pass filter responsible for smoothing the pulsating voltage at the phase nodes. The output filter also must provide the transient energy until the regulator can respond. Because it has a low bandwidth compared to the switching frequency, the output filter limits the system transient response. The output capacitors must supply or sink load current while the current in the output inductors increases or decreases to meet the demand.

In high-speed converters, the output capacitor bank is usually the most costly (and often the largest) part of the circuit. Output filter design begins with minimizing the cost of this part of the circuit. The critical load parameters in choosing the output capacitors are the maximum size of the load step, ΔI, the load-current slew rate, di/dt, and the maximum allowable output-voltage deviation under transient loading, ΔV Capacitors are characterized according to their capacita nce, ESR, and ESL (equivalent series inductance).

At the beginning of the load transient, the output capacitors supply all of the transient current. The output voltage will initially deviate by an amount approximated by the voltage drop across the ESL. As the load current increases, the voltage drop across the ESR increases linearly until the load current reaches its final value. The capacitors selected must have sufficiently low ESL and ESR so that the total output-voltage deviation is less than the allowable maximum. Neglecting the contribution of inductor current and regulator response, the output voltage initially deviates by an amount as shown in Equation 54:

ΔVESL

-----

⋅ ESR ΔI⋅+≈

The filter capacitor must have sufficiently low ESL and ESR so that ΔV < ΔV

MAX

Most capacitor solutions rely on a mixture of high frequency capacitors with relatively low capacitance in combination with bulk capacitors having high capacitance but limited high-frequency performance. Minimizing the ESL of the high-frequency capacitors allows them to support the output voltage as the current increases. Minimizing the ESR of the bulk capacitors allows them to supply the increased current with less output voltage deviation.

MAX

(EQ. 54)

⎛⎞

NV⋅

–

⎝⎠

------------------------------------------------------------------- -

⋅≥

ESR

⋅⋅

SVINVP-P MAX()

OUT

⋅

OUT

(EQ. 55)

Since the capacitors are supplying a decreasing portion of the load current while the regulator recovers from the transient, the capacitor voltage becomes slightly depleted. The output inductors must be capable of assuming the entire load current before the output voltage decreases more than ΔV

. This places an upper limit on inductance.

MAX

Equation 56 gives the upper limit on L for the cases when the trailing edge of the current transient causes a greater output-voltage deviation than the leading edge. Equation 57 addresses the leading edge. Normally, the trailing edge dictates the selection of L because duty cycles are usually less than 50%. Nevertheless, both inequalities should be evaluated, and L should be selected based on the lower of the two results. In each equation, L is the per-channel inductance, C is the total output capacitance, and N is the number of active channels.

2NCV

⋅⋅⋅

---------------------------------

1.25

-----------------------------

()

ΔI

NC⋅⋅

()

ΔI

ΔV

⋅⋅≤

ΔV

MAX

ΔIESR⋅()–⋅≤

MAX

ΔI ESR⋅()– VINVO–

⎛⎞ ⎝⎠

(EQ. 56)

(EQ. 57)

Switching Frequency

There are a number of variables to consider when choosing the switching frequency, as there are considerable effects on the upper MOSFET loss calculation. These effects are outlined in “MOSFETs” on page 25, and they establish the upper limit for the switching frequency. The lower limit is established by the requirement for fast transient response and small output-voltage ripple as outlined in “Output Filter Design” on page 31. Choose the lowest switching frequency that allows the regulator to meet the transient-response requirements.

Switching frequency is determined by the selection of the frequency-setting resistor, R are provided to assist in selecting the correct value for R

[]

RT10

10.61 1.035 fS()log⋅()–

. Figure 24 and Equation 58

(EQ. 58)

The ESR of the bulk capacitors also creates the majority of the output-voltage ripple. As the bulk capacitors sink and source the inductor AC ripple current (see “Interleaving” on page 11 and Equation 3), a voltage develops across the bulk capacitor ESR equal to I

(ESR). Thus, once the output

C(P-P)

capacitors are selected, the maximum allowable ripple voltage, V

P-P(MAX)

, determines the lower limit on the

inductance.

FN6878.0

ISL6323A

100

(kΩ)

10k 100k 1M 10M

SWITCHING FREQUENCY (Hz)

FIGURE 24. RT vs SWITCHING FREQUENCY

Input Capacitor Selection

The input capacitors are responsible for sourcing the AC component of the input current flowing into the upper MOSFETs. Their RMS current capacity must be sufficient to handle the AC component of the current drawn by the upper MOSFETs which is related to duty cycle and the number of active phases.

0.3 I

= 0

L(P-P)

= 0.25 I

)

RMS/

0.2

0.1

L(P-P)

= 0.5 I = 0.75 I

phase and two-phase designs respectively. Use the same approach for selecting the bulk capacitor type and number.

0.3

= 0

(P-P)

)

(P-P)

= 0.25 I

DUTY CYCLE (V

RMS/

0.2

0.1

INPUT-CAPACITOR CURRENT (I

00.4 1.00.2 0.6 0.8

= 0.5 I

(P-P)

= 0.75 I

)

IN/VO

FIGURE 26. NORMALIZED INPUT-CAPACITOR RMS

CURRENT FOR 3-PHASE CONVERTER

Low capacitance, high-frequency ceramic cap acitors are needed in addition to the input bulk capacitors to suppress leading and falling edge voltage spike s. The spikes result from the high current slew rate produced by the upper MOSFET turn on and off. Select low ESL ceramic capacitors and place one as close as possible to each upper MOSFET drain to minimize board parasitics and maximize suppression.

0.3

)

RMS/

0.2

INPUT-CAPACITOR CURRENT (I

00.4 1.00.2 0.6 0.8 DUTY CYCLE (V

O/VIN

)

FIGURE 25. NORMALIZED INPUT- CAP ACITOR RMS CURRENT

vs DUTY CYCLE FOR 4-PHASE CONVERTER

For a four-phase design, use Figure 25 to determine the input-capacitor RMS current requirement set by the duty cycle, maximum sustained output current (I of the peak-to-peak inductor current (I

), and the ratio

) to IO. Select a

L,PP

bulk capacitor with a ripple current rating which will minimize the total number of input capacitors required to support the RMS current calculated.

The voltage rating of the capacitors should also be at least 1.25 times greater than the maximum input voltage. Figures 26 and 27 provide the same input RMS current information for three-

0.1

= 0

(P-P)

= 0.5 I

(P-P)

INPUT-CAPACITOR CURRENT (I

(P-P)

00.4 1.00.2 0.6 0.8

= 0.75 I

DUTY CYCLE (V

IN/VO

)

FIGURE 27. NORMALIZED INPUT-CAPACITOR RMS

CURRENT FOR 2-PHASE CONVERTER

Layout Considerations

MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. These voltage spikes can degrade efficiency, radiate noise into the circuit and lead to device overvoltage stress. Careful component selection, layout, and placement minimizes these voltage spikes. Consider, as an example, the turnoff transition of the upper PWM MOSFET. Prior to turnoff, the upper MOSFET

FN6878.0

ISL6323A

was carrying channel current. During the turnoff, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET. Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes.

There are two sets of critical components in a DC/DC converter using a ISL6323A controller. The power components are the most critical because they switch large amounts of energy. Next are small signal components that connect to sensitive nodes or supply critical bypassing current and signal coupling.

The power components should be placed first, which include the MOSFET s, input and output capacitors, and the inductors. It is important to have a symmetrical layout for each power train, preferably with the controller located equidistant from each. Symmetrical layout allows heat to be dissipated equally across all power trains. Equidistant placement of the controller to the CORE and NB power trains it controls through the integrated drivers helps keep the gate drive traces equally short, resulting in equal trace impedances and similar drive capability of all sets of MOSFETs.

When placing the MOSFETs try to keep the source of the upper FETs and the drain of the lower FETs as close as thermally possible. Input high-frequency capacitors, C should be placed close to the drain of the upper FETs and the source of the lower FETs. Input bulk cap acitors, CBULK, case size typically limits following the same rule as the high-frequency input capacitors. Place the input bulk capacitors as close to the drain of the upper FETs as possible and minimize the distance to the source of the lower FETs.

Locate the output inductors and output capacitors between the MOSFETs and the load. The high-frequency output decoupling capacitors (ceramic) should be placed as close as practicable to the decoupling target, making use of the shortest connection paths to any internal planes, such as vias to GND next or on the capacitor solder pad.

The critical small components include the bypass capacitors (C surrounding the controller including the feedback network and current sense components. Locate the VCC/PVCC bypass capacitors as close to the ISL6323A as possible. It is especially important to locate the components associated with the feedback circuit close to their respective controller pins, since they belong to a high-impedance circuit loop, sensitive to EMI pick-up.

) for VCC and PVCC, and many of the components

FILTER

ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. Keep the metal runs from the PHASE terminal to output inductors short. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring.

Routing UGATE, LGATE, and PHASE Traces

Great attention should be paid to routing the UGATE, LGATE, and PHASE traces since they drive the power train MOSFETs using short, high current pulses. It is important to size them as large and as short as possible to reduce their overall impedance and inductance. They should be sized to carry at least one ampere of current (0.02” t o 0 . 0 5 ” ) . G o i n g between layers with vias should also be avoided, but if so, use two vias for interconnection when possible.

Extra care should be given to the LGATE traces in particular since keeping their impedance and inductance lo w helps to significantly reduce the possibility of shoot-through. It is also important to route each channels UGATE and PHASE traces in as close proximity as possible to reduce their inductances.

Current Sense Component Placement and Trace Routing

One of the most critical aspects of the ISL6323A regulator layout is the placement of the inductor DCR current sense components and traces. The R-C current sense components must be placed as close to their respective ISEN+ and ISEN- pins on the ISL6323A as possible.

The sense traces that connect the R-C sense components to each side of the output inductors should be routed on the bottom of the board, away from the noisy switching components located on the top of the board. These traces should be routed side by side, and they should be very thin traces. It’s important to route these traces as far away from any other noisy traces or planes as possible. These traces should pick up as little noise as possible.

Thermal Management

For maximum thermal performance in high current, high switching frequency applications, connecting the thermal GND pad of the ISL6323A to the ground plane with multiple vias is recommended. This heat spreading allows the part to achieve its full thermal potential. It is also recommended that the controller be placed in a direct path of airflow if possible to help thermally manage the part.

A multi-layer printed circuit board is recommended. Figure 28 shows the connections of the critical components for the converter. Note that capacitors C represent numerous physical capacitors. Dedicate one solid layer, usually the one underneath the component side of the board, for a ground plane and make all critical component

and C

could each

OUT

FN6878.0

FILTER

ISL6323A

FILTER

BOOT

+12V

3_2

1_1

1_2

V_CORE

BULK

CPU

LOAD

2_2

2_1

4_2

FB COMP ISEN3+ ISEN3PWM3

APA

DVC

+5V

VCC

OFS

NC NC

RSET

VFIXEN SEL SVD SVC VID4 VID5 PWROK

+5V

SET

VDDPWRGD GND

VSEN

BOOT1

UGATE1

PHASE1

LGATE1

ISEN1-

ISEN1+

PVCC1_2

BOOT2

UGATE2

PHASE2

LGATE2

ISEN2-

ISEN2+

RGND

ISEN4+

ISEN4-

BOOT

+12V

BOOT

3_1

BOOT1 UGATE1 PHASE1

LGATE1

PWM1

PGND

VCC

PVCC

GND

+12V

FILTER

ISL6614

+12V

BOOT

BOOT2

UGATE2 PHASE2

4_1

PWM2

LGATE2

PWM4

OFF ON

+12V

2_NB

EN1

EN2

COMP_NB

FB_NB

C_NB

ISL6323A

PVCC_NB

BOOT_NB

UGATE_NB

PHASE_NB

LGATE_NB

ISEN_NB-

ISEN_NB+

FB_NB

FILTER

BOOT_NB

+12V

1_NB

2_NB

BULK

LOAD

V_NB

KEY

HEAVY TRACE ON CIRCUIT PLANE LAYER

ISLAND ON POWER PLANE LAYER

ISLAND ON CIRCUIT PLANE LAYER

VIA CONNECTION TO GROUND PLANE

RED COMPONENTS: LOCATE CLOSE TO IC TO MINIMIZE CONNECTION PATH

BLUE COMPONENTS: LOCATE NEAR LOAD (MINIMIZE CONNECTION PATH)

MAGENTA COMPONENTS: LOCATE CLOSE TO SWITCHING TRANSISTORS (MINIMIZE CONNECTION PATH)

FIGURE 28. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS

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Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality

Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implicat ion or oth erwise u nde r any p a tent or p at ent r ights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see www.intersil.com

FN6878.0

Package Outline Drawing

L48.7x7

48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 4, 10/06

7.00

PIN 1

INDEX AREA

ISL6323A

44X

5.5

0.50 48

PIN #1 INDEX AREA

(4X) 0.15

( 6 . 80 TYP )

( 4 . 30 )

TOP VIEW

TYPICAL RECOMMENDED LAND PATTERN

7.00

0 . 90 ± 0 . 1

( 44X 0 . 5 )

( 48X 0 . 23 )

( 48X 0 . 60 )

48X 0 . 40± 0 . 1

BOTTOM VIEW

SIDE VIEW

0 . 2 REF

0 . 00 MIN. 0 . 05 MAX.

DETAIL "X"

BASE PLANE

4. 30 ± 0 . 15

M0.10 C AB

0.23 +0.07 / -0.05

SEE DETAIL "X"

0.10

SEATING PLANE

C0.08

NOTES:

Dimensions are in millimeters.1. Dimensions in ( ) for Reference Only.

Dimensioning and tolerancing conform to AMSE Y14.5m-1994.

Unless otherwise specified, tolerance : Decimal ± 0.05

Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip.

Tiebar shown (if present) is a non-functional feature.

The configuration of the pin #1 identifier is optional, but must be

6. located within the zone indicated. The pin #1 indentifier may be

either a mold or mark feature.

FN6878.0

intersil ISL6323A DATA SHEET

Specifications and Main Features

Frequently Asked Questions

User Manual