intersil HSP50210 DATA SHEET

®

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HSP50210

Data Sheet July 2, 2008

Digital Costas Loop

The Digital Costas Loop (DCL) performs many of the
baseband processing tasks required for the demodulation of
BPSK, QPSK, 8-PSK, OQPSK, FSK, AM and FM
waveforms. These tasks include matched filtering, carrier
tracking, symbol synchronization, AGC, and soft decision
slicing. The DCL is designed for use with the HSP50110
Digital Quadrature Tuner to provide a two chip solution for
digital down conversion and demodulation.

The DCL processes the In-phase (I) and Quadrature (Q)
components of a baseband signal which have been digitized
to 10 bits. As shown in the block diagram, the main signal
path consists of a complex multiplier, selectable matched
filters, gain multipliers, cartesian-to-polar converter, and soft
decision slicer. The complex multiplier mixes the I and Q
inputs with the output of a quadrature NCO. Following the
mix function, selectable matched filters are provided, which
perform integrate and dump or root raised cosine filtering
(α ~ 0.40). The matched filter output is routed to the slicer,
which generates 3-bit soft decisions, and to the cartesian-to­polar converter, which generates the magnitude and phase
terms required by the AGC and Carrier Tracking Loops.

The PLL system solution is completed by the HSP50210
error detectors and second order Loop Filters that provide
carrier tracking and symbol synchronization signals. In
applications where the DCL is used with the HSP50110,
these control loops are closed through a serial interface
between the two parts. To maintain the demodulator
performance with varying signal power and SNR, an internal
AGC loop is provided to establish an optimal signal level at
the input to the slicer and to the cartesian-to-polar converter.

FN3652.5

Features

• Clock Rates Up to 52MHz

• Selectable Matched Filtering with Root Raised Cosine or
Integrate and Dump Filter

• Second Order Carrier and Symbol Tracking Loop Filters

• Automatic Gain Control (AGC)

• Discriminator for FM/FSK Detection and Discriminator
Aided Acquisition

• Swept Acquisition with Programmable Limits

• Lock Detector

• Data Quality and Signal Level Measurements

• Cartesian-to-Polar Converter

• 8-Bit Microprocessor Control - Status Interface

• Designed to Work With the HSP50110 Digital Quadrature
Tuner

• 84 Lead PLCC

• Pb-Free Available (R oH S compliant)

Applications

• Satellite Receivers and Modems

• BPSK, QPSK, 8-PSK, OQPSK, FSK, AM and FM
Demodulators

• Digital Carrier Tracking

• Related Products: HSP50110 Digital Quadrature Tuner,
D/A Converters HI5721, HI5731, HI5741

• HSP50110/210EVAL Digital Demod Evaluation Board

Block Diagram

CARRIER

TRACK

CONTROL

HI/LO

I SER OR

(9-0)

I

IN

SERCLK

OR CLK

Q SER OR

Q

(9-0)

IN

SYMBOL

TRACK

CONTROL

CONTROL/

STATUS

BUS

(COF)

LEVEL

DETECT

(SOF)

COS

10

10

13

NCO

SIN

SYMBOL

TRACKING

LOOP FILTER

1

CARRIER ACQ/TRK

LOOP FILTER

I

RRC

FILTER

Q

RRC

FILTER

CONTROL

INTERFACE

CARRIER PHASE

ERROR DETECT

LOOP

FILTER

INTEGRATE/

DUMP

INTEGRATE/

DUMP

SYMBOL

PHASE
ERROR

DETECT

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

1-888-INTERSIL or 1-888-468-3774

LEVEL

DETECT

8

CARTESIAN

8

| Intersil (and design) is a registered trademark of Intersil Americas Inc.

All other trademarks mentioned are the property of their respective owners.

LOCK

DETECT

MAGNITUDE

8

PHASE

TO

POLAR

Copyright Intersil Americas Inc. 2000, 2008. All Rights Reserved

8

SLICER

3

3

Q

I

DATA PATH MULTIPLEXER

LKINT

THRESH
A

OUT(9-0)

10

10

B
OUT(9-0)

SMBLCLK

OEA

OEB

Pinout

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IIN6

IIN7

IIN8

IIN9

SERCLK

GND

SSYNC

HSP50210

HSP50210

(84 LD PLCC)

TOP VIEW

HI/LO

ISER

QSER

VCC

THRESH

SLOCLK

OEA

AOUT8

AOUT9

GND

AOUT7

AOUT5

AOUT6

AOUT4

757677787980818283841234567891011

IIN5
IIN4
IIN3
IIN2

GND

IIN1
IIN0

SYNC

QIN9
QIN8
QIN7

QIN6
QIN5
QIN4

VCC
QIN3
QIN2
QIN1
QIN0

SOFSYNC

SOF

12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32

C1

C2

C3

C4

C5

C6

C7

A0

A1

WR

COF

COFSYNC

GND

VCC

A2

RD

C0

FZ-ST

FZ-CT

LKINT

535251504948474645444342414039383736353433

GND

74
73
72
71
70
69
68
67
66
65

64
63
62
61
60
59
58
57
56
55
54

AOUT3
AOUT2
AOUT1
AOUT0
SMBLCLK
VCC
CLK
GND
BOUT9
BOUT8

BOUT7
BOUT6
BOUT5
GND
BOUT4
BOUT3
BOUT2
BOUT1
BOUT0

OEB

VCC

Ordering Information

PART NUMBER PART MARKING TEMP. RANGE (°C) PACKAGE PKG. DWG. #

HSP50210JC-52 HSP50210JC-52 0 to +70 84 Ld PLCC N84.1.15
HSP50210JC-52Z (Note) HSP50210JC-52Z 0 to +70 84 Ld PLCC (Pb-free) N84.1.15
HSP50210JI-52 HSP50210JI-52 -40 to +85 84 Ld PLCC N84.1.15
HSP50210JI-52Z (Note) HSP50210JI-52Z -40 to +85 84 Ld PLCC (Pb-free) N84.1.15

NOTE: These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100%
matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.

2

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

Pin Description

NAME TYPE DESCRIPTION

VCC - +5V Power Supply.

GND - Ground.

IIN9-0 I In-Phase Parallel Input. Data may be two’s complement or offset binary format (see Table 15). These inputs are

sampled by CLK when the SYNC

QIN9-0 I Quadrature Parallel Input. Data may be two’s complement or offset binary format (see Table 15). These inputs are

sampled by CLK when the SYNC

SYNC

COF O Carrier Offset Frequency. The frequency term generated by the Carrier Tracking Loop Filter is output serially via this

COFSYNC O Carrier Offset Frequency Sync. This signal is asserted one CLK or SLOCLK cycle before the MSB of the serial data

SOF O Sampler Offset Frequency. Sample frequency correction term generated by the Symbol Tracking Loop Filter is

SOFSYNC O Sampler Offset Frequency Sync. This signal is asserted one CLK or SLOCLK cycle before the MSB of the serial data

A2-0 I Address Bus. The address on these pins specify a target register for reading or writing (see “Microprocessor

C7-0 I/O Microprocessor Interface Data Bus. This bi-directional bus is used for reading and writing to the processor interface.

WR

RD

I Data Sync. When SYNC is asserted “Low”, data on IIN9-0 and QIN9-0 is clocked into the processing pipeline by the

rising edge of CLK.

pin. The new offset frequency is shifted out MSB first by CLK or SLOCLK starting with the clock cycle after the
assertion of COFSYNC.

word. (Programmable Polarity, see Table 42 on page 42, Bit 11).

output serially via this pin. The frequency word is shifted out MSB first by CLK or SLOCLK starting with the clock
cycle after assertion of SOFSYNC.

word. (Programmable Polarity, see Table 42 on page 42, Bit 12).

Interface” on page 27). A0 is the LSB.

These are the data I/O pins for the processor interface. C0 is the LSB.
I Write. This is the write strobe for the processor interface (see “Microprocessor Interface” on page 27).
I Read. This is the read enable for the processor interface (see “Microprocessor Interface” on page 27).

signal is active Low. IIN9 is the MSB. See “Input Controller” on page 6.

signal is active Low. QIN9 is t heMSB. “Input Controller” on page 6.

FZ_ST I Freeze Symbol Tracking Loop. Asserting this pin “high” zeroes the sampling error into the Symbol Tracking Loop

FZ_CT I Freeze Carrier Tracking Loop. Asserting this pin “high” zeroes the carrier Phase Error input to the Carrier Tracking

LKINT O Lock Detect Interrupt. This pin is asserted “high” for at least 4 CLK cycles when the Lock Detector Integration cycle

THRESH

SLOCLK O Slow Clock. Optional serial clock used for outputting data from the Carrier and Symbol Tracking Loop Filters. The

ISER I In-Phase Serial Input. Serial data input for In-Phase Data. Data on this pin is shifted in MSB first and is synchronous

QSER I Quadrature Serial Input. Serial data input for Quadrature Data. Data on this pin is shifted in MSB first and is

SSYNC I Serial Word Sync. This input is asserted “high” one CLK before the first data bit of the serial word (see Figure 2).
SERCLK I Serial Clock. May be asynchronous to other clocks. Used to clock in serial data (see “Input Controller” on page 6).
AOUT9 -0 O A Output. Data on this output depend on the configuration of Output Selector. AOUT9 is the MSB (see Table 43 on

BOUT9-0 O B Output. Data on this output depend on the configuration of Output Selector. BOUT9 is the MSB (see T able 43 page 44).

SMBLCLK O Symbol Clock. 50% duty cycle clock aligned with soft bit decisions (see Figure 19).

Filter (see “Symbol Tracking Loop Filter” on page 17).

Loop Filter.

is finished (see “Lock Detector” on page 23). Used as an interrupt for a pr ocessor. The Lock Detect Interrupt may
be asserted “high” longer than 4 CLK cycles, depending on the Lock Detector mode.

O Threshold Exceeded. This output is asserted “low” when the magnitude out of the Cartesian to Polar converter

exceeds the programmable Power Detect Threshold (see Table 16 on page33 and “AGC” on page 10).

clock is programmable and has a 50% duty cycle. Note: Not used when the HSP50110 is used with the HSP50210
(see Table 42 page 42).

to SERCLK (see “Input Controller” on page 6).

synchronous to SERCLK (see “Input Controller” on page 6).

page 44).

3

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

Pin Description (Continued)

NAME TYPE DESCRIPTION

OEA I A Output Enable. This pin is the three-state control pin for the AOUT9-0. When OEA is high, the AOUT9-0 is high

impedance.

OEB

HI/LO

CLK I System Clock. Asynchronous to the processor interface and serial inputs.

I B Output Enable. This pin is the three-state control pin for the BOUT9-BOUT0. When OEB is high, the AOUT9-0 is

high impedance.

0 HI/LO. The output of the Input Level Detector is provided on this pin (see “Input Level Detector” o n page 6). This

signal can be externally averaged and used to control the gain of an amplifier to close an AGC loop around the A/D
converter. This type of AGC sets the level based on the median value on the input.

4

FN3652.5

July 2, 2008

AGC

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LOOP

HI/LO

SYNC

5

IIN9-0

QIN9-0

SSYNC

SERCLK

ISER

QSER

LEVEL

DETECT

INPUT CONTROLLER

SYNTHESIZER/

I

Q

MIXER

COS

NCO

MATCHED FILTERING

M
U
X

M
U
X

SIN

RRC

RRC

M
U
X

M
U
X

FILTER

I AND D

I AND D

GAIN ERROR

DETECT

M
U
X

M
U
X

CARTESIAN

TO

POLAR

I2+Q

Q

TAN-1( )

SLICER

2

I

THRESH

SMBLCLK

SYMBOL TRACKING

SOFSYNC

SOF

COFSYNC

COF

SLOCLK

C7-0

A2-0

CLK

FRZ_ST
FRZ_CT

July 2, 2008

FN3652.5

8

WR

RD

SERIAL

OUTPUT

FORMATTER

MICROPROCESSOR

INTERFACE

FROM
LOCK

DETECTOR

ACQUISITION

CONTROL

FIGURE 1. FUNCTIONAL BLOCK DIAGRAM OF THE HSP50210

2ND ORDER LOOP

FILTER

CARRIER TRACKING

2ND ORDER LOOP

FILTER

SYMBOL PHASE
ERROR DETECT

CARRIER PHASE

ERROR DETECT

DISCRIMINATOR

FREQUENCY

ERROR DETECT

LOCK

DETECT

AOUT9-0

BOUT9-0

OEA

OEB

d

dt

LKINT

HSP50210

HSP50210

www.BDTIC.com/Intersil

Functional Description

The HSP50210 Digital Costas Loop (DCL) contains most of
the baseband processing functions needed to implement a
digital Costas Loop Demodulator. These functions include
LO generation/mixing, matched filtering, AGC, carrier phase
and frequency error detection, timing error detection, carrier
loop filtering, bit sync loop filtering, lock detection,
acquisition/tracking control, and soft decision slicing for
forward error correction algorithms. While the DCL is
designed to work with the HSP50110 Digital Quadrature
Tuner (DQT) as a variable rate PSK demodulator for satellite
demodulation, functions on the chip are common to many
communications receivers.

The DCL provides the processing blocks for the three
tracking loops commonly found in a data demodulator: the
Automatic Gain Control (AGC) loop, the Carrier Tracking
Loop, and a Symbol Tracking Loop. The AGC loop adjusts
for input signal power variations caused by path loss or
signal-to-noise variations. The carrier tracking loop removes
the frequency and phase uncertainties in the carrier due to
oscillator inaccuracies and doppler. The symbol tracking
loop removes the frequency and phase uncertainties in the
data and generates a recovered clock synchronous with the
received data. Each loop consists of an error detector , a loop
filter, and a frequency or gain adjustment/control. The AGC
loop is internal to the DCL, while the symbol and carrier
tracking loops are closed external to the DCL. When the
DCL is used together with the HSP50110, the tracking loops
are closed around the baseband filtering to center the signal
in the filter bandwidth. In addition, the AGC function is
divided between the two chips with the HSP50110 providing
the coarse AGC, and the HSP50210 providing the fine or
final AGC.

A top level block diagram of the HSP50210 is shown in
Figure 1. This diagram shows the major blocks and the
multiplexers used to reconfigure the data path for various
architectures.

Input Controller

In-Phase (I) and Quadrature (Q) data enters the part through
the Input Controller. The 10-bit data enters in either serial or
parallel fashion using either two’s complement or offset
binary format. The input mode and binary format is set in the
Data Path Configuration Control Register, bits 14 and 15
(see Table 15 on page 32).

If Parallel Input mode is selected, I and Q data are clocked
into the part through IIN0-9 and QIN0-9 respectively. Data
enters the processing pipeline when the input enable
(SYNC

) is sampled “low” by the processing clock (CLK). The
enable signal is pipelined with the data to the various
processing elements to minimize pipeline delay where
possible. As a result, the pipeline delay through the AGC,
Carrier Tracking, and Symbol Tracking Loop Filters is
measured in CLKs; not input data samples.

If serial input mode is selected, the I and Q data enters via
the ISER and QSER pins using SERCLK and SSYNC. The
beginning of a serial word is designated by asserting
SSYNC ‘high’ one SERCLK prior to the first data bit, as
shown in Figure 2. On the following SERCLKs, data is
shifted into the register until all 10 bits have been input. Data
shifting is then disabled and the contents of the register are
held until the next assertion of SSYNC. The assertion of a
SSYNC transfers data into the processing pipeline, and the
Shift Register is enabled to accept new data on the following
SERCLK. When data is transferred to the processing
pipeline by SSYNC, a processing enable is generated which
follows the data through the pipeline. This enable allows the
delay through processing elements (like the loop filters) to be
minimized since their pipeline delay is expressed in CLKs
not SSYNC periods. Note: SSYNC should not be
asserted for more than one SERCLK cycle.

SERCLK

SSYNC

ISER/QSER

NOTE: Data must be loaded MSB first.

FIGURE 2. SERIAL INPUT TIMING FOR ISER AND QSER INPUTS

MSB

SSYNC LEADS 1st DATA BIT

MSB

Input Level Detector

The Input Level Detector generates a one-bit error signal for
an external IF AGC filter and amplifier. The error signal is
generated by comparing the magnitude of the input samples
to a user programmable threshold. The HI/LO pin is then
driven “high” or “low” depending on the relationship of its
magnitude to the threshold. The sense of the HI/LO pin is
programmable so that a magnitude exceeding the threshold
can either be represented as a “high” or “low” logic state.
The Input Level Detector (HI/LO output) threshold and the
sense are set by the Data Path Configuration Control
Register bits 16 to 23 and 13 (see Table 15 page 32).

Note: The Inpu t Level Detector is typically not used in
applications which use the HSP50210 with the
HSP50110.

The high/low outputs can be integrated by an external loop
filter to close an AGC loop. Using this method, the gain of
the loop forces the median magnitude of the input samples
to the threshold. When the magnitude of half of the samples
is above the threshold (and half is below), the error signal is
integrated to zero by the loop filter. The magnitude of the
complex input is estimated using Equation 1:

Mag (I, Q) I 0.375 Q if I Q and>×+=
Mag (I, Q) Q 0.375 I if Q I>×+=

(EQ. 1)

6

FN3652.5

July 2, 2008

REGISTER ENABLE RATE

www.BDTIC.com/Intersil

@ = SYNC RATE

= TWICE SYMBOL RATE

*

! = SYMBOL RATE

BLANK = CLK RATE

MID AND END

SYMBOL SAMPLES

TO SYMBOL TRACKING

I

MID

I

END

Q

MID

Q

END

D

HI/LO

MATCHED FILTERING

7

REG
REG

DETECT

LEVEL

IIN9-0QIN9-0

NCO MIXER

COMPLEX

MULTIPLY

BYPASS

MIXER

BYPASS

R
E
G

R
E

M

G

U
X

R

R

E

E

G

G
R

R

E

E

G

G

@

@

RRC

15 TAP RRC

15 TAP RRC

M
U

R

R

X

E

E

G

G
R

R

E

E

G

G

R

R

L

E

E

I

G

G

M

R

R

I

E

E

T

G

G

DUMP

+

DUMP

+

M
U
X

DATA DE-SKEW

OQPSK

“0”

R

R

E

E

G

G

@

*

“0”

M
U

X

R
E
G

M
U
X

@

SIN

COS
REG REG
REG REG

SIN/COS

ROM

REG

+

REG

CF
REGISTER

ROOT RAISED COSINE

(RRC)

AGC LOOP FILTER

L

I

R

M

E

I

G

UPPER

GAIN
LIMIT

T

LOWER
GAIN
LIMIT

@ OR !

INTEGRATE AND DUMP

R
E

+

G

LOOP GAIN
EXPONENT

FALSE LOCK

COMPARE

S
H

I
F
T

*

TWO SAMPLE

SUMMER

S
H

F
T

R
E

I

G

*

S

R

H

E

I

G

F
T

LOOP GAIN

MANTISSA

REG

R
E
G

BYPASS

I AND D

+

M
U

X

+

HOLD AGC

M

U
X

E
M
U
X

D
E
M
U
X

M
U
X

AGC ERROR DETECT

“0”

AGC THRESHOLD

GAIN
ERROR

POWER
THRESHOLD

CARTESIAN TO

POLAR

2

DELAY

I2+Q

Q

-1

TAN

( )

DELAY

I

PHASE OUT AT

MAG OUT AT

-

+

COMPARE

SOFT

DECISION

SLICER

TEST

M
U
X

5

REG

5

REG

@ OR

@ OR !

R
E
G

R
E
G

R
E
G

R
E
G

O

R

U

E

**

T

G

P

R

U

! OR ! OR

E

T

G

S
E
L
E

8

C
T

8

AOUT9-0 BOUT9-0

R
E
G

R
E
G

HSP50210

*

TO
CARRIER
TRACKING
AND
DISCRIMINATOR

THRESH

R
E
G

FROM CARRIER TRACKING

LOOP FILTER

July 2, 2008

FN3652.5

FIGURE 3. MAIN DATA PATH

HSP50210

www.BDTIC.com/Intersil

NCO/Mixer

The NCO/Mixer performs a complex multiply between the
baseband input and the output of a quadrature NCO
(Numerically Controlled Oscillator). When the HSP50210
(DQT) is used with the HSP50110 (DCL), the NCO/Mixer
shortens the Carrier Tracking Loop (i.e., minimizes pipeline
delay around the loop) while providing wide loop
bandwidths. This becomes important when operating at
symbol rates near the maximum range of the part.

There are three configurations possible for closing the
Carrier Tracking Loop when the DQT and the DCL are used
together. The first configuration utilizes the NCO on the DQT
and bypasses the NCO in the DCL. The Data Path
Configuration Control Register (see Table 15 on page 32),
Bit 10, and Carrier Loop Filter Control Register #1
(see Table 21 on page 34), Bit 6, are used to bypass the
DCL NCO/Mixer and route the Loop filter outputs,
respectively. The DQT provides maximum flexibility in NCO
control with respect to frequency and phase offsets.

The second configuration feeds the lead Carrier Loop filter
term to the DCL NCO/Mixer, and the lag Loop filter Term to
the DQT NCO. This reduces the loop transport delay while
maintaining wide loop bandwidths and reasonable loop
damping factors. This configuration is especially useful in
SATCOM applications with medium to high symbol rates.
The Carrier Loop Filter Control Register #1, Bit 5 is where
the lead/lag destination is set.

The final configuration feeds both the lead and lag Carrier
Loop Filter terms back to the DCL NCO/Mixer. This provides
the shortest transport delay. The DCL NCO/Mixer provides
only for frequency/phase control from the Carrier Loop filter.
The center frequency of this NCO/Mixer is set to the average
of the Upper and Lower Carrier Loop Limits programmable
parameters. These parameters are set in the two control
registers bearing their names (see Tables 23 and 24 on
page 35).

The NCO/Mixer uses a complex multiplier to multiply the
baseband input by the output of a quadrature NCO. This
operation is represented by Equations 2 and 3:

I

OUTIIN

Q

OUTIIN

Equation 3 illustrates how the complex multiplier implicitly
performs the summing function when the DCL is configured
as a modulator. The quadrature outputs of the NCO are
generated by driving a sine/cosine look-up table with the
output of a phase accumulator, as shown in Figure 3 on
page 7. Each time the phase accumulator is clocked, its sum
is incremented by the contents of the Carrier Frequency (CF)
Register. As the accumulator sum increments from 0 to 2
the SIN/COS ROM produces quadrature outputs whose
phase advances from 0 to 360°. The CF Register contains a

ωC()cos Q

ωC()sin Q

IN

IN

ωC()sin–=

ωC()cos+=

(EQ. 2)

(EQ. 3)

32

32-bit phase increment, which is updated with the output of
Carrier Tracking Loop. Large phase increments take fewer
clocks to step through the sine wave cycle, which results in a
higher frequency NCO output.

The CF Register sets the NCO frequency using Equation 4:

F

CfCLK

CF INT F

where f
complement hexadecimal value loaded into the Carrier
Frequency Register. As an example, if the CF Register is
loaded with a value of 4000 0000 (Hex), and the CLK
frequency is 40MHz, the NCO would produce quadrature
terms with a frequency of 10MHz. When CF is a negative
value, a clockwise cos/sin vector rotation is produced. When
CF is positive, a counterclockwise vector rotation is
produced.

Note: The NCO is set to a fixed frequency by programming
the upper and lower limits of the Carrier Tracking Loop Filter
to the same value and zeroing the lead gain.

×=

32

⁄()2

[]H=

CfCLK

is the CLK frequency, and CF is the 32-bit two’s

CLK

32

CF()2⁄

(EQ. 4)

Matched Filtering

The HSP50210 provides two selectable matched filters: a
Root Raised Cosine Filter (RRC) and an Integrate and Dump
(I and D) filter. These are shown in Figure 3. The RRC filter
is provided for shaped data pulses and the I and D filter is
provided for square wave data. The filters may be cascaded
for better adjacent channel rejection for square wave data. If
these two filters do not meet baseband filtering
requirements, then they can be bypassed and an external
digital filter (such as the HSP43168 Dual FIR Filter or the
HSP43124 Serial I/O Filter) used to implement the desired
matched filter. The desired filter configuration is set in the
Data Path Configuration Control Register, bits 1 through 7
(see Table 15 on page 32).

The sample rate of the baseband input depends on the
symbol rate and filtering configuration chosen. In
configurations which bypass both filters or use only the RRC
Filter, the input sample rate must be twice the symbol rate. In
configurations which use the I and D Filter, the input sample
rate is decimated by the I and D Filter, down to two samples
per symbol. I and D configurations support input sample
rates up to 32x the input symbol rate.

The RRC filter is a fixed coefficient 15 Tap FIR filter. It has
~40% excess bandwidth beyond Nyquist, which equates to
α = ~0.4 shape factor. The filter frequency response is
shown in Figures 4 and 5. In addition, the 9-bit filter
coefficients are listed as integer values in Table 1. The noise

,

equivalent bandwidth of the RRC filter and other filter
configurations possible with the HSP50110/210 chipset are
given in Appendix A.

8

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

0

-20

-40

-60

-80

NORMALIZED MAGNITUDE (dB)

-100
0

f

CLK

10

FREQUENCY (NORMALIZED TO INPUT SAMPLE RATE)

2f

CLK

10

3f

CLK

10

4f

CLK

10

f

CLK

2

FIGURE 4. RRC FILTER IN HSP50210

0

-0.18

-0.36
SHOWN BELOW

-0.54

-0.72

NORMALIZED MAGNITUDE (dB)

-0.90

0

0

-0.07

-0.14

-0.21

-0.28

NORMALIZED MAGNITUDE (dB)

-0.35
0

FREQUENCY (NORMALIZED TO INPUT SAMPLE RATE)

ENLARGED FOR CLARITY

f

CLK

20

2f

CLK

25

3f

f

CLK

f

CLK

25

40

CLK

40

3f

CLK

25

f

CLK

10

4f

CLK

25

5f

CLK

40

f

CLK

5

3f

CLK

20

FIGURE 5. PASSBAND RIPPLE OF RRC FILTER IN HSP50210

TABLE 1. ROOT RAISED COSINE COEFFICIENTS

COEFFICIENT INDEX COEFFICIENT

02
1-2
21
38
4-16
5-14
686
7 160
886

9-14
10 -16
11 8
12 1
13 -2
14 2

The I and D filter consists of an accumulator, a
programmable shifter and a two sample summer, as shown
in Figure 3. The programmable shifter is provided to
compensate for the gain introduced by the accumulator (see
Table 15). The accumulator provides Integrate and Dump
Filtering for decimation factors up to 16. The two sample
summer provides the moving average required for an
additional decimation factor of 2. A decimation factor of 1
(bypass), 2, 4, 8, 16, or 32 may be selected. At the maximum
decimation rate, a baseband signal sampled at 32x the
symbol rate can be filtered.

The output of the two sample summer is demultiplexed into
two sample streams at the symbol rate. The demultiplexed
data streams from the I and Q processing paths are fed to
the Symbol Tracking Block and Soft decision slicer. The
multiplexed data streams on I and Q are provided as one of
the selectable inputs for the Cartesian-to-Polar Converter.

Cartesian/Polar Converter

The Cartesian/Polar Converter maps samples on the I and Q
processing paths to their equivalent phase/magnitude
representation. The magnitude conversion is equivalent to
Equation 5:

Mag (I, Q) 0.81()∗I2Q2+()=

where 0.81 is the gain of the conversion process. The
magnitude output is an 8-bit unsigned value ranging from 0.0
to 1.9922.

9

(EQ. 5)

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

The phase conversion is equivalent to Equation 6:

Phase (I, Q) tan

-1

where tan

( ) is the arctangent function. The phase

1–

QI⁄(),=

(EQ. 6)

conversion output is an 8-bit two’s complement output,
which ranges from -1.0 to 0.9922 (80 to 7f HE X,
respectively). The -1 to almost 1 range of the phase output
represents phase values from -π to π, respectively. An
example of the I/Q to phase mapping is shown in Figures 6A
through 6C. The phase and magnitude values may be output
via the Output Selector bits 0 through 3 (see Tab le 43).

1.0

0.5

0

MAGNITUDE

-0.5

-1.0

-π

FIGURE 6A. I INPUT TO CARTESIAN/POLAR CONVERTER

1.0

0.5

0

MAGNITUDE

-0.5

-1.0

-π

FIGURE 6B. Q INPUT TO CARTESIAN/POLAR CONVERTER

1.0

0

INPUT PHASE

0

INPUT PHASE

π/2 π-π/2

π/2 π-π/2

The I/Q data path selected for input to the Cartesian-to-Polar
converter determines the input data rate of the AGC and
carrier tracking loops. If the I/Q data path out of the Integrate
and Dump Filter is selected, the AGC is fed with magnitude
values produced by the end-symbol samples. Magnitude
values produced by midsymbol samples are not used
because these samples occur on symbol transitions, resulting
in poor signal magnitude estimates. The Carrier Tracking
block is fed with phase values generated from both the end
and mid-symbol samples. The carrier tracking loop filte r,
however, is only fed with Ph ase Erro r terms gen erate d by the
end symbol samples. If the input of the I and D is selected for
input to the coordinate converter , the control loops are fed with
data at the I/Q data rate. The desired dat a p a th input to the
Cartesian to Polar converter is specified in the Data Path
Configuration Control Register , Bit 8 (see Table 15 on
page 32).

AGC

The AGC loop operates on the main data path (I and Q) and
performs three signal level adjusting functions:

1. Maximizing dynamic range

2. Compensating for SNR variations

3. Maintaining an optimal level into the Soft Decision Slicer.

The AGC Loop Block Diagram, shown in Figure 7, consists
of an Error Detector, a Loop Filter, and Signal Gain Adjusters
(multipliers). The AGC Error Detector generates an error
signal by subtracting the programmable AGC threshold from
the magnitude output of the Cartesian to Polar Converter.
This difference signal is scaled (gain adjusted via multiplier
and shifter), then filtered (integrated) by the AGC Loop Filter
to generate the gain correction to the I and Q signals at the
multipliers. If a fixed gain is desired, set the upper and lower
limits equal.

The AGC responds to the magnitude of the sum of all the
signals in the bandpass of the narrowest filter preceding the
Cartesian to Polar Coordinate Converter. This filter may be
the Integrate and Dump filter shown in Figure 7 on page 12,
the RRC filter upstream in the HSP50210 data path, or some
other filter outside the DCL chip. The magnitude signal
usually contains several components:

0.5

2. The noise comp onent, and

3. Interfering signals component.

1. The signal of interest component,

0

At high SNR’s the signal of interest is significantly greater
than the other components. At lower SNR’s, components 2

-0.5

OUTPUT VOLTAGE

or 3 may become greater than the signal of interest.
Narrowing the filter bandwidth is the primary technique

-1.0

-π

0

INPUT PHASE

π/2 π-π/2

used to mitigate magnitude contributions of component 3.
This will also improve the SNR by reducing the magnitude
contributions of element 2. Consideration of the range of
signal amplitudes expected into the HSP50210, in

FIGURE 6C. CARTESIAN/POLAR CONVERTER PHASE OUTPUT

conjunction with a gain distribution analysis, will provide the

10

FN3652.5

July 2, 2008

A

HSP50210

www.BDTIC.com/Intersil

necessary insight to set the signal level into the Soft
Decision Slicer to yield opti mum perf or ma nce .

Note: Failure to consider the variations due to noise or
interfering signals, can result in signal limiting in the
HSP50210 processing algorithms, which will degrade the
system Bit Error Rate performance.

The AGC Loop is configured by the Power Detect Threshold
and AGC Loop Parameters Control Registers (see Tables 16
and 17 on page 33). Seven programmable parameters must be
set to configure the AGC Loop and its status outputs. Two
parameters, the Power Threshold and the AGC Threshold are
associated with the Error Detector and are represented in 8-bit
fractional unsigned binary format: 2

02-12-22-32-42-52-62-7

.
While the format provides a range from 0 to 1.9961 for the
thresholds, the Cartesian-to-Polar Converter scales the I
and Q input magnitudes by 0.81. Thus, if a full scale (±1)
complex (I and Q) input signal is presented to the converter,
the output will be √(0.81)

2

+ (0.81)2 = 1.1455. The AGC
Threshold parameter value is the desired magnitude of the
signal as it enters the Soft Decision Slicer. It is the parameter
that will determine the error signal in the AGC loop. The
Power Threshold, on the other hand, determines only the
power threshold at which the THRESH

signal is asserted. If
the signal magnitude exceeds the threshold, then the
THRESH

is asserted. This may be used for signal detection,
power detection or external AGC around the A/D converter.
The AGC Threshold parameter is set in the AGC Loop
Parameters Control Register, Bits 16 through 23 (see
Table 17 on page 33). The Power Threshold parameter is
set in the Power Detect Threshold Control Register, Bits 0
through 7 (see Table 16 on page 33). Note that these two
threshold parameters are not required to be set to identical
or even related values, since they perform independent
functions.

The Enable AGC parameter sets the AGC Error Detector
output to zero if asserted and to normal error detection
output when not asserted. This control bit is set in the AGC
Loop Parameter Control Register, Bit 31 (see Tables 17 on
page 33). This bit is used to disable the AGC loop.

The remaining AGC parameters determine the AGC loop
characteristics: gain tracking, tracking rate and tracking limits.
The AGC Loop gain is set via two parameters: AGC Loop Gain
Exponent and AGC Loop Gain Mantissa. In general, the higher
the loop gain, the faster signal level acquisition and tracking,
but this must be tempered by the specific signal characteristics
of the application and the remaining programmable loop
parameters. For the HSP50210, the AGC Loop Gain provides
for a variable attenuation of t he input to the loop filter. The AGC
gain mantissa is a 4-bit value which provides error signal
scaling from 0.000 to 0.9375, with a resolution of 0.0625.
Table 2 on page 11 details the discrete set of decimal values
possible for the AGC Loop Gain mantissa. The exponent

-7

provides a shift factor scaling from 2

to 2

-14

. Table 3 on

page 11 details the discrete set of decimal values possible for

the AGC Loop Gain Exponent. When combined, the exponent
and mantissa provide a loop gain defined as Equation 7:

GC Loop Gain: G

AGC

M()24–()[]2

7E+()–

()[]=

(EQ. 7)

where M is a binary number with a range from 0 to 15 and E
is a 3-bit binary value from 0 to 7. M and E are the
parameters set in the AGC Loop Parameters Control
Register, Bits 24 through 30 (see Table 17 on page 33). The
composite range of the AGC loop Gain is 0.0000 to
[0.9375][2 to 7]. This will scale the AGC error signal to a
range of 0.000 to (1.1455)(0.9375)(2 to 7) = 1.07297(2 to 7).

TABLE 2. AGC LOOP GAIN BINARY MANTISSA TO DECIMAL

SCALED MANTISSA MAPPING

BINARY

CODE

(MMMM)

0000 0.0000 1000 0.5000
0001 0.0625 1001 0.5625
0010 0.1250 1010 0.6250
0011 0.1875 1011 0.6875
0100 0.2500 1100 0.7500
0101 0.3125 1101 0.8125
0110 0.3750 1110 0.8750
0111 0.4375 1111 0.9375

TABLE 3. AGC LOOP BINARY EXPONENT TO SCALED

BINARY CODE

(EEE)

000 0 2
001 1 2
010 2 2
011 3 2
100 4 2
101 5 2
110 6 2
111 7 2

DECIMAL

SCALED

MANTISSA

DECIMAL EXPONENT MAPPING

DECIMAL/HEX

EXPONENT

BINARY

CODE

(MMMM)

DECIMAL SCALED

DECIMAL

SCALED

MANTISSA

EXPONENT

-7

-8

-9

-10

-11

-12

-13

-14

11

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

READ

REG

AGC GAIN = (1.0 + M) x 2

GAIN

ADJUST

G

AGC

I

Q

AGC LOOP FILTER

AGC

AGC

UPPER

E

LOWER

LIMIT †

LIMIT †

L

R

I

E

M

G

I

T

1.0000 TO 15.8572 = G

(0 TO 24dB)

AGC LOOP

GAIN

MANTISSA †

-7

TO 2

(2

S

R

H

I

E

+

0.000 TO 1.07297(2

L

I

M

I

T

F

G

T

AGC

CART/POLAR INPUT SELECT†

AGC LOOP

EXPONENT †

-14

(0.000 TO 0.9375)

)

R

E

G

-7

)

I AND D FILTER

I AND D FILTER

FIGURE 7. AGC LOOP BLOCK DIAGRAM

The AGC Loop Filter integrates the scaled error signal to
provide a correction control term to the multipliers in the I and
Q path. The loop filter accumulator has internal upper and
lower limiters. The upper eight bits of the accumulator output
map to an exponent and mantissa format that is used to set
these upper and lower limits. The format, illustrated in Figure
8, is used for the AGC Upper Limit, AGC Lower Limit and the
Correction Control Term (AGC output). This format should not
be confused with the similar format used for the AGC Loop
Gain. The input to the AGC Loop Filter is included in Figure 8
to show the relative weighting of the input to output of the loop
filter. The loop filter input is represented as the eleve n letter
“G”s. Lower case “e” and “m” detail the format for the AGC
Upper and Lower Limits. This change in type case should help
keep the AGC Limits and AGC Gain formats from being
confused. The AGC Upper and Lower Limits are set in the
AGC Loop Parameters Control Register, Bit s 0 throu gh15,
(see Table 17). This 6-bit unsigned mantissa format provides
for an AGC output control range from 0.0000 to 0.9844, with a
resolution of 0.015625. The 2-bit exponent format provides an
AGC output control range from 1 to 8. The decimal values for
each of the 64 binary mantissa values is detailed in Table 4,
while Table 5 details the decimal value for the 4 exponent
values.

GAIN

M
U
X

“0”

ENABLE AGC †

The AGC Output is implemented in the multiplier according
to Equations 8 and 9.

where m and e are the binary values for mantissa and
exponent found in Ta bles 4 and 5.

Note: This format is identical to the format used to program
the AGC Upper and Lower Limits, but in this usage it is not a
programmed value. It is a representation of the digital AGC
output number, which is presented to the Gain Adjuster
(multipliers) to correct the gain of the I and Q data signals in
the main data path.

These equations yield a composite (mantissa and
exponent) AGC output range of 0.0000 to 1.9844(2
is a logarithmic range from 0dB to 24dB. Figure 9 has
graphed the results of Equations 8 and 9 for both the linear
and logarithmic equations. Figure 9 also has a linear
estimate of the logarithmic equation. This linear
approximation will be used in calculating the AGC response
time.

M
U

X

Out

Out

AGC ERROR DETECT

COMPARE

POWER

THRSHLD †

+

GAIN
ERROR

AGC THRSHLD †

CARTESIAN TO POLAR

G

1.0

AGC linear–

AGC dB–

I2+Q

TAN

20 log 1.0 m

R
E

THRESH

G

-

1.64

-----------=

-1

( )

dcloutlvl agc thresh

where dcloutlvl is the

2

2

Q
I

magnitude output expressed
in dB from Full Scale (dBFS)

0.8
MAGNITUDE

(0 TO 1.1455)

PHASE

=

† Indicates a microprocessor control signal.

1.0 m

+()2e()=

AGC

+()2e()[]=

AGC

3

) which

1.64

⎛⎞

------------

⎝⎠

2

(EQ. 8)

(EQ. 9)

1 20 .2-1 2-2 2-3 2-4 2-5 2-6 2-7 2-8 2-9 2-10 2-11 2-12 2-13 2-14 2-15 2-16 2-17 2-18

2
ee.mmmmmm

GGGG GGGGGG G

FIGURE 8. AGC OUTPUT AND AGC LIMITS BIT WEIGHTING

12

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

TABLE 4. AGC GAIN MANTISSA TO DECIMAL MAPPING

DECIMAL

VALUE

BINARY CODE

(MMMMMM

TABLE 5. AGC GAIN EXPONENT TO DECIMAL MAPPING

BINARY CODE

AGC

000000 0.000000 100000 0.500000
000001 0.015625 100001 0.515625
000010 0.031250 100010 0.531250
000011 0.046875 100011 0.546875
000100 0.062500 100100 0.562500
000101 0.078125 100101 0.578125
000110 0.093750 100110 0.593750
000111 0.109375 100111 0.609375
001000 0.125000 101000 0.625000
001001 0.140625 101001 0.640625
001010 0.156250 101010 0.656250
001011 0.171875 101011 0.671875
001100 0.187500 101100 0.687500
001101 0.203125 101101 0.703125
001110 0.218750 101110 0.718750

001111 0.234375 101111 0.734375
010000 0.250000 110000 0.750000
010001 0.265625 110001 0.765625
010010 0.281250 110010 0.781250
010011 0.296875 110011 0.796875
010100 0.312500 110100 0.812500
010101 0.328125 110101 0.828125
010110 0.343750 110110 0.843750
010111 0.359375 110111 0.859375
011000 0.375000 111000 0.875000
011001 0.390625 111001 0.890625
011010 0.406250 111010 0.906250
011011 0.421875 111011 0.921875
011100 0.437500 111100 0.937500
011101 0.453125 111101 0.953125

011110 0.468750 111110 0.968750

011111 0.484375 111111 0.984375

00 0 2
01 1 2
10 2 2
11 3 2

OF AGC

)

MANTISSA

BINARY CODE

(MMMMMM

DECIMAL/HEX

EXPONENT

AGC

DECIMAL SCALED

DECIMAL

VALUE

OF AGC

)

MANTISSA

EXPONENT

0
1
2
3

224

240

24

18

12

6

0

256

16

12

LINEAR ESTIMATE IN dB

8

4

DATA PATH GAIN (LINEAR)

1
0

0

1632486480

(8 MSBs OF LOOP FILTER ACCUMULATOR)

FIGURE 9. GAIN CONTROL TRANSFER FUNCTION

GAIN dB

96

112

128

144

GAIN CONTROL WORD

160

176

192

GAIN

LINEAR

208

There are two techniques for setting a fixed gain for the
AGC. The first is to set Control Word 2 Bit 31 = 1. This
precludes any error update of present AGC gain value. The
second is to set the upper and lower AGC limits to the
desired gain using Figure 9. The upper and lower limits
have the same value for this case.

The HSP50210 provides two mechanisms for monitoring
signal strength. The first, which involved the THRESH
signal, has already been described. The second
mechanism is via the Microprocessor Interface. The 8 most
significant bits of the AGC loop filter output can be read by
a microprocessor. Refer to the “Microprocessor Interface”
on page 27 for details of how to read this value. This AGC
value has the format describ ed i n Fi gur e 8.

AGC Bit Weighting and Loop Response

The AGC loop response is a function of the programmable
gain, the bit weightings inherent in the connection of each
element of the loop, the AGC Loop filter limits and the
magnitude of the input gain error step. Table 6 on page 14
details the bit weighting between each element of the AGC
Loop from the error detector through the weighting at the
gain adjuster in the signal path. The AGC Loop Gain sets the
growth rate of the sum in the loop filter accumulator. The
Loop filter output growth rate determines how quickly the
AGC loop traces the transfer function shown previously in
Figure 9. To calculate the rate at which the AGC can adjust
over a given period of time, a gain step is introduced to the
gain error detector and the amount of change that is
observed between clocks at the AGC Level Adjusters
(multipliers) is the AGC response time in dB per symbol.
This AGC loop will respond immediately with the greatest
correction term, then asymptotically approach zero
correction.

We begin calculation of the loop response with a full scale
error detector input of ±1. This error input is scaled by the
Cartesian to Polar converter, the error detecto r and th e AGC

GAIN (dB)

13

FN3652.5

July 2, 2008

HSP50210

www.BDTIC.com/Intersil

Loop Gain, accumulated in the loop filter , limited and output to
the gain adjusters. The AGC loop tries to make the error
correction as quickly as possible, but is limited by the AGC
Loop Gain and potentially, the AGC limits. The maximum
AGC response is the maximum gain adjustment made in any
given clock cycle. This involves applying maximum Loop gain

TABLE 6. AGC BIT WEIGHTING

AGC

ACCUM

BIT

POSITION

22 Shifter → E1 12
21 Shifter → E0 6
20 Multiplier → M-1 3
19 M-2 1.5
18 M-3 0.75
17 M -4 0.375
16 M -5 0.1875
15 Multiplier → 1 M -6 0.09375
14
13 1 G -8 0.02344
12 2 G -9 0.01172
11 3 G -10 0.00586
10 4 G -11 0.00293

9 5 G -12 0.00146
8 8(S) = 1(S) 0. 12(S) 12(S) = 1 1 6 G -13 0.000732
77= 0• x1111= 0•
6 6 = 1 x 10 10 = 1 1 G -15 0.000183
5 5 = 2 x 9 9 = 2 2 G -16 0.0000916
4 4 = 3 x 8 8 = 3 3 G -17 0.0000458
3 3 = 4 7 7 = 4 4 G -18 0.0000229
2 2 = 5 6 6 = 5 5 -19 0.0000114
1 1 = 6 5 5 = 6 6 -20 0.00000572
0 0 = 7 4 -21 0.00000286

GAIN

ERROR

INPUT

GAIN

ERROR

BIT

WEIGHT

AGC LOOP

FILTER GAIN

(MANTISSA)

AGC LOOP

FILTER

GAIN

MULTIPLIER

(OUTPUT)

3
2
1
0

and setting the AGC limits as wide as possible. A calculation
using only exponent terms of the various gains will be
sufficient to yield a rough order of magnitude of the range of
the AGC Loop response. The results are shaded in the last
column of Table 6 on page 14 and provided in detail in
Equations 10 and 11.

AGC

LOOP

FILTER

GAIN BITS

KEPT

(rnd) SHIFT = 0 SHIFT = 7

0• • 0-7 0.04688

0• G -14 0.000366

AGC

OUTPUT

AND AGC

LIMITS BIT

WEIGHT

RESOLUTION

AGC GAIN

(dB)

AGC Response

AGC Response

where (0.5) is the MSB of the 0.81 scaling in the Cartesian-to-Polar Coo rdinate Con verter, (0.5) is the MSB of the mantissa of the
Loop Gain, (2

= Input (Cartesian to Polar Converter Gain)(Error Detector Gain)(AGC Loop Gain)(AGC Output Weighting)

MAX

MAX

-7)

10.5()± 0.5()27–()24() 129–()24()± 0.04688dB symbol time⁄===

is the maximum shift gain, and 24 is the maximum loop filter gain.

A similar procedure is used to calculate the minimum AGC response rate.

AGC Response

MIN

10.5()± 0.5()2

14–

()24() 12

16–

()24()± 0.000366dB symbol time⁄===

Thus, the expected range for the AGC rate is approximately 0.0004 to 0.0469dB/symbol time.

14

(EQ. 10)

(EQ. 11)

FN3652.5

July 2, 2008

SYNTHESIZER/

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MIXER

G = 1.0, 0.5 (NOTE 1)

PART

INPUT

(NOTE 4)

0

BINARY

POINT

-2

-1

2

AGC GAIN

MANTISSA

1.0 TO 1.9844

(0.0156 STEPS)

G = 1.0 - 1.9844*2

RRC

FILTER

G = 1.0, 1.13 (NOTE 2)

G

1

0

-2

-1

2

-2

0

2

-1

2

AGC

HSP50210

EXPONENT

0

3

TO 2

2

3

ACCUMULATOR

G = 1 TO 16

L

I

8

M

/

I

T

5

-2

4

2

3

2

2

2

1

2

0

2

-1

2

INT/DUMP

0

-2

-1

2

INTEGRATE AND

DUMP FILTER

INT/DUMP

SHIFTER

0 TO 2-4

G = 2

4

-2

3

2

2

2

1

2

0

2

-1

2

SAMPLE PAIR

SUMMER

G = 0.5, 1.0 (NOTE 3)

L

I

M

I

T

4

-2

3

2

2

2

1

2

0

2

-1

2

0

-2

-1

2

INPUT TO
SOFT DECISION
SLICER
AND
SYMBOL TRACKING
BLOCK

0

-2

-1

2

-9

2

INPUT TO CARTESIAN-TO-POLAR CONVERTER

-10

2

RND

IF AGC OUTPUT SELECTED

-10

2
RND

-9

2

-7

2

RND

INPUT TO CARTESIAN-TO-POLAR CONVERTER

-7

2

IF INT/DUMP OUTPUT SELECTED

-11

2

-6

2

-6

2

-7

2

NOTES:

1. If the Mixer is enabled, the result of the complex multiply is scaled by two (G = 0.5). If the mixer is bypassed, the data passes unmodified (G = 1.0).

2. If the Root Raised Cosine Filter is enabled, a gain of G = 1.13 is introduced. If the RRC filters bypassed, the gain is unity.

-7

3. If the integrate and Dump Filter is bypassed the Sample Pair summer has a gain of G = 1.0 and the 2

-bit position is set to 1. If the integrate

and dump is enabled, the sample pair sum is scaled by one half (G = 0.5).

4. The negative sign on the MSBs indicates use of 2’s complement data format.

FIGURE 10. GAIN DISTRIBUTION AND INTERMEDIATE BIT WEIGHTINGS

Gain Distribution

The gain distribution in the DCL is shown in Figure 10.
These gains consist of a combination of fixed,
programmable, and adaptive gains. The fixed gains are
introduced by processing elements such as the Mixer and
Square Root of Root Raised Cosine Filter. The adaptive
gains are set to compensate for variations in input signal
strength.

The main signal path, with processing block gains and path
bit weightings, is shown in Figure 10. The quadrature inputs
to the HSP50210 are 10-bit fractional two’s complement
numbers with relative bit weightings, as shown in Figure 10.

Following the AGC, the signal path is limited to 8 bits and
passed through the Integrate and Dump Filter en route to the
Soft Decision Slicer and Symbol Tracking Block. The I and D
Filter uses an accumulator together with a sample pair summer
to achieve the desired decimation rate. The I and D shifter is
provided to compensate for the gain introduced by the I and D
Accumulator. The accumulator introduces gain equal to the
decimation factor R, and the shifter gain can be set to 1/R. For
example, if the I and D Filter decimation of 16 is chosen, the I
and D Accumulator will accumulate 8 samples before dumping,
which produces a gain of 8. Thus, for unity gain, the I and D
Shifter would be set for a gain of 2

-3

. The Sample Pair Summer

is unity gain since its output is scaled by one-half.

The first element in the processing chain is the Mixer, which
scales the quadrature outputs of the complex multiplier by
1/2 providing a gain of G = 0.5. If the Mixer is bypassed, the
signal is passed unmodified with a gain of 1.0. Following the
mixer, the quadrature signal is passed to the fixed coefficient
RRC filtering block, which has a gain of 1.13 if enabled and

1.0 if bypassed. Next, the AGC supplies gain to maintain an
optimal signal level at the input to the Soft Decision Slicer,
Cartesian-to-Polar Converter, and the Symbol Tracking
Loop. The gain supplied by the AGC ranges from 1.0 to

1.9844*2

3

.

Symbol Tracking

The symbol tracking loop adjusts the baseband sampling
frequency to force sampling of the baseband waveform at
optimal points for data decisions. The key elements of this loop
are the Sampling Error Detector and Symbol Tracking Loop
Filter shown in Figure 11. The output of these two blocks is a
frequency correction term which is used to adjust the baseband
sample frequency external to the HSP50210. In typical
applications, the frequency correction term is fed back to the
HSP50110 to adjust baseband sampling via the Resampling
NCO (see HSP501 10 Datasheet).

15

FN3652.5

July 2, 2008

REGISTER ENABLE RATE

www.BDTIC.com/Intersil

! = SYMBOL RATE

BLANK = CLK RATE

SYMBOL TRACK
LOOP FILTER

LEAD GAIN

16

FRZ_ST

MID AND END

SYMBOL SAMPLES

END

MID

!

END

MID

R
E
G

SAMPLING ERROR DETECTOR

I

I

Q

Q

DATA

DECISION

MID-SYMBOL

DATA

DECISION

MID-SYMBOL

TRANSITION

DETECT

TRANSITION

MID-POINT

TRANSITION

DETECT

TRANSITION

MID-POINT

LEAD

MANTISSA

‘0’ ‘-1’‘1’

MUX

ACQ

ZERO

LEAD

“0”

LEAD

MANTISSA

TRACK

REG REG

MUX

-

+

“0”

‘0’ ‘-1’‘1’

‘0’

MUX

SINGLE/
DOUBLE

RAIL

SAMPLING

MUX

-

+

+

MUX

ERROR

INVERT

INVERT

ERROR

!

ERROR

ACCUM.

“0”

ZERO

LAG

REG REG

LAG

MANTISSA

ACQ

MUX MUX

MUX

LAG
MANTISSA
TRACK

LEAD

EXPONENT

ACQ

REG

REG REG

LAG

EXPONENT

ACQ

MUX

SHIFT SHIFT

MUX

LEAD
EXPONENT
TRACK

REG

+

MUX

LOAD
ACC

LAG
EXPONENT
TRACK

REG

LIMIT

ACC LIMITS

UPPER/LOWER

LAG

ACCUMULATOR

+

SERIAL

OUTPUT

FORMATTER

SOFSYNC

REG

TO
μP
INTERFACE

SOF

HSP50210

July 2, 2008

FN3652.5

FIGURE 11. SYMBOL TRACKING

LAG GAIN