Fairchild RC5050, RC5051 User Manual

www.fairchildsemi.com

Application Note 50

Implementing the RC5050 and RC5051 DC-DC
Converters on Pentium

®

Pro Motherboards

Introduction

This document describes how to implement a switching volt­age regulator using an RC5050 or an RC5051 high speed
controller, a power inductor, a Schottky diode, appropriate
capacitors, and external power MOSFETs. This regulator
forms a step down DC-DC converter that can deliver up to

14.5A of continuous load current at voltages ranging from

1.3V to 3.5V. A specific application circuit, design consider­ations, component selection, PCB layout guidelines, and per­formance evaluations are covered in detail.

In the past 10 years, microprocessors have ev olved at such an
exponential rate that a modern chip can rival the computing
power of a mainframe computer. Such evolution has been
possible because of the increasing numbers of transistors that
processors integrate. Pentium CPUs, for example, integrate
well over 5 million transistors on a single piece of silicon.

To integrate so many transistors on a piece of silicon, their
physical geometry has been reduced to the sub-micron level.
As a result of each geometry reduction, the corresponding
operational voltage for each transistor has also been reduced.
The changing CPU voltage demands the design of a pro­grammable power supply—a design that is not completely
re-engineered with every change in CPU voltage.

The voltage range of the CPU has shown a downw ards trend
for the past 5 years: from 3.3V for the Pentium, to 3.1V for
the Pentium Pro, and to 1.8V for future processors. With this
trend in mind, Raytheon Electronics has designed the
RC5050 and RC5051 controllers. These controllers integrate
the necessary programmability to address the changing
power supply requirements of lower voltage CPUs.

Previous generations of DC-DC converter controllers were
designed with fixed output voltages adjustable only with a
set of external resistors. In a high volume production envi­ronment (such as with personal computers), however, a CPU
voltage change requires a CPU board re-design to accommo­date the new voltage requirement. The 5-bit DAC in the
RC5050 and the RC5051 reads the voltage ID code that is
programmed into modern processors and provides the appro­priate CPU voltage. In this manner, the PC board does not
have to be re-designed each time the CPU voltage changes.
The CPU can thus automatically configure its own required
supply voltage.

Intel Pentium Pro Processor Power
Requirements

Refer to Intel’s AP-523 Application Note, Pentium® Pro
Processor Power Distribution Guidelines, November 1995

(order number 242764-001), as a basic reference. The speci­fications contained in this document have been modified
slightly from the original Intel document to include updated
specifications for more recent processors. Please contact
Intel Corporation for specific details.

Input V oltages

A v ailable inputs are +12V ±5% and +5V ±5%. Either one or
both of these inputs can be used by the DC-DC converter.
The input voltage requirements for Raytheon’s RC5050
and RC5051 DC-DC converters are listed in Table 1.

Table 1. Input Voltage Requirements

MOSFET

Part # Vcc for IC

RC5050
RC5051

+5V ±5% +5V ±5% 12V ±5% or

Drain

Pentium Pro DC Power Requirements

Refer to Table 2, Intel Pentium Pro and OverDrive® Proces­sor Power Specifications. For a motherboard designs without
a standard VRM (Voltage Regulator Module) socket, the
on-board DC-DC converter must supply a minimum of

13.9A of current @2.5V and 12.4A of current @3.3V. For a
Flexible Motherboard design, the on-board DC-DC con­verter must supply 14.5A maximum ICCP.

DC V oltage Regulation

As indicated in Table 2, the voltage level supplied to the
CPU must be within ±5% of its nominal setting. Voltage reg­ulation limits must include:

• Output load ranges specified in Table 2

• Output ripple/noise

• DC output initial voltage set point

• Temperature and w arm up drift (Ambient +10°C to +50°C
at full load with a maximum rate of change of 5°C per 10
minutes minimum but no more than 10°C per hour)

• Output load transient with:
Slew rate >30A/µs at converter pins
Range: 0.3A - ICCP Max (as defined in Table 2).

MOSFET

Gate Bias

+5V ±5%

Rev. 1.1.0

AN50 APPLICATION NOTE

Table 2. Intel Pentium Pro and OverDrive® Processor Power Specifications

Voltage

Specification,

CPU Model, Features

VCCP (VDC)

150MHz, 256K L2 Cache 3.1 ±5% 9.9 29.2
166MHz, 512K L2 Cache 3.3 ±5% 11.2 35.0
180MHz, 256K L2 Cache 3.3 ±5% 10.1 31.7
200MHz, 256K L2 Cache 3.3 ±5% 11.2 35.0
200MHz, 512K L2 Cache 3.3 ±5% 12.4 37.9

OverDrive Processors

150Mhz

2.5 ±5% 11.2
180Mhz
200Mhz

Flexible Motherboard

Notes:

1. Maximum power values are measured at typical V

2. Flexible motherboard specifications are recommendations only. Actual specifications are subject to change.

2

2.4-3.5 ±5% 14.5 45.0

P to take into account the thermal time constant of the CPU package.

CC

Maximum

Current,
ICCP (A)

12.5

13.9

Maximum Thermal

Design power

1

(W)

26.7

29.7

32.9

Output Ripple and Noise

Ripple and noise are defined as periodic or random signals
over the frequency band of 20Mhz at the output pins. Output
ripple and noise requirements of ±13mV must be met
throughout the full load range and under all specified input
voltage conditions.

Efficiency

The efficiency of the DC-DC converter must be greater than
80% at maximum output current and greater than 40% at low
current draw.

Processor Voltage Identification

There are four voltage identification Pins, VID3-VID0, on
the Pentium Pro processor package which can be used to
support automatic selection of the power supply voltage.
These pins are internally unconnected or are shorted to
ground (VSS). The logic status of the VID pins defines the
voltage required by the processor. In order to address future
low voltage microprocessors, the RC5050 and RC5051
include a VID4 input bit to extend the output voltage range
as low as 1.3V. The output voltage programming codes are
presented in Table 3. A “1” refers to an open pin and a ‘0’
refers to a short to ground.

Table 3. Output Voltage Programming
Codes

VID4 VID3 VID2 VID1 VID0 V

0 1 1 1 1 1.30V
0 1 1 1 0 1.35V
0 1 1 0 1 1.40V
0 1 1 0 0 1.45V
0 1 0 1 1 1.50V
0 1 0 1 0 1.55V

OUT

to CPU

Table 3. Output Voltage Programming
Codes

Note:

1. 0 = processor pin is tied to GND

(continued)

VID4 VID3 VID2 VID1 VID0 V

OUT

0 1 0 0 1 1.60V
0 1 0 0 0 1.65V
0 0 1 1 1 1.70V
0 0 1 1 0 1.75V
0 0 1 0 1 1.80V
0 0 1 0 0 1.85V
0 0 0 1 1 1.90V
0 0 0 1 0 1.95V
0 0 0 0 1 2.00V
0 0 0 0 0 2.05V
1 1 1 1 1 No CPU
1 1 1 1 0 2.1V
1 1 1 0 1 2.2V
1 1 1 0 0 2.3V
1 1 0 1 1 2.4V
1 1 0 1 0 2.5V
1 1 0 0 1 2.6V
1 1 0 0 0 2.7V
1 0 1 1 1 2.8V
1 0 1 1 0 2.9V
1 0 1 0 1 3.0V
1 0 1 0 0 3.1V
1 0 0 1 1 3.2V
1 0 0 1 0 3.3V
1 0 0 0 1 3.4V
1 0 0 0 0 3.5V

1 = processor pin is open.

to CPU

2

APPLICATION NOTE AN50

I

L

VINV

OUT

–( )T

ON

L1

-----------------------------------------------=

V

OUT

V

IN

T

ON

T

S

-----------

 

 

=

I/O Controls

In addition to the Voltage Identification, there are several sig­nals that control the DC-DC converter or provide feedback
from the DC-DC converter to the CPU. They are Power­Good (PWRGD), Output Enable (OUTEN), and Upgrade
Present (UP#). These signals will be discussed later.

RC5050 and RC5051 Description

Simple Step-Down Converter

S1

V

IN

Figure 1. Simple Buck DC-DC Converter

Figure 1 illustrates a step-down DC-DC converter with no
feedback control. The derivation of the basic step-down con­verter is the basis for the design equations for the RC5050
and RC5051. Referring to Figure 1, the basic operation
begins by closing the switch S1. When S1 is closed, the input
voltage V

is impressed across inductor L1. The current

IN

flowing in this inductor is given by the following equation:

where T

is the duty cycle (the time when S1 is closed).

ON

When S1 opens, the diode D1 conducts the inductor
current and the output current is delivered to the load accord­ing to the following equation:

V

-------------------------------------------=

I

L

–( )

OUTTSTON

L1

whereTS is the overall switching period and (TS - TON) is the
time during which S1 is open.

D1

L1

C1 RL Vout

65-5050-06

+

–

The RC5050 and RC5051 Controllers

The RC5050 is a programmable non-synchronous DC-DC
controller IC. The RC5051 is a synchronous version of the
RC5050. When designed around the appropriate external
components, either of these devices can be configured to
deliver more than 14.5A of output current. The RC5050 and
RC5051 utilize both current-mode and voltage-mode PWM
control to create an integrated step-down voltage regulator.
The key differences between the RC5050 and RC5051 are
listed in Table 4.

Table 4. RC5050 and RC5051 Differences

RC5051 RC5050

Operation Synchronous Non-Synchronous
Package 20-SOIC 20-SOIC
Output Enable/

Yes Yes

Disable

Main Control Loop

Refer to the RC5051 Block Diagram illustrated in Figure 2.
The control loop of the regulator contains two main sections;
the analog control block and the digital control block. The
analog section consists of signal conditioning amplifiers
feeding into a set of comparators which provide the inputs to
the digital control block. The signal conditioning section
accepts inputs from the IFB (current feedback) and VFB
(voltage feedback) pins and sets up two controlling signal
paths. The voltage control path amplifies the VFB signal and
presents the output to one of the summing amplifier inputs.
The current control path takes the difference between the
IFB and VFB pins and presents the resulting signal to
another input of the summing amplifier. These two signals
are then summed together with the slope compensation input
from the oscillator. This output is then presented to a
comparator, which provides the main PWM control signal to
the digital control block.

The additional comparators in the analog control section set
the point at which the current limit comparator disables the
output drive signals to the external power MOSFETs.

By solving these two equations, we can arrive at the basic
relationship for the output voltage of a step-down con v erter:

In order to obtain a more accurate approximation for V
we must also include the forward voltage VD across diode
D1 and the switching loss, VSW. After taking into account
these factors, the new relationship becomes:

T

ON

-----------

V

OUT

VINVDVSW–+( )

V

–=

D

T

S

where VSW= MOSFET switching loss

= IL • R

DS,ON

OUT

The digital control block takes the comparator inputs and the
main clock signal from the oscillator to provide the appropri­ate pulses to the HIDRV and LODRV output pins. These
pins control the external power MOSFETs. The digital sec­tion utilizes high speed Schottky transistor logic, allowing
the RC5050 and the RC5051 to operate at clock speeds as

,

high as 1MHz.

High Current Output Drivers

The RC5051 contains two identical high current output
drivers that utilize high speed bipolar transistors in a
push-pull configuration. Each driver is capable of
delivering 1A of current in less than 100ns. Each driver’s
power and ground are separated from the chip’s power and
ground for additional switching noise immunity.

3

AN50 APPLICATION NOTE

+12V

RC5051

OSC

–
+

–
+

+5V

–
+

VREF

VID0

5-BIT

DAC

VID2 RSEL

VID1

VID3

1.24v

REFERENCE

Figure 2. RC5051 Block Diagram

The HIDRV driver has a power supply, VCCQP, supplied
from a 12V source as illustrated in Figure 2. The resulting
voltage is sufficient to provide the gate to source voltage to
the external MOSFET that is required to achieve a low
R

. Since the low side synchronous FET is referenced

DS,ON

to ground, there is no need to boost the gate drive voltage,
and its VCCP power pin can be tied to VCC.

Internal Voltage Reference

The reference included in the RC5050 and RC5051 is a pre­cision band-gap voltage reference. The internal resistors are
precisely trimmed to provide a near zero temperature coeffi­cient (TC). Added to the reference input is the resulting out­put from an integrated 5-bit DAC—provided in accordance
to the Pentium Pro specification guidelines. These guidelines
require the DC-DC converter output to be directly program­mable via a 4-bit voltage identification (VID) code. This
code scales the reference voltage from 2.0V (no CPU) to

3.5V in 100mV increments. To target future generations of
low-voltage processors, the RC5050 and RC5051 incorpo­rate a VID4 pin to allo w additional programmability between

1.3V and 2.05V. For guaranteed stable operation under all
operating conditions, a 0.1µF of decoupling capacitance
should be connected to the VREF pin. No load should be
imposed on this pin.

Power Good (PWRGD)

The RC5050 and RC5051 Power Good function is designed
in accordance with the Pentium Pro DC-DC converter speci­fication to provide a constant voltage monitor on the VFB
pin. The circuit compares the VFB signal to the VREF volt-

–
+

DIGITAL

CONTROL

POWER

GOOD

PWRGD

VO

65-5051-01

age and outputs an active-low interrupt signal to the CPU
when the power supply voltage exceeds ±12% of nominal.
The Power Good flag provides no other control function to
the RC5050 or the RC5051.

Output Enable (OUTEN)

The DC-DC converter accepts an open collector signal for
controlling the output voltage. The low state disables the out­put voltage. When disabled, the PWRGD output is in the lo w
state.

Upgrade Present (UP#)

Intel specifications state that the DC-DC converter should
accept an open collector signal, used to indicate the presence
of an upgrade processor. The typical state is high (that is, a
standard processor is in the system). When in the low or
ground state (an OverDrive processor is present), the output
voltage must be disabled unless the conv erter can supply the
requirements of the OverDrive processor . When disabled, the
PWRGD output must be in the low state. Because the
RC5050 and RC5051 can supply the requirements of the
OverDrive processor, the #UP signal is not required.

Over-Voltage Protection

The RC5050 and RC5051 constantly monitor the output
voltage for protection against over voltage conditions. If the
voltage at the VFB pin e xceeds 20% of the selected program
voltage, an over-voltage condition is assumed and the chip
disables the output drive signal to the external MOSFET(s).

4

APPLICATION NOTE AN50

Short Circuit Protection

A current sense methodology is implemented to disable the
output drive signal to the MOSFET(s) when an over-current
condition is detected. The voltage drop created by the output
current flowing across a sense resistor is presented to an
internal comparator. When the voltage developed across the
sense resistor exceeds the comparator threshold voltage, the
chip reduces the output drive signal to the MOSFET(s).

The DC-DC converter returns to normal operation after the
fault has been removed, for either an over-voltage or a short
circuit condition.

Oscillator

The RC5050 and RC5051 oscillator section uses a fixed cur­rent capacitor charging configuration. An external capacitor
(C

) is used to preset the oscillator frequency between

EXT

200KHz and 1MHz. This scheme allows maximum flexibil­ity in setting the switching frequency and in choosing exter­nal components.

In general, a lower operating frequency decreases the peak
ripple current flowing in the output inductor, thus allowing
the use of a smaller inductor value. Unfortunately, operation
at lower frequencies increases the amount of energy storage
that must be provided by the bulk output capacitors during
load transients due to slower loop response of the controller.

In addition, the efficiency losses due to switching of the
MOSFETs increase as the operating frequency is increased.
Thus, efficiency is optimized at lower operating frequencies.
An operating frequency of 300 kHz was chosen to optimize
efficiency while maintaining excellent regulation and tran­sient performance under all operating conditions.

Design Considerations and
Component Selection

Figure 3 shows a typical non-synchronous application using
the RC5050. Figure 4 illustrates the synchronous applica­tion using the RC5051.

VREF

GND

+12V

+5V

C4

0.1µF

C7

0.1µF

L2

2.5µH

VID4

VID3

VID2

VID1

VID0

C1

1000 µF

C2

1000 µF

12
13
14
15
16
17
18
19
20

C3

1000µF

RC5050

C

EXT

100pF

C5

0.1µ F

1011

9
8
7
6
5
4
3
2
1

C10

0.1µ F

R5
47

C6

4.7µF

ENABLE

C12

1µF

D1

1N4691

IRF7413

R6
10K

M1

C11

0.1µ F

C8

0.1µ F

DS1
MBR2015CTL

VCC

PWRGD

M2

IRF7413

C9

0.1µF

L1

1.3µ H

R

6mΩ

SENSE

F

µ

1500

C13

1500 µF

C14

C15

VO

1500µF

1500µF

C16

Figure 3. Non-Synchronous DC-DC Converter Application Schematic Using the RC5050

5

AN50 APPLICATION NOTE

+12V

+5V

C4

0.1µF

VREF

GND

C7

0.1µF

L2

2.5µH

VID4
VID3

VID2

VID1

VID0

C1

1000 µF

C2

1000 µF

12
13
14
15
16
17
18
19
20

C3

1000µF

RC5051

C

100pF

C5

R5

0.1µ F

47

1011
9
8
7
6
5
4
3
2
1

EXT

C10

0.1µ F

C6

4.7µF

ENABLE

C12

1µF

D1

1N4691

IRF7413

10K

IRF7413

M3

R6

M1

C11

0.1µ F

C8

0.1µ F

VCC

PWRGD

M2

M4

IRF7413

0.1µF

IRF7413

1.3µ H

DS1

1N5817

C9

R

L1

6mΩ

SENSE

F

µ

1500

C13

1500 µF

C14

C15

VO

1500 µF

1500µF

C16

Figure 4. Synchronous DC-DC Converter Application Schematic Using the RC5051

6

APPLICATION NOTE AN50

MOSFET Selection Cosiderations

• Power package with low Thermal Resistance

• Drain current rating of 20A minimum

MOSFET Selection

• Drain-Source voltage > 15V.
This application requires N-channel Logic Level Enhance­ment Mode Field Effect Transistors. Desired characteristics
are as follows:

The on-resistance (R

) is the primary parameter for

DS,ON

MOSFET selection. It determines the power dissipation
within the MOSFET and, therefore, significantly affects the

• Low Static Drain-Source On-Resistance,

R

< 37 mΩ (lower is better)

DS,ON

efficiency of the DC-DC converter. Table 5 is a selection
table for MOSFETs.

• Low gate drive voltage, VGS ≤ 4.5V

Table 3. MOSFET Selection Table

R

Manufacturer & Model # Conditions

Fuji

V

= 4V, ID = 17.5A TJ = 25°C 25 37 TO-220 Φ

GS

2SK1388
Siliconix

V

= 4.5V, ID = 5A TJ = 25°C 16.5 20 SO-8

GS

SI4410DY
National Semiconductor

V

= 5V, ID = 40A TJ = 25°C 13 15 TO-220 Φ

GS

1

TJ = 125°C 37 —

TJ = 125°C 28 34

NDP706AL
NDP706AEL TJ = 125°C 20 24
National Semiconductor V
NDP603AL T
National Semiconductor V

= 4.5V, ID = 10A TJ = 25°C 31 40 TO-220 Φ

GS

= 125°C 42 54 ΦJC= 2.5

J

= 5V, ID = 24A TJ = 25°C 22 25 TO-220 ΦJA= 62.5

GS

NDP606AL TJ = 125°C 33 40 Φ
Motorola V

= 5V, ID = 37.5A TJ = 25°C 6 9 TO-263 Φ

GS

MTB75N03HDL TJ = 125°C 9.3 14 (D2 PAK) Φ
Int. Rectifier V

= 5V, ID = 31A TJ = 25°C — 28 TO-220 Φ

GS

IRLZ44 TJ = 125°C — 46 Φ
Int. Rectifier V

= 4.5V, ID = 28A TJ = 25°C — 19 TO-220 Φ

GS

IRL3103S TJ = 125°C — 31 Φ
Intl Rectifier V
IRF7413 SMD

= 4.5V,

GS

ID = 3.7A

TA = 25°C 18 SO-8 Φ

DS, ON

(mΩ)

Package

(SMD)

Thermal

ResistanceTyp. Max.

= 75

JA

Φ

= 50

JA

= 62.5

JA

Φ

= 1.5

JC

= 62.5

JA

= 1.5

JC

= 62.5

JA

= 1.0

JC

= 62.5

JA

= 1.0

JC

= 62.5

JA

= 1.0

JC

= 50

JA

Note:

1. R

values at Tj = 125°C for most devices were extrapolated from the typical operating curves supplied by the

DS,ON

manufacturers and are approximations only.

7

AN50 APPLICATION NOTE

Two MOSFETs in parallel.

We recommend two MOSFETs used in parallel instead of
one single MOSFET. The following significant advantages
are realized using two MOSFETs in parallel:

• Significant reduction of Power dissipation.

Maximum current of 14A with one MOSFET:

P

MOSFET

= (I2 R

)(Duty Cycle) =

DS,ON

(14)2(0.050*)(3.3+0.4)/(5+0.4-0.35) = 7.2 W
With two MOSFETs in parallel:

P

MOSFET

= (I2 R

)(Duty Cycle) =

DS,ON

(14/2)2(0.037*)(3.3+0.4)/(5+0.4-0.35) = 1.3W/FET

* Note: R

25°C. R
using a single MOSFET. When using two MOSFETs in parallel, the
temperature effects should not cause the R
listed maximum value of 37mΩ.

increases with temperature. Assume R

DS,ON

can easily increase to 50mΩ at high temperature when

DS,ON

DS,ON

DS,ON

to rise above the

• Less heat sink required.

With power dissipation down to around one watt and with
MOSFETs mounted flat on the motherboard, considerable
less heat sink is required. The junction-to-case thermal
resistance for the MOSFET package (TO-220) is typically
at 2°C/W and the motherboard serves as an excellent heat
sink.

• Higher current capability.

With thermal management under control, this on-board
DC-DC converter is able to deliver load currents up to

14.5A with no performance or reliability concerns.

MOSFET Gate Bias

MOSFET can be biased by one of two methods: Charge
Pump and 12V Gate Bias.

• Method 1. Charge pump (or Boostrap) method.

Figure 5 employs a charge pump to provide gate bias.
Capacitor CP is the charge pump deployed to boost the
voltage of the RC5050 output driver. When the MOSFET
switches off, the source of the MOSFET is at -0.6V.
VCCQP is charged through the Schottky diode to 4.5V.
Thus, the capacitor CP is charged to 5V. When the MOS­FET turns on, the source of the MOSFET voltage is equal
to 5V. The capacitor voltage follows, and hence provides a
voltage at VCCQP equal to 10V. The Schottky diode is
required to provide the charge path when the MOSFET is
off, and reverses bias when the VCCQP goes to 10V. The
charge pump capacitor, CP, needs to be a high Q, high fre­quency capacitor. A 1µF ceramic capacitor capacitor is
recommended here.

= 25mΩ at

+5V

DS2

PWM/PFM

Control

VCCQP

HIDRV

CP

M1

DS1

L1

RS

VO

CB

65-AP50-01

Figure 5. Charge Pump Configuration

• Method 2. 12V Gate Bias.

Figure 6 illustrates how a 12V source can be used to bias
the VCCQP. A 47 Ω resistor is used to limit the transient
current into the VCCQP pin and a 1µF capacitor filter is
used to filter the VCCQP supply. This method provides a
higher gate bias voltage (VGS) to the MOSFET, and there­fore reduces the R

of the MOSFET and reduces the

DS,ON

power loss due to the MOSFET. Figure 7 shows how
R

reduces dramatically with VGS increases. A 6.2V

DS,ON

Zener diode (D1) is placed to clamp the voltage at VCCQP
to a maximum of 12V and ensure that the absolute maxi­mum voltage of the IC will not be exceeded

+5V

47Ω

+12V

D1

1µF

6.2V
M1

L1

RS

DS1

R(DS)Fuji
R(DS)7060
R(DS)706A
R(DS)-706AEL

(V)

GS

VO

CB

65-AP50-02

0.1

0.09

0.08

0.07

0.06

(Ω)

0.05

DS,ON

0.04

R

0.03

0.02

0.01

VCCQP

HIDRV

PWM/PFM

Control

Figure 6. 12V Gate Bias Configuration

0

1.5 2 2.5 3 3.5 4 5 6 7 8 9 10 11

Gate-Source Voltage, V

Figure 7. R

vs. VGS for Selected MOSFETs

DS,ON

8

APPLICATION NOTE AN50

Converter Efficiency

Losses due to parasitic resistance in the switches, coil, and
sense resistor dominate at high load-current level. The major
loss mechanisms under heavy loads, in usual order of impor­tance, are:

• MOSFET I2R Losses

• Coil Losses

• Sense Resistor Losses

Calculation of Converter Efficiency Under Heavy Loads

• gate-charge losses

• diode-conduction losses

• transition losses

• Input Capacitor losses

• losses due to the operating supply current of the IC.

P

Efficiency

P

LOSS

where , where

PD

COILIOUT

PD

SENSERIOUT

PD

GATEqGATE

PD

DIODE

PD

TRAN

PD

CAPIRMS

PD

IC

OUT

------------­p

IN

PD

MOSFET

PD

MOSFETIOUT

2

×=

VfID× 1 Dutycycle–( )=

2

V

C

× I

IN

-------------------------------------------------------------=

2

ESR×=

VCCICC×=

R

2

I

×

OUTVOUT

--------------------------------------------------------= =
I

OUTVOUTPLOSS

PD

+ + + + + + +=

COIL

2

× DutyCycle×= DutyCycle

COIL

R

×=

SENSE

, where q

f 5V××=

R

PD

DS,ON

+×

SENSER

GATE

PD

GATE

is the gate charge and f is the switching frequency

× f×

RSS

I

DRIVE

LOAD

, where C

is the reverse transfer capacitance of the high-side MOSFET.

RSS

PD

DIODE

PD

TRAN

PD

CAP

V

+

OUTVD

------------------------------------------=
V

–+

INVDVSW

PD

Example

DutyCycle

3.3 0.5+

------------------------------ 0.73= =

5 0.5 0.3–+

IC

PD

MOSFET

PD

COIL

PD

SENSER

PD

GATE

PD

DIODE

PD

TRAN

PD

CAP

PD

IC

PD

LOSS

Efficiency

1020.030× 0.73× 2.19W= =

1020.010× 1W= =

1020.0065× 0.65W= =

CV f× 5V× 1.75nf 9 1–( )V 285Khz 5V××× 0.019W= = =

0.5 10 1 0.73–( )× 1.35W= =

52400pf× 10× 285khz×

----------------------------------------------------------------

0.7A

0.010W∼=

7.5 2.5–( )20.015× 0.37W= =

0.2W=

2.19W 1.0W 0.65W 0.019W 1.35W 0.010W 0.37W 0.2W+ + + + + + + 5.789W= =

3.3 10×

---------------------------------------

3.3 10 5.815+×

85%≈=∴

9

AN50 APPLICATION NOTE

Selecting the Inductor

The inductor is one of the most critical components to be
selected for a DC-DC converter application. The critical
parameters are inductance (L), maximum DC current (IO),
and DC coil resistance (Rl). The inductor core material is a
crucial factor in determining the amount of current the
inductor is able to withstand. As with all engineering
designs, tradeoffs exist between various types of core materi­als. In general, Ferrites are popular due to low cost, low EMI
properties, and high frequency (>500KHz) characteristics.
Molypermalloy powder (MPP) materials exhibit good satu­ration characteristics, low EMI, and low hysteresis losses,
but tend to be expensive and more effectively utilized at
operating frequencies below 400KHz. Another critical
parameter is the DC winding resistance of the inductor. This
value should typically be reduced as much as possible, as the
power loss in the DC resistance degrades the efficiency of
the converter by the relationship: P

loss

of the inductor is a function of the oscillator duty cycle
(TON) and the maximum inductor current (IPK). IPK can be
calculated from the relationship:

VINVSW– VD–

 

I

PKIMIN

-----------------------------------------

+=

 

L

T

ON

Where TON is the maximum duty cycle and VD is the
forward voltage of diode DS1.

2

= I

x Rl. The value

O

When designing the external current sense circuitry, pay
careful attention to the output limitations during normal
operation and during a fault condition. If the short circuit
protection threshold current is set too low, the DC-DC con­verter may not be able to continuously deliver the maximum
CPU load current. If the threshold level is too high, the out­put driver may not be disabled at a safe limit and the result­ing power dissipation within the MOSFET(s) may rise to
destructive levels.

The following is the design equation used to set the short cir­cuit threshold limit:

V

th

R

SENSE

I

SCIinductor

Where I
I

load, max

You must also take into account the current (Ipk -I

--------

, where: I

I

SC

≥ I

and I

pk

min

Load, max

= Output short circuit current=

SC

IpkI

----------------------------+=

are peak ripple current and

= maximum output load current.

–( )

min

2

), or the

min

ripple current flowing through the inductor under normal
operation. Figure 8 illustrates the inductor current waveform
for the RC5050 DC-DC converter at maximum load.

Ipk

Then the inductor value can be calculated using the
relationship:

L

 

I

–

PKIMIN

Where VSW (R

DS,ON

T

ON

x IO) is the drain-to-source voltage of

– VO–

V

INVSW

 

-----------------------------------------

=

M1 when it is switched on.

Implementing Short Circuit Protection

Intel currently requires all power supply manufacturers to
provide continuous protection against short circuit condi­tions that may damage the CPU. T o address this requirement,
Raytheon Electronics has implemented a current sense meth­odology to limit the power delivered to the load in the event
of overcurrent. The voltage drop created by the output cur­rent across a sense resistor is presented to one terminal of an
internal comparator with hysterisis. The other comparator
terminal has the threshold voltage, nominally of 120mV.
Table 6 states the limits for the comparator threshold of the
Switching Regulator.

Table 6. RC5050 Short Circuit Comparator
Threshold Voltage

Short Circuit Comparator

V

threshold

Typical 120
Minimum 100
Maximum 140

(mV)

I

(Ipk-I

)/2

min

I

Imin

T

ON

T=1/f

Figure 8. Typical DC-DC Converter

Inductor Current Waveform

T

OFF

s

LOAD, MAX

t

The calculation of this ripple current is as follows:

IpkI

–( )

min

---------------------------­2

V

– V

INVSW

-----------------------------------------------------

–( )

L

OUT

V

+( )

OUTVD

----------------------------------------------­V

– VD+( )

INVSW

T×=

where:
V

= input voltage to converter,

IN

V

= voltage across switcher MOSFET = I

SW

LOAD

x R

DS,ON

VD = Forward Voltage of the Schottky diode,
T = the switching period of the converter = 1/fS, and
fS = switching frequency.

For an input voltage of 5V, output voltage of 3.3V, L equals

1.3µH and a switching frequency of 285KHz (using
C

= 100pF), the inductor current can be calculated at

EXT

approximately 1A:

I

–( )

pkImin

---------------------------­2

3.3 0.5+( )

---------------------------------------------------------

5.0 14.5 0.037×– 0.5+

5.0 14.5 0.037×– 3.3–( )

--------------------------------------------------------------

1.3 10

6–

×

1

-----------------------

× 2A=

285 10

3

×

×=

,

10

APPLICATION NOTE AN50

Therefore, for load current of 14.5A, the peak current
through the inductor, Ipk, is found to be approximately

15.5A:

IPKI

–( )

min

ISCI

≥ I

inductor

Load, max

-----------------------------+ 14.5 2+ 16.5A= = =
2

Therefore, the short circuit detection threshold must be at
least 16.5A.

Table 7. Comparison of Sense Resistors

Discrete Iron

Motherboard

Description

Tolerance

Trace Resistor

±29% ±5%

Factor (TF)
Size

(L x W x H)

2" x 0.2" x 0.001"

(1 oz Cu trace)

Power capability >50A/in 1 watt

Temperature

+4,000 ppm +30 ppm ±75 ppm ±30 ppm ±20 ppm

Coefficient
Cost

@10,000 piece

Low included in

motherboard

Alloy

Resistor (IRC)

(±1% available)

0.45" x 0.065" x

0.200"

(3W and 5W

available)

$0.31 $0.47 $0.09 $0.09

The next step is to determine the value of the sense resistor.
Including sense resistor tolerance, the sense resistor value
can be approximated as follows

R

SENSE

V

th,min

---------------­I

SC

1 TF–( )×

V

th,min

-----------------------------------

1.0 I

+

Load,max

1 TF–( )×= =

Where TF = Tolerance Factor for the sense resistor.

Table 7 describes tolerance, size, power capability, tempera­ture coefficient and cost of various type of sense resistors.

Discrete Metal

Strip Surface

Mount Resistor

(Dale)

Discrete MnCu

Alloy Wire

Resistor

Discrete

CuNi Alloy

Wire Resistor

(Copel)

±1% ±10% ±10%

0.25" x 0.125" x

0.025"

0.200" x 0.04" x

0.160"

0.200" x 0.04" x

0.100"

1 watt 1 watt 1 watt

Refer to Appendix A for Directory of component suppliers

Based on the Tolerance Factor in the above table, for an
embedded PC trace resistor and for I

V

th,min

R

SENSE

100mV

---------------------------------

2.0A 14.5A+

----------------------------------------

2.0A I

+

Load, max

1 29%–( )× 4.3mΩ=

For a discrete resistor and for I

V

th,min

R

SENSE

100mV

---------------------------------

2.0A 14.5A+

----------------------------------------

2.0A I

+

Load, max

1 5%–( )× 5.8mΩ=

1 TF–( )× = =

load, max

1 TF–( )× = =

load,max

= 14.5A:

= 14.5A:

For user convenience, Table 8 lists the recommended values
for sense resistors for various load currents using embedded
PC trace resistors and discrete resistors.

Table 8. R

I

Load,max

(A)

for Various Load Currents

sense

R

SENSE

PC Trace

Resistor (mΩ)

R

SENSE

Discrete

Resistor (mΩ)

10.0 5.9 7.9

11.2 5.4 7.2

12.4 4.9 6.6

13.9 4.5 6.0

14.0 4.4 5.9

14.5 4.3 5.8

Discrete Sense Resistor

Discrete Iron Alloy resistors come in variety of tolerances
and power ratings, and are most ideal for precision imple­mentation. MnCu Alloy wire resistors or CuNi Alloy wire
resistors are ideal for low cost implementations.

11

AN50 APPLICATION NOTE

R ρ

L

W t

×

-------------

×=

Embedded Sense Resistor (PC Trace Resistor)

Embedded PC trace resistors have the advantage of near zero
cost implementation. However, the value of the PC trace
resistor has large variations. Embedded resistors have 3
major error sources: the sheet resistivity of the inner layer,
the mismatch due to L/W, and the temperature variation of
the resistor. All three error sources must be considered for
laying out embedded sense resistors.

• Sheet resistivity.

For 1 ounce copper, the thickness variation is typically

1.15 mil to 1.35 mil. Therefore error due to sheet resistiv­ity is (1.35 - 1.15)/1.25 = 16%

• Mismatch due to L/W.

Percent error in L/W is dictated by geometry and the
power dissipation capability of the sense resistor. The
sense resistor must be able to handle the load current and
therefore requires a minimum width which is calculated as
follows.

I

L

----------=

W

0.05

where: W = minimum width required for proper power
dissipation (mils) and I

= Load Current in Amps.

L

For 15A of load current, minimum width required is
300mils, which reflects a 1% L/W error.

• Thermal Consideration.

Due to I2R power losses the surface temperature of the
resistor will increase leading to a higher value. In addition,
ambient temperature
variation will add the change in resistor value:

R R

where: R20 is the resistance at 20°C,

1 α20T 20–( )]+[=

20

α

= 0.00393/ °C, T

20

is the operating temperature, and R is the desired value.

where:
ρ = Resistivity(µΩ-mil),

L = Length(mils),
W = Width(mils), and

W

L

t

t = Thickness(mils).
For 1oz copper, t = 1.35 mils, ρ = 717.86 µΩ-mil,

1 L/1 W = 1 Square ( ■ ).
For example, you can layout a 5.30mΩ embedded sense

resistor using the equations above:

I

10

L

----------

W

L

0.05

R W× t×

------------------------

ρ

---------- 200mils= = =

0.05

0.00530 200× 1.35×

--------------------------------------------------- 2000mils= = =

717.86

L/W = 10 ■
Therefore, to model 5.30mΩ embedded sense resistor, you

need W = 200 mils and L = 2000 mils. Refer to Figure 9.

1 1 1 1 1 1 1 1 1 1

W = 200 mils

L = 2000

Figure 9. 5.30mΩ Sense Resistor (10 ■)

You can also implement the sense resistor in the following
manner. Each corner square is counted as 0.6 square since
current flowing through the corner square does not flow
uniformly and it is concentrated towards the inside edge, as
shown in Figure 10.

1 1 1 1 1 1

.6 .6

1 1

.8

For temperature T = 50°C, the %R change = 12%.

Table 9 is the summary of the tolerance for the Embedded
PC Trace Resistor.

Table 9. Summary PC Trace Resistor Tolerance

Tolerance due to Sheet Resistivity variation 16%
Tolerance due to L/W error 1%
Tolerance due to temperature variation 12%
Total Tolerance for PC Trace Resistor 29%

Design Rules for Using an Embedded Resistor

The basic equation for laying an embedded resistor is:

12

Figure 10. 5.30mΩ Sense Resistor (10 ■)

A Design Example Combining an Embedded Resistor
and a Discrete Resistor

For low cost implementation, the embedded PC trace resistor
is most desirable. However, its wide tolerance (29%) pre­sents a challenge. In addition, requirements for the CPU
change frequently, and, thus, the maximum load current may
be subject to change. Combining embedded resistors with
discrete resistors may be a desirable option. Figure 11 shows
a design that provides flexibility with a solution to address
wide tolerances.

In this design, you have the option to choose an embedded
or a discrete MnCu sense resistor. To use the discrete sense
resistor, populate R21 with a shorting bar (zero Ohm resis­tor) for proper Kelvin connection and add the MnCu sense
resistor. To use the embedded sense resistor, on the other
hand, populate R22 with a shorting bar for Kelvin connec-

APPLICATION NOTE AN50

Embedded Sense Resistor

IFBH

MnCu Discrete
Resistor

IFBL

Figure 11. Short Circuit Sense Resistor Design Using a PC Trace Resistor and an Optional Discrete Sense Resistor

R21 R22

Output Power
Plane (Vout)

R-∆r

R

R+∆r

tion. The embedded sense resistor allo ws the user to choose a
plus or a minus delta resistance tap to offset any large sheet
resistivity change. In this design, the center tap yields 6mΩ,
the left tap yields 6.7mΩ, and the right tap yields 5.3mΩ.

RC5050 and RC5051 Short Circuit Current
Characteristics

The RC5050 and RC5051 short circuit current characteristic
includes a hysteresis function that prevents the DC-DC con­verter from oscillating in the event of a short circuit. Figure
12 shows the typical characteristic of the DC-DC converter
circuit with a 6mΩ sense resistor. The converter exhibits a
normal load regulation characteristic until the voltage across
the resistor exceeds the internal short circuit threshold of
120mV. At this point, the internal comparator trips and
signals the controller to turn off the gate drive to the power
MOSFET. This causes a drastic reduction in output voltage
as the load regulation collapses into the short circuit control
mode. The output voltage does not return to its nominal
value the output current is reduced to a value within the safe
range for the DC-DC converter.

3.5

3.0

2.5

2.0

1.5

1.0

Output Voltage

0.5
0

0 5 10 15 20 25

Output Current

Power Dissipation Consideration During a
Short Circuit Condition

The RC5050 and RC5051 controllers respond to an output
short circuit by drastically changing the duty cycle of the
gate drive signal to the power MOSFET. In doing this, the
power MOSFET is protected from stress and from eventual
failure. Figure 13A shows the gate drive signal of a typical
RC5050 operating in continuous mode with a load current of
10A. The duty cycle is set by the ratio of the input voltage to
the output voltage. If the input voltage is 5V, and the output
voltage is 3.1V, the ratio of Vout/ Vin is 62%. Figure 13B
shows the result of a RC5050 going into its short circuit
mode with a duty cycle approximately of 20%. Calculating
the power in the MOSFET at each condition on the graph
(Figure 12) shows how the protection works. The power dis­sipated in the MOSFET at normal operation for a load cur­rent of 14.5A, is given by:

2

14.5

 

P

I2RON× DutyCycle

D

----------

 

2

for each MOSFET.

The power dissipated in the MOSFET at short circuit
condition for a peak short current of 20A, is given by:

P

D

 

2

.037× .2 × 0.74W==

2

20

 

------

for each MOSFET.

These calculations show that the MOSFET is not being
over-stressed during a short circuit condition.

× .62 1.2W=×=×=

˙

.037

Figure 12. RC5050 Short Circuit Characteristic

13

AN50 APPLICATION NOTE

Figure 13A. V

Operation Condition with V

Output Waveform for Normal

CCQP

= 3.3V@10A

out

P

D Diode,

14 0.45× 0.8× 5W≈

I

VF× 1 DutyCycle–( )× = =

F ave,

Thus, for the Schottky diode, the thermal dissipation during
a short circuit is greatly magnified. This requires that the
thermal dissipation of the diode be properly managed by an
appropriate heat sink. To protect the Schottky from being
destroyed in the event of a short circuit, you should limit the
junction temperature to less than 130°C. You can find the
required thermal resistance using the equation for maximum
junction temperature:

T

–

J max( )TA

-------------------------------=

P

D

R

ΘJA

Assuming that the ambient temperature is 50°C,

T

–

ΘJA

------------------------------­P

D

R

J max( )TA

130 50–

--------------------- 16°C W⁄= = =

5

Thus, you need to provide a heat sink that gives the Schottky
diode a thermal resistance of 16°C/W or lower to protect the
device during an indefinite short.

In summary, with proper heat sink, the Schottk y diode is not
over-stressed during a short circuit condition.

Figure 13B. V

Output Shorted to Ground

Power dissipation on the Schottky diode during a short cir­cuit condition must also be considered. During normal oper­ation, the Schottky diode dissipates power while the power
MOSFET is off. The power dissipated in the diode during
normal operation, is given by:

P

D Diode,

14.5 0.5V× 1 0.62–( ) 2.75W=×

IFVF× 1 DutyCycle–( )× = =

During a short circuit, the duty cycle dramatically reduces to
around 20%. The forward current in the short circuit condi­tion decays exponentially through the inductor. The power
dissipated in the diode during short circuit condition, is
approximately given by:

1

-----------–

I

F ending,

F ave,

20A 7.9A+( ) 2⁄ 14A≈ ≈

I

L R⁄

Isce

× 20A e

Output Waveform for

CCQP

1.5µs

--------------–

1.3µs

× 7.9A≈==

Schottky Diode Selection

The application circuit diagram of Figure 3 shows a Schottky
diode, DS1. In non-synchronous mode, DS1 is used as a fly­back diode to provide a constant current path for the inductor
when M1 is turned off. Table 10 shows the characteristics of
several Schottky diodes. Note that MBR2015CTL has a v ery
low forward voltage drop. This diode is ideal for applications
where the output voltage is required to be less than 2.8V.

Table 10. Schottky Diode Selection Table

Manufacturer

Model # Conditions

Philips
PBYR1035

Motorola
MBR2035CT

Motorola
MBR1545CT

Motorola
MBR2015CTL

IF = 20A; Tj = 25°C
IF = 20A; Tj = 125°C

IF = 20A; Tj = 25°C
IF = 20A; Tj = 125°C

IF = 15A; Tj = 25°C
IF = 15A; Tj = 125°C

IF = 20A; Tj = 25°C
IF = 20A; Tj = 150°C

Forward Voltage

V

F

< 0.84v
< 0.72v

< 0.84v
< 0.72v

< 0.84v
< 0.72v

< 0.58v
< 0.48v

Output Filter Capacitors

Output ripple performance and transient response are func­tions of the filter capacitors. Since the 5V supply of a PC
motherboard may be located several inches away from the
DC-DC converter, the input capacitance may play an impor­tant role in the load transient response of the RC5050 and
RC5051. The higher input capacitance, the more charge stor­age is available for improving current transfer through the

14

APPLICATION NOTE AN50

FET. Low Equiv alent Series Resistance (ESR) capacitors are
best suited for this type of application. Incorrect selection
can hinder the converter's overall performance. The input
capacitor should be placed as close to the drain of the FET as
possible to reduce the effect of ringing caused by long trace
lengths.

The ESR rating of a capacitor is a difficult number to
quantify. ESR is defined as the resonant impedance of the
capacitor. Since the capacitor is actually a complex imped­ance device having resistance, inductance, and capacitance,
it is natural for this device to have a resonant frequenc y. As a
rule, the lower the ESR, the better suited the capacitor is for
use in switching power supply applications. Many capacitor
manufacturers do not supply ESR data. A useful estimate of
the ESR can be obtained using the following equation:

ESR

DF

-------------=
2πfC

where DF is the dissipation factor of the capacitor, f is the
operating frequency, and C is the capacitance in farads.

With this in mind, correct calculation of the output capaci­tance is crucial to the performance of the DC-DC converter.
The output capacitor determines the overall loop stability,
output voltage ripple, and load transient response. The calcu­lation is as follows:

I

∆T×

O

C µF( )

--------------------------------------=
∆V I

ESR×–

O

For I

= 12.2A (0-13A load step) and ∆V = 100mV, the bulk

O

capacitance required can be approximated as follows:

C µF( )

∆T×

O

-------------------------------------­∆V I

ESR×–

O

12.2A 2µs×

--------------------------------------------------------------- 2870µF= = =
100mV 12.2A 7.5mΩ×–

I

Because the control loop response of the controller is not
instantaneous, the initial load transient must be supplied
entirely by the output capacitors. The initial voltage deviation
is determined by the total ESR of the capacitors used and the
parasitic resistance of the output traces. For a detailed analysis
of capacitor requirements in a high-end microprocessor
system, please refer to Application Bulletin 5.

Input Filter

The DC-DC converter should include an input inductor
between the system +5V supply and the converter input as
described below. This inductor serves to isolate the +5V
supply from the noise in the switching portion of the
DC-DC converter, and to limit the inrush current into the
input capacitors during power up. A value of 2.5µH is rec-
ommended, as illustrated in Figure 14.

5V Vin

0.1µF

2.5µH

1000µF, 10V
Electrolytic

65-AP42-17

where ∆V is the maximum voltage deviation due to load

Figure 14. Input Filter

transients, ∆T is the reaction time of the power source (loop
response time for the RC5050 and RC5051 isapproximately
2µs), and I

is the output load current.

O

Bill of Material

Table 11 is the Bill of Material for the Application Circuits
of Figure 3 and Figure 4.

Table 11. Bill of Materials for a 13A Pentium Pro Klamath Application

Quantity Reference Manufacturer Part

Order #

7 C4, C5, C7,

C8, C9, C10,

Panasonic
ECU-V1H104ZFX

0.1µF 50V capacitor

C11

1 C6 Panasonic

4.7µF 16V capacitor

ECSH1CY475R

1 Cext Panasonic

120pF capacitor

ECU-V1H121JCG

1 C12 Panasonic

1µF 16V capacitor

ECSH1CY105R

3 C1, C2, C3 United Chemi-con

LXF16VB102M

4 C13, C14,

C15, C16

1 DS1

(note 1)

Sanyo
6MV1500GX

Motorola
MBR2015CT

1000µF 6.3V electrolytic
capacitor 10mm x 20mm

1500µF 6.3V electrolytic
capacitor 10mm x 20mm

Shottky diode, 15A Vf < 0.52V @ I

1 D1 Motorola 1N4691 6.2V Zener Diode

Description Requirements and

Comments

ESR < 0.047 Ω

ESR < 0.047 Ω

= 10A

f

15

AN50 APPLICATION NOTE

Table 11. Bill of Materials for a 13A Pentium Pro Klamath Application (continued)

Quantity Reference Manufacturer Part

Order #

1 L1 Pulse Engineering

1.3µH inductor

Description Requirements and

Comments

PE-53680

1 L2* Pulse Engineering

PE-53681

2-4
(note 2)

M1-M4 International Rectifier

IRF7413

1 Rsense Coppel

2.5µH inductor *Optional—helps
reduce ripple on 5v line

N-Channel Logic Level
Enhancement Mode MOSFET

R

DS,ON

V

= 4.5V, ID = 5A

GS

6 mΩ, 1W

CuNi Wire resistor

1 R5 Panasonic

47Ω 5% resistors

ERJ-6GEY050Y

1 R6 Panasonic

10KΩ 5% resistor

ERJ-6ENF10.0KY

U1 Raytheon

RC5050M or RC5051M

Refer to Appendix A for Directory of component suppliers.
Notes:

1. When used in synchronous mode, a 1A schottky diode such as the 1N5817 should be substituted for the MBR2015CT.

2. A target R

value of 10mΩ should be used for each output driver switch. Refer to Table 3 for alternative MOSFETs.

DS,ON

PCB Layout Guidelines and
Considerations

PCB Layout Guidelines

• Placement of the MOSFETs relative to the RC5050 is
critical. Place the MOSFETs (M1 & M2) so that the trace
length of the HIDRV pin from the RC5050 to the FET
gates is minimized. A long lead length on this pin would
cause high amounts of ringing due to the inductance of the

Programmable DC-DC
converter

trace and the large gate capacitance of the FET. This noise
radiates all throughout the board, and, because it is
switching at such a high voltage and frequency, it is very
difficult to suppress.

Figure 15 shows an example of good placement for the
MOSFETs in relation to the RC5050. In addition, this fig­ure shows an example of problematic placement for the
MOSFETs.

< 18mΩ

16

M1

M2

Good layout

RC5050

11
12
13

14
15
16
17
18
19
20

10

9
8

7
6
5
4
3
2
1

“Quiet" Pins=

Figure 15. Placement of the MOSFETs

Bad layout

RC5050

11
12
13

14
15
16
17
18
19
20

10

9
8

7
6
5
4
3
2
1

M1

M2

APPLICATION NOTE AN50

In general, all of the noisy switching lines should be kept
away from the quiet analog section of the RC5050. That is,
traces that connect to pins 12 and 13 (HIDRV and
VCCQP) should be kept far away from the traces that con­nect to pins 1 through 5, and pin 16.

• Place the 0.1µF decoupling capacitors as close to the

RC5050 pins as possible. Extra lead length negates their
ability to suppress noise.

• Each VCC and GND pin should have its own via to the

appropriate plane. This helps to provide isolation between
pins.

• Surround the CEXT timing capacitor with a ground trace.

Be sure to place a ground or power plane under the
capacitor for further noise isolation to provide additional
shielding to the oscillator pin 1 from the noise on the
PCB. In addition, place this capacitor as close to the
RC5050 pin 1 as possible.

• Place the MOSFETs, inductor and Schottky as close

together as possible for the same reasons on the first bullet
above. Place the input bulk capacitors as close to the
drains of MOSFETs as possible. In addition, placement of
a 0.1µF decoupling capacitor right on the drain of each
MOSFET helps to suppress some of the high frequency
switching noise on the input of the DC-DC converter.

• Place the output bulk capacitors as close to the CPU as
possible to optimize their ability to supply instantaneous
current to the load in the event of a current transient.
Additional space between the output capacitors and the
CPU allows the parasitic resistance of the board traces to
degrade the DC-DC conv erter’ s performance under se vere
load transient conditions, causing higher voltage
deviation. For more detailed information regarding
capacitor placement, refer to Application Bulletin AB-5.

• The traces that run from the RC5050 IFB (pin 4) and VFB
(pin 5) pins should be run next to each other and Kelvin
connected to the sense resistor. Running these lines
together prevents some of the common mode noise that is
presented to the RC5050 feedback input. Try, as much as
possible, to run the noisy switching signals (HIDRV &
VCCQP) on one layer, but use the inner layers for power
and ground only. If the top layer is being used to route all
of the noisy switching signals, use the bottom layer to
route the analog sensing signals VFB and IFB.

Example of a PC Motherboard Layout and
Gerber File.

This section shows a reference design for motherboard
implementation of the RC5050 along with the Layout Gerber
File and Silk Screen. The actual PCAD Gerber File can be
obtained from Raytheon Electronics local Sales Office or
from the Semiconductor Division Marketing department at
415-966-7819.

17

AN50 APPLICATION NOTE

Guidelines for Debugging and
Performance Evaluations

Debugging Your First Design Implementation

1. Note the setting of the VID pins to know what voltage is
to be expected.

2. Do not connect any load to the circuit. While monitoring
the output voltage, apply power to the part with current
limiting at the power supply. This ensures that no cata­strophic shorts are present.

3. If proper voltage is not achieved go to "Procedures "
below.

4. When you have proper voltage, increase the current lim­iting of the power supply to 16A.

5. Apply load at 1A increments. An active load (HP6060B
or equivalent) is suggested.

6. In case of poor regulation refer to "Procedures" below.

Procedures

1. If there is no voltage at the output and the circuit is not
drawing current look for openings in the connections,
check the circuitry versus schematic, and check the
power supply pins at the device to make sure that volt­age(s) are applied.

2. If there is no voltage at the output and the circuit is
drawing excessive current (>100mA) with no load,
check for possible shorts. Determine the path of the
excessive current and which devise is drawing it—this
current may be drawn by peripheral components.

3. If the output voltage comes close to the expected value,
check the VID inputs at the device pins. The part is fac­tory set to correspond to the VID inputs.

4. Premature shut down can be caused by an inappropriate
value of the sense resistor. See the “Sense Resistor” sec­tion.

5. Poor load regulation can be due to many causes. Check
the voltages and signals at the critical pins.

6. The VREF pin should be at the voltage set by the VID
pins. If the power supply pins and the VID pins are
correct the VREF should have the correct voltage.

7. Next check the oscillator pin. You should see a saw tooth
wave at the frequency set by the external capacitor.

8. When the VREF and CEXT pins are checked and
correct and the output voltage is incorrect, look at the
waveform at VCCQP. This pin should be swinging from
ground to +12V (in the +12V application), and from
slightly below +5V to about +10V (charge pump appli­cation). If the VCCQP pin is noisy, with ripples/over­shoots riding on it this may make the converter not to
function correctly.

9. Next, look at HIDRV pin. This pin directly drives the
gate of the FET. It should provide a gate drive (Vgs) of
about 5V when turning the FET on. A careful study of
the layout is recommended. Refer to the “PCB Layout
Guidelines” section.

10. Past experience shows that the most frequent errors are
incorrect components, improper connections, and poor
layout.

Performance Evaluation

This section shows a sample ev aluation results as a reference
guide for evaluating a DC-DC Converter using the RC5050
on a Pentium Pro motherboard.

Load Regulation

VID I

10100 0.5 3.0904

Load Regulation 0.5A – 9.9A 0.70%

VID I

10010 0.5 3.2805

Load Regulation 0.5A – 12.4A 0.64%

(A) V

load

1.0 3.0825

2.0 3.0786

3.0 3.0730

4.0 3.0695

5.0 3.0693

6.0 3.0695

7.0 3.0695

8.0 3.0694

9.0 3.0694

9.9 3.0691

(A) V

load

1.0 3.2741

2.0 3.2701

3.0 3.2642

4.0 3.2595

5.0 3.2597

6.0 3.2606

7.0 3.2611

8.0 3.2613

9.0 3.2611

10.0 3.2607

11.0 3.2599

12.0 3.2596

12.4 3.2596

out

out

(V)

(V)

18

APPLICATION NOTE AN50

VID I

(A) V

load

out

(V)

11010 0.5 2.505

1.0 2.504

2.0 2.501

3.0 2.496

4.0 2.493

5.0 2.493

6.0 2.492

7.0 2.492

8.0 2.491

9.0 2.490

10.0 2.989

11.0 2.488

12.0 2.486

13.0 2.485

13.9 2.484

Load Regulation 0.5A – 13.9A 0.84%

Note:
Load regulation is expected to be typically around 0.8%. The
load regulation performance for this device under evaluation
is excellent.

Output Voltage Load Transients Due to Load Current Step

This test is performed using Intel P6.0/P6S/P6T Voltage
Transient Tester.

Low to High
Current Step

0.5A-9.9A -76.0mV Refer to
Attachment
A for Scope
Picture

High to Low
Current Step

9.9A-0.5A +70mV Refer to
Attachment
B for Scope
Picture

Low to High
Current Step

0.5A-12.4A -97.6mV Refer to
Attachment
C for Scope
Picture

High to Low
Current Step

12.4A-0.5A +80.0mV Refer to
Attachment
D for Scope
Picture

Low to High
Current Step

0.5A-13.9A -99.2mV Refer to
Attachment
E for Scope
Picture

High to Low
Current Step

13.9A-0.5A +105.2mV Refer to
Attachment
F for Scope
Picture

Note:
Transient voltage is recommended to be less than 4% of the

output voltage. The performance of the device under evalua­tion is significantly better than a typical VRM.

19

AN50 APPLICATION NOTE

Input Ripple and Power on Input Rush Current

Power on Input Rush Current was not measured on the moth­erboard because we did not want to cut the 5V trace and

I

= 9.9A Input Ripple

load

Voltage = 15mV

Refer to Attach­ment G for Scope
Picture

Note:
Excellent input ripple voltage. Input ripple voltage is recom-

mended to be less than 5% of the output voltage.

Component Case Temperature

Case Temperature

I

load

Device Description

Q3A MOSFET

insert a current probe in series with the supply. However,
with the input filter design, the Input Rush Current is well
within specification.

Case Temperature

= 9.9A

(°C)

I

load

= 12.4A
(°C)

57 63 66.3

K1388

Q3B MOSFET

58 64 66.6

K1388

L1 Inductor,

53 56 61.2

Unknown

Q2 Schottky Diode

66 70 87

2048CT
IC Raytheon’s RC5050 52 54 58
Cin Input Cap. 1000µF 38.2 36.8 39
Cout Output Cap.

35 34.8 38.2

1500µF

Case Temperature

I

= 13.9A

load

(°C)

Note:

The values for case temperatures are within guidelines. That is, case temperatures for all components should be below

105°C @25°C Ambient.

Evaluation Summary

The on-board DC-DC converter is fully functional. It has
excellent load regulation, transient response, and input volt­age ripple.

Attachment A

Attachment B

20

APPLICATION NOTE AN50

Attachment C

Attachment D

Attachment E

Attachment F

Attachment G

21

AN50 APPLICATION NOTE

Summary

This application note covers many aspects of the RC5050
and RC5051 for implementation of a DC-DC converter a on
Pentium Pro motherboard. A detailed discussion includes
the processor power requirements, a description of the
RC5050 and RC5051, design considerations and compo­nents selection, layout guidelines and considerations, guide­lines for debugging, and performance evaluations.

RC5050 Evaluation Board

Raytheon Electronics provides an evaluation board to verify­ing system level performance of the RC5050. The e v aluation
board serves as a guide to performance expectations when
using the supplied external components and PCB layout.

Call Raytheon Electronics local Sales Office or the Market­ing department at 415-966-7819 for an evaluation board.

22

APPLICATION NOTE AN50

Appendix A

Directory of Component Suppliers

Dale Electronics, Inc.
E. Hwy. 50, PO Box 180
Yankton, SD 57078-0180
PH: [phone]

National Semiconductor
2900 Semiconductor Drive
Santa Clara, CA 95052-8090
PH: [phone]

Fuji Electric
Collmer Semiconductor Inc.
14368 Proton Rd.
Dallas, Texas 75244
PH: [phone]

General Instrument
Power Semiconductor Division
10 Melville Park Road
Melville, NY 11747
PH: [phone]

Hoskins Manufacturing Co.
(Copel Resistor Wire)
10776 Hall Road
Hamburg, MI 48139-0218
PH: [phone]

Intel Corp.
5200 NE Elam Young Pkwy.
Hillsboro, OR. 97123
PH: [phone] Tech. Support
for Power Validator

International Rectifier
233 Kansas St.
El Segundo, CA 90245
PH: [phone]

IRC Inc.
PO Box 1860
Boone, NC 28607
PH: [phone]

Motorola Semiconductors
PO Box 20912
Phoenix, Arizona 85036
PH:[phone]

Nihon Inter Electronics Corp.
Quantum Marketing Int’l, Inc.
12900 Rolling Oaks Rd.
Caliente, CA 93518
PH: [phone]

Panasonic Industrial Co.
6550 Katella Avenue
Cypress, CA 90630
PH: [phone]

Pulse Engineering
12220 World Trade Drive
San Diego, CA 92128
PH: [phone]

Sanyo Energy USA
2001 Sanyo Avenue
San Diego, CA 92173
PH: [phone]

Siliconix
Temic Semiconductors
2201 Laurelwood Road
Santa Clara, CA 95056-1595
PH: [phone]

Sumida Electric USA
5999 New Wilke Road Suite #110
Rolling Meadows, IL 60008
PH: [phone]

Xicon Capacitors
PO Box 170537
Arlington, Texas 76003
PH:[phone]

23

AN50 APPLICATION NOTE

LIFE SUPPORT POLICY

FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:

1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonable expected to result in a significant injury of the
user.

Fairchild Semiconductor
Corporation

Americas

Customer Response Center
Tel:[phone]

www.fairchildsemi.com

Fairchild Semiconductor
Europe

Fax: [phone]

Email: [e-mail]
Deutsch Tel: [phone]
English Tel: [phone]
Italy Tel: [phone]

2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.

Fairchild Semiconductor
Hong Kong Ltd.

13th Floor, Straight Block,
Ocean Center, 5 Canto Rd.
Tsimshatsui, Kowloon
Hong Kong
Tel:[phone]
Fax:[phone]

National Semiconductor
Japan Ltd.

Tel:[phone]
Fax:[phone]

2/98 0.0m

 1998 Fairchild Semiconductor Corporation

Stock#AN30000050