Diodes AP6502 User Manual

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Page 1

P6502

Y (%

Description

The AP6502 is a 340kHz switching frequency external compensated

synchronous DC/DC buck converter. It has integrated low R

high and low side MOSFETs.

The AP6502 enables continues load current of up to 2A with

DSON

Pin Assignments

( Top View )

COMP

efficiency as high as 95%.

The AP6502 features current mode control operation, which enables

fast transient response times and easy loop stabilization.

SO-8EP

GND

The AP6502 simplifies board layout and reduces space requirements

with its high level of integration and minimal need for external

components, making it ideal for distributed power architectures.

NEW PRODUCT

The AP6502 is available in a standard Green SO-8 and SO-8EP

package with exposed PAD for improved thermal performance and is

RoHS compliant.

Features

• VIN 4.7V to 18V

• 2A Continuous Output Current, 3A Peak

• V

• 340kHz Switching Frequency

• Programmable Soft-Start

• Enable Pin

• Protection

• Totally Lead-Free & Fully RoHS Compliant (Notes 1 & 2)

• Halogen and Antimony Free. “Green” Device (Note 3)

Adjustable from 0.925V to 16V

OUT

 OCP

 Thermal Shutdown

Applications

• Gaming Consoles

• Flat Screen TV Sets and Monitors

• Set Top Boxes

• Distributed power systems

• Home Audio

• Consumer Electronics

• Network Systems

• FPGA, DSP and ASIC Supplies

• Green Electronics

Figure 1 Package Pin Out

Notes: 1. No purposely added lead. Fully EU Directive 2002/95/EC (RoHS) & 2011/65/EU (RoHS 2) compliant.

2. See http://www.diodes.com for more information about Diodes Incorporated’s definitions of Halogen- and Antimony-free, "Green" and Lead-free.

3. Halogen- and Antimony-free "Green” products are defined as those which contain <900ppm bromine, <900ppm chlorine (<1500ppm total Br + Cl) and <1000ppm antimony compounds.

Typical Application Circuit

340kHz 18V 2A SYNCHRONOUS DC/DC BUCK CONVERTER

100

V = 5V

V = 12V

)

IEN

EFFI

V = 3.3V

OUT

L = 10µ H

0 0.4 0.8 1.2 1.6 2

LOAD CURRENT (A)

Effici ency vs. Load Current

AP6502

Document Number: DS35423 Rev. 9 - 2

Figure 2 Typical Application Circuit

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Page 2

Pin Descriptions

Number

Name

1 BS

2 IN

3 SW

4 GND Ground (Connect the exposed pad to Pin 4).

5 FB

NEW PRODUCT

6 COMP

7 EN

8 SS

EP EP EP exposed thermal pad connect to Pin 4 GND, not applicable in the SO-8 package.

Functional Block Diagram

P6502

Function

High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET switch. Connect a 0.01µF or greater capacitor from SW to BS to power the high side switch.

Power Input. IN supplies the power to the IC, as well as the step-down converter switches. Drive IN with a 4.7V to 18V power source. Bypass IN to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input Capacitor.

Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch.

Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage divider connected to it from the output voltage. The feedback threshold is 0.925V. See Setting the Output Voltage.

Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC network from COMP to GND. In some cases, an additional capacitor from COMP to GND is required. See Compensation Components.

Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on the regulator; low to turn it off. Attach to IN with a 100k pull up resistor for automatic startup.

Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the softstart period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the soft-start feature, leave SS floating.

COMP

AP6502

Document Number: DS35423 Rev. 9 - 2

1.1V

0.3 V

0.925 V

0.9V

2.5V

ERROR

AMPLIFIER

LOCKOUT

COMPARATOR

OVP

OSCILLATOR

100/340 KHz

EN OK

SHUTDOWN

RAMP

CLK

6uA

Figure 3 Functional Block Diagram

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CURRENT

SENSE

AMPLIFIER

Logic

CURRENT

COMPARATOR

INTERNAL

REGULATORS

100m

disable

IN < 4.10V

GND

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Page 3

Absolute Maximum Ratings (Note 4) (@T

Symbol Parameter Rating Unit

VIN

VSW

VBS

VFB

VEN

COMP

TST

NEW PRODUCT

ESD Susceptibility (Note 5)

HBM Human Body Model 3 kV

MM Machine Model 250 V

Notes: 4. Stresses greater than the 'Absolute Maximum Ratings' specified above may cause permanent damage to the device. These are stress ratings only; functional operation of the device at these or any other conditions exceeding those indicated in this specification is not implied. Device reliability may be affected by exposure to absolute maximum rating conditions for extended periods of time.

5. Semiconductor devices are ESD sensitive and may be damaged by exposure to ESD events. Suitable ESD precautions should be taken when handling and transporting these devices.

Thermal Resistance (Note 6) (@T

Symbol Parameter Rating Unit

JA

JC

Note: 6. Test condition: SO-8: Device mounted on 1"x1" FR-4 substrate PCB, 2oz copper, with minimum recommended pad layout. SO-8EP: Device mounted on 1" x 1" FR-4 substrate PC board, 2oz copper, with minimum recommended pad on top layer and thermal vias to bottom layer ground plane.

Supply Voltage

Switch Node Voltage

Bootstrap Voltage

Feedback Voltage -0.3 to +6 V

Enable/UVLO Voltage -0.3 to +6 V

Comp Voltage -0.3 to +6 V

Storage Temperature -65 to +150 °C

Junction Temperature +150 °C

Lead Temperature +260 °C

= +25°C, unless otherwise specified.)

Junction to Ambient

Junction to Case

Recommended Operating Conditions (Note 7) (@T

Symbol Parameter Min Max Unit

VIN

Note: 7. The device function is not guaranteed outside of the recommended operating conditions.

AP6502

Document Number: DS35423 Rev. 9 - 2

Supply Voltage

Operating Ambient Temperature Range

= +25°C, unless otherwise specified.)

-0.3 to +20

-1.0 to V

SO-8EP 74

SO-8 126

SO-8EP 16

SO-8 28

= +25°C, unless otherwise specified.)

4.7

-40

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+0.3

-0.3 to VSW +6

18 V

+85 °C

P6502

°C/W

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Electrical Characteristics (V

Symbol Parameter Test Conditions Min Typ Max Unit

IIN

DS(ON)1

DS(ON)2

LIMIT

Shutdown Supply Current

Supply Current (Quiescent)

High-Side Switch On-Resistance (Note 8) 130 m

Low-Side Switch On-Resistance (Note 8) 130 m

HS Current Limit Minimum duty cycle 4.4 A

LS Current Limit From Drain to Source 0.9 A

High-Side Switch Leakage Current

AVEA Error Amplifier Voltage Gain

NEW PRODUCT

GEA

GCS

FSW

FFB

MAX

TON

VFB

(Note 8)

Error Amplifier Transconductance

COMP to Current Sense Transconductance

Oscillator Frequency

Fold-back Frequency

Maximum Duty Cycle

Minimum On Time 130 ns

Feedback Voltage

Feedback Overvoltage Threshold 1.1 V

EN_Rising

EN Rising Threshold 0.7 0.8 0.9 V

EN Lockout Threshold Voltage 2.2 2.5 2.7 V

EN Lockout Hysteresis 220 mV

INUV

Vth

VIN Under Voltage Threshold Hysteresis

HYS

Under Voltage Threshold Rising

Soft-Start Current

Soft-Start Period

TSD

Note: 8. Guaranteed by design

Thermal Shutdown (Note 8) 160 °C

AP6502

Document Number: DS35423 Rev. 9 - 2

= 12V, @TA = +25°C, unless otherwise specified.)

VEN = 0V

VEN = 2.0V, VFB = 1.0V

= 0V, VSW = 0V,

= 12V

800 V/V

I

= ±10µA

2.8 A/V

= 0.75V

= 0V

VFB = 800mV

TA = -40°C to +85°C

3.80 4.05 4.40 V

250 mV

= 0V

= 0.1µF

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P6502

0.3 3.0 µA

0.6 1.5 mA

0 10 A

1000 µA/V

300 340 380 kHz

0.30

fSW

90 %

900 925 950 mV

6 A

15 ms

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OLT

Typical Performance Graphs (V

0.6

= 12V, @TA = +25°C, unless otherwise specified.)

0.074

P6502

NEW PRODUCT

0.58

0.56

0.54

0.52

0.5

QUIESCENT SUPPLY CURRENT (mA)

0.48 0 5 10 15 20

Quiescent Supply Current vs. Input Voltage

6.2

5.8

5.6

5.4

5.2

CURRENT LIMIT (A)

4.8

-60 -40 -20 0 20 40 60 80 100

0.92

0.918

0.916

E (V)

0.914

0.912

INPUT VOLTAGE (V)

TEMPERATURE (C)

Current Limit vs. Temperature

0.064

0.054

0.044

0.034

0.024

0.014

SHUTDOWN SUPPLY CURRENT (µA)

0.004 0 5 10 15 20

OUTPUT VOLTAGE (V)

Shutdown Supply Current vs. Input Voltage

3.33

3.329

3.328

3.327

3.326

3.325

3.324

3.323

3.322

3.321

3.32 4101520

375

370

365

INPUT VOLTAGE (V)

V = 12V

INPUT VOLTAGE (V)

Line Regulation

0.91

K V

0.908

0.906

FEEDBA

0.904

0.902

0.9

-60 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C)

Feedback Voltage vs. Temperature

AP6502

Document Number: DS35423 Rev. 9 - 2

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360

355

OSCILLATOR FREQUENCY (Khz)

350

-60 -40 -20 0 20 40 60 80 100 TEMPERATURE (°C)

Oscillator Frequency vs. Temperature

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CIENCY

Typical Performance Graphs (cont.) (V

NEW PRODUCT

(%)

IEN

EFFI

V = 1.2V

OUT

L = 3.3µH

0 0.4 0.8 1.2 1.6 2

LOAD CURRENT (A)

Efficiency vs. Load Current

100

V = 12V

= 12V, V

= 3.3V, (@TA = +25°C, unless otherwise specified.)

OUT

P6502

V = 5V

V = 12V

(%)

IEN

EFFI

V = 1.8V

OUT

L = 3.3µH

0 0.4 0.8 1.2 1.6 2

LOAD CURRENT (A)

Efficiency vs. Load Current

(%)

EFFI

V = 12V

V = 5V

OUT

L = 10µH

020.4 0.8 1.2 1.6 LOAD CURRENT (A)

Efficiency vs. Load Current

AP6502

Document Number: DS35423 Rev. 9 - 2

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Typical Performance Characteristics

(VIN = 12V, V

= 3.3V, L = 10µH, C1 = 22µF, C2 = 47µF, @TA = +25°C, unless otherwise specified.)

OUT

Steady State Test no load

Steady State Test 2A

P6502

Startup Through Enable_no load

NEW PRODUCT

Time -2µs/div

Startup Through Enable 2A

Time -2ms/div

Load Transient Test 1.0A to 2.0A

Time -2µs/div

Shutdown Through Enable_no load

Time -10ms/div

Short Circuit Test

Time -10ms/div

Shutdown Through Enable 2A

Time -5ms/div

Short Circuit Recovery

Time -100µs/div

AP6502

Document Number: DS35423 Rev. 9 - 2

Time -20µs/div

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Time -20µs/div

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Applications Information

Theory of Operation

The AP6502 is a 2A current mode control, synchronous buck regulator with built in power MOSFETs. Current mode control assures excellent

line and load regulation and a wide loop bandwidth for fast response to load transients. Figure 3 depicts the functional block diagram of

AP6502.

The operation of one switching cycle can be explained as follows. At the beginning of each cycle, HS (high-side) MOSFET is off. The error

amplifier (EA) output voltage is higher than the current sense amplifier output, and the current comparator’s output is low. The rising edge of

the 340kHz oscillator clock signal sets the RS Flip-Flop. Its output turns on HS MOSFET. The current sense amplifier is reset for every

switching cycle.

When the HS MOSFET is on, inductor current starts to increase. The current sense amplifier senses and amplifies the inductor current. Since

the current mode control is subject to sub-harmonic oscillations that peak at half the switching frequency, ramp slope compensation is utilized.

This will help to stabilize the power supply. This ramp compensation is summed to the current sense amplifier output and compared to the

error amplifier output by the PWM comparator. When the sum of the current sense amplifier output and the slope compensation signal

NEW PRODUCT

exceeds the EA output voltage, the RS Flip-Flop is reset and HS MOSFET is turned off.

For one whole cycle, if the sum of the current sense amplifier output and the slope compensation signal does not exceed the EA output, then

the falling edge of the oscillator clock resets the Flip-Flop. The output of the error amplifier increases when feedback voltage (VFB) is lower

than the reference voltage of 0.925V. This also increases the inductor current as it is proportional to the EA voltage.

If in one cycle the current in the power MOSFET does not reach the COMP set current value, the power MOSFET will be forced to turn off.

When the HS MOSFET turns off, the synchronous LS MOSFET turns on until the next clock cycle begins. There is a “dead time” between the

HS turn off and LS turn on that prevents the switches from “shooting through” from the input supply to ground.

The voltage loop is compensated through an internal transconductance amplifier and can be adjusted through the external compensation

components.

Enable

Above the ‘EN Rising Threshold’, the internal regulator is turned on and the quiescent current can be measured above this threshold. The

enable (EN) input allows the user to control turning on or off the regulator. To enable the AP6502, EN must be pulled above the ‘EN Lockout

Threshold Voltage’ and to disable the AP6502, EN must be pulled below ‘EN Lockout Threshold Voltage - EN Lockout Hysteresis’

(2.2V-0.22V =1.98V).

External Soft Start

Soft start is traditionally implemented to prevent the excess inrush current. This in turn prevents the converter output voltage from

overshooting when it reaches regulation. The AP6502 has an internal current source with a soft start capacitor to ramp the reference voltage

from 0V to 0.925V. The soft start current is 6uA. The soft start sequence is reset when there is a Thermal Shutdown, Under Voltage Lockout

(UVLO) or when the part is disabled using the EN pin.

External Soft Start can be calculated from the formula below:

Where;

= Soft Start Current

C = External Capacitor

DV = change in feedback voltage from 0V to maximum voltage

DT = Soft Start Time

I =

Current Limit Protection

In order to reduce the total power dissipation and to protect the application, AP6502 has cycle-by-cycle current limiting implementation. The

voltage drop across the internal high-side MOSFET is sensed and compared with the internally set current limit threshold. This voltage drop is

sensed at about 30ns after the HS turns on. When the peak inductor current exceeds the set current limit threshold, current limit protection is

activated. During this time the feedback voltage (VFB) drops down. When the voltage at the FB pin reaches 0.3V, the internal oscillator shifts

the frequency from the normal operating frequency of 340kHz to a fold-back frequency of 102kHz. The current limit is reduced to 70% of

nominal current limit when the part is operating at 102kHz. This low fold-back frequency prevents runaway current.

AP6502

Document Number: DS35423 Rev. 9 - 2

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P6502

Page 9

Applications Information (cont.)

Under Voltage Lockout (UVLO)

Under Voltage Lockout is implemented to prevent the IC from insufficient input voltages. The AP6502 has a UVLO comparator that monitors

the input voltage and the internal bandgap reference. If the input voltage falls below 4.0V, the AP6502 will latch an under voltage fault. In this

event the output will be pulled low and power has to be re-cycled to reset the UVLO fault.

Over Voltage Protection

When the AP6502 FB pin exceeds 20% of the nominal regulation voltage of 0.925V, the over voltage comparator is tripped and the COMP

pin and the SS pin are discharged to GND, forcing the high-side switch off.

Thermal Shutdown

The AP6502 has on-chip thermal protection that prevents damage to the IC when the die temperature exceeds safe margins. It implements a

thermal sensing to monitor the operating junction temperature of the IC. Once the die temperature rises to approximately +160°C, the thermal

protection feature gets activated. The internal thermal sense circuitry turns the IC off thus preventing the power switch from damage.

NEW PRODUCT

A hysteresis in the thermal sense circuit allows the device to cool down to approximately +120°C before the IC is enabled again through soft

start. This thermal hysteresis feature prevents undesirable oscillations of the thermal protection circuit.

Setting the Output Voltage

The output voltage can be adjusted from 0.925V to 16V using an external resistor divider. Table 1 shows a list of resistor selection for

common output voltages. Resistor R1 is selected based on a design tradeoff between efficiency and output voltage accuracy. For high

values of R1 there is less current consumption in the feedback network. However the trade off is output voltage accuracy due to the bias

current in the error amplifier. R1 can be determined by the following equation:

⎜

0.925

⎝

⎛

OUT

−⋅= 1

⎞ ⎟

⎟ ⎠

P6502

When output voltage is low, network as shown in Figure 4 is recommended.

Table 1 – Resistor Selection for Common Output Voltages

(V)

OUT

5 45.3 10

3.3 26.1 10

2.5 16.9 10

1.8 9.53 10

1.2 3 10

R1 (kΩ) R2 (kΩ)

Figure 4 Feedback Divider Network

Compensation Components

The AP6502 has an external COMP pin through which system stability and transient response can be controlled. COMP pin is the output of

the internal trans-conductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the

characteristics of the control system. The DC gain of the voltage feedback loop is given by:

OUT

is the

VEA

January 2013

Where V

error amplifier voltage gain.

is the feedback voltage (0.925V), R

AP6502

Document Number: DS35423 Rev. 9 - 2

A ×××=

LOAD

VDC

LOAD

is the load resistor value, GCS is the current sense trans-conductance and A

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VEA

Page 10

Applications Information (cont.)

Compensation Components (cont.)

The control loop transfer function incorporates two poles one is due to the compensation capacitor (C3) and the output resistor of error

amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at:

OUT

/C2

<×

××π

A3C2

××π

R2C2

××π

RC/R3

(kΩ)

VEA

LOAD

3R3C2

fc3R

×××π

CC/C3

(nF)

fs1.02C2

OUT

(µH)

, to below one fourth of the

Where G

One zero is present due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:

NEW PRODUCT

The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency

where the feedback loop has the unity gain is crucial.

A rule of thumb is to set the crossover frequency to below one-tenth of the switching frequency. Use the following procedure to optimize the

compensation components:

1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation:

Where f

2. Choose the compensation capacitor (C3) to achieve the desired phase margin set the compensation zero, f

crossover frequency to provide sufficient phase margin. Determine the C3 value by the following equation:

is the error amplifier trans-conductance.

××π

is the crossover frequency, which is typically less than one tenth of the switching frequency.

fc2C2

Where R3 is the compensation resistor value.

CIN/C1

OUT

(V)

(µF)

1.2 22 47 3.24 6.8 3.3

1.8 22 47 6.8 6.8 3.3

2.5 22 47 6.8 6.8 10

3.3 22 47 6.8 6.8 10 5 22 47 6.8 6.8 10

12 22 47 6.8 6.8 15

Table 2 – Recommended Component Selection

OUT

(µF)

Inductor

Calculating the inductor value is a critical factor in designing a buck converter. For most designs, the following equation can be used to

calculate the inductor value;

)

OUT

I

Where

And

Choose the inductor ripple current to be 30% of the maximum load current. The maximum inductor peak current is calculated from:

AP6502

Document Number: DS35423 Rev. 9 - 2

is the inductor ripple current.

is the buck converter switching frequency.

I +=

L(MAX)

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LOAD

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−⋅

OUT

I

fLI

⋅⋅

P6502

January 2013

Page 11

Applications Information (cont.)

Inductor (cont.)

Peak current determines the required saturation current rating, which influences the size of the inductor. Saturating the inductor decreases

the converter efficiency while increasing the temperatures of the inductor and the internal MOSFETs. Hence choosing an inductor with

appropriate saturation current rating is important.

A 1µH to 10µH inductor with a DC current rating of at least 25% percent higher than the maximum load current is recommended for most

applications.

For highest efficiency, the inductor’s DC resistance should be less than 200m. Use a larger inductance for improved efficiency under light

load conditions.

Input Capacitor

The input capacitor reduces the surge current drawn from the input supply and the switching noise from the device. The input capacitor has

to sustain the ripple current produced during the on time on the upper MOSFET. It must hence have a low ESR to minimize the losses.

NEW PRODUCT

The RMS current rating of the input capacitor is a critical parameter that must be higher than the RMS input current. As a rule of thumb,

select an input capacitor which has RMs rating that is greater than half of the maximum load current.

Due to large dI/dt through the input capacitors, electrolytic or ceramics should be used. If a tantalum must be used, it must be surge

protected. Otherwise, capacitor failure could occur. For most applications, a 4.7µF ceramic capacitor is sufficient.

Output Capacitor

The output capacitor keeps the output voltage ripple small, ensures feedback loop stability and reduces the overshoot of the output voltage.

The output capacitor is a basic component for the fast response of the power supply. In fact, during load transient, for the first few

microseconds it supplies the current to the load. The converter recognizes the load transient and sets the duty cycle to maximum, but the

current slope is limited by the inductor value.

Maximum capacitance required can be calculated from the following equation:

ESR of the output capacitor dominates the output voltage ripple. The amount of ripple can be calculated from the equation below:

ESR*

)

V)V V(

−+

out

L(I

I

out

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inductor

I

inductor

Vout =

capacitor

An output capacitor with ample capacitance and low ESR is the best option. For most applications, a 22µF ceramic capacitor will be

sufficient.

Where

V is the maximum output voltage overshoot.

AP6502

Document Number: DS35423 Rev. 9 - 2

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P6502

January 2013

Page 12

Applications Information (cont.)

PC Board Layout

This is a high switching frequency converter. Hence attention must be paid to the switching currents interference in the layout. Switching

current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and

circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces.

NEW PRODUCT

P6502

AP6502SP-13 is exposed at the bottom of the package and must be soldered directly to a well designed thermal pad on the PCB. This will

help to increase the power dissipation. This is not applicable for the AP6502S-13.

External Bootstrap Diode

It is recommended that an external bootstrap diode be added when the input voltage is no greater than 5V or the 5V rail is available in the

system. This helps to improve the efficiency of the regulator. This solution is also applicable for D > 65%. The bootstrap diode can be a low

cost one such as BAT54 or a schottky that has a low Vf.

Recommended Diodes:

Part Number

B130 30V, 1A Diodes Inc

SK13 30V, 1A Diodes Inc

Voltage/Current

Rating

Figure 7—External Bootstrap

Compensation Components

Vendor

AP6502

Document Number: DS35423 Rev. 9 - 2

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Page 13

Ordering Information

P6502

Device

Package

Code

Packaging

Quantity Part Number Suffix

13” Tape and Reel

AP6502SP-13 SP SO-8EP 2500/Tape & Reel -13

AP6502S-13 S SO-8 2500/Tape & Reel -13

NEW PRODUCT

Marking Information

Package Outline Dimensions (All dimensions in mm.)

(1) SO-8EP

9° (All si des)

AP6502

Document Number: DS35423 Rev. 9 - 2

4° ± 3°

Bottom View

www.diodes.com

Exposed Pad

Gauge Plane

Seating Plane

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SO-8EP (SOP-8L-EP)

Dim Min Max Typ

A 1.40 1.50 1.45

A1 0.00 0.13 -

b 0.30 0.50 0.40 C 0.15 0.25 0.20 D 4.85 4.95 4.90

E 3.80 3.90 3.85 E0 3.85 3.95 3.90 E1 5.90 6.10 6.00

e - - 1.27 F 2.75 3.35 3.05 H 2.11 2.71 2.41 L 0.62 0.82 0.72 N - - 0.35 Q 0.60 0.70 0.65

All Dimensions in mm

January 2013

Page 14

Package Outline Dimensions (cont.) (All dimensions in mm.)

(2) SO-8

NEW PRODUCT

Detail ‘A’

Suggested Pad Layout (All dimensions in mm.)

(1) SO-8EP

(2) SO-8

AP6502

Document Number: DS35423 Rev. 9 - 2

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0.254 Gauge Plane

Seating Plane

7°~9

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Detail ‘A’

Dim Min Max

Dimensions

C 1.270

X 0.802 X1 3.502 X2 4.612

Y 1.505 Y1 2.613 Y2 6.500

Dimensions Value (in mm)

X 0.60

Y 1.55 C1 5.4 C2 1.27

SO-8

A - 1.75 A1 0.10 0.20 A2 1.30 1.50 A3 0.15 0.25

b 0.3 0.5 D 4.85 4.95 E 5.90 6.10

E1 3.85 3.95

e 1.27 Typ

h - 0.35

L 0.62 0.82

All Dimensions in mm

0° 8°

Value

(in mm)

P6502

January 2013

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DIODES INCORPORATED MAKES NO WARRANTY OF ANY KIND, EXPRESS OR IMPLIED, WITH REGARDS TO THIS DOCUMENT, INCLUDING, BUT NOT LIMITED TO, THE IMPLIED WARRANTIES OF MERCHANTABILITY AND FITNESS FOR A PARTICULAR PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS OF ANY JURISDICTION).

Diodes Incorporated and its subsidiaries reserve the right to make modifications, enhancements, improvements, corrections or other changes without further notice to this document and any product described herein. Diodes Incorporated does not assume any liability arising out of the application or use of this document or any product described herein; neither does Diodes Incorporated convey any license under its patent or trademark rights, nor the rights of others. Any Customer or user of this document or products described herein in such applications shall assume all risks of such use and will agree to hold Diodes Incorporated and all the companies whose products are represented on Diodes Incorporated website, harmless against all damages.

Diodes Incorporated does not warrant or accept any liability whatsoever in respect of any products purchased through unauthorized sales channel. Should Customers purchase or use Diodes Incorporated products for any unintended or unauthorized application, Customers shall indemnify and hold Diodes Incorporated and its representatives harmless against all claims, damages, expenses, and attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized application.

NEW PRODUCT

Products described herein may be covered by one or more United States, international or foreign patents pending. Product names and markings noted herein may also be covered by one or more United States, international or foreign trademarks.

This document is written in English but may be translated into multiple languages for reference. Only the English version of this document is the final and determinative format released by Diodes Incorporated.

Diodes Incorporated products are specifically not authorized for use as critical components in life support devices or systems without the express written approval of the Chief Executive Officer of Diodes Incorporated. As used herein:

A. Life support devices or systems are devices or systems which:

1. are intended to implant into the body, or

labeling can be reasonably expected to result in significant injury to the user.

B. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or to affect its safety or effectiveness.

Customers represent that they have all necessary expertise in the safety and regulatory ramifications of their life support devices or systems, and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of Diodes Incorporated products in such safety-critical, life support devices or systems, notwithstanding any devices- or systemsrelated information or support that may be provided by Diodes Incorporated. Further, Customers must fully indemnify Diodes Incorporated and its representatives against any damages arising out of the use of Diodes Incorporated products in such safety-critical, life support devices or systems.

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2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the

IMPORTANT NOTICE

LIFE SUPPORT

P6502

AP6502

Document Number: DS35423 Rev. 9 - 2

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January 2013