Datasheets isl6327%202, isl6327 Datasheet

Page 1

ISL6327

Data Sheet May 5, 2008

Enhanced 6-Phase PWM Controller with 8-Bit VID Code and Differential Inductor DCR or Resistor Current Sensing

The ISL6327 controls microprocessor core voltage regulation by driving up to 6 synchronous-rectified buck channels in parallel. Multiphase buck converter architecture uses interleaved timing to multiply channel ripple frequency and reduce input and output ripple currents. Lower ripple results in fewer components, lower component cost, reduced power dissipation, and smaller implementation area.

Microprocessor loads can generate load transients with extremely fast edge rates. The ISL6327 utilizes Intersil’s proprietary Active Pulse Positioning (APP) and Adaptive Phase Alignment (APA) modulation scheme to achieve the extremely fast transient response with fewer output capacitors.

Today’s microprocessors require a tightly regulated output voltage position versus load current (droop). The ISL6327 senses the output current continuously by utilizing patented techniques to measure the voltage across the dedicated current sense resistor or the DCR of the output inductor. Current sensing provides the needed signals for precision droop, channel-current balancing, and overcurrent protection. A programmable integrated temperature compensation function is implemented to effectively compensate the temperature variation of the current sense element. The current limit function provides the overcurrent protection for the individual phase.

A unity gain, differential amplifier is provided for remote voltage sensing. Any potential difference between remote and local grounds can be completely eliminated using the remote-sense amplifier. Eliminating ground differences improves regulation and protection accuracy. The thresholdsensitive enable input is available to accurately coordinate the start-up of the ISL6327 with any other voltage rail. Dynamic-VID™ technology allows seamless on-the-fly VID changes. The offset pin allows accurate voltage offset settings that are independent of VID setting.

FN9276.4

Features

• Proprietary Active Pulse Positioning and Adaptive Phase Alignment Modulation Scheme

• Precision Multiphase Core Voltage Regulation

- Differential Remote Voltage Sensing

- ±0.5% System Accuracy Over Life, Load, Line and

Temperature

- Adjustable Precision Reference-Voltage Offset

• Precision Resistor

or DCR Current Sensing

- Accurate Load-Line Programming

- Accurate Channel-Current Balancing

- Differential Current Sense

• Microprocessor Voltage Identification Input

- Dynamic VID™ T echnology

- 8-Bit VID Input with Selectable VR11 code and

Extended VR10 Code at 6.25mV Per Bit

- 0.5V to 1.600V Operation Range

• Thermal Monitoring

• Integrated Programmable Temperature Compensation

• Overcurrent Protection and Channel Current Limit

• Overvoltage Protection with OVP Output Indication

• 2, 3, 4, 5 or 6 Phase Operation

• Adjustable Switching Frequency up to 1MHz Per Phase

• Package Option

- QFN Compliant to JEDEC PUB95 MO-220 QFN - Quad

Flat No Leads - Product Outline

- QFN Near Chip Scale Package Footprint; Improves

PCB Efficiency, Thinner in Profile

• Pb-Free (RoHS Compliant)

Ordering Information

PART

NUMBER

(Note)

ISL6327CRZ* ISL6327CRZ 0 to +70 48 Ld 7x7 QFN L48.7x7 ISL6327IRZ* ISL6327IRZ -40 to +85 48 Ld 7x7 QFN L48.7x7 *Add “-T” suffix for tape and reel. Please refer to TB347 for details on

reel specifications. NOTE: These Intersil Pb-free plastic packaged products employ

special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.

PART

MARKING

TEMP.

(°C)

PACKAGE

(Pb-Free)

PKG.

DWG. #

CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.

1-888-INTERSIL or 1-888-468-3774

| Intersil (and design) is a registered trademark of Intersil Americas Inc.

All other trademarks mentioned are the property of their respective owners.

Page 2

ISL6327

Pinout

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0

VRSEL

OFS

IOUT

DAC

ISL6327

(48 LD QFN)

TOP VIEW

VR_HOT

48 47 46 45 44 43 42 41 40 39 38 37 1 2

3 4 5 6 7 8 9

10 11 12

13 14 15 16 17 18 19 20 21 22 23 24

VR_RDY

VR_FAN

GND

EN_VTT

EN_PWR

ISEN6-

ISEN6+

PWM6

PWM3

ISEN3+

ISEN3-

ISEN1-

ISEN1+

PWM1

PWM4

ISEN4+

ISEN4-

ISEN2-

ISEN2+

PWM2

COMP

IDROOP

VDIFF OVP

VSEN

RGND

VCC

TCOMP

ISEN5-

PWM5

ISEN5+

REF

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May 5, 2008

Page 3

ISL6327 Block Diagram

VDIFF

VR_RDY

OVP

ISL6327

VCC

RGND

VSEN

OFS

REF

DAC

VRSEL

VID7

VID6

OVP

+175MV

OFFSET

SOFT-START

AND

FAULT LOGIC

OVP

DRIVE

POWER-ON

CLOCK AND

RAMP

GENERATOR

RESET (POR)

APP AND APA

MODULATOR

APP AND APA

MODULATOR

APP AND APA

MODULATOR

APP AND APA

MODULATOR

THREE-STATE

0.875V EN_VTT

0.875V EN_PWR

PWM1

PWM2

PWM3

PWM4

VID5

VID4

VID3 VID2 VID1

VID0

COMP

IOUT

IDROOP

DYNAMIC

VID D/A

APP AND APA

MODULATOR

E/A

CHANNEL CURRENT

BALANCE AND

CURRENT LIMIT

OC2

OC1

I_TOT

I_TRIP

1 N

THERMAL

MONITORING

∑

APP AND APA

MODULATOR

CHANNEL

DETECT

TEMPERATURE

COMPENSATION

TEMPERATURE

COMPENSATION

GAIN

CHANNEL CURRENT

SENSE

PWM5

PWM6

ISEN1+ ISEN1-

ISEN2+ ISEN2ISEN3+

ISEN3ISEN4+

ISEN4ISEN5+ ISEN5-

ISEN6+ ISEN6-

GND

TM VR_HOTVR_FAN

TCOMP

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May 5, 2008

Page 4

ISL6327

Typical Application - 6-Phase Buck Converter with DCR Sensing and External TCOMP

VTT

VR_RDY

VID7 VID6 VID5 VID4

VID3 VID2

VID1 VID0

VRSEL

OVP

VR_FAN

VR_HOT

+5V

IOUT

NTC2

NETWORK

FB COMP

IDROOP VDIFF VSEN

RGND EN_VTT

ISL6327

IOUT

TM TCOMP

OFS

EXTERNAL TCOMP COMPENSATION

+5V

REF

DAC

VCC

GND

PWM6

ISEN6-

ISEN6+

PWM4

ISEN4-

ISEN4+

PWM2

ISEN2-

ISEN2+

PWM1

ISEN1-

ISEN1+

PWM3

ISEN3-

ISEN3+

PWM5

ISEN5-

ISEN5+

EN_PWR

FSOFS

VIN

+5V

PWM

GND

PWM

GND

PWM

GND

EN PWM

PWM

VCC

GND

VCC

ISL6609 DRIVER

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

VIN

µP

LOAD

VIN

NTC

+5V

PWM

GND

VCC

ISL6609 DRIVER

BOOT

UGATE

PHASE

LGATE

VIN

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May 5, 2008

Page 5

ISL6327

Typical Application - 6-Phase Buck Converter with DCR Sensing and Integrated TCOMP

VTT

VR_RDY

VID7 VID6 VID5 VID4 VID3

VID2 VID1

VID0

VRSEL

OVP

VR_FAN

VR_HOT

+5V

IOUT

+5V

IDROOP VDIFF VSEN

RGND EN_VTT

IOUT

TCOMP

OFS

COMP

ISL6327

EN_PWR

FSOFS

REF

DAC

VCC

GND

PWM6

ISEN6-

ISEN6+

PWM4

ISEN4-

ISEN4+

PWM2

ISEN2-

ISEN2+

PWM1

ISEN1-

ISEN1+

PWM3

ISEN3-

ISEN3+

PWM5

ISEN5-

ISEN5+

VIN

+5V

PWM

GND

PWM

GND

PWM

GND

PWM

VCC

GND

VCC

ISL6609 DRIVER

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

BOOT

UGATE

PHASE

LGATE

VIN

μP

LOAD

VIN

NTC

+5V

PWM

GND

VCC

ISL6609 DRIVER

BOOT

UGATE

PHASE

LGATE

VIN

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May 5, 2008

Page 6

ISL6327

Absolute Maximum Ratings

Supply Voltage, VCC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+6V

All Pins. . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V to V

ESD Rating

Human Body Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>2kV

Machine Model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .>200V

Charged Device Model. . . . . . . . . . . . . . . . . . . . . . . . . . . . >1.5kV

+ 0.3V

Thermal Information

Thermal Resistance (Typical, Notes 1, 2) θJA (°C/W) θJC (°C/W)

48 Ld QFN Package. . . . . . . . . . . . . . . 32 3.5

Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C

Maximum Storage Temperature Range. . . . . . . . . .-65°C to +150°C

Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below

http://www.intersil.com/pbfree/Pb-FreeReflow.asp

Operating Conditions

Supply Voltage, VCC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5V ±5%

Ambient Temperature (ISL6327CRZ) . . . . . . . . . . . . . 0°C to +70°C

Ambient Temperature (ISL6327IRZ) . . . . . . . . . . . . .-40°C to +85°C

CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and result in failures not covered by warranty.

NOTES:

is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See

1. θ

Tech Brief TB379.

2. For θ

, the “case temp” location is the center of the exposed metal pad on the package underside.

Electrical Specifications Operating Conditions: VCC = 5V, Unless Otherwise Specified

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

VCC SUPPLY CURRENT

Nominal Supply VCC = 5VDC; EN_PWR = 5VDC; R

ISEN1 = ISEN2 = ISEN3 = ISEN4 = ISEN5 = ISEN6 = -70µA

Shutdown Supply VCC = 5VDC; EN_PWR = 0VDC; R

POWER-ON RESET AND ENABLE

POR Threshold VCC Rising 4.3 4.5 4.7 V

VCC Falling 3.7 3.9 4.2 V

EN_PWR Threshold Rising 0.850 0.875 0.910 V

Hysteresis - 130 - mV Falling 0.720 0.745 0.775 V

EN_VTT Threshold Rising 0.850 0.875 0.910 V

Hysteresis - 130 - mV Falling 0.720 0.745 0.775 V

REFERENCE VOLTAGE AND DAC

System Accuracy of ISL6327CRZ (VID = 1V to 1.6V), T

System Accuracy of ISL6327CRZ (VID = 0.5V to 1V), T

System Accuracy of ISL6327IRZ (VID = 1V to1.6V), T

System Accuracy of ISL6327IRZ (VID = 0.5V to 1V), T

VID Pull-up -60 -40 -20 µA VID Input Low Level --0.4V VID Input High Level 0.8 - - V VRSEL Input Low Level --0.4V VRSEL Input High Level 0.8 - - V DAC Source Current -47mA

= 0°C to +70°C

= -40°C to +85°C

(Note 3) -0.5 - 0.5 %VID

(Note 3) -0.9 - 0.9 %VID

(Note 3) -0.6 - 0.6 %VID

(Note 3) -1 - 1 %VID

= 100kΩ,

= 100kΩ -1421mA

-1826mA

FN9276.4

May 5, 2008

Page 7

ISL6327

Electrical Specifications Operating Conditions: VCC = 5V, Unless Otherwise Specified (Continued)

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

DAC Sink Current - - 300 µA REF Source Current 45 50 55 µA REF Sink Current 45 50 55 µA

PIN-ADJUSTABLE OFFSET

Voltage at OFS Pin Offset resistor connected to ground 380 400 420 mV

Voltage below VCC, offset resistor connected to VCC 1.55 1.60 1.65 V

OSCILLATORS

Accuracy of Switching Frequency Setting R Adjustment Range of Switching Frequency (Note 4) 0.08 - 1.0 MHz Soft-Start Ramp Rate (Notes 5, 6) R Adjustment Range of Soft-Start Ramp Rate (Note 4) 0.625 - 6.25 mV/µs

PWM GENERATOR

Sawtooth Amplitude -1.25- V

ERROR AMPLIFIER

Open-Loop Gain R Open-Loop Bandwidth C Slew Rate C Maximum Output Voltage 3.8 4.3 4.9 V Output High Voltage @ 2mA 3.6 - - V Output Low Voltage @ 2mA --1.8V

REMOTE-SENSE AMPLIFIER

Bandwidth (Note 4) - 20 - MHz Output High Current VSEN - RGND = 2.5V -500 - 500 µA Output High Current VSEN - RGND = 0.6V -500 - 500 µA

PWM OUTPUT

PWM Output Voltage LOW Threshold I PWM Output Voltage HIGH Threshold I

CURRENT SENSE AND OVERCURRENT PROTECTION

Sensed Current Tolerance ISEN1 = ISEN2 = ISEN3 = ISEN4 = ISEN5 = ISEN6 = 60µA 57 60 63 µA Overcurrent Trip Level for Average Current 72 85 98 µA Peak Current Limit for Individual Channel 100 120 140 µA Maximum Voltage at IDROOP and IOUT

Pins

THERMAL MONITORING

TM Input Voltage for VR_FAN Trip 1.55 1.65 1.75 V TM Input Voltage for VR_FAN Reset 1.85 1.95 2.05 V TM Input Voltage for VR_HOT Trip 1.3 1.4 1.5 V TM Input Voltage for VR_HOT Reset 1.55 1.65 1.75 V Leakage Current of VR_HOT With external pull-up resistor connected to VCC - - 30 µA VR_HOT Low Voltage With 1.25kΩ resistor pull-up to VCC, I

= 100kΩ 225 250 275 kHz

= 100kΩ - 1.563 - mV/µs

= 10kΩ to ground (Note 4) - 96 - dB

= 100pF, RL = 10kΩ to ground (Note 4) - 80 - MHz

= 100pF (Note 4) - 25 - V/µs

= ±500µA - - 0.5 V

LOAD

= ±500µA 4.3 - - V

LOAD

1.97 2.0 2.03 V

= 4mA - - 0.4 V

VR_HOT

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May 5, 2008

Page 8

ISL6327

Electrical Specifications Operating Conditions: VCC = 5V, Unless Otherwise Specified (Continued)

PARAMETER TEST CONDITIONS MIN TYP MAX UNITS

Leakage Current of VR_FAN With external pull-up resistor connected to VCC - - 30 µA VR_FAN Low Voltage With 1.25kΩ resistor pull-up to VCC, I

VR READY AND PROTECTION MONITORS

Leakage Current of VR_RDY With externally pull-up resistor connected to VCC - - 30 µA VR_RDY Low Voltage I Undervoltage Threshold VDIFF Falling 48 50 52 %VID VR_RDY Reset Voltage VDIFF Rising 58 60 62 %VID Overvoltage Protection Threshold Before valid VID 1.250 1.275 1.300 V

After valid VID, the voltage above VID 150 175 200 mV Overvoltage Protection Reset Hysteresis -100- mV OVP Output Low Voltage IOVP = 4mA - - 0.4 V

NOTES:

3. These parts are designed and adjusted for accuracy with all errors in the voltage loop included.

4. Limits established by characterization and are not production tested.

5. During soft-start, VDAC rises from 0 to 1.1V first and then ramp to VID voltage after receiving valid VID input.

6. Soft-start ramp rate is determined by the adjustable soft-start oscillator frequency at the speed of 6.25mV per cycle.

= 4mA - - 0.4 V

VR_RDY

= 4mA - - 0.4 V

VR_FAN

Functional Pin Description

VCC - Supplies the power necessary to operate the chip. The controller starts to operate when the voltage on this pin exceeds the rising POR threshold and shuts down when the voltage on this pin drops below the falling POR threshold. Connect this pin directly to a +5V supply.

GND - Bias and reference ground for the IC. The bottom metal base of ISL6327 is the GND.

EN_PWR - This pin is a threshold-sensitive enable input for the controller. Connecting the 12V supply to EN_PWR through an appropriate resistor divider provides a means to synchronize power-up of the controller and the MOSFET driver ICs. When EN_PWR is driven above 0.875V, the ISL6327 is active depending on status of EN_VTT, the internal POR, and pending fault states. Driving EN_PWR below 0.745V will clear all fault states and prime the ISL6327 to soft-start when re-enabled.

EN_VTT - This pin is another threshold-sensitive enable input for the controller. It’s typically connected to VTT output of VTT voltage regulator in the computer mother board. When EN_VTT is driven above 0.875V, the ISL6327 is active depending on status of ENLL, the internal POR, and pending fault states. Driving EN_VTT below 0.745V will clear all fault states and prime the ISL6327 to soft-start when re-enabled.

FS - Use this pin to set up the desired switching frequency. A resistor, placed from FS to ground will set the switching frequency. The relationship between the value of the resistor and the switching frequency will be described by an approximate equation.

SS - Use this pin to set-up the desired start-up oscillator frequency. A resistor, placed from SS to ground will set up the soft-start ramp rate. The relationship between the value of the resistor and the soft-start ramp up time will be described by an approximate equation.

VID7, VID6, VID5, VID4, VID3, VID2, VID1 and VID0 -

These are the inputs to the internal DAC that generates the reference voltage for output regulation. Connect these pins either to open-drain outputs with or without external pull-up resistors or to active-pull-up outputs. All VID pins have 40µA internal pull-up current sources that diminish to zero as the voltage rises above the logic-high level. These inputs can be pulled up externally as high as VCC plus 0.3V.

VRSEL - VRSEL is the pin used to select the internal VID code. When it is connected to GND, the extended VR10 code is selected. VRSEL pin has 40µA internal pull-up

current sources that diminish to zero as the voltage rises

above the logic-high level. When it’s floated or pulled to high, VR1 1 code is selected. This input can be pulled up as high

as VCC plus 0.3V.

VDIFF, VSEN, and RGND - VSEN and RGND form the precision differential remote-sense amplifier. This amplifier converts the differential voltage of the remote output to a single-ended voltage referenced to local ground. VDIFF is the amplifier’s output and the input to the regulation and protection circuitry. Connect VSEN and RGND to the sense pins of the remote load. VDIFF is connected to FB through a resistor.

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ISL6327

FB and COMP - The inverting input and the output of the error amplifier respectively. FB can be connected to VDIFF through a resistor. A properly chosen resistor between VDIFF and FB can set the load line (droop), when IDROOP pin is tied to FB pin. The droop scale factor is set by the ratio of the ISEN resistors and the inductor DCR or the dedicated current sense resistor. COMP is tied back to FB through an external R-C network to compensate the regulator.

DAC and REF - The DAC pin is the output of the precision internal DAC reference. The REF pin is the positive input of the Error Amp. In typical applications, a 1kΩ, 1% resistor is used between DAC and REF to generate a precision offset voltage. This voltage is proportional to the offset current determined by the offset resistor from OFS to ground or VCC. A capacitor is used between REF and ground to smooth the voltage transition during Dynamic VID™ operations.

PWM1, PWM2, PWM3, PWM4, PWM5, PWM6 - Pulse width modulation outputs. Connect these pins to the PWM input pins of the Intersil driver IC. The number of active channels is determined by the state of PWM3, PWM4, PWM5, and PWM6. For 2-phase operation, connect PWM3 to VCC; similarly, PWM4 for 3-phase, PWM5 for 4-phase, and PWM6 for 5-phase operation.

TABLE 1. PHASE FIRING SEQUENCE

CONFIGURATION PHASE SEQUENCE

6-Phase 1 - 2 - 3 - 4 - 5 - 6 5-Phase 1 - 2 - 3 - 4 - 5 4-Phase 1 - 2 - 4 - 3 3-Phase 1 - 2 - 3

ISEN1+, ISEN1-; ISEN2+, ISEN2-; ISEN3+, ISEN3-; ISEN4+, ISEN4-; ISEN5+, ISEN5-; ISEN6+, ISEN6- - The

ISEN+ and ISEN- pins are current sense inputs to individual differential amplifiers. The sensed current is used for channel current balancing, overcurrent protection, and droop regulation. Inactive channels should have their respective current sense inputs left open (for example, open ISEN6+ and ISEN6- for 5-phase operation).

For DCR sensing, connect each ISEN- pin to the node between the RC sense elements. Tie the ISEN+ pin to the other end of the sense capacitor through a resistor, R

ISEN

. The voltage across the sense capacitor is proportional to the inductor current. Therefo re, the sense current is proportional to the inductor current, and scaled by the DCR of the inductor and R

ISEN

VR_RDY - VR_RDY indicates that the soft-start is completed and the output voltage is within the regulated range around VID setting. It is an open-drain logic output. When OCP or OVP occurs, VR_RDY will be pulled to low. It will also be pulled low if the output voltage is below the undervoltage threshold.

OFS - The OFS pin provides a means to program a DC offset current for generating a DC offset voltage at the REF input. The offset current is generated via an external resistor and precision internal voltage references. The polarity of the offset is selected by connecting the resistor to GND or VCC. For no offset, the OFS pin should be left unterminated.

TCOMP - Temperature compensation scaling inpu t. The voltage sensed on the TM pin is utilized as the temperature input to adjust IDROOP and the overcurrent protection limit to effectively compensate for the temperature coefficient of the current sense element. To implement the integrated temperature compensation, a resistor divider circuit is needed with one resistor being connected from TCOMP to VCC of the controller and another resistor being connected from TCOMP to GND. Changing the ratio of the resistor values will set the gain of the integrated thermal compensation. When integrated temperature compensation function is not used, connect TCOMP to GND.

OVP - The Overvoltage protection output indication pin. This pin can be pulled to VCC and is latched when an overvoltage condition is detected. When the OVP indication is not used, keep this pin open.

IDROOP - IDROOP is the output pin of the sensed average channel current, which is proportional to the load current. In the application, which does not require loadline, leave this pin open. In the application which requires load line, connect this pin to FB so that the sensed average current will flow through the resistor between FB and VDIFF to create a voltage drop, which is proportional to the load current.

IOUT - IOUT has the same output as IDROOP with additional OCP adjustment function. In actual application, a resistor needs to be placed between IOUT and GND to ensure the proper operation. The voltage at IOUT pin will be proportional to the load current. If the voltage is higher than 2V, ISL6327 will go into the OCP mode, this means it will shut down first and then hiccup. The additional OCP trip level can be adjusted by changing the resistor value.

TM - TM is an input pin for VR temperature measurement. Connect this pin through NTC thermistor to GND and a resistor to VCC of the controller. The voltage at this pin is reverse proportional to the VR temperature. ISL6327 monitors the VR temperature based on the voltage at the TM pin and the output signals at VR_HOT and VR_FAN.

VR_HOT - VR_HOT is used as an indication of high VR temperature. It is an open-drain logic output. It will be open when the measured VR temperature reaches a certain level.

VR_FAN - VR_FAN is an output pin with open-drain logic output. It will be open when the measured VR temperature reaches a certain level.

FN9276.4

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Page 10

ISL6327

Operation

Multiphase Power Conversion

Microprocessor load current profiles have changed to the point that the advantages of multiphase power conversion are impossible to ignore. The technical challenges associated with producing a single-phase converter which is both cost-effective and thermally viable, have forced a change to the cost-saving approach of multiphase. The ISL6327 controller helps reduce the complexity of implementation by integrating vital functions and requiring minimal output components. The block diagrams on page 3, page 4 and page 5 provide top level views of multiphase power conversion using the ISL6327 controller.

Interleaving

The switching of each channel in a multiphase converter is timed to be symmetrically out-of-phase with each of the other channels. In a 3-phase converter, each channel switches 1/3 cycle after the previous channel and 1/3 cycle before the following channel. As a result, the three-phase converter has a combined ripple frequency three times greater than the ripple frequency of any one phase. In addition, the peak-to-peak amplitude of the combined inductor current is reduced in proportion to the number of phases (Equations 1 and 2). The increased ripple frequency and the lower ripple amplitude mean that the designer can use less per-channel inductance and lower total output capacitance for any performance specification.

Figure 1 illustrates the multiplicative effect on output ripple frequency. The three channel currents (IL1, IL2, and IL3) combine to form the AC ripple current and the DC load current. The ripple component has three times the ripple frequency of each individual channel current. Each PWM pulse is triggered 1/3 of a cycle after the start of the PWM pulse of the previous phase. The DC components of the inductor currents combine to feed the load.

IL1 + IL2 + IL3, 7A/DIV

IL3, 7A/DIV

PWM3, 5V/DIV

IL2, 7A/DIV

PWM2, 5V/DIV

IL1, 7A/DIV

PWM1, 5V/DIV

1µs/DIV

FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS

FOR 3-PHASE CONVERTER

T o understand the reduction of the ripple current amplitude in the multiphase circuit, examine the equation representing an individual channel’s peak-to-peak inductor current.

VINV

–()V

OUT

------------------------------------------------------=

LfSV

In Equation 1, V

and V

OUT

are the input and the output

OUT

(EQ. 1)

voltages respectively, L is the single-channel inductor value, and f

is the switching frequency.

INPUT-CAPACITOR CURRENT 10A/DIV

CHANNEL 3 INPUT CURRENT 10A/DIV

CHANNEL 2 INPUT CURRENT 10A/DIV

CHANNEL 1 INPUT CURRENT 10A/DIV

1µs/DIV

FIGURE 2. CHANNEL INPUT CURRENTS AND INPUT-

CAPACITOR RMS CURRENT FOR 3-PHASE CONVERTER

The output capacitors conduct the ripple component of the inductor current. In the case of multiphase converters, the capacitor current is the sum of the ripple currents from each of the individual channels. Compare Equation 1 to the expression for the peak-to-peak current after the summation of N symmetrically phase-shifted inductor currents in Equation 2. Peak-to-peak ripple current decreases by an amount proportional to the number of channels. Output voltage ripple is a function of capacitance, capacitor equivalent series resistance (ESR), and inductor ripple current. Reducing the inductor ripple current allows the designer to use fewer or less costly output capacitors.

VINNV

–()V

OUT

(EQ. 2)

CP P–()

------------------------------------------------------------= LfSV

Another benefit of interleaving is to reduce the input ripple current. The input capacitance is determined in part by the maximum input ripple current. Multiphase topologies can improve the overall system cost and size by lowering the input ripple current and allowing the designer to reduce the cost of input capacitance. The example in Figure 2 illustrates the input currents from a three-phase converter combining to reduce the total input ripple current.

The converter depicted in Figure 2 delivers 36A to a 1.5V load from a 12V input. The RMS input capacitor current is 5.9A. Compare this to a single-phase conve rter also step ping down 12V to 1.5V at 36A. The single-phase converter has 11.9A

FN9276.4

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ISL6327

RMS input capacitor current. The single-phase converter must use an input capacitor bank with twice the RMS current capacity as the equivalent three-phase converter.

Figures 19, 20 and 21 in the section titled “Input Capacitor Selection” on page 27 can be used to determine the input capacitor RMS current based on the load current, the duty cycle, and the number of channels. They are provided as aids in determining the optimal input capacitor solution. Figure 22 shows the single phase input-capacitor RMS current for comparison.

PWM Modulation Scheme

The ISL6327 adopts Intersil's proprietary Active Pulse Positioning (APP) modulation scheme to improve the transient performance. APP control is a unique dual-edge PWM modulation scheme with both PWM leading and trailing edges being independently moved to provide the best response to the transient loads. The PWM frequency, however, is constant and set by the external resistor between the FS pin and GND.

To further improve the transient response, the ISL6327 also implements Intersil's proprietary Adaptive Phase Alignment (APA) technique. APA, with sufficiently large load step currents, can turn on all phases together.

With both APP and APA control, ISL6327 can achieve excellent transient performance and reduce the demand on the output capacitors.

Under the steady state conditions the operation of the ISL6327 PWM modulator appears to be that of a conventional trailing edge modulator. Conventional analysis and design methods can therefore be used for steady state and small signal operation.

PWM Operation

The timing of each converter is set by the number of active channels. The default channel setting for the ISL6327 is six. The switching cycle is defined as the time between PWM pulse termination signals of each channel. The cycle time of the pulse termination signal is the inverse of the switching frequency set by the resistor between the FS pin and ground. The PWM signals command the MOSFET drivers to turn on/off the channel MOSFETs.

In the default 6-phase operation, the PWM2 pulse hap pens 1/6 of a cycle after PWM1, the PWM3 pulse happens 1/6 of a cycle after PWM2, the PWM4 pulse happens 1/6 of a cycle after PWM3, the PWM5 pulse happens 1/6 of a cycle after PWM4, and the PWM6 pulse happens 1/6 of a cycle after PWM5.

Connecting the PWM4 to VCC selects 3-phase operation and the pulse times are spaced in 1/3 cycle increments. Connecting the PWM3 to VCC selects 2-phase operation and the pulse times are spaced in 1/2 cycle increments.

Switching Frequency

The switching frequency is determined by the selection of the frequency-setting resistor, RT, which is connected from FS pin to GND (see the figures labelled Typical Applications on page 4 and page 5). Equation 3 is provided to assist in selecting the correct resistor value.

2.5X10

where f

------------------------- f

600–=

is the switching frequency of each phase.

(EQ. 3)

Current Sensing

ISL6327 senses the current continuously for fast response. ISL6327 supports inductor DCR sensing, or resistive sensing techniques. The associated channel current sense amplifier uses the ISEN inputs to reproduce a signal proportional to the inductor current, I I

, is used for the current balance, the load-line

SEN

regulation, and the overcurrent protection. The internal circuitry, shown in Figures 3 and 4, represents

one channel of an N-channel converter. This circuitry is repeated for each channel in the converter, but may not be active depending on the status of the PWM3, PWM4, PWM5, and PWM6 pins, as described in “PWM Operation” on page 11.

INDUCTOR DCR SENSING

An inductor’s winding is characteristic of a distributed resistance as measured by the DCR (Direct Current Resistance) parameter. Consider the inductor DCR as a separate lumped quantity, as shown in Figure 3. The channel current I

, flowing through the inductor, will also

pass through the DCR. Equation 4 shows the S-domain (equivalent voltage across the inductor V

LIL

sL DCR+⋅()⋅=

A simple RC network across the inductor extracts the DCR voltage, as shown in Figure 3.

The voltage on the capacitor V proportional to the channel current I

⎛⎞

-------------

⋅ 1+

⎝⎠

DCR

---------------------------------------------------------------------

sRC 1+⋅()

DCR I

⋅()⋅

. The sensed current,

, can be shown to be

, see Equation 5.

(EQ. 4)

(EQ. 5)

The ISL6327 works in 2, 3, 4, 5, or 6 phase configuration. Connecting the PWM6 to VCC selects 5-phase operation and the pulse times are spaced in 1/5 cycle increments. Connecting the PWM5 to VCC selects 4-phase operation and the pulse times are spaced in 1/4 cycle increments.

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ISL6327

ISL6609

PWM(n)

ISL6327 INTERNAL CIRCUIT

CURRENT

SENSE

SEN

DCR

----------------- -

ISEN

INDUCTOR

ISEN-(n)

ISEN+(n)

ILs()

DCR

VC(s)

FIGURE 3. DCR SENSING CONFIGURATION

The same capacitor C between ISEN- and ISEN+ signals. Select the proper C

OUT

R (PTC)

ISEN(n)

OUT

keep the time constant of R to 27ns.

Equation 7 shows the ratio of the channel current to the sensed current I

SENIL

ISL6327 INTERNAL CIRCUIT

SENSE

-----------------------

⋅=

ISEN

CURRENT

SENSE

SEN

is needed to match the time delay

and CT (R

ISEN

SENSE

--------------------------= R

ISEN

ISEN-(n)

ISEN+(n)

x CT) close

OUT

ISEN(n)

(EQ. 7)

If the RC network components are selected such that the RC time constant (= R*C) matches the inductor time constant (= L/DCR), the voltage across the capacitor V

is equal to

the voltage drop across the DCR, i.e., proportional to the channel current.

With the internal low-offset current amplifier, the capacitor voltage V Therefore the current out of ISEN+ pin, I

is replicated across the sense resistor R

, is proportional

SEN

ISEN

to the inductor current. Because of the internal filter at ISEN- pin, one capacitor C

is needed to match the time delay between the ISEN- and ISEN+ signals. Select the proper C constant of R

and CT (R

ISEN

to keep the time

x CT) close to 27ns.

ISEN

Equation 6 shows that the ratio of the channel current to the sensed current I

is driven by the value of the sense

SEN

resistor and the DCR of the inductor.

DCR

SENIL

----------------- -

⋅=

ISEN

(EQ. 6)

RESISTIVE SENSING

For accurate current sense, a dedicated current-sense resistor R

in series with each output inductor can

SENSE

serve as the current sense element (see Figure 4). This technique is more accurate, but reduces overall converter efficiency due to the additional power loss on the current sense element R

SENSE

FIGURE 4. SENSE RESISTOR IN SERIES WITH INDUCTORS

The inductor DCR value will increase as the temperature increases. Therefore the sensed current will increase as the temperature of the current sense element increases. In order to compensate the temperature effect on the sensed current signal, a Positive Temperature Coefficient (P TC) resistor can be selected for the sense resistor R

, or the integrated

ISEN

temperature compensation function of ISL6327 should be utilized. The integrated temperature compensation function is described in “Temperature Compensation” on page 21.

Channel-Current Balance

The sensed current In from each active channel are summed together and divided by the number of active channels. The resulting average current I total load current. Channel current balance is achieved by comparing the sensed current of each channel to the average current to make an appropriate adjustment to the PWM duty cycle of each channel with Intersil’s patented current-balance method.

Channel current balance is essential in achieving the thermal advantage of multiphase operation. With good current balance, the power loss is equally dissipated over multiple devices and a greater area.

provides a measure of the

AVG

FN9276.4

May 5, 2008

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ISL6327

Voltage Regulation

The compensation network shown in Figure 5 assures that the steady-state error in the output voltage is limited only to the error in the reference voltage (output of the DAC) and offset errors in the OFS current source, remote-sense and error amplifiers. Intersil specifies the guaranteed tolerance of the ISL6327 to include the combined tolerances of each of these elements.

The output of the error amplifier , V sawtooth waveforms to generate the PWM signals. The PWM signals control the timing of the Intersil MOSFET drivers and regulate the converter output to the specified reference voltage. The internal and external circuitries, which control the voltage regulation, are illustrated in Figure 5.

The ISL6327 incorporates an internal differential remote-sense amplifier in the feedback path. The amplifier removes the voltage error encountered when measuring the output voltage relative to the local controller ground reference point resulting in a more accurate means of sensing output voltage. Connect the microprocessor sense pins to the non-inverting input, VSEN, and inverting input, RGND, of the remote-sense amplifier. The remote-sense output, V

, is connected to the inverting input

DIFF

of the error amplifier through an external resistor. A digital-to-analog converter (DAC) generates a reference

voltage based on the state of logic signals at pins VID7 through VID0. The DAC decodes the 8-bit logic signal (VID) into one of the discrete voltages shown in T able 3. Each VID input offers a 45µA pull-up to an internal 2.5V source for use with open-drain outputs. The pull-up current diminishes to zero above the logic threshold to protect voltage-sensitive output devices. External pull-up resistors can augment the pull-up current sources in case the leakage into the driving device is greater than 45µA.

EXTERNAL CIRCUIT ISL6327 INTERNAL CIRCUIT

REF

DROOP

OUT

FIGURE 5. OUTPUT VOLT AGE AND LOAD-LINE

COMP

DAC

REF

IDROOP

VDIFF

VSEN

RGND

REGULATION WITH OFFSET ADJUSTMENT

AVG

, is compared to the

COMP

ERROR AMPLIFIER

DIFFERENTIAL REMOTE-SENSE AMPLIFIER

COMP

Load-Line Regulation

Some microprocessor manufacturers require a precisely controlled output resistance. This dependence of the output voltage on the load current is often termed “droop” or “load line” regulation. By adding a well controlled output impedance, the output voltage can effectively be level shifted in a direction which works to achieve the load-line regulation required by these manufacturers.

In other cases, the designer may determine that a more cost-effective solution can be achieved by adding droop. Droop can help to reduce the output-voltage spike that results from the fast changes of the load-current demand.

The magnitude of the spike is dictated by the ESR and ESL of the output capacitors selected. By positioning the no-load voltage level near the upper specification limit, a larger negative spike can be sustained without crossing the lower limit. By adding a well controlled output impedance, the output voltage under load can effectively be level shifted down so that a larger positive spike can be sustained without crossing the upper specification limit.

TABLE 2. VR10 VID TABLE (WITH 6.25mV EXTENSION)

VID4

VID3

VID2

VID1

VID0

VID5

VID6

400mV

200mV

100mV

010101 010101 010110 010110 010111 010111 011000 011000 011001 011001 011010 011010 011011 011011 011100 011100 011101 011101 011110 011110 011111 011111 100000 100000

50mV

25mV

12.5mV

VOLTAGE

6.25mV

11.6 0 1.59375 1 1.5875 0 1.58125 1 1.575 0 1.56875 1 1.5625 0 1.55625

11.55 0 1.54375 1 1.5375 0 1.53125 1 1.525 0 1.51875 1 1.5125 0 1.50625

11.5 0 1.49375 1 1.4875 0 1.48125 1 1.475 0 1.46875 1 1.4625 0 1.45625

(V)

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May 5, 2008

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ISL6327

TABLE 2. VR10 VID TABLE (WITH 6.25mV EXTENSION) (Continued)

VID4

VID3

400mV

200mV

VID2

100mV

VID1

50mV

VID0

25mV

VID5

12.5mV

VID6

6.25mV

VOLTAGE

(V)

10000111.45 100001 100010 100010 100011 100011 100100 100100 100101 100101 100110 100110 100111 100111 101000 101000 101001 101001 101010 101010 101011 101011 101100 101100 101101 101101 101110 101110 101111 101111 110000 110000 110001 110001 110010 110010 110011 110011 110100 110100 110101

0 1.44375 1 1.4375 0 1.43125 1 1.425 0 1.41875 1 1.4125 0 1.40625

11.4 0 1.39375 1 1.3875 0 1.38125 1 1.375 0 1.36875 1 1.3625 0 1.35625

11.35 0 1.34375 1 1.3375 0 1.33125 1 1.325 0 1.31875 1 1.3125 0 1.30625

11.3 0 1.29375 1 1.2875 0 1.28125 1 1.275 0 1.26875 1 1.2625 0 1.25625

11.25 0 1.24375 1 1.2375 0 1.23125 1 1.225 0 1.21875 1 1.2125 0 1.20625

11.2

TABLE 2. VR10 VID TABLE (WITH 6.25mV EXTENSION) (Continued)

VID4

VID3

VID2

VID1

VID0

VID5

VID6

400mV

200mV

100mV

50mV

25mV

12.5mV

6.25mV

VOLTAGE

(V)

1101010 1.19375 110110 110110 110111 110111 111000 111000 111001 111001 111010 111010 111011 111011 111100 111100 111101 111101 111110 111110 111111 111111 000000 000000 000001 000001 000010 000010 000011 000011 000100 000100 000101 000101 000110 000110 000111 000111 001000 001000 001001 001001

1 1.1875 0 1.18125 1 1.175 0 1.16875 1 1.1625 0 1.15625

11.15 0 1.14375 1 1.1375 0 1.13125 1 1.125 0 1.11875 1 1.1125 0 1.10625

11.1 0 1.09375 1OFF 0OFF 1OFF 0OFF 1 1.0875 0 1.08125 1 1.075 0 1.06875 1 1.0625 0 1.05625

11.05 0 1.04375 1 1.0375 0 1.03125 1 1.025 0 1.01875 1 1.0125 0 1.00625 11 0 0.99375 1 0.9875 0 0.98125 1 0.975 0 0.96875

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Page 15

ISL6327

TABLE 2. VR10 VID TABLE (WITH 6.25mV EXTENSION) (Continued)

VID4

VID3

400mV

200mV

0010101 0.9625 001010 001011 001011 001100 001100 001101 001101 001110 001110 001111 001111 010000 010000 010001 010001 010010 010010 010011 010011 010100 010100

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOLTAGE

00000000OFF 00000001OFF

000000101.60000

000000111.59375

000001001.58750

000001011.58125

000001101.57500

000001111.56875

000010001.56250

000010011.55625

000010101.55000

000010111.54375

000011001.53750

000011011.53125

000011101.52500

000011111.51875

VID2

100mV

VID1

VID0

50mV

25mV

TABLE 3. VR11 VID 8-BIT

VID5

12.5mV

VID6

VOLTAGE

6.25mV

0 0.95625

10.95 0 0.94375 1 0.9375 0 0.93125 1 0.925 0 0.91875 1 0.9125 0 0.90625

10.9 0 0.89375 1 0.8875 0 0.88125 1 0.875 0 0.86875 1 0.8625 0 0.85625

10.85 0 0.84375 1 0.8375 0 0.83125

(V)

TABLE 3. VR11 VID 8-BIT (Continued)

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOLTAGE

000100001.51250

000100011.50625

000100101.50000

000100111.49375

000101001.48750

000101011.48125

000101101.47500

000101111.46875

000110001.46250

000110011.45625

000110101.45000

000110111.44375

000111001.43750

000111011.43125

000111101.42500

000111111.41875

001000001.41250

001000011.40625

001000101.40000

001000111.39375

001001001.38750

001001011.38125

001001101.37500

001001111.36875

001010001.36250

001010011.35625

001010101.35000

001010111.34375

001011001.33750

001011011.33125

001011101.32500

001011111.31875

001100001.31250

001100011.30625

001100101.30000

001100111.29375

001101001.28750

001101011.28125

001101101.27500

001101111.26875

001110001.26250

001110011.25625

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May 5, 2008

Page 16

ISL6327

TABLE 3. VR11 VID 8-BIT (Continued)

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOLTAGE

001110101.25000

001110111.24375

001111001.23750

001111011.23125

001111101.22500

001111111.21875

010000001.21250

010000011.20625

010000101.20000

010000111.19375

010001001.18750

010001011.18125

010001101.17500

010001111.16875

010010001.16250

010010011.15625

010010101.15000

010010111.14375

010011001.13750

010011011.13125

010011101.12500

010011111.11875

010100001.11250

010100011.10625

010100101.10000

010100111.09375

010101001.08750

010101011.08125

010101101.07500

010101111.06875

010110001.06250

010110011.05625

010110101.05000

010110111.04375

010111001.03750

010111011.03125

010111101.02500

010111111.01875

011000001.01250

011000011.00625

011000101.00000

011000110.99375

TABLE 3. VR11 VID 8-BIT (Continued)

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOLTAGE

011001000.98750

011001010.98125

011001100.97500

011001110.96875

011010000.96250

011010010.95625

011010100.95000

011010110.94375

011011000.93750

011011010.93125

011011100.92500

011011110.91875

011100000.91250

011100010.90625

011100100.90000

011100110.89375

011101000.88750

011101010.88125

011101100.87500

011101110.86875

011110000.86250

011110010.85625

011110100.85000

011110110.84375

011111000.83750

011111010.83125

011111100.82500

011111110.81875

100000000.81250

100000010.80625

100000100.80000

100000110.79375

100001000.78750

100001010.78125

100001100.77500

100001110.76875

100010000.76250

100010010.75625

100010100.75000

100010110.74375

100011000.73750

100011010.73125

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ISL6327

TABLE 3. VR11 VID 8-BIT (Continued)

VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VOLTAGE

100011100.72500

100011110.71875

100100000.71250

100100010.70625

100100100.70000

100100110.69375

100101000.68750

100101010.68125

100101100.67500

100101110.66875

100110000.66250

100110010.65625

100110100.65000

100110110.64375

100111000.63750

100111010.63125

100111100.62500

100111110.61875

101000000.61250

101000010.60625

101000100.60000

101000110.59375

101001000.58750

101001010.58125

101001100.57500

101001110.56875

101010000.56250

101010010.55625

101010100.55000

101010110.54375

101011000.53750

101011010.53125

101011100.52500

101011110.51875

101100000.51250

101100010.50625

101100100.50000 11111110OFF 11111111OFF

Figure 5 shows a current proportional to the average current of all active channels, I line regulation resistor R across R

is proportional to the output current, effectively

, flows from FB through a load-

AVG

. The resulting voltage drop

creating an output voltage droop with a steady-state value defined using Equation 8:

DROOPIAVGRFB

(EQ. 8)

The regulated output voltage is reduced by the droop voltage V

. The output voltage as a function of load current is

DROOP

derived by combining Equation 8 with the appropriate sample current expression defined by the current sense method employed in Equation 9.

⎛⎞

OUT

OUTVREFVOFS

Where V

–

is the reference voltage, V

REF

-------------

–=

⎜⎟ ⎝⎠

programmed offset voltage, I of the converter, R the ISEN+ pin, and R

is the sense resistor connected to

ISEN

is the feedback resistor, N is the

active channel number, and R

OUT

----------------- -R R

ISEN

OFS

is the total output current

is the DCR, or R

(EQ. 9)

is the

SENSE

depending on the sensing method. Therefore the equivalent loadline impedance (i.e. Droop

impedance) is equal to:

------------

----------------- -=

ISEN

(EQ. 10)

Output-Voltage Offset Programming

The ISL6327 allows the designer to accurately adjust the offset voltage. When a resistor, R OFS to VCC, the voltage across it is regulated to 1.6V. This causes a proportional current (I R

is connected to ground, the voltage across it is

OFS

regulated to 0.4V, and I between DAC and REF, R product (I

OFS

x R

OFS

flows out of OFS. A resistor

OFS

REF

) is equal to the desired offset voltage.

These functions are shown in Figure 6. Once the desired output offset voltage has been determined,

use Equation 11 to set R

OFS

For Positive Offset (connect R

1.6 R

REF

OFS

------------------------------

OFFSET

For Negative Offset (connect R

0.4 R

REF

------------------------------

OFS

OFFSET

, is connected between

OFS

) to flow into OFS. If

OFS

, is selected so that the

to VCC):

OFS

to GND):

OFS

(EQ. 11)

(EQ. 12)

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ISL6327

DYNAMIC

VID D/A

E/A

1.6V +

VCC

FIGURE 6. OUTPUT VOLTAGE OFFSET PROGRAMMING

0.4V

GND

ISL6327

DAC

VCC

GND

OFS

REF

OFS

Dynamic VID

Modern microprocessors need to make changes to their core voltage as part of the normal operation. They direct the core voltage regulator to do this by making changes to the VID inputs during the regulator operation. The power management solution is required to monitor the DAC inputs and respond to on-the-fly VID changes in a controlled mann er. Supervising the safe output voltage transition within the DAC range of the processor without discontinuity or disruption is a necessary function of the core-voltage regulator.

In order to ensure the smooth transition of output voltage during VID change, a VID step change smoothing network, composed of R R

is based on the desired offset voltage as detailed in

REF

and C

, can be used. The selection of

REF

“Output-Voltage Offset Programming” on page 17. The selection of C

is based on the time duration for 1 bit VID

REF

change and the allowable delay time. Assuming the microprocessor controls the VID change at 1-bit

every T R

REF

REFRREFTVID

, the relationship between the time constant of

VID

and C

network and T

REF

is given by Equation 13.

VID

(EQ. 13)

Operation Initialization

Prior to converter initialization, proper conditions must exist on the enable inputs and VCC. When the conditions are met, the controller begins soft-start. Once the output voltag e is within the proper window of operation, VR_RDY asserts logic high.

Enable and Disable

While in shutdown mode, the PWM outputs are held in a high-impedance state to assure the drivers remain off. The following input conditions must be met before the ISL6327 is released from shutdown mode.

1. The bias voltage applied at VCC must reach the internal power-on reset (POR) rising threshold. Once this threshold is reached, proper operation of all aspects of the ISL6327 is guaranteed. Hysteresis between the rising and falling thresholds assure that once enabled, the ISL6327 will not inadvertently turn off unless the bias voltage drops substantially (see “Electrical Specifications” on page 6).

2. The ISL6327 features an enable input (EN_PWR) for power sequencing between the controller bias voltage and another voltage rail. The enable comparator holds the ISL6327 in shutdown until the voltage at EN_PW R rises above 0.875V. The enable comparator has about 130mV of hysteresis to prevent bounce. It is important that the driver ICs reach their POR level before the ISL6327 becomes enabled. The schematic in Figure 7 demonstrates sequencing the ISL6327 with the ISL66xx family of Intersil MOSFET drivers, which require 12V bias.

3. The voltage on EN_VTT must be higher than 0.875V to enable the controller. This pin is typically connected to the output of VTT VR.

ISL6327 INTERNAL CIRCUIT

POR

CIRCUIT

SOFT-START

AND

FAULT LOGIC

FIGURE 7. POWER SEQUENCING USING THRESHOLD-

SENSITIVE ENABLE (EN) FUNCTION

ENABLE

COMPARATOR

0.875V

When all conditions above are satisfied, ISL6327 begins the soft-start and ramps the output voltage to 1.1V first. After remaining at 1.1V for some time, ISL6327 reads the VID code at VID input pins. If the VID code is valid, ISL6327 will regulate the output to the final VID setting. If the VID code is OFF code, ISL6327 will shut down, and cycling VCC, EN_PWR or EN_VTT is needed to restart.

EXTERNAL CIRCUIT

VCC

EN_PWR

EN_VTT

+12V

10kΩ

910Ω

FN9276.4

May 5, 2008

Page 19

ISL6327

Soft-Start

ISL6327 based VR has 4 periods during soft-start as shown in Figure 8. After VCC, EN_VTT and EN_PWR reach their POR/enable thresholds, The controller will have fixed delay period t start ramp until the output voltage reaches 1.1V VBOOT voltage. Then, the controller will regulate the VR voltage at

1.1V for another fixed period t ISL6327 reads the VID signals. If the VID code is valid, ISL6327 will initiate the second soft-start ramp until the voltage reaches the VID voltage minus offset voltage.

. After this delay period, the VR will begin first soft-

. At the end of tD3 period,

VOUT, 500mV/DIV

tD3t

tD1

FIGURE 8. SOFT-START WAVEFORMS

EN_VTT

VR_RDY

500µs/DIV

After the DAC voltage reaches the final VID setting, VR_RDY will be set to high with the fixed delay t typical value for t

is 85µs.

. The

Fault Monitoring and Protection

The ISL6327 actively monitors output voltage and current to detect fault conditions. Fault monitors trigger protective measures to prevent damage to a microprocessor load. One common power good indicator is provided for linking to external system monitors. The schematic in Figure 9 outlines the interaction between the fault monitors and the VR_RDY signal.

VR_RDY Signal

The VR_RDY pin is an open-drain logic output to indicate that the soft-start period is completed and the output voltage is within the regulated range. VR_RDY is pulled low during shutdown and releases high after a successful soft-start and a fix delay time, t undervoltage, overvoltage, or overcurrent condition is detected, or the controller is disabled by a reset from EN_PWR, EN_VTT, POR, or VID OFF-code.

50%

. VR_RDY will be pulled low when an

VR_RDY

The soft-start time is the sum of the 4 periods as shown in Equation 14:

SStD1tD2tD3tD4

+++=

(EQ. 14)

is a fixed delay with the typical value as 1.36ms. tD3 is determined by the fixed 85µs plus the time to obtain valid VID voltage. If the VID is valid before the output reaches the

1.1V, the minimum time to validate the VID input is 500ns. Therefore the minimum t

is about 86µs.

During tD2 and tD4, ISL6327 digitally controls the DAC voltage change at 6.25mV per step. The time for each step is determined by the frequency of the soft-start oscillator which is defined by the resistor R soft-start ramp times t

from SS pin to GND. The two

and tD4 can be calculated based on

Equations 15 and 16:

1.1xR

----------------------- -

6.25x25

------------------------------------------------

For example, when VID is set to 1.5V and the R 100kΩ, the first soft-start ramp time t second soft-start ramp time t

VID

6.25x25

μs()=

1.1–()xR

μs()=

will be 256µs.

(EQ. 15)

(EQ. 16)

is set at

will be 704µs an d th e

85µA

AVG

VDIFF

DAC

SOFT-START, FAULT

AND CONTROL LOGIC

VID + 0.175V

FIGURE 9. VR_RDY AND PROTECTION CIRCUITRY

Undervoltage Detection

The undervoltage threshold is set at 50% of the VID voltage. When the output voltage at VSEN is below the undervoltage threshold, VR_RDY gets pulled low. When the output voltage comes back to 60% of the VID voltage, VR_RDY will return back to high.

Overvoltage Protection

Regardless of the VR being enabled or not, the ISL6327 overvoltage protection (OVP) circuit will be active after its POR. The OVP thresholds are different under different operation conditions. When VR is not enabled and before the 2nd soft-start, the OVP threshold is 1.275V. Once the controller detects a valid VID input, the OVP trip point will be changed to the VID voltage plus 175mV.

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Two acti on s are taken by the ISL6327 to protect the microprocessor load when an overvoltage condition occurs.

ISL6327

At the inception of an overvoltage event, all PWM outputs are commanded low instantly (less than 20ns). This causes the Intersil drivers to turn on the lower MOSFETs and pull the output voltage below a level to avoid damaging the load. When the VDIFF voltage falls below the DAC plus 75mV, PWM signals enter a high-impedance state. The Intersil drivers respond to the high-impedance input by turning off both upper and lower MOSFET s. If the overvoltage condition reoccurs, the ISL6327 will again command the lower MOSFETs to turn on. The ISL6327 will continue to protect the load in this fashion as long as the overvoltage condition occurs.

Once an overvoltage condition is detected, normal PWM operation ceases until the ISL6327 is reset. Cycling the voltage on EN_PWR, EN_VTT or VCC below the POR falling threshold will reset the controller. Cycling the VID codes will not reset the controller.

Overcurrent Protection

ISL6327 has two levels of overcurrent protection. Each phase is protected from a sustained overcurrent condition by limiting its peak current, while the combined phase currents are protected on an instantaneous basis.

In instantaneous protection mode, the ISL6327 utilizes the sensed average current I condition. See “Channel-Current Balance” on page 12 for more detail on how the average current is measured. The average current is continually compared with a constant 85µA reference current as shown in Figure 9. Once the average current exceeds the reference current, a comparator triggers the converter to shutdown.

At the beginning of overcurrent shutdown, the controller places all PWM signals in a high-impedance state within 20ns commanding the Intersil MOSFET driver ICs to turn off both upper and lower MOSFET s. The system remains in this state a period of 4096 switching cycles. If the controller is still enabled at the end of this wait period, it will attempt a soft-start. If the fault remains, the trip-retry cycles will continue indefinitely (as shown in Figure 10) until either controller is disabled or the fault is cleared. Note that the energy delivered during trip-retry cycling is much less than during full-load operation, so there is no thermal hazard during this kind of operation.

to detect an overcurrent

AVG

OUTPUT CURRENT

OUTPUT VOLTAGE

FIGURE 10. OVERCURRENT BEHAVIOR IN HICCUP MODE.

= 500kHz

2ms/DIV

The overcurrent protection level for the above two OCP modes can be adjusted by changing the value of current sensing resistors. In addition, ISL6327 can also adjust the average OCP threshold level by adjusting the value of the resistor from IOUT to GND. This provides additional safety for the voltage regulator.

The following equation can be used to calculate the value of the resistor R

IOUT

based on the desired OCP level I

IOUT

-------------------------------

AVG OCP2,

AVG, OCP2

(EQ. 17)

Current Sense Output

The ISL6327 has 2 current sense output pins IDROOP and IOUT; They are identical. In typical application, IDROOP pin is connected to FB pin for the application where load line is required. IOUT pin was designed for load current measurement. As shown in the typical application schematics on page 4 and page 5, load current information can be obtained by measuring the voltage at IOUT pin with a resistor connecting IOUT pin to the ground. When the programmable temperature compensation function of ISL6327 is properly used, the output current at IOUT pin is proportional to the load current as shown in Figure 11.

V_IOUT, 200mV/DIV

For the individual channel overcurrent protection, the ISL6327 continuously compares the sensed current signal of each channel with the 120µA reference current. If one channel current exceeds the reference current, ISL6327 will pull PWM signal of this channel to low for the rest of the switching cycle. This PWM signal can be turned on next cycle if the sensed channel current is less than the 120µA reference current. The peak current limit of individual channel will not trigger the converter to shutdown.

FIGURE 11. VOL TAGE A T IOUT PIN WITH A NTC NETWORK

PLACED BETWEEN IOUT TO GROUND WHEN LOAD CURRENT CHANGES

50A

100A

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ISL6327

Thermal Monitoring (VR_HOT/VR_FAN)

There are two thermal signals to indicate the temperature status of the voltage regulator: VR_HOT and VR_FAN. Both VR_FAN and VR_HOT are open-drain outputs, and external pull-up resistors are required. Those signals are valid only after the controller is enabled.

VR_FAN signal indicates that the temperature of the voltage regulator is high and more cooling airflow is needed. VR_HOT signal can be used to inform the system that the temperature of the voltage regulator is too high and the CPU should reduce its power consumption. VR_HOT signal may be tied to the CPU’s PROCHOT# signal.

The diagram of thermal monitoring function block is shown in Figure 12. One NTC resistor should be placed close to the power stage of the voltage regulator to sense the operational temperature, and one pull-up resistor is needed to form the voltage divider for TM pin. As the temperature of the power stage increases, the resistance of the NTC will reduce, resulting in the reduced voltage at TM pin. Figure 13 shows the TM voltage over the temperature for a typical design with a recommended 6.8kΩ NTC (P/N: NTHS0805N02N6801 from Vishay) and 1kΩ resistor RTM1. We recommend using those resistors for the accurate temperature compensation.

100

(%)

0 20 40 60 80 100 120 140

TEMPERATURE (°C)

FIGURE 13. THE RATIO OF TM VOLTAGE TO NTC

TEMPERATURE WITH RECOMMENDED PARTS

0.39*VCC

0.33*VCC

0.28*VCC

VR_FAN

There are two comparators with hysteresis to compare the TM pin voltage to the fixed thresholds for VR_FAN and VR_HOT signals respectively. VR_FAN signal is set to high when TM voltage is lower than 33% of VCC voltage, and is pulled to GND when TM voltage increases to above 39% of V

voltage. VR_HOT is set to high when TM voltage goes

below 28% of VCC voltage, and is pulled to GND when TM voltage goes back to above 33% of VCC voltage. Figure 14 shows the operation of those signals.

VCC

VR_FAN

TM1

NTC

FIGURE 12. BLOCK DIAGRAM OF THERMAL MONITORING

FUNCTION

0.33V

0.28V

VR_HOT

T1 T2 T3

TEMPERATURE

FIGURE 14. VR_HOT AND VR_FAN SIGNAL vs TM VOLTAGE

Based on the NTC temperature characteristics and the desired threshold of VR_HOT signal, the pull-up resistor RTM1 of TM pin is given by Equation 18:

TM1

NTC(T3)

2.75xR

NTC T3()

(EQ. 18)

is the NTC resistance at the VR_HOT threshold

temperature T3. The NTC resistance at the set point T2 and release point T1 of

VR_FAN signal can be calculated using Equations 19 and 20:

NTC T2()

NTC T1()

1.267xR

1.644xR

NTC T3()

(EQ. 19)

(EQ. 20)

With the NTC resistance value obtained from Equations 19 and 20, the temperature value T2 and T1 can be found from the NTC datasheet.

Temperature Compensation

ISL6327 supports inductor DCR sensing, or resistive sensing techniques. The inductor DCR have the positive temperature coefficient, which is about +0.38%/°C. Because the voltage across inductor is sensed for the output current information, the sensed current has the same po si tive temperature coefficient as the inductor DCR.

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ISL6327

In order to obtain the correct current information, there should be a way to correct the temperature impact on the current sense component. ISL6327 provides two methods: integrated temperature compensation and external temperature compensation.

Integrated Temperature Compensation

When TCOMP voltage is equal or greater than VCC/15, ISL6327 will utilize the voltage at TM and TCOMP pins to compensate the temperature impact on the sensed current. The block diagram of this function is shown in Figure 15.

TM1

Non-linear

NTC

TC1

TCOMP

TC2

FIGURE 15. BLOCK DIAGRAM OF INTEGRATED

A/D

D/A

4-bit

A/D

TEMPERATURE COMPENSATION

Channel current sense

Over current protection

Droop, Iout &

When the TM NTC is placed close to the current sense component (inductor), the temperature of the NTC will track the temperature of the current sense component. Therefore, the TM voltage can be utilized to obtain the temperature of the current sense component.

Based on VCC voltage, ISL6327 converts the TM pin voltage to a 6-bit TM digital signal for temperature compensation. With the non-linear A/D converter of ISL6327, TM digital signal is linearly proportional to the NTC temperature. For accurate temperature compensation, the ratio of the TM voltage to the NTC temperature of the practical design should be similar to that in Figure 13.

Depending on the location of the NTC and the air-flowing, the NTC may be cooler or hotter than the current sense component. TCOMP pin voltage can be utilized to correct the temperature difference between NTC and the current sense component. When a different NTC type or different voltage divider is used for the TM function, TCOMP voltage can also be used to compensate for the difference between the recommended TM voltage curve in Figure 14 and that of the actual design. According to the VCC voltage, ISL6327 converts the TCOMP pin voltage to a 4-bit TCOMP digital signal as TCOMP factor N.

sen6

sen5

sen4

sen3

sen2

sen1

TCOMP factor N is an integer between 0 and 15. The integrated temperature compensation function is disabled for N = 0. For N = 4, the NTC temperature is equal to the temperature of the current sense component. For N <4, the NTC is hotter than the current sense component. The NTC is cooler than the current sense component for N >4. When N >4, the larger TCOMP factor N, the larger the difference between the NTC temperature and the temperature of the current sense component.

ISL6327 multiplexes the TCOMP factor N with the TM digital signal to obtain the adjustment gain to compensate the temperature impact on the sensed channel current. The compensated channel current signal is used for droop and overcurrent protection functions.

Design Procedure

1. Properly choose the voltage divider for TM pin to match the TM voltage vs temperature curve with the recommended curve in Figure 13.

2. Run the actual board under the full load and the desired cooling condition.

3. After the board reaches the thermal steady state, record the temperature (T (inductor) and the voltage at TM and VCC pins.

4. Use Equation 21 to calculate the resistance of the TM NTC, and find out the corresponding NTC temperature T

from the NTC datasheet.

NTC

NTC T

()

NTC

5. Use Equation 22 to calculate the TCOMP fa ctor N:

209x T

()

CSC

--------------------------------------------------------

3xT

NTC

6. Choose an integral number close to the above result for the TCOMP factor. If this factor is higher than 15, use N = 15. If it is less than 1, use N = 1.

7. Choose the pull-up resistor R

8. If N = 15, do not need the pull-down resistor R otherwise obtain R

NxR

TC1

TC2

-----------------------

15 N–

9. Run the actual board under full load again with the proper resistors to TCOMP pin.

10. Record the output voltage as V1 immediately after the output voltage is stable with the full load; record the output voltage as V2 after the VR reaches the thermal steady state.

11. If the output voltage increases over 2mV as the temperature increases, i.e. V2 - V1 >2mV, reduce N and redesign R

; if the output voltage decreases over 2mV

TC2

as the temperature increases, i.e. V1 - V2 >2mV, increase N and redesign R

The design spreadsheet is available for those calculations.

) of the current sense component

CSC

VTMxR

TM1

------------------------------- -= VCCV–

T–

NTC

400+

4+=

TC1

using Equation 23:

TC2

(typical 10kΩ);

(EQ. 21)

(EQ. 22)

TC2

(EQ. 23)

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ISL6327

External Temperature Compensation

By pulling the TCOMP pin to GND, the integrated temperature compensation function is disabled. And one external temperature compensation network, shown in Figure 16, can be used to cancel the temperature impact on the droop (i.e. load line).

ISL63 2 7

INTERNAL

CIRCUIT

COMP

IDROOP

VDIFF

FIGURE 16. EXTERNAL TEMPERATURE COMPENSATION

The sensed current will flow out of IDROOP pin and develop the droop voltage across the resistor (R VDIFF pins. If R

resistance reduces as the temperature

) between FB and

increases, the temperature impact on the droop can be compensated. An NTC resistor can be placed close to the power stage and used to form R

. Due to the non-linear

temperature characteristics of the NTC, a resistor network is needed to make the equivalent resistance between FB and VDIFF pin reverse proportional to the temperature.

The external temperature compensation network can only compensate the temperature im pact on the droop, while it has no impact to the sensed current inside ISL6327. Therefore, this network cannot compensate for the temperature impact on the overcurrent protection function.

General Design Guide

This design guide is intended to provide a high-level explanation of the steps necessary to create a multiphase power converter. It i s assume d that the reader is familiar with many of the basic skills and techniques referenced below. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts for all common microprocessor applications.

Power Stages

The first step in designing a multiphase converter is to determine the number of phases. This determination depends heavily on the cost analysis which in turn depends on system constraints that differ from one design to the next. Principally, the designer will be concerned with whether components can be mounted on both sides of the circuit board; whether through-hole components are permitted; and the total board space available for power-supply circuitry. Generally speaking, the most economical solutions are

those in which each phase handles between 15A and 20A. All surface-mount designs will tend toward the lower end of this current range. If through-hole MOSFETs and inductors can be used, higher per-phase currents are possible. In cases where board space is the limiting constraint, current can be pushed as high as 40A per phase, but these designs require heat sinks and forced air to cool the MOSFETs, inductors, and heat-dissipating surfaces.

MOSFETS

The choice of MOSFETs depends on the current each MOSFET will be required to conduct; the switching frequency; the capability of the MOSFETs to dissipate heat; and the availability and nature of heat sinking and air flow.

LOWER MOSFET POWER CALCULATION

The calculation for heat dissipated in the lower MOSFET is simple, since virtually all of the heat loss in the lower MOSFET is due to current conducted through the channel resistance (r continuous output current; I current (see Equation 1); d is the duty cycle (V

). In Equation 24, IM is the maximum

DS(ON)

is the peak-to-peak inductor

P-P

OUT/VIN

); and

L is the per-channel inductance.

LOW 1,

DS ON()

⎛⎞

⎜⎟

----- N

⎝⎠

1d–()

LPP,

--------------------------------+=

1d–()

(EQ. 24)

An additional term can be added to the lower-MOSFET loss equation to account for additional loss accrued during the dead time when inductor current is flowing through the lower MOSFET body diode. This term is dependent on the diode forward voltage at I the length of dead times, t

, V

; the switching frequency, fS; and

D(ON)

and td2, at the beginning and the

end of the lower-MOSFET conduction interval respectively.

LOW 2,

DON()fS

⎛⎞

----- -

-------- -+

⎝⎠

⎛⎞

⎜⎟

----- -

⎝⎠

-------- -

–

(EQ. 25)

Thus the total maximum power dissipated in each lower MOSFET is approximated by the summation of P P

LOW,2

LOW,1

and

UPPER MOSFET POWER CALCULATION

In addition to r

losses, a large portion of the upper

DS(ON)

MOSFET losses are due to currents conducted across the input voltage (V

) during switching. Since a substantially

higher portion of the upper-MOSFET losses are dependent on switching frequency , the power cal culation is more complex. Upper MOSFET losses can be divided into separate components involving the upper-MOSFET switching times; the lower-MOSFET body-diode reverse-recovery charge, Q and the upper MOSFET r

conduction loss.

DS(ON)

When the upper MOSFET turns off, the lower MOSFET does not conduct any portion of the inductor current until the voltage at the phase node falls below ground. Once the lower MOSFET begins conducting, the current in the upper MOSFET falls to zero as the current in the lower MOSFET

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Page 24

ISL6327

ramps up to assume the full inductor current. In Equation 26, the required time for this commutation is t approximated associated power loss is P

≈

UP 1,VIN

⎛⎞

----- -

⎝⎠

⎛⎞

-------- -+ 2

----

⎜⎟

⎝⎠

and the

UP,1

(EQ. 26)

At turn-on, the upper MOSFET begins to conduct and this transition occurs over a time t approximate power loss is P

≈

UP 2,VIN

⎛⎞

⎜⎟

----- N

⎝⎠

⎛⎞

-------- -

----

⎜⎟

–

⎝⎠

. In Equation 27, the

UP,2

(EQ. 27)

A third component involves the lower MOSFET’s reverse recovery charge, Qrr. Since the inductor current has fully commutated to the upper MOSFET before the lower MOSFET’s body diode can draw all of Q

, it is conducted

through the upper MOSFET across VIN. The power dissipated as a result is P

UP 3,

VINQrrf

and is approximately

UP,3

(EQ. 28)

Finally, the resistive part of the upper MOSFET’s is given in Equation 29 as P

UP 4,rDS ON()

UP,4

⎛⎞

⎜⎟

----- N

⎝⎠

d+≈

---------12

(EQ. 29)

The total power dissipated by the upper MOSFET at full load can now be approximated as the summa ti on of the resu l ts from Equations 26, 27, 28 and 29. Since the power equations depend on MOSFET parameters, choosing the correct MOSFETs can be an iterative process involving repetitive solutions to the loss equations for different MOSFETs and different switching frequencies.

Current Sensing Resistor

The resistors connected to the Isen+ pins determine the gains in the load-line regulation loop and the channel-current balance loop as well as setting the overcurrent trip point. Select values for these resistors by the Equation 30.

OCP

ISEN

where R

----------------------85 10

ISEN

pin, N is the active channel number, R

------------- -=

–

is the sense resistor connected to the ISEN+

is the resistance of

the current sense element, either the DCR of the inductor or R desired overcurrent trip point. Typically, I

depending on the sensing method, and I

SENSE

can be chosen

OCP

to be 1.3 times the maximum load current of the specific application.

With integrated temperature compensation, the sensed current signal is independent on the operational temperature of the power stage, i.e. the temperature effect on the current sense element R temperature compensation function. R

is cancelled by the integrated

in Equation 30

should be the resistance of the current sense element at the room temperature.

OCP

(EQ. 30)

is the

When the integrated temperature compensation function is disabled by pulling the TCOMP pin to GND, the sensed current will be dependent on the operational temperature of the power stage, since the DC resistance of the current sense element may be changed according to the operational temperature. R

in Equation 30 should be the maximum DC

resistance of the current sense element at all the operational temperature.

In certain circumstances, it may be necessary to adjust the value of one or more ISEN resistors. When the components of one or more channels are inhibited from effectively dissipating their heat so that the affected channels run hotter than desired, choose new, smaller values of RISEN for the affected phases (see the section titled “Channel-Current Balance” o n page 12). Choose R

in proportion to the

ISEN,2

desired decrease in temperature rise in order to cause proportionally less current to flow in the hotter phase.

ΔT

ISEN 2,

ISEN

In Equation 31, make sure that ΔT

----------= ΔT

2 1

is the desired temperature

rise above the ambient temperature, and ΔT

is the measured

(EQ. 31)

temperature rise above the ambient temperature. While a single adjustment according to Equation 31 is usually sufficient, it may occasionally be necessary to adjust R

ISEN

two or more times to achieve optimal thermal balance between all channels.

Load-Line Regulation Resistor

The load-line regulation resistor is labelled RFB in Figure 5. Its value depends on the desired loadline requirement of the application.

The desired loadline can be calculated using Equation 32:

DROOP

------------------------ -= I

is the full load current of the specific application,

FL DROOP

--------------------------------- -=

is the desired voltage droop under the full

, the loadline regulation

ISEN

depending on the sensing method.

SENSE

is the sense

is the

(EQ. 32)

(EQ. 33)

where I and VR load condition.

Based on the desired loadline R resistor can be calculated using Equation 33:

where N is the active channel number, R resistor connected to the ISEN+ pin, and R resistance of the current sense element, either the DCR of the inductor or R

If one or more of the current sense resistors are adjusted for thermal balance, as in Equation 31, the load-line regulation resistor should be selected based on the average value of the current sensing resistors, as given in Equation 34:

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ISL6327

---------- R

ISEN(n)

∑

ISEN n()

is the current sensing resistor connected to

(EQ. 34)

where R the nth ISEN+ pin.

Compensation

The two opposing goals of compensating the voltage regulator are stability and speed. Depending on whether the regulator employs the optional load-line regulation as described in Load-Line Regulation, there are two distinct methods for achieving these goals.

COMPENSATING LOAD-LINE REGULATED CONVERTER

The load-line regulated converter behaves in a similar manner to a peak-current mode controller because the two poles at the output-filter L-C resonant frequency split with the introduction of current information into the control loop. The final location of these poles is determined by the system function, the gain of the current signal, and the value of the compensation components, R

Since the system poles and zero are affected by the values of the components that are meant to compensate them, the solution to the system equation becomes fairly complicated. Fortunately there is a simple approximation that comes very close to an optimal solution. Treating the system as though it were a voltage-mode regulator by compensating the L-C poles and the ESR zero of the voltage-mode approximation yields a solution that is always stable with very close to ideal transient performance.

C2 (OPTIONAL)

DROOP

and CC.

COMP

IDROOP

VDIFF

ISL6327

------------------- f

Case 1:

Case 2:

Case 3:

2π LC

CRFB

0.75V

------------------------------------=

2π VPPRFBf

------------------2π LC

CRFB

-------------------------------------------------------------=

2π()2f

------------------------------> 2π C ESR()

CRFB

0.75VINESR()C

-------------------------------------------------=

2π V

2π f0VppLC

----------------------------------- -=

0.75V

------------------------------<≤

2π C ESR()

VPP2π()2f

--------------------------------------------=

0.75 V

0.75V

VPPRFBLC

2π f0VppL

------------------------------------------=

0.75 V

PPRFBf0

ESR()

(EQ. 35)

In Equation 35, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent-series resistance of the bulk output-filter capacitance; and V

is the peak-to-

peak sawtooth signal amplitude as described in “Electrical Specifications” on page 6.

The optional capacitor C

, is sometimes needed to bypass

noise away from the PWM comparator (see Figure 18). Keep a position available for C

, and be prepared to install a high

frequency capacitor of between 22pF and 150pF in case any leading-edge jitter problem is noted.

Once selected, the compensation values in Equation 35 assure a stable converter with reasonable transient performance. In most cases, transient performance can be improved by making adjustments to R value of R

while observing the transient performance on an

. Slowly increase the

oscilloscope until no further improvement is noted. Normally, C

will not need adjustment. Keep the value of CC from

Equation 35 unless some performance issue is noted.

FIGURE 17. COMPENSATION CONFIGURATION FOR

LOAD-LINE REGULATED ISL6327 CIRCUIT

The feedback resistor, R

, has already been chosen as

outlined in “Load-Line Regulation Resistor” on page 24. Select a target bandwidth for the compensated system, f

The target bandwidth must be large enough to assure adequate transient performance, but smaller than 1/3 of the per-channel switching frequency. The values of the compensation components depend on the relationships of f to the L-C pole frequency and the ESR zero frequency. For each of the three cases in Equation 35, there are a separate set of equations for the compensation components.

COMPENSATION WITHOUT LOAD-LINE REGULATION

The non load-line regulated converter is accurately modeled as a voltage-mode regulator with two poles at the L-C resonant frequency and a zero at the ESR frequency. A type III controller, as shown in Figure 18, provides the necessary compensation.

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ISL6327

FIGURE 18. COMPENSATION CIRCUIT FOR ISL6327 BASED

CONVERTER WITHOUT LOAD-LINE REGULATION

COMP

IDROOP

VDIFF

ISL6327

The first step is to choose the desired bandwidth, f0, of the compensated system. Choose a frequency high enough to assure adequate transient performance but not higher than 1/3 of the switching frequency. The type-III compensator has an extra high-frequency pole, f

. This pole can be used for

added noise rejection or to assure adequate attenuation at the error-amplifier high-order pole and zero frequencies. A good general rule is to choose f higher if desired. Choosing f

= 10f0, but it can be

to be lower than 10f0 can

cause problems with too much phase shift below the system bandwidth.

In the solutions to the compensation equations, there is a single degree of freedom. For the solutions presented in Equation 36, R

is selected arbitrarily. The remaining

compensation components are then selected according to Equation 36.

R1R

C ESR()

-----------------------------------------=

LC C ESR()–

-----------------------------------------= R

0.75V

----------------------------------------------------------------------- -=

2π()2f0fHFLCRFBV

⎛⎞

VPP2π

⎝⎠

---------------------------------------------------------------------=

0.75V

----------------------------------------------------------------------- -=

2π()

f0fHFLCRFBV

f0fHFLCR

⎛⎞

2π fHFLC 1–

⎝⎠

⎛⎞

2π fHFLC 1–

⎝⎠

PP–

(EQ. 36)

PP–

In Equation 36, L is the per-channel filter inductance divided by the number of active channels; C is the sum total of all output capacitors; ESR is the equivalent-series resistance of the bulk output-filter capacitance; and V

is the peak-to-

peak sawtooth signal amplitude as described in “Electrical Specifications” on page 6.

Output Filter Design

The output inductors and the output capacitor bank together form a low-pass filter responsible for smoothing the pulsating voltage at the phase nodes. The output filter also must provide the transient energy until the regulator can respond. Because it has a low bandwidth compared to the switching frequency, the output filter necessarily limits the system transient response. The output capacitor must supply or sink load current while the current in the output inductors increases or decreases to meet the demand.

In high-speed converters, the output capacitor bank is usually the most costly (and often the largest) part of the circuit. Output filter design begins with minimizing the cost of this part of the circuit. The critical load parameters in choosing the output capacitors are the maximum size of the load step, ΔI; the load-current slew rate, di/dt; and the maximum allowable output-voltage deviation under transient loading, ΔV their capacitance, ESR, and ESL (equivalent series inductance).

At the beginning of the load transient, the output capacitors supply all of the transient current. The output voltage will initially deviate by an amount approximated by the voltage drop across the ESL. As the load current increases, the voltage drop across the ESR increases linearly until the load current reaches its final value. The capacitors selected must have sufficiently low ESL and ESR so that the total outputvoltage deviation is less than the allowable maximum. Neglecting the contribution of inductor current and regulator response, the output voltage initially deviates by an amount:

ΔV ESL()

The filter capacitor must have sufficiently low ESL and ESR so that ΔV < ΔV

Most capacitor solutions rely on a mixture of high-frequency capacitors with relatively low capacitance in combination with bulk capacitors having high capacitance but limited high-frequency performance. Minimizing the ESL of the high-frequency capacitors allows them to support the output voltage as the current increases. Minimizing the ESR of the bulk capacitors allows them to supply the increased current with less output voltage deviation.

The ESR of the bulk capacitors also creates the majority of the output-voltage ripple. As the bulk capacitors sink and source the inductor AC ripple current (see “Interleaving” on page 10 and Equation 2), a voltage develops across the bulk-capacitor ESR equal to I output capacitors are selected, the maximum allowable ripple voltage, V inductance.

Since the capacitors are supplying a decreasing portion of the load current while the regulator recovers from the

. Capacitors are characterized according to

MAX

-----ESR()ΔI+≈ dt

MAX

(ESR). Thus, once the

C(P-P)

P-P(MAX)

, determines the lower limit on the

(EQ. 37)

FN9276.4

May 5, 2008

Page 27

⎛⎞

L ESR()

⎝⎠

------------------------------------------------------------≥

–

INNVOUT

fSVINV

PPMAX()–

OUT

(EQ. 38)

transient, the capacitor voltage becomes slightly depleted. The output inductors must be capable of assuming the entire load current before the output voltage decreases more than ΔV

. This places an upper limit on inductance.

MAX

Equation 39 gives the upper limit on L for the cases when the trailing edge of the current transient causes a greater output-voltage deviation than the leading edge. Equation 40 addresses the leading edge. Normally, the trailing edge dictates the selection of L because duty cycles are usually less than 50%. Nevertheless, both inequalities should be evaluated, and L should be selected based on the lower of the two results. In each equation, L is the per-channel inductance, C is the total output capacitance, and N is the number of active channels.

2NCV

--------------------- ΔV

()

ΔI

()

1.25

≤

------------------------- - ΔV

()

ΔI

MAX

ΔI ESR()–≤

ΔIESR()– VINVO–

⎛⎞ ⎝⎠

(EQ. 39)

(EQ. 40)

Switching Frequency

There are a number of variables to consider when choosing the switching frequency, as there are considerable effects on the upper-MOSFET loss calculation. These effects are outlined in “MOSFETs” on page 23, and they establish the upper limit for the switching frequency. The lower limit is established by the requirement for fast transient response and small output-voltage ripple as outlined in “Output Filter Design” on page 26. Choose the lowest switching frequency that allows the regulator to meet the transient-response requirements.

ISL6327

0.3

)

RMS

0.2

0.1 I

= 0

L(P-P)

= 0.5 I

L(P-P)

INPUT-CAPACITOR CURRENT (I

00.4 1.00.2 0.6 0.8

FIGURE 19. NORMALIZED INPUT- CAP ACITOR RMS CURRENT

vs DUTY CYCLE FOR 2-PHASE CONVERTER

0.3 I

= 0

L(P-P)

)

RMS/

0.2

0.1

INPUT-CAPACITOR CURRENT (I

FIGURE 20. NORMALIZED INPUT- CAP ACITOR RMS CURRENT

= 0.25 I

L(P-P)

00.4 1.00.2 0.6 0.8

vs DUTY CYCLE FOR 3-PHASE CONVERTER

= 0.75 I

DUTY CYCLE (V

DUTY CYCLE (VO/VIN)

L(P-P)

O/VIN

= 0.5 I = 0.75 I

)

Switching frequency is determined by the selection of the frequency-setting resistor, R

(see the figures labelled

“Typical Application” on page 4 and page 5). Equation 3 is provided to assist in selecting the correct value for R

Input Capacitor Selection

The input capacitors are responsible for sourcing the AC component of the input current flowing into the upper MOSFETs. Their RMS current capacity must be sufficient to handle the AC component of the current drawn by the upper MOSFETs that is related to duty cycle and the number of active phases.

For a two phase design, use Figure 19 to determine the input-capacitor RMS current requirement given the duty cycle, maximum sustained output current (I of the per-phase peak-to-peak inductor current (I Select a bulk capacitor with a ripple current rating which will minimize the total number of input capacitors required to support the RMS current calculated. The voltage rating of the capacitors should also be at least 1.25 times greater than the maximum input voltage.

), and the ratio

L,PP

) to IO.

Figures 20 and 21 provide the same input RMS current information for three and four phase designs respectively. Use the same approach to selecting the bulk capacitor type and number as described previously.

Low capacitance, high-frequency ceramic capacitors are needed in addition to the bulk capacitors to suppress leading and falling edge voltage spikes. They result from the high current slew rates produced by the upper MOSFETs turning on and off. Select low ESL ceramic capacitors and place one as close as possible to each upper MOSFET drain to minimize board parasitic impedances and maximize suppression.

MULTIPHASE RMS IMPROVEMENT

Figure 22 is provided as a reference to demonstrate the dramatic reductions in input-capacitor RMS current upon the implementation of the multiphase topology. For example, compare the input RMS current requirements of a two-phase converter versus that of a single phase. Assume both converters have a duty cycle of 0.25, maximum sustained output current of 40A, and a ratio of I single phase converter would require 17.3A

to IO of 0.5. The

L(P-P)

RMS

current

FN9276.4

May 5, 2008

Page 28

ISL6327

capacity while the two-phase converter would only require

10.9A

. The advantages become even more pronounced

RMS

when output current is increased and additional phases are added to keep the component cost down relative to the single phase approach.

0.3 I

= 0

L(P-P)

)

RMS/

0.2

0.1

INPUT-CAPACITOR CURRENT (I

= 0.25 I

L(P-P)

00.4 1.00.2 0.6 0.8

FIGURE 21. NORMALIZED INPUT -CAP ACITOR RMS CURRENT

vs DUTY CYCLE FOR 4-PHASE CONVERTER

0.6

)

RMS/

0.4

L(P-P)

DUTY CYCLE (V

= 0.5 I = 0.75 I

O/VIN

)

Component Placement

Within the allotted implementation area, orient the switching components first. The switching components are the most critical because they carry large amounts of energy and tend to generate high levels of noise. Switching component placement should take into account power dissipation. Align the output inductors and MOSFETs such that spaces between the components are minimized while creating the PHASE plane. Place the Intersil MOSFET driver IC as close as possible to the MOSFETs they control to reduce the parasitic impedances due to trace length between critical driver input and output signals. If possible, duplicate the same placement of these components for each phase.

Next, place the input and output capacitors. Position one highfrequency ceramic input capacitor next to each upper MOSFET drain. Place the bulk input capacitors as close to th e upper MOSFET drains as dictated by the component size and dimensions. Long distances between input capacito rs and MOSFET drains result in too much trace inductance and a reduction in capacitor performance. Locate the output capacitors between the inductors and the load, while keeping them in close proximity to the microprocessor socket.

The ISL6327 can be placed off to one side or centered relative to the individual phase switching components. Routing of sense lines and PWM signals will guide final placement. Critical small signal components to place close to the controller include the ISEN resistors, R feedback resistor, and compensation components.

resistor,

Bypass capacitors for the ISL6327 and ISL66XX driver bias

0.2

INPUT-CAPACITOR CURRENT (I

= 0

L(P-P)

= 0.5 I

L(P-P)

00.4 1.00.2 0.6 0.8

= 0.75 I

DUTY CYCLE (V

O/VIN

)

FIGURE 22. NORMALIZED INPUT-CAP ACITOR RMS

CURRENT vs DUTY CYCLE FOR SINGLE-PHASE CONVERTER

Layout Considerations

The following layout strategies are intended to minimize the impact of board parasitic impedances on con verter performance and to optimize the heat-dissipating capabili ties of the printed-circuit board. These sections highlight some important practices which should not be overlooked during the

supplies must be placed next to their respective pins. Trace parasitic impedances will reduce their effectiveness.

Plane Allocation and Routing

Dedicate one solid layer, usually a middle layer, for a ground plane. Make all critical component ground connections with vias to this plane. Dedicate one additional layer for power planes; breaking the plane up into smaller islands of common voltage. Use the remaining layers for signal wiring.

Route phase planes of copper filled polygons on the top and bottom once the switching component placement is set. Size the trace width between the driver gate pins and the MOSFET gates to carry 4A of current. When routing components in the switching path, use short wide traces to reduce the associated parasitic impedances.

layout process.

All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.

Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality

Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implicat ion or oth erwise u nde r any p a tent or p at ent r ights of Intersil or its subsidiaries.

For information regarding Intersil Corporation and its products, see www.intersil.com

FN9276.4

May 5, 2008

Page 29

Package Outline Drawing

L48.7x7

48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE Rev 4, 10/06

7.00

PIN 1

INDEX AREA

ISL6327

44X

5.5

0.50 48

PIN #1 INDEX AREA

(4X) 0.15

( 6 . 80 TYP )

( 4 . 30 )

TOP VIEW

TYPICAL RECOMMENDED LAND PATTERN

7.00

0 . 90 ± 0 . 1

( 44X 0 . 5 )

( 48X 0 . 23 )

( 48X 0 . 60 )

48X 0 . 40± 0 . 1

BOTTOM VIEW

SIDE VIEW

0 . 2 REF

DETAIL "X"

0 . 00 MIN. 0 . 05 MAX.

BASE PLANE

4. 30 ± 0 . 15

M0.10 C AB

0.23 +0.07 / -0.05

SEE DETAIL "X"

0.10

SEATING PLANE

C0.08

NOTES:

Dimensions are in millimeters.1.

Dimensions in ( ) for Reference Only.

Dimensioning and tolerancing conform to AMSE Y14.5m-1994.

Unless otherwise specified, tolerance : Decimal ± 0.05

Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip.

Tiebar shown (if present) is a non-functional feature.

The configuration of the pin #1 identifier is optional, but must be

6. located within the zone indicated. The pin #1 indentifier may be

either a mold or mark feature.

FN9276.4

May 5, 2008

Datasheets isl6327%202, isl6327 Datasheet

Specifications and Main Features

Frequently Asked Questions

User Manual