a
TOP VIEW
(Not to Scale)
8
V+
1
VREF
7
OVER
2
SET HIGH
6
UNDERSET LOW
3
5
HEATERGND
4
Airflow and Temperature Sensor
TMP12*
FEATURES
Temperature Sensor Includes 100 Ω Heater
Heater Provides Power IC Emulation
Accuracy 63°C typ. from 240°C to 1100°C
Operation to 1150°C
5 mV/°C Internal Scale-Factor
Resistor Programmable Temperature Setpoints
20 mA Open-Collector Setpoint Outputs
Programmable Thermal Hysteresis
Internal 2.5 V Reference
Single 5 V Operation
400 µA Quiescent Current (Heater OFF)
Minimal External Components
APPLICATIONS
System Airflow Sensor
Equipment Over-Temperature Sensor
Over-Temperature Protection
Power Supply Thermal Sensor
Low-Cost Fan Controller
GENERAL DESCRIPTION
The TMP12 is a silicon-based airflow and temperature sensor
designed to be placed in the same airstream as heat generating
components that require cooling. Fan cooling may be required
continuously, or during peak power demands, e.g. for a power
supply, and if the cooling systems fails, system reliability and/or
safety may be impaired. By monitoring temperature while emulating a power IC, the TMP12 can provide a warning of cooling
system failure.
The TMP12 generates an internal voltage that is linearly proportional to Celsius (Centigrade) temperature, nominally
15 mV/°C. The linearized output is compared with voltages
from an external resistive divider connected to the TMP12’s
2.5 V precision reference. The divider sets up one or two reference voltages, as required by the user, providing one or two
temperature setpoints. Comparator outputs are open-collector
transistors able to sink over 20 mA. There is an on-board hysteresis generator provided to speed up the temperature-setpoint
output transitions, this also reduces erratic output transitions in
noisy environments. Hysteresis is programmed by the external
resistor chain and is determined by the total current drawn from
the 2.5 V reference. The TMP12 airflow sensor also incorporates a precision, low temperature coefficient 100 Ω heater
resistor that may be connected directly to an external 5 V supply. When the heater is activated it raises the die temperature in
FUNCTIONAL BLOCK DIAGRAM
HYSTERESIS
VREF
SET
HIGH
SET
LOW
GND
CURRENT
CURRENT
MIRROR
VOLTAGE
REFERENCE
AND
SENSOR
WINDOW
COMPARATOR
1kΩ
+
-
I
HYS
+
+
-
HYSTERESIS
VOLTAGE
V+
OVER
UNDER
HEATER
100Ω
PINOUTS
DIP And SO
the DIP package approximately 20°C above ambient (in still
air). The purpose of the heater in the TMP12 is to emulate a
power IC, such as a regulator or Pentium CPU which has a high
internal dissipation.
When subjected to a fast airflow, the package and die temperatures of the power device and the TMP12 (if located in the
same airstream) will be reduced by an amount proportional to
the rate of airflow. The internal temperature rise of the TMP12
may be reduced by placing a resistor in series with the heater, or
reducing the heater voltage.
The TMP12 is intended for single 5 V supply operation, but will
operate on a 12 V supply. The heater is designed to operate from
5 V only. Specified temperature range is from 240°C to 1125°C,
operation extends to 1150°C at 5 V with reduced accuracy.
The TMP12 is available in 8-pin plastic DIP and SO packages.
*Protected by U.S. Patent No. 5,195,827.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
© Analog Devices, Inc., 1995
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700 Fax: 617/326-8703
TMP12–SPECIFICATIONS
(VS = 15 V, 240°C ≤ TA ≤ 1125°C unless otherwise noted.)
Parameter Symbol Conditions Min Typ Max Units
ACCURACY
Accuracy (High, Low Setpoints) T
Accuracy (High, Low Setpoints) T
Internal Scale Factor T
Power Supply Rejection Ratio PSRR 4.5 V ≤ 1V
Linearity T
Repeatability T
Long Term Stability T
= 125°C 62 63 °C
A
= 240°C to 1100°C 63 65 °C
A
= 240°C to 1100°C 14.9 15 15.1 mV/°C
A
= 240°C to 1125°C 0.5 °C
A
= 240°C to 1125°C 0.3 °C
A
= 1125°C for 1 k Hrs 0.3 °C
A
≤ 5.5 V 0.1 0.5 °C/V
S
SETPOINT INPUTS
Offset Voltage V
OS
Output Voltage Drift TCV
Input Bias Current I
B
OS
0.25 mV
3 µV/°C
25 100 nA
VREF OUTPUT
Output Voltage VREF T
Output Voltage VREF T
= 125°C, No Load 2.49 2.50 2.51 V
A
= 240°C to 1100°C, 2.5 60.015 V
A
No Load
Output Drift TC
Output Current, Zero Hysteresis I
VREF
Hysteresis Current Scale Factor SF
VREF
HYS
210 ppm/°C
7 µA
5 µA/°C
OPEN-COLLECTOR OUTPUTS
I
Output Low Voltage V
Output Low Voltage V
Output Leakage Current I
Fall Time t
OL
OL
OH
HL
= 1.6 mA 0.25 0.4 V
SINK
I
= 20 mA 0.6 V
SINK
V
= 12 V 1 100 µA
S
See Test Load 40 ns
HEATER
T
Resistance R
H
Temperature Coefficient T
Maximum Continuous Current I
H
= 125°C 97 100 103 Ω
A
= 240°C to 1125°C 100 ppm/°C
A
See Note 1 60 mA
POWER SUPPLY
Supply Range 1V
Supply Current I
NOTES
1
Guaranteed but not tested.
2
TMP12 is specified for operation from a 5 V supply. However, operation is allowed up to a 12 V supply, but not tested at 12 V. Maximum heater supply is 6 V.
S
SY
I
SY
Unloaded at 15 V 400 600 µA
Unloaded at 112 V
2
4.5 5.5 V
450 µA
Specifications subject to change without notice.
TEST LOAD
1kΩ
20pF
REV. 0–2–
TMP12
WAFER TEST LIMITS
(VS = 15 V, GND = O V, TA = 125°C, unless otherwise noted.)
Parameter Symbol Conditions Min Typ Max Units
ACCURACY
Accuracy (High, Low Setpoints) T
Internal Scale Factor T
= 125°C 63 °C
A
= 125°C 14.9 15 15.1 mV/°C
A
SETPOINT INPUTS
Input Bias Current I
B
100 nA
VREF OUTPUT
Output Voltage VREF T
= 125°C, No Load 2.49 2.51 V
A
OPEN-COLLECTOR OUTPUTS
I
Output Low Voltage V
Output Leakage Current I
OL
OH
= 1.6 mA 0.4 V
SINK
V
= 12 V 100 µA
S
HEATER
Resistance R
H
T
= 125°C 97 100 103 Ω
A
POWER SUPPLY
Supply Range 1V
Supply Current I
NOTE
Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed
for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.
SY
S
Unloaded at 15 V 600 µA
4.5 5.5 V
DICE CHARACTERISTICS
Die Size 0.078 3 0.071 inch, 5,538 sq. mils
(1.98 3 1.80 mm, 3.57 sq. mm)
Transistor Count: 105
8
7
6
5
1. VREF
2. SET HIGH INPUT
3. SET LOW INPUT
4. GND
5. HEATER
UNDER OUTPUT
6.
OVER OUTPUT
7.
8. V1
1
2
3
4
For additional DICE ordering information, refer to databook.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the TMP12 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. 0
–3–
TMP12
ABSOLUTE MAXIMUM RATINGS*
Supply Voltage. . . . . . . . . . . . . . . . . . . . . . . . 20.3 V to 115 V
Heater Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 V
Setpoint Input Voltage . . . . . . . . . . . 20.3 V to [(V1) 10.3 V]
Reference Output Current . . . . . . . . . . . . . . . . . . . . . . . . 2 mA
Open-Collector Output Current . . . . . . . . . . . . . . . . . . 50 mA
Open-Collector Output Voltage . . . . . . . . . . . . . . . . . . . 115 V
Operating Temperature Range . . . . . . . . . . 255°C to 1150°C
Dice Junction Temperature . . . . . . . . . . . . . . . . . . . . . 1175°C
Storage Temperature Range . . . . . . . . . . . . 265°C to 1160°C
Lead Temperature(Soldering, 60 sec) . . . . . . . . . . . . . 1300°C
Package Type Θ
JA
8-Pin Plastic DIP (P) 103
8-Lead SOIC (S) 158
NOTES
1
ΘJA is specified for device in socket (worst case conditions).
2
ΘJA is specified for device mounted on PCB.
CAUTION
Θ
1
43 °C/W
2
43 °C/W
JC
Units
1. Stresses above those listed under “Absolute Maximum
Ratings” may cause permanent damage to the device. This is a
stress rating only and functional operation at or above this
specification is not implied. Exposure to the above maximum
rating conditions for extended periods may affect device reliability.
2. Digital inputs and outputs are protected, however, permanent
damage may occur on unprotected units from high-energy
electrostatic fields. Keep units in conductive foam or packaging
at all times until ready to use. Use proper antistatic handling
procedures.
3. Remove power before inserting or removing units from their
sockets.
ORDERING GUIDE
FUNCTIONAL DESCRIPTION
The TMP12 incorporates a heating element, temperature sensor, and two user-selectable setpoint comparators on a single
substrate. By generating a known amount of heat, and using the
setpoint comparators to monitor the resulting temperature rise,
the TMP12 can indirectly monitor the performance of a
system’s cooling fan.
The TMP12 temperature sensor section consists of a bandgap
voltage reference which provides both a constant 2.5 V output
and a voltage which is proportional to absolute temperature
(VPTAT). The VPTAT has a precise temperature coefficient of
5 mV/K and is 1.49 V (nominal) at 125°C. The comparators
compare VPTAT with the externally set temperature trip points
and generate an open-collector output signal when one of their
respective thresholds has been exceeded.
The heat source for the TMP12 is an on-chip 100 Ω low
tempco thin-film resistor. When connected to a 5 V source, this
resistor dissipates:
2
PD =
V
R
52 V
100 Ω
= 0.25 W ,=
which generates a temperature rise of about 32°C in still air for
the SO packaged device. With an airflow of 450 feet per minute
(FPM), the temperature rise is about 22°C. By selecting a temperature setpoint between these two values, the TMP12 can provide
a logic-level indication of problems in the cooling system.
A proprietary, low tempco thin-film resistor process, in conjunction with production laser trimming, enables the TMP12 to
provide a temperature accuracy of 63°C (typ) over the rated
temperature range. The open-collector outputs are capable of
sinking 20 mA, allowing the TMP12 to drive small control relays directly. Operating from a single 15 V supply, the quiescent
current is only 600 µA (max), without the heater resistor current.
Temperature Package Package
Model/Grade Range
1
Description Option
TMP12FP XIND Plastic DIP N-8
TMP12FS XIND SOIC SO-8
TMP12GBC 125°C Die
NOTE
1
XIND = 240°C to 1125°C
REV. 0–4–
TMP12
35
V = 5V
SO–8 SOLDERED TO .5 "
30
25
a. 0 FPM
20
b. 250 FPM
c. 450 FPM
15
d. 600 FPM
10
5
0
JUNCTION TEMPERATURE RISE ABOVE AMBIENT – °C
0
HEATER RESISTOR POWER DISSIPATION – mW
50 100 150 200
.3" Cu PCB
a
AIR FLOW RATES
b
d
c
250
Figure 1. SOIC Junction Temperature Rise vs. Heater
Dissipation
25
V = 5V
PDIP SOLDERED TO 2"
20
a. 0 FPM
b. 250 FPM
15
c. 450 FPM
d. 600 FPM
10
5
0
JUNCTION TEMPERATURE RISE ABOVE AMBIENT – °C
0 25050 100 150 200
HEATER RESISTOR POWER DISSIPATION – mW
1.31" Cu PCB
b
a
AIR FLOW RATES
c
d
140
TRANSITION FROM 100°C STIRRED
BATH TO FORCED
120
V = 5V, NO LOAD, HEATER OFF
SO–8 SOLDERED TO .5" .3" Cu PCB
100
PDIP SOLDERED TO 2" 1.31" Cu PCB
80
60
TIME CONSTANT – sec
40
20
0
0 700100
25°C AIR
a. PDIP PCB
b. SOIC PCB
a
b
200 300 400 500 600
AIR VELOCITY – FPM
Figure 4. Package Thermal Time Constant in Forced Air
120
TRANSITION FROM STILL 25°C
110
100
90
80
70
60
50
40
30
JUNCTION TEMPERATURE – °C
20
10
0
02
AIR TO STIRRED
a
100°C BATH
V = 5V, NO LOAD, HEATER OFF
SO–8 SOLDERED TO .5" .3" Cu PCB
b
PDIP SOLDERED TO 2" 1.31" Cu PCB
a. SO–8 PCB
b. PDIP PCB
468101214161820
TIME – sec
Figure 2. PDIP Junction Temperature Rise vs. Heater
Dissipation
70
a. SO–8, HTR @ 5V
65
b. PDIP, HTR @ 5V
60
c. SO–8, HTR @ 3V
d. PDIP, HTR @ 3V
55
50
45
40
35
30
25
20
V = 5V RHEATER TO EXTERNAL
15
JUNCTION TEMPERATURE – °C
SUPPLY TURNED ON @ t = 5 sec
SO–8 SOLDERED TO .5" .3" COPPER PCB
10
PDIP SOLDERED TO 2" 1.31 COPPER PCB
5
0
0 10 20 30 40 50 60 70 80 90 100 110 120 130
TIME – sec
a
b
c
d
Figure 3. Junction Temperature Rise in Still Air
REV. 0 –5–
Figure 5. Thermal Response Time in Stirred Oil Bath
102
101.5
100.5
HEATER RESISTANCE – Ω
101
100
99.5
98.5
99
98
-75
-25 25 75 125 175
TEMPERATURE – °C
V+ = +5V
Figure 6. Heater Resistance vs. Temperature
TMP12
2.52
V = 5V, NO LOAD, HEATER OFF
2.515
2.51
2.505
2.5
REFERENCE VOLTAGE – V
2.495
2.49
-75 175
-25 25 75 125
TEMPERATURE – °C
Figure 7. Reference Voltage vs. Temperature
5
START-UP VOLTAGE DEFINED AS OUTPUT
READING BEING WITHIN
4.5
4
3.5
START-UP SUPPLY VOLTAGE – V
3
-75 175
-25 25 75 125
TEMPERATURE – °C
5 °C OF OUTPUT AT 5V
NO LOAD, HEATER OFF
6
5
4
3
2
1
0
-1
-2
-3
ACCURACY ERROR – °C
-4
-5
-6
-50 -25 25 75 125
b
0 50 100
TEMPERATURE – °C
a. MAXIMUM LIMIT
b. ACCURACY ERROR
a
c. MINIMUM LIMIT
Figure 10. Accuracy Error vs. Temperature
500
450
400
350
300
250
200
150
SUPPLY CURRENT – µA
100
50
0
081
234
Ta = 25°C, NO LOAD, HEATER OFF
5
SUPPLY VOLTAGE – V
c
67
Figure 8. Start-up Voltage vs. Temperature
500
475
450
425
400
375
SUPPLY CURRENT – µA
350
325
300
-75 175
V = 5V, NO LOAD, HEATER OFF
-25 25 75 125
TEMPERATURE – °C
Figure 9. Supply Current vs. Temperature
Figure 11. Supply Current vs. Supply Voltage
0.5
V = 4.5 TO 5.5V
NO LOAD, HEATER OFF
-25 25 75 125
TEMPERATURE – °C
POWER SUPPLY REJECTION – °C/V
0.4
0.3
0.2
0.1
0
-75
Figure 12. VPTAT Power Supply Rejection vs.
Temperature
175
REV. 0–6–
TMP12
40
38
36
34
32
30
28
V
26
24
22
OPEN COLLECTOR SINK CURRENT – mA
20
-75
-25
= 1V, V = 5V
OL
25 75 125 175
TEMPERATURE – °C
Figure 13. Open-Collector Output Sink Current vs.
Temperature
APPLICATIONS INFORMATION
A typical application for the TMP12 is shown in Figure 15. The
TMP12 package is placed in the same cooling airflow as a
high-power dissipation IC. The TMP12’s internal resistor produces a temperature rise which is proportional to air flow, as
shown in Figure 16. Any interruption in the airflow will produce
an additional temperature rise. When the TMP12 chip temperature exceeds a user-defined setpoint limit, the system controller
can take corrective action, such as: reducing clock frequency,
shutting down unneeded peripherals, turning on additional fan
cooling, or shutting down the system.
PGA
SOCKET
PGA
PACKAGE
POWER I.C.
AIR FLOW
PC BOARD
TMP12
Figure 15. Typical Application
65
60
55
a
b
700
a. LOAD = 10mA
600
b. LOAD = 5mA
c. LOAD = 1mA
500
400
300
200
100
OPEN–COLLECTOR OUTPUT VOLTAGE – mV
0
-75 175
-25 25 75 125
V = 5V
TEMPERATURE – °C
a
b
c
Figure 14. Open-Collector Voltage vs. Temperature
Temperature Hysteresis
The temperature hysteresis at each setpoint is the number
of degrees beyond the original setpoint temperature that
must be sensed by the TMP12 before the setpoint comparator will be reset and the output disabled. Hysteresis
prevents “chatter” and “motorboating” in feedback control
systems. For monitoring temperature in computer systems,
hysteresis prevents multiple interrupts to the CPU which
can reduce system performance.
Figure 17 shows the TMP12’s hysteresis profile. The hysteresis is programmed, by the user, by setting a specific load
current on the reference voltage output VREF. This output
current, I
, is also called the hysteresis current. I
REF
REF
is mirrored internally by the TMP12, as shown in the functional
block diagram, and fed to a buffer with an analog switch.
OUTPUT
VOLTAGE
OVER, UNDER
LO
HI
HYSTERESIS
LOW
HYSTERESIS HIGH =
HYSTERESIS LOW
HYSTERESIS
HIGH
50
45
DIE TEMPERATURE (°C)
40
35
a. TMP12 DIE TEMP NO AIR FLOW
b. HIGH SET POINT
c. LOW SET POINT
d. TMP12 DIE TEMP MAX AIR FLOW
e. SYSTEM AMBIENT TEMPERATURE
50 100 150 200 2500
TMP12 PD (mW)
c
d
e
Figure 16. Choosing Temperature Setpoints
REV. 0 –7–
T
SETLOW
TEMPERATURE
T
SETHIGH
Figure 17. TMP12 Hysteresis Profile
After a temperature setpoint has been exceeded and a comparator tripped, the hysteresis buffer output is enabled. The
result is a current of the appropriate polarity which generates a hysteresis offset voltage across an internal 1 kΩ
resistor at the comparator input. The comparator output
remains “on” until the voltage at the comparator input,
now equal to the temperature sensor voltage VPTAT
summed with the hysteresis effect, has returned to the programmed setpoint voltage. The comparator then returns
TMP12
LOW, deactivating the open-collector output and disabling the
hysteresis current buffer output. The scale factor for the programmed hysteresis current is:
I = I
= 5 µA/°C 1 7 µA
VREF
Thus, since VREF = 2.5 V, a reference load resistance of 357 kΩ
or greater (output current of 7 µA or less) will produce a tem-
perature setpoint hysteresis of zero degrees. For more details, see
the temperature programming discussion below. Larger values of
load resistance will only decrease the output current below 7 µA,
but will have no effect on the operation of the device. The
amount of hysteresis is determined by selecting an appropriate
value of load resistance for VREF, as shown below.
Programming the TMP12
The basic thermal monitoring application only requires a simple
three-resistor ladder voltage divider to set the high and low
setpoints and the hysteresis. These resistors are programmed in
the following sequence:
1. Select the desired hysteresis temperature.
2. Calculate the hysteresis current, I
VREF
3. Select the desired setpoint temperatures.
4. Calculate the individual resistor divider ladder values needed
to develop the desired comparator setpoint voltages at the
Set High and Set Low inputs.
The hysteresis current is readily calculated, as shown above. For
example, to produce 2 degrees of hysteresis I
to 17 µA. Next, the setpoint voltages V
SETHIGH
should be set
VREF
and V
SETLOW
are
determined using the VPTAT scale factor of 5 mV/K = 5 mV/
(°C 1 273.15), which is 1.49 V for 125°C. Finally, the divider
resistors are calculated, based on the setpoint voltages.
The setpoint voltages are calculated from the equation:
= (T
V
SET
This equation is used to calculate both the V
V
values. A simple 3-resistor network, as shown in Figure
SETLOW
1 273.15)(5 mV/°C)
SET
SETHIGH
and the
18, determines the setpoints and hysteresis value. The equations
used to calculate the resistors are:
R1 (kΩ) = (V
R2 (kΩ) = (V
R3 (kΩ) = V
(
VREF
– V
SETHIGH
(V
– V
SETHIGH
SETLOW
V
SETLOW
2 V
REF
SETHIGH
SETLOW/IVREF
= 2.5 V
VREF
) / I
VREF
V
SETHIGH
) / I
VREF
V
SETLOW
/ I
VREF
SETHIGH
2 V
SETLOW
= R1
= R2
= R3
GND
)/I
VREF
I
VREF
)/I
1
2
3
4
= (2.5 V 2 V
VREF
TMP12
SETHIGH
V+
8
7
OVER
6
UNDER
5
HEATER
)/I
VREF
Figure 18. TMP12 Setpoint Programming
For example, setting the high setpoint for 180°C, the low
setpoint for 155°C, and hysteresis for 3°C produces the
following values:
I
HYS
= I
= (3°C 3 5 µA/°C) 1 7 µA = 15 µA 1 7 µA =
VREF
22 µA
V
SETHIGH
= (T
1 273.15)(5 mV/°C) = (80°C 1
SETHIGH
273.15)(5 mV/°C) = 1.766 V
V
SETLOW
= (T
1 273.15)(5 mV/°C) = (55°C 1 273.15)
SETLOW
(5 mV/°C) = 1.641 V
)/I
R1 (kΩ) = (VREF 2 V
SETHIGH
= (2.5 V 2 1.766 V)/
VREF
22 µA = 33.36 kΩ
R2 (kΩ) = (V
SETHIGH
2 V
SETLOW
)/I
= (1.766 V 2 1.641 V)/
VREF
22 µA = 5.682 kΩ
R3 (kΩ) = V
SETLOW/IVREF
= (1.641 V)/22 µA = 74.59 kΩ
The total of R1 1 R2 1 R3 is equal to the load resistance
needed to draw the desired hysteresis current from the
reference, or I
VREF
.
The nomograph of Figure 19 provides an easy method of
determining the correct VPTAT voltage for any temperature.
Simply locate the desired temperature on the appropriate scale
(K, °C or °F) and read the corresponding VPTAT value from
the bottom scale.
218 248 273 298 323 348 373 398
K
–55 –25
°C
–67 –25 0 32 50 77 100 150 200 212 257
°F
1.09 1.24 1.365 1.49 1.615 1.74 1.865 1.99
VPTAT
–18
0 25 50 75 100 125
Figure 19. Temperature 2 VPTAT Scale
The formulas shown above are also helpful in understanding the
calculations of temperature setpoint voltages in circuits other
than the standard two-temperature thermal/airflow monitor. If a
setpoint function is not needed, the appropriate comparator input should be disabled. SETHIGH can be disabled by tying it
to V1 or VREF, SETLOW by tying it to GND. Either output
can be left disconnected.
Selecting Setpoints
Choosing the temperature setpoints for a given system is an empirical process, because of the wide variety of thermal issues in
any practical design. The specific setpoints are dependent on
such factors as airflow velocity in the system, adjacent component location and size, PCB thickness, location of copper
ground planes, and thermal limits of the system.
The TMP12’s temperature rise above ambient is proportional to
airflow (Figures 1, 2 and 16). As a starting point, the low
setpoint temperature could be set at the system ambient temperature (inside the enclosure) plus one half of the temperature
rise above ambient (at the actual airflow in the system). With
this setting, the low limit will provide a warning either if the fan
output is reduced or if the ambient temperature rises (for example, if the fan’s cool air intake is blocked). The high setpoint
could then be set for the maximum system temperature to provide a final system shutdown control.
REV. 0–8–
TMP12
Measuring the TMP12 Internal Temperature
As previously mentioned, the TMP12’s VPTAT generator represents the chip temperature with a slope of 5 mV/K. In some cases,
selecting the setpoints is made easier if the TMP12’s internal
VPTAT voltage (and therefore the chip temperature) is known.
For example, the case temperature of a high power microprocessor
can be monitored with a thermistor, thermocouple, or other measurement method. The case temperature can then be correlated
with the TMP12’s temperature to select the setpoints.
The TMP12’s VPTAT voltage is not available externally, so indirect methods must be used. Since the VPTAT voltage is applied to
the internal comparators, measuring the voltage at which the digital
output changes state will reflect the VPTAT voltage.
A simple method of measuring the TMP12 VPTAT is shown in
Figure 20. To measure VPTAT, adjust potentiometer R1 until
the LED turns ON. The voltage at Pin 2 of the TMP12 will
then match the TMP12’s internal VPTAT.
VPTAT
R1
R1
200K
200K
TMP12
1
VREF
2
SET
OVER
HIGH
3
SET
UNDER
LOW
4
GND
HEATER
+5V
8
V+
7
6
NC
5
330
+5V
LED
+5V
Figure 20. Measuring VPTAT with a Potentiometer
The method described in Figure 20 can be automated by replacing the discrete resistors with a digital potentiometer. The
improved circuit, shown in Figure 21, permits the VPTAT voltage to be monitored with a microprocessor or other digital
controller. The AD8402-100 provides two 100 kΩ potentiom-
µC
INTERFACE
OVER
+5V
eters which are adjusted to 8-bit resolution via a 3-wire serial interface. The controller simply sweeps the wiper of
potentiometer 1 from the A1 terminal to the B1 terminal
(digital value = 0), while monitoring the comparator output
at Pin 7 of the TMP12. When Pin 7 goes low, the voltage
at Pin 2 equals the VPTAT voltage. This Circuit sweeps
Pin 2's voltage from maximum to minimum, so that the
TMP12's setpoint hystersis will not affect the reading.
The circuit of Figure 21 provides approximately 1°C of
resolution. The two potentiometers divide VREF by two,
and the 8-bit potentiometer further divides VREF by 256,
so the resolution is:
=
2.5 V
2
28
VREF
Resolution = = 4.9 mV
2
2N
where VREF is the voltage reference output (Pin 1 of the
TMP12) and N is the resolution of the AD8402. Since the
VPTAT has a slope of 5 mV/K, the AD8402 provides 1°C
of resolution. The adjustment range of this circuit extends
from VREF/2 (i.e. 1.25 V, or 223°C) to VREF 2 1 LSB
(i.e. 2.5 V 2 4.9 mV, or 226°C). The VPTAT is therefore:
VPTAT = 1.25 V + (Digital Count 4.9 mV)
where Digital Count is the value sent to the AD8402 which
caused the setpoint 1 output to go LOW.
A third way to measure the VPTAT voltage is to close a
feedback loop around one of the TMP12’s comparators.
This causes the comparator to oscillate, and in turn forces
the voltage at the comparator input to equal the VPTAT
voltage. Figure 22 is a typical circuit for this measurement.
An OP193 operational amplifier, operating as an integrator,
provides additional loop-gain to ensure that the TMP12
comparator will oscillate.
6
10 11
A1 13
W1
B1 14
A2 3
W2 4
B2 2
12
NC
SDI
CLK
CS
AD8402–100
9
8
7
V
RSSHDN
DD
SERIAL
DATA
INTERFACE
DGNDAGND
15
Figure 21. Measuring VPTAT with a Digital Potentiometer
REV. 0 –9–
VREF
1
2
3
4
TEMPERATURE
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
TMP12
VPTAT
100
8
7
NC
6
5
+5V
TMP12
∆T = P
DISS
JA = 0.012 W 158°C/W = 1.9°C
due to IR voltage drops and coupling of external noise sources.
In any case, a 0.1 µF capacitor for power supply bypassing is
Understanding Error Sources
The accuracy of the VPTAT sensor output is well characterized
and specified, however preserving this accuracy in a thermal
monitoring control system requires some attention to minimizing the various potential error sources. The internal sources of
setpoint programming error include the initial tolerances and
temperature drifts of the reference voltage VREF, the setpoint
comparator input offset voltage and bias current, and the hysteresis current scale factor. When evaluating setpoint programming
errors, remember that any VREF error contribution at the comparator inputs is reduced by the resistor divider ratios. Each
comparator’s input bias current drops to less than 1 nA (typ)
when the comparator is tripped. This change accounts for some
setpoint voltage error, equal to the change in bias current multi-
always recommended at the chip
Safety Considerations in Heating and Cooling System Design
Designers should anticipate potential system fault conditions
that may result in significant safety hazards which are outside
the control of and cannot be corrected by the TMP12-based circuit. Governmental and Industrial regulations regarding safety
requirements and standards for such designs should be observed
where applicable.
Self-Heating Effects
In some applications the user should consider the effects of selfheating due to the power dissipated by the open-collector outputs,
which are capable of sinking 20 mA continuously. Under full load,
the TMP12 open-collector output device is dissipating:
plied by the effective setpoint divider ladder resistance to ground.
The thermal mass of the TMP12 package and the degree of
thermal coupling to the surrounding circuitry are the largest factors in determining the rate of thermal settling, which ultimately
which in a surface-mount SO package accounts for a temperature increase due to self-heating of:
determines the rate at which the desired temperature measurement accuracy may be reached (see graph in Figure 3). Thus,
one must allow sufficient time for the device to reach the final
temperature. The typical thermal time constant for the SOIC
plastic package is approximately 70 seconds in still air. Therefore, to reach the final temperature accuracy within 1%, for a
temperature change of 60 degrees, a settling time of 5 time constants, or 6 minutes, is necessary. Refer to Figure 4.
External error sources to consider are the accuracy of the external
programming resistors, grounding error voltages, and thermal gradients. The accuracy of the external programming resistors directly
impacts the resulting setpoint accuracy. Thus, in fixed-temperature
applications the user should select resistor tolerances appropriate
to the desired programming accuracy. Since setpoint resistors are
typically located in the same air flow as the TMP12, resistor temperature drift must be taken into account also.
This effect can be
minimized by selecting good quality components, and by keeping all components in close thermal proximity. Careful circuit
board layout and component placement are necessary to minimize common thermal error sources. Also, the user should take
This increase is for still air, of course, and will be reduced at
high airflow levels. However, the user should still be aware that
self-heating effects can directly affect the accuracy of the
TMP12. For setpoint 2, self-heating will add to the setpoint
temperature (that is, in the above example the TMP12 will
switch the setpoint 2 output off 1.9 degrees early). Self-heating
will not affect the temperature at which setpoint 1 turns on, but
will add to the hysteresis. Several circuits for adding external
driver transistors and other buffers are presented in following
sections of this data sheet. These buffers will reduce self-heating
and improve accuracy.
Buffering the Voltage Reference
The reference output VREF is used to generate the temperature
setpoint programming voltages for the TMP12. Since the hysteresis is set by the reference current, external circuits which draw
current from the reference will increase the hysteresis value.
care to keep the bottom of the setpoint programming divider
ladder as close to GND (Pin 4) as possible to minimize errors
P
DISS
= 0.6 V
.
0.020 A = 12 mW
1
NC
VREF
2
SET
HIGH
3
SET
LOW
4
GND
UNDER
HEATER
TMP12
+5V
OVER
NC
+5V
5k
200k
300k
+5V
~1.5V
130k
8
V+
7
6
5
1uF
OP193
Figure 22. An Analog Measurement Circuit for VPTAT
10k
VPTAT
0.1UF
REV. 0–10–
TMP12
The on-board VREF output buffer is typically capable of 500 µA
output drive into as much as 50 pF load (max). Exceeding this
load will affect the accuracy of the reference voltage, could cause
thermal sensing errors due to excess heat build-up, and may induce
oscillations. External buffering of VREF with a low-drift voltage
follower will ensure optimal reference accuracy. Amplifiers which
offer low drift, low power consumption, and low cost appropriate
to this application include the OP284, and members of the OP113
and OP193 families.
With excellent drift and noise characteristics, VREF offers a good
voltage reference for data acquisition and transducer excitation applications as well. Output drift is typically better than 210 ppm/°C,
with 315 nV/Hz (typ) noise spectral density at 1 kHz.
Preserving Accuracy Over Wide Temperature Range Operation
The TMP12 is unique in offering both a wide-range temperature
sensor and the associated detection circuitry needed to implement
a complete thermostatic control function in one monolithic device.
The voltage reference, setpoint comparators, and output buffer
amplifiers have been carefully compensated to maintain accuracy
over the specified temperature ranges in this application. Since the
TMP12 is both sensor and control circuit, in many applications the
external components used to program and interface the device are
subjected to the same temperature extremes. Thus, it is necessary
to place components in close thermal proximity minimizing large
temperate differentials, and to account for thermal drift errors
where appropriate, such as resistor matching temperature coefficients, amplifier error drift, and the like. Circuit design with the
TMP12 requires a slightly different perspective regarding the thermal behavior of electronic components.
PC Board Layout Considerations
The TMP12 also requires a different perspective on PC board layout. In many applications, wide traces and generous ground planes
are used to extract heat from components. The TMP12 is slightly
different, in that ideal path for heat is via the cooling system air
flow. Thus, heat paths through the PC traces should be minimized.
This constraint implies that minimum pad sizes and trace widths
should be specified in order to reduce heat conduction. At the
same time, analog performance should not be compromised. In
particular, the bottom of the setpoint resistor ladder should be
located as close to GND as possible, as discussed in the Understanding Error Sources section of this data sheet.
Thermal Response Time
The time required for a temperature sensor to settle to a
specified accuracy is a function of the thermal mass of the
sensor, and the thermal conductivity between the sensor and
the object being sensed. Thermal mass is often considered
equivalent to capacitance. Thermal conductivity is commonly
specified using the symbol Q, and is the inverse of thermal
resistance. It is commonly specified in units of degrees per
watt of power transferred across the thermal joint. Figures 3
and 5 illustrate the typical RC time constant response to a
step change in ambient temperature. Thus, the time required
for the TMP12 to settle to the desired accuracy is dependent
on the package selected, the thermal contact established in
the particular application, and the equivalent thermal conductivity of the heat source. For most applications, the
settling-time is probably best determined empirically.
Switching Loads with the Open-Collector Outputs
In many temperature sensing and control applications some
type of switching is required. Whether it be to turn on a
heater when the temperature goes below a minimum value
or to turn off a motor that is overheating, the open-collector
outputs can be used. For the majority of applications, the
switches used need to handle large currents on the order of
1 Amp and above. Because the TMP12 is accurately measuring temperature, the open-collector outputs should
handle less than 20 mA of current to minimize self-heating.
Clearly, the trip point outputs should not drive the equipment directly. Instead, an external switching device is
required to handle the large currents. Some examples of
these are relays, power MOSFETs, thyristors, IGBTs, and
Darlington transistors.
This section shows a variety of circuits where the TMP12
controls a switch. The main consideration in these circuits,
such as the relay in Figure 23, is the current required to activate the switch.
+12V
TEMPERATURE
VREF
1
R1
2
R2
3
R3
4
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
TMP12
VPTAT
100
8
IN4001
OR EQUIV
7
6
140
5
Ω
2604-12-311
NC
+12 V
COTO
MOTOR
SHUTDOWN
Figure 23. Reed Relay Drive
It is important to check the particular relay you choose to
ensure that the current needed to activate the coil does not
exceed the TMP12’s recommended output current of
20 mA. This is easily determined by dividing the relay coil
voltage by the specified coil resistance. Keep in mind that
the inductance of the relay will create large voltage spikes
that can damage the TMP12 output unless protected by a
commutation diode across the coil, as shown. The relay
shown has contact rating of 10 Watts maximum. If a relay
capable of handling more power is desired, the larger contacts will probably require a commensurably larger coil,
with lower coil resistance and thus higher trigger current.
As the contact power handling capability increases, so does
the current needed for the coil, In some cases an external
driving transistor should be used to remove the current load
on the TMP12 as explained in the next section.
REV. 0 –11–
TMP12
IRGBC40S
NC
2N1711
V+
4.7kΩ
4.7kΩ
MOTOR
CONTROL
+5V
TEMPERATURE
SENSOR &
VOLTAGE
REFERENCE
HYSTERESIS
GENERATOR
WINDOW
COMPARATOR
VPTAT
VREF
100
1
2
3
4
8
5
7
6
TMP12
NC = NO CONNECT
Power FETs are popular for handling a variety of high current
dc loads. Figure 24 shows the TMP12 driving a P-channel
MOSFET transistor for a simple heater circuit. When the output transistor turns on, the gate of the MOSFET is pulled down
to approximately 0.6 V, turning it on. For most MOSFETs a
gate-to-source voltage or Vgs on the order of -2 V to -5 V is sufficient to turn the device on. Figure 25 shows a similar circuit
for turning on an N-channel MOSFET, except that now the
gate to source voltage is positive. For this reason an external
transistor must be used as an inverter so that the MOSFET will
turn on when the trip point pulls down.
1
2
3
4
VREF
TEMPERATURE
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
VPTAT
100
V+
8
7
6
5
NC
+5V
2.4kΩ (12V)
1.2kΩ (6V)
5%
TMP12
NC = NO CONNECT
Figure 24. Driving a P-Channel MOSFET
VREF
1
2
3
4
TEMPERATURE
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
TMP12
VPTAT
100
V+
8
4.7kΩ 4.7kΩ
NC
7
6
5
+5V
2N1711
IRFR9024
OR EQUIV
HEATING
ELEMENT
HEATING
ELEMENT
IRF130
Figure 26. Driving an IGBT
The last class of high power devices discussed here are Thyristors,
which include SCRs and Triacs. Triacs are a useful alternative to
relays for switching ac line voltages. The 2N6073A shown in Figure 27 is rated to handle 4 A (rms). The opto-isolated MOC3021
Triac shown features excellent electrical isolation from the noisy ac
line and complete control over the high power Triac with only a
few additional components.
TEMPERATURE
1
2
3
4
VREF
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
VPTAT
100
V+ = 5V
8
300Ω
NC
7
1
2
6
5
MOC3011
34
+5V
LOAD
150Ω
6
5
2N6073A
AC
TMP12
NC = NO CONNECT
Figure 27. Controlling the 2N6073A Triac
Figure 25. Driving an N-Channel MOSFET
Isolated Gate Bipolar Transistors (IGBTs) combine many of the
benefits of power MOSFETs with bipolar transistors and are
used for a variety of high power applications. Because IGBTs
have a gate similar to MOSFETs, turning on and off the devices
is relatively simple as shown in Figure 26. The turn on voltage
for the IGBT shown (IRGB40S) is between 3.0 and 5.5 volts.
This part has a continuous collector current rating of 50 A and a
maximum collector to emitter voltage of 600 V, enabling it to
work in very demanding applications.
REV. 0–12–
High Current Switching
As mentioned earlier, internal dissipation due to large loads on
the TMP12 outputs will cause some temperature error due to
self-heating. External transistors buffer the load from the
TMP12 so that virtually no power is dissipated in the internal
transistors and minimal self-heating occurs. This section shows
several examples using external transistors. The simplest case
uses a single transistor on the output to invert the output signal
is shown in Figure 28. When the open-collector of the TMP12
turns “ON” and pulls the output down, the external transistor
Q1’s base will be pulled low, turning off the transistor. Another
transistor can be added to re-invert the signal as shown in Figure
29. Now, when the output of the TMP12 is pulled down, the
first transistor, Q1, turns off and its collector goes high, which
turns Q2 on, pulling its collector low. Thus, the output taken
from the collector of Q2 is identical to the output of the TMP12.
By picking a transistor that can accommodate large amounts of
current, many high power devices can be switched.
TEMPERATURE
VREF
1
2
3
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
VPTAT
V+
8
4.7kΩ
7
6
2N1711
Q1
I
C
TMP12
TEMPERATURE
VREF
1
2
3
4
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
TMP12
Figure 29. Second Transistor Maintains Polarity of TMP12
Output
An example of a higher power transistor is a standard
Darlington configuration as shown in Figure 30. The part chosen, TIP-110, can handle 2 A continuous which is more than
enough to control many high power relays. In fact the
Darlington itself can be used as the switch, similar to
MOSFETs and IGBTs.
VPTAT
100
4.7kΩ
V+
4.7kΩ
2N1711
Q1
8
7
6
5
Q2
I
C
2N1711
4
HYSTERESIS
GENERATOR
5
100
TMP12
Figure 28. An External Transistor Minimizes Self-Heating
TEMPERATURE
VREF
1
2
3
4
SENSOR &
VOLTAGE
REFERENCE
WINDOW
COMPARATOR
HYSTERESIS
GENERATOR
VPTAT
100
8
7
6
5
TMP12
Figure 30. Darlington Transistor Can Handle Large Currents
V+
4.7kΩ
+5V
4.7kΩ
2N1711
+12V
TIP-110
RELAY
MOTOR
SWITCH
I
C
REV. 0 –13–
TMP12
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Pin Epoxy DIP
0.160 (4.06)
0.115 (2.93)
0.2440 (6.20)
0.2284 (5.80)
0.0098 (0.25)
0.0040 (0.10)
0.0500 (1.27) BSC
0.210
(5.33)
MAX
8
1
0.430 (10.92)
0.022 (0.558)
0.014 (0.356)
8
1
0.1968 (5.00)
0.1890 (4.80)
0.348 (8.84)
0.0192 (0.49)
0.0138 (0.35)
5
0.280 (7.11)
0.240 (6.10)
4
0.070 (1.77)
0.045 (1.15)
0.015
(0.381) TYP
SEATING
0.100
PLANE
(2.54)
BSC
8-Pin SOIC
5
0.1574 (4.00)
0.1497 (3.80)
4
0.102 (2.59)
0.094 (2.39)
SEATING
PLANE
0.130
(3.30)
MIN
0.325 (8.25)
0.300 (7.62)
0.015 (0.381)
0.008 (0.204)
0°- 15°
0.0196 (0.50)
0.0099 (0.25)
0.0098 (0.25)
0.0075 (0.19)
× 45°
0.195 (4.95)
0.115 (2.93)
0.0500 (1.27)
0.0160 (0.41)
C2074-10-10/95PRINTED IN U.S.A.
0°8°
REV. 0–14–
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