Datasheet TK65224MTL, TK65221MTL, TK65218MTL Datasheet (TOKO)

ADVANCED INFORMATION

V

IN

GND

SW

LOI

V

OUT

CONTROL

CIRCUIT

OSCILLATOR

Vref UVLO

LBI

20P

ADVANCED INFORMATION

TK652xx

STEP-UP VOLTAGE CONVERTER WITH VOLTAGE MONITOR

FEATURES

■ Guaranteed 0.9 V Operation

■ Very Low Quiescent Current

■ Internal Bandgap Reference

■ High Efficiency MOS Switching

■ Low Output Ripple

■ Laser-Trimmed Output Voltage

■ Low Output Voltage Monitor

■ Low Battery Monitor

■ Undervoltage Lockout

■ Regulation by Pulse Burst Modulation (PBM)

DESCRIPTION

The TK652xx low power step-up DC-DC converter is designed for portable battery powered systems, capable of operating from a single battery cell down to 0.9 V. The TK652xx provides the power switch and the control circuit for a boost converter. The converter takes a DC input and boosts it up to a regulated 1.8, 2.1, or 2.4 V.

The output voltage is laser-trimmed. Two internal detectors monitor the output voltage and battery voltage. The Low Output Indicator (LOI) generates an active low when the battery voltage falls below 90 % of the output voltage. The Low Battery Indicator (LBI) generates an active low when the battery voltage falls below 1.1 V. These outputs can be used to notify the microprocessor or power monitor circuit of a fault condition. An internal Undervoltage Lockout (UVLO) circuit is utilized to prevent the inductor switch from remaining in the “on” mode when the battery voltage is too low to permit normal operation. Pulse Burst Modulation (PBM) is used to regulate the voltage at the V

pin of the IC. PBM is the process in which an

OUT

oscillator signal is gated or not gated to the switch drive each period. The decision is made just before the start of each cycle and is based on comparing the output voltage to an internally-generated bandgap reference. The decision

APPLICATIONS

■ Battery Powered Systems

■ Cellular Telephones

■ Pagers

■ Personal Communications Equipment

■ Portable Instrumentation

■ Portable Consumer Equipment

■ Radio Control Systems

is latched, so the duty ratio is not modulated within a cycle. The average duty ratio is effectively modulated by the “bursting” and skipping of pulses which can be seen at the SW pin of the IC. Special care has been taken to achieve high reliability through the use of Oxide, double Nitride passivation. The TK652xx is available in a miniature SOT-23L-6 surface mount package.

Customized levels of accuracy in oscillator frequency and output voltage are available.

TK652xx

V

IN

LBI

SW

BLOCK DIAGRAM

LOI

GND

V

OUT

ORDERING INFORMATION

Voltage Code

VOLTAGE CODE

18 = 1.8 V 21 = 2.1 V 24 = 2.4 V

January 1999 TOKO, Inc. Page 1

TK652xxM

Tape/Reel Code

TAPE/REEL CODE

TL: Tape Left

TK652xx

ADVANCED INFORMATION

ABSOLUTE MAXIMUM RATINGS

All Pins Except SW and GND .................................... 6 V

SW Pin ....................................................................... 9 V

Power Dissipation (Note 1) ................................ 400 mW

Storage Temperature Range ................... -55 to +150 °C

Operating Temperature Range ...................-20 to +80 °C

Junction Temperature ...........................................150 °C

TK652xx ELECTRICAL CHARACTERISTICS

Over operating temperature range and supply voltage range, unless otherwise specified.

LOBMYSRETEMARAPSNOITIDNOCTSETNIMPYTXAMSTINU

V

NI

f

CSO

V

∆V ∆V

D V

Note 1: Derate at 0.8 mW/oC for operation above TA = 25 oC ambient temperature, when heat conducting copper foil path is maximized on the printed

)GER(TUO

)DAOL(TUO

)ENIL(TUO

CSO

IBL

circuit board. When this is not possible, a derating factor of 1.6 mW/ °C must be used.

egatloVylppuS09.006.1V

ycneuqerFrotallicsOlanretnIT

VfodlohserhTnoitalugeR

TUO

VfonoitalugeRdaoL

VfonoitalugeReniL

)GER(TUO

A

T

A

V

NI

∆V

rotallicsOfooitaRytuDemit-nOT

A

dlohserhTrotacidnIyrettaBwoLT

A

C°52=273869zHk C°52=%3-V

,V3.1=I

TUO

NI

I,V07.0=

TUO

Am4ot0=005Vm

Am1=

02-002Vm

TUO

%3+V

C°52=448445%

I,C°52=

TUO

Am1=50.101.151.1V

GENERAL CIRCUIT

V

LOI

300 kΩ

OUTSW

I

I(V

)

OUT

V

OUT

C

V

IN

I(VIN)

I

B

V

IN

L

LBI GND

D

Page 2 January 1999 TOKO, Inc.

ADVANCED INFORMATION

FINAL TEST CIRCUIT

TK652xx

300 kΩ

LOI

300 kΩ

GND

V

OUTSW

R

OF

15

++

C

U

10 µF

C

10 µF

I

OUT

V

OUT

D

C

N

10 µF

I

B

V

IN

L = 95 µF

Note: Inductor L: Toko A682AE-014 or equivalent Diode D: LL101 Capacitors CN:CD:CU: Panasonic TE series, ECS-TOJY106R

V

IN

LBI

C

S

220 pF

R

1 K

D

S

Above is the Final Test Circuit through which each of the production parts must pass. In this test circuit, the part is tested against the specification limits in the data sheet (the min. and max. values in the Electrical Characteristics) at room temperature, and is rejected if the tested values are outside the minimum (min.) and maximum (max.) values.

The Bench Test Circuits shown on the following pages are the circuits used most of the time to measure the typical (typ.) values in the Electrical Characteristics section, and make the Typical Performance graphs.

Note: In measuring the oscillator frequency and the Max I skipping” mode was achieved.

on the bench, the converter was loaded until “no pulse

OUT

January 1999 TOKO, Inc. Page 3

TK652xx

ADVANCED INFORMATION

TK65218 ELECTRICAL CHARACTERISTICS

Over operating temperature range and supply voltage range, unless otherwise specified.

LOBMYSRETEMARAPSNOITIDNOCTSETNIMPYTXAMSTINU

I

)Q(B

V(INI)VotnitnerruCtnecseiuQ

V(I

)VotnitnerruCtnecseiuQ

TUO

∆f

/∆T

CSO

V ∆V

R

TUO

)GER(TUO

/∆T

)IOL(TUO

)NO(WS

FFE)3,2setoN(ycneiciffEretrevnoC

V

VLU

V

NI

)3etoN(tnerruCyrettaBdaoLoN

T

A

V

niP

NI

niPVNI,V3.1=I

TUO

VfodlohserhTnoitalugeR

TUO

VfoytilibatSerutarepmeT

V

TUO

V

)IOL(TUO

noitisnarTIOLgniruDV

siseretsyHdlohserhTT

niPWSfoecnatsiser-nOV

egatloVtuokcoLegatlovrednUT

NI

T

A

rotallicsOfoytilibatSerutarepmeTV

NI

T

A

V

)GER(TUO

NI

A

TUO

V

NI

A

V

NI

V

NI

,V3.1=I

TUO

C°52=

,V3.1=I

TUO

C°52=

TUO

,V3.1=

,Am0=

,Am4ot0=

0406Aµ

1102Aµ

Am0=2102Aµ

gnippikSesluPoN

1.0C°/%

C°52=647.1008.1548.1V

,V3.1=I

TUO

T,V3.1=

A

Am1=001C°/mpp

C°52=55.126.186.1V

C°52=52Vm

V=

,T

)GER(TUO

A

I,V3.1=

TUO

C°52=0.1mho

,Am6=

lioC37D,Hµ001=L

67%

)4etoN(,C°52=74.097.0V

T,V9.0=

A

T,V1.1=

A

,C°52=

lioCFD3,Hµ59=L

,C°52=

lioCFD3,Hµ59=L

50.7Am

60.31Am

ImumixaM

I

)XAM(TUO

TUO

)3,1setoN(

Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, ECS-TOJY106R

V

R

N

1 K

I

B

IN

I(VIN)

C

N

10 µF

L = 95 µH

C

B

10 µF

retrevnoCrof

V

NI

T,V3.1=

A

lioCFD3,Hµ59=L

V

NI

T,V9.0=

A

lioC37D,Hµ93=L

V

NI

T,V1.1=

A

lioC37D,Hµ93=L

BENCH TEST CIRCUIT

300 kΩ

V

IN

LBI GND

C

S

220 pF

R

S

1 K

LOI

300 kΩ

V

OUTIND

I(V

)

OUT

D

C

10 µF

O

I

OUT

,C°52=

70.51Am

0.71Am

0.82Am

Note 1: Maximum load current depends on inductor value and input voltages. Note 2: Output ripple depends on filter capacitor values, ESRs and the inductor value. Note 3: When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Note 4: Regulation not guaranteed.

V

OUT

Page 4 January 1999 TOKO, Inc.

TK65218

V

(V)

V

(V)

V

(V)

ADVANCED INFORMATION

TYPICAL PERFORMANCE CHARACTERISTICS

TK652xx

OSCILLATOR FREQUENCY

TEMPERATURE

91

87

(kHz)

83

OSC

f

79

75

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE

LOAD CURRENT

1.9

L = 95 µH TOKO P/N: A682AE-014 (3DF SERIES)

TA = 25 °C

1.8

V

1.7

IN

= 0.9 V

1.3 V

OUT

1.6

1.1 V

VS.

1.6 V

OUTPUT REGULATION VOLTAGE

TEMPERATURE

1.800

1.795

(V)

1.790

OUT(REG)

V

1.785

1.780

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE

LOAD CURRENT

1.9

L = 100 µH TOKO P/N: 636CY-101M (D73F SERIES)

TA = 25 °C

1.8

V

1.7

IN

= 0.9 V

1.3 V

OUT

1.6

1.1 V

VS.

1.6 V

VS.

BATTERY CURRENT

INPUT VOLTAGE

200

TA = 25 °C

160

NO LOAD

120

(µA)

B

I

80

40

0

0.1 0.5 1.0 1.5 VIN (V)

OUTPUT VOLTAGE

LOAD CURRENT

1.9

L = 39 µH TOKO P/N: 636CY-390M (D73 SERIES)

TA = 25 °C

1.8 V

= 0.9 V

IN

1.3 V

1.7

OUT

1.1 V

1.6

VS.

1.6 V

1.5 1 10 100

I

(mA)

OUT

EFFICIENCY VS. LOAD CURRENT

80

L = 95 µF TOKO P/N: A682AE-014 (3DF SERIES) SMALL COIL

75

TA = 25 °C

70

65

EFF (%)

1.1 V

60

VIN = 0.9 V

55

50

0.1 1 10 100

I

(mA)

OUT

1.6 V

1.3 V

1.5 1 10 100

I

(mA)

OUT

EFFICIENCY

85

TA = 25 °C

VS.

LOAD CURRENT

80

75

70

VIN = 0.9 V

EFF (%)

65

60

55

1.1 V

L = 100 µF TOKO P/N: A636CY-101M (D73 SERIES) LARGER COIL

0.1 1 10 100 I

(mA)

OUT

1.6 V

1.3 V

1.5 1 10 100

I

(mA)

OUT

MAXIMUM OUTPUT CURRENT

INDUCTOR VALUE

100

NO PULSE SKIPPING MODE TA = 25 °C

(mA)

80

60

1.3

40

OUT(MAX)

I

20

0

1.1 V

VIN = 0.9 V

0 40 80 120 160

INDUCTOR VALUE (µH)

VS.

1.6 V

January 1999 TOKO, Inc. Page 5

TK652xx

ADVANCED INFORMATION

TK65221 ELECTRICAL CHARACTERISTICS

Over operating temperature range and supply voltage range, unless otherwise specified.

LOBMYSRETEMARAPSNOITIDNOCTSETNIMPYTXAMSTINU

I

)Q(B

V(INI)VotnitnerruCtnecseiuQ

V(I

)VotnitnerruCtnecseiuQ

TUO

∆f

/∆T

CSO

V ∆V

R

TUO

)GER(TUO

/∆T

)IOL(TUO

)NO(WS

FFE)3,2setoN(ycneiciffEretrevnoC

V

VLU

V

NI

)3etoN(tnerruCyrettaBdaoLoN

T

A

V

niP

NI

niPVNI,V3.1=I

TUO

VfodlohserhTnoitalugeR

TUO

VfoytilibatSerutarepmeT

V

TUO

V

)IOL(TUO

noitisnarTIOLgniruDV

siseretsyHdlohserhTT

niPWSfoecnatsiser-nOV

egatloVtuokcoLegatlovrednUT

NI

T

A

rotallicsOfoytilibatSerutarepmeTV

NI

T

A

V

)GER(TUO

NI

A

TUO

V

NI

A

V

NI

V

NI

,V3.1=I

TUO

C°52=

,V3.1=I

TUO

C°52=

TUO

,V3.1=

,Am0=

,Am4ot0=

5456Aµ

5.2102Aµ

Am0=5.4132Aµ

gnippikSesluPoN

1.0C°/%

C°52=830.2001.2361.2V

,V3.1=I

TUO

T,V3.1=

A

Am1=001C°/mpp

C°52=38.198.159.1V

C°52=83Vm

V=

,T

)GER(TUO

A

,V3.1=I

TUO

C°52=0.1mho

,Am6=

lioC37D,Hµ001=L

87%

)4etoN(,C°52=84.097.0V

T,V9.0=

A

T,V1.1=

A

,C°52=

lioCFD3,Hµ59=L

,C°52=

lioCFD3,Hµ59=L

40.6Am

50.11Am

ImumixaM

I

)XAM(TUO

TUO

)3,1setoN(

Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, ECS-TOJY106R

I

B

V

IN

R

N

1 K

C

10 µF

L = 95 µH

C

10 µF

retrevnoCrof

V

NI

V

NI

V

NI

T,V3.1=

A

T,V9.0=

A

T,V1.1=

A

BENCH TEST CIRCUIT

300 kΩ

LOI

300 kΩ

V

OUTIND

I(V

OUT

B

N

I(VIN)

V

IN

LBI GND

C

220 pF

D

S

R

S

1 K

,C°52=

lioCFD3,Hµ59=L

,C°52=

lioC37D,Hµ93=L

,C°52=

lioC37D,Hµ93=L

60.31Am

0.51Am

0.42Am

Note 1: Maximum load current depends on inductor value and input voltages. Note 2: Output ripple depends on filter capacitor values, ESRs and the inductor value. Note 3: When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Note 4: Regulation not guaranteed.

I

C

10 µF

OUT

V

OUT

O

)

Page 6 January 1999 TOKO, Inc.

TK65221

V

(V)

V

(V)

V

(V)

ADVANCED INFORMATION

TYPICAL PERFORMANCE CHARACTERISTICS

TK652xx

OSCILLATOR FREQUENCY

TEMPERATURE

91

87

(kHz)

83

OSC

f

79

75

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE

2.2

L = 95 µH TOKO P/N: A682AE-014 (3DF SERIES)

LOAD CURRENT

TA = 25 °C

2.1

OUT

2.0

1.9

V

IN

= 0.9 V

1.3 V

1.6 V

1.1 V

VS.

OUTPUT REGULATION VOLTAGE

TEMPERATURE

2.100

2.095

(V)

2.090

OUT(REG)

V

2.085

2.080

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE

LOAD CURRENT

2.2

L = 100 µH TOKO P/N: 636CY-101M (D73F SERIES)

TA = 25 °C

2.1

2.0

V

IN

= 0.9 V

1.3 V

OUT

1.9

1.1 V

VS.

1.6 V

VS.

BATTERY CURRENT

INPUT VOLTAGE

200

TA = 25 °C

160

NO LOAD

120

(µA)

B

I

80

40

0

0.1 0.5 1.0 1.5 VIN (V)

OUTPUT VOLTAGE

LOAD CURRENT

2.2

L = 39 µH TOKO P/N: A636CY-390M (D73 SERIES)

TA = 25 °C

2.1

2.0

OUT

1.9

V

IN

= 0.9 V

1.3 V

1.1 V

VS.

1.6 V

1.8 1 10 100

I

(mA)

OUT

EFFICIENCY VS. LOAD CURRENT

80

L = 95 µF TOKO P/N: A682AE-014 (3DF SERIES) SMALL COIL

75

TA = 25 °C

70

65

1.1 V

1.3 V

EFF (%)

60

VIN = 0.9 V

55

50

0.1 1 10 100

I

(mA)

OUT

1.6 V

1.8 1 10 100

I

(mA)

OUT

EFFICIENCY VS. LOAD CURRENT

85

TA = 25 °C

80

75

70

EFF (%)

1.1 V

65

VIN = 0.9 V

60

55

L = 100 µF TOKO P/N: A636CY-101M (D73 SERIES) LARGER COIL

0.1 1 10 100 I

(mA)

OUT

1.6 V

1.3 V

1.8 1 10 100

I

(mA)

OUT

MAXIMUM OUTPUT CURRENT

INDUCTOR VALUE

100

NO PULSE SKIPPING

80

(mA)

60

40

OUT(MAX)

I

20

0

1.3

1.1 V

VIN = 0.9 V

MODE TA = 25 °C

0 40 80 120 160

INDUCTOR VALUE (µH)

1.6 V

January 1999 TOKO, Inc. Page 7

VS.

TK652xx

ADVANCED INFORMATION

TK65224 ELECTRICAL CHARACTERISTICS

Over operating temperature range and supply voltage range, unless otherwise specified.

LOBMYSRETEMARAPSNOITIDNOCTSETNIMPYTXAMSTINU

I

)Q(B

V(I

)VotnitnerruCtnecseiuQ

NI

V(I

)VotnitnerruCtnecseiuQ

TUO

∆f

/∆T

CSO

V ∆V

TUO

V ∆V

R

FFE)3,2setoN(ycneiciffEretrevnoC

V

VLU

/∆T

)IOL(TUO

)NO(WS

V

NI

)3etoN(tnerruCyrettaBdaoLoN

T

A

V

niP

NI

niPVNI,V3.1=I

TUO

)GER(TUO

VfodlohserhTnoitalugeR

TUO

VfoytilibatSerutarepmeT

V

TUO

V

)IOL(TUO

noitisnarTIOLgniruDV

siseretsyHdlohserhTT

niPWSfoecnatsiser-nOV

egatloVtuokcoLegatlovrednUT

NI

T

A

rotallicsOfoytilibatSerutarepmeTV

NI

T

A

V

)GER(TUO

NI

A

TUO

V

NI

A

V

NI

V

NI

,V3.1=I

TUO

C°52=

,V3.1=I

TUO

C°52=

TUO

,V3.1=

C°52=823.2004.2274.2V

,V3.1=I

TUO

T,V3.1=

A

C°52=05Vm

V=

,T

)GER(TUO

,V3.1=I

TUO

T,V9.0=

A

T,V1.1=

A

,Am0=

,Am4ot0=

5658Aµ

0253Aµ

Am0=2204Aµ

gnippikSesluPoN

1.0C°/%

Am1=001C°/mpp

C°52=70.261.252.2V

A

C°52=0.1mho

,Am6=

lioC37D,Hµ001=L

08%

)4etoN(,C°52=94.097.0V

,C°52=

lioCFD3,Hµ59=L

,C°52=

lioCFD3,Hµ59=L

57.30.5Am

5.40.9Am

ImumixaM

I

)XAM(TUO

TUO

)3,1setoN(

Inductor L: Toko A682AE-014 or equivalent Diode D: LL103A or equivalent Capacitors CN:CO:CB: Panasonic TE series, ECS-TOJY106R

I

B

V

IN

retrevnoCrof

V

NI

V

NI

V

NI

BENCH TEST CIRCUIT

R

N

1 K

C

10 µF

L = 95 µH

C

10 µF

I(VIN)

N

B

V

IN

LBI GND

C

220 pF

D

S

R

S

1 K

LOI

V

OUTIND

T,V3.1=

T,V9.0=

T,V1.1=

300 kΩ

I(V

A

,C°52=

lioCFD3,Hµ59=L

,C°52=

lioC37D,Hµ93=L

,C°52=

lioC37D,Hµ93=L

5.50.21Am

0.31Am

0.12Am

Note 1: Maximum load current depends on inductor value and input voltages. Note 2: Output ripple depends on filter capacitor values, ESRs and the inductor value. Note 3: When using specified Toko inductor and Schottky diode with VF = 0.45 V @ 100 mA. Note 4: Regulation not guaranteed.

I

C

10 µF

OUT

V

OUT

O

OUT

)

Page 8 January 1999 TOKO, Inc.

TK65224

V

(V)

ADVANCED INFORMATION

TYPICAL PERFORMANCE CHARACTERISTICS

TK652xx

OSCILLATOR FREQUENCY

TEMPERATURE

91

87

(kHz)

83

OSC

f

79

75

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE VS.

2.5

L = 95 µH TOKO P/N: A682AE-014 (3DF SERIES)

LOAD CURRENT

TA = 25 °C

2.4

V

(V)

OUT

V

2.3

2.2

IN

= 0.9 V

1.3 V

1.1 V

1.6 V

VS.

OUTPUT REGULATION VOLTAGE

TEMPERATURE

2.400

2.395

(V)

2.390

OUT(REG)

V

2.385

2.380

-50 0 50 100 TEMPERATURE (°C)

OUTPUT VOLTAGE VS.

LOAD CURRENT

2.5

L = 100 µH TOKO P/N: 636CY-101M (D73 SERIES)

TA = 25 °C

2.4

V

(V)

OUT

V

2.3

2.2

IN

= 0.9 V

1.3 V

1.1 V

1.6 V

VS.

BATTERY CURRENT

INPUT VOLTAGE

200

TA = 25 °C

160

NO LOAD

120

(µA)

B

I

80

40

0

0.1 0.5 1.0 1.5 VIN (V)

OUTPUT VOLTAGE

LOAD CURRENT

2.5

L = 39 µH TOKO P/N: 636CY-390M (D73 SERIES)

TA = 25 °C

2.4

V

2.3

OUT

2.2

IN

= 0.9 V

1.3 V

1.1 V

VS.

1.6 V

2.1 1 10 100

I

(mA)

OUT

EFFICIENCY VS. LOAD CURRENT

80

L = 95 µF TOKO P/N: A682AE-014 (3DF SERIES) SMALL COIL

75

1.3 V

70

65

VIN = 0.9 V

TA = 25 °C

EFF (%)

60

1.1 V

55

50

0.1 1 10 100

I

(mA)

OUT

1.6 V

2.1 1 10 100

I

(mA)

OUT

EFFICIENCY VS. LOAD CURRENT

80

TA = 25 °C

80

75

VIN = 0.9 V

70

EFF (%)

65

60

55

L = 100 µF TOKO P/N: A636CY-101M (D73 SERIES) LARGER COIL

0.1 1 10 100 I

(mA)

OUT

1.3 V

1.6 V

1.1 V

2.1 1 10 100

I

(mA)

OUT

MAXIMUM OUTPUT CURRENT

INDUCTOR VALUE

100

NO PULSE SKIPPING MODE TA = 25 °C

(mA)

80

60

40

20

0

1.3

1.1 V

VIN = 0.9 V

OUT(MAX)

I

0 40 80 120 160

INDUCTOR VALUE (µH)

VS.

1.6 V

January 1999 TOKO, Inc. Page 9

TK652xx

ADVANCED INFORMATION

SINGLE-CELL APPLICA TION

The TK652xx is a boost converter control IC with the power MOSFET switch built into the device. It operates from a single battery cell and steps up the output voltage to a regulated 1.8, 2.1 and 2.4 V. The device operates at a fixed nominal clock frequency of 83 kHz.

In its simplest form, a boost power converter using the TK652xx requires only three external components: an inductor, a diode, and a capacitor.

The analysis is easier to follow when referencing the simple boost circuit below.

V

IN

LBI GND

LOI

V

OUTSW

V

+

OUT

FIGURE 1: SIMPLE BOOST CONVERTER

filtering component values (consult the “Ripple and Noise Considerations” section) can be determined if needed or desired.

The TK652xx runs with a fixed oscillator frequency and it regulates by applying or skipping pulses to the internal power switch. This regulation method is called Pulse Burst Modulation (PBM).

ANALYSIS OF SWITCHING CYCLE

I

PEAK

di/dt = VIN / L

t (on)

di/dt = - (V

t (off)

+ Vf - VIN)/ L

OUT

t (deadtime)

THEORY OF OPERATION

The converter operates with one terminal of an inductor connected to the DC input and the other terminal connected to the switch pin of the IC. When the switch is turned on, the inductor current ramps up. When the switch is turned off (or “lets go” of the inductor), the voltage flies up as the inductor seeks out a path for its current. A diode, also connected to the switching node, provides a path of conduction for the inductor current to the boost converter’s output capacitor. The TK652xx monitors the voltage of the output capacitor and has a 1.8, 2.1 and 2.4 V threshold at which the converter switching becomes deactivated. So the output capacitor charges up to 1.8, 2.1 and 2.4 V and regulates there, provided that no more current is drawn from the output than the inductor can provide. The primary task,

then, in designing a boost converter with the TK652xx is to determine the inductor value (and its peak current rating to prevent inductor core saturation problems) which will provide the amount of current needed to guarantee that the output voltage will be able to maintain regulation up to a specified maximum load current. Secondary necessary tasks also include choosing

the diode and the output capacitor. Then the snubber and

Above is the input or inductor current waveform over a switching cycle.

From an oscillator standpoint, the switching cycle consists of only an on-time and an off-time. But from an inductor current standpoint, the switching cycle breaks down into three important sections: on-time, off-time, and deadtime. The on-time of the switch and the inductor current are synonymous. During the on-time, the inductor current increases. During the off-time, the inductor current decreases as it flows into the output. When the inductor current reaches zero, that marks the end of the inductor current off-time. For the rest of the cycle, the inductor current remains at zero. Since no energy is being either stored or delivered, that remaining time is called “deadtime.” This mode of the inductor current decaying to zero every cycle is called “discontinuous mode.” In summary, energy is stored in the inductor during on-time, delivered to the output during off-time, and remains at zero during deadtime. The output current of the boost converter comes from the

Page 10 January 1999 TOKO, Inc.

ADVANCED INFORMATION

SINGLE-CELL APPLICATION (CONT.)

TK652xx

second half of the input current triangle waveform (averaged over the period or multiplied by the frequency) given by the equation:

I

= [IPK x t(off)] x f / 2

OUT

and:

IPK = (VIN / L) x t(on) = V

IN

D / f L

and:

t(off) = IPK / [(V = (V = V

D / f L) / [(V

IN

D / f (V

IN

+ VF - VIN) / L]

OUT

+ VF - VIN)

OUT

+ VF - VIN) / L]

therefore:

I

= (VIN)2 (D)2 / 2 f L (V

OUT

+ VF - VIN)

OUT

which derives Equation 1 of the next section.

INDUCTOR SELECTION

It is under the condition of lowest input voltage that the boost converter output current capability is the lowest for a given inductance value. Three other significant parameters with worst-case values for calculating the inductor value are: highest switching frequency, lowest duty ratio (of the switch on-time to the total switching period), and highest diode forward voltage. Other parameters which can affect the required inductor value, but for simplicity will not be considered in this first analysis are: the series resistance of the DC input source (i.e., the battery), the series resistance of the internal switch, the series resistance of the inductor itself, ESR of the output capacitor, input and output filter losses, and snubber power loss.

The converter reaches maximum output current capability when the switch runs at the oscillator frequency, without pulses being skipped. The output current of the boost converter is then given by the equation:

2

+ VF - VIN)

(1)

I

OUT

=

2

f

L (V

(VIN)2 (D)

OUT

where “VIN” is the input voltage, “D” is the on-time duty ratio of the switch, “f ” is the switching (oscillator) frequency, “L” is the inductor value, “V

” is the output voltage, and “VF”

OUT

is the diode forward voltage. It is important to note that Equation 1 makes the assumption stated in Equation 2:

VIN ≤ (V

+ VF)(1 - D)

OUT

(2) The implication from Equation ) is that the inductor will

operate in discontinuous mode. Now, plugging in worst case conditions, the inductor value

can be determined by simply transforming the above equation in terms of “L”:

L

(MIN)

=

2

f

V

(MAX) IOUT(MAX)

IN(MIN)

[V

OUT(MIN)

2

D

(MIN)

+ V

2

F(MAX)

- V

IN(MIN)

] (3) where “V

” is best approximated by the diode forward

F(MAX)

voltage at about two-thirds of the peak diode current value. The peak diode current is the same as the peak input current, the peak switch current, and the peak inductor current. The formula is:

VIN D

IPK =

f

L

(4) Some reiteration is implied because “L” is a function of “VF”

which is a function of “IPK” which, in turn, is a function of “L”. The best way into this loop is to first approximate “VF”, determine “L”, determine “IPK”, and then determine a new “VF”. Then, if necessary, reiterate.

When selecting the actual inductor, it is necessary to make sure the peak current rating of the inductor (i.e., the current which causes the core to saturate) is greater than the maximum peak current the inductor will encounter. To determine the maximum peak current, use Equation 4 again, using the maximum values for “VIN” and “D”, and minimum values for “f ” and “L”.

It may also be necessary when selecting the inductor to check the rms current rating of the inductor. Whereas peak

January 1999 TOKO, Inc. Page 11

TK652xx

ADVANCED INFORMATION

SINGLE-CELL APPLICATION (CONT.)

current rating is determined by core saturation, rms current rating is determined by wire size and power dissipation in the wire resistance. The inductor rms current is given by:

IPK f L

I

= I

L(RMS)

(5) where “IPK” is the same maximized value that was just used

to check against inductor peak current rating, and the term in the numerator within the radical that is added to the [on-time] duty ratio, “D”, is the off-time duty ratio.

PK

D +

V

OUT

+ VF - V

3

IN

Toko America, Inc. can offer a miniature matched magnetic solution

sizes to accommodate varying power level requirements. The following series of Toko inductors work especially well with the TK652xx : 10RF, 12RF, 3DF, D73, and D75. The 5CA series can be used for isolated-output applications, although such design objectives are not considered here.

OTHER CONVERTER COMPONENTS

In choosing a diode, parameters worthy of consideration are: forward voltage, reverse leakage, and capacitance. The biggest efficiency loss in the converter is due to the diode forward voltage. A Schottky diode is typically chosen to minimize this loss. Possible choices for Schottky diodes are: LL103A from ITT MELF case; 1N5017 from Motorola (through hole case); MBR0530 from Motorola (surface mount) or 15QS02L from Nihon EC (surface mount).

Reverse leakage current is generally higher in Schottkys than in pin-junction diodes. If the converter spends a good deal of the battery lifetime operating at very light load (i.e., the system under power is frequently in a standby mode), then the reverse leakage current could become a substantial fraction of the entire average load current, thus degrading battery life. So don’t dramatically oversize the Schottky diode if this is the case.

Diode capacitance isn’t likely to make much of an undesirable contribution to switching loss at this relatively low switching frequency. It can, however, increase the snubber (look in the “Ripple and Noise Considerations” section) dissipation requirement.

The output capacitor, the capacitor connected from the diode cathode to ground, has the function of averaging the

in a wide range of inductor values and

current pulses delivered from the inductor while holding a relatively smooth voltage for the converter load. Typically, the ripple voltage cannot be made smooth enough by this capacitor alone, so an output filter is used. In any case, to minimize the dissipation required by the output filter, the output capacitor should still be chosen with consideration to smoothing the voltage ripple. This implies that its Equivalent Series Resistance (ESR) should be low. This usually means choosing a larger size than the smallest available for a given capacitance. To determine the peak ripple voltage on the output capacitor for a single switching cycle, multiply the ESR by the peak current which was calculated in Equation 4. ESR can be a strong function of temperature, being worse case when cold. The capacitance should be capable of integrating a current pulse with little ripple. Typically, if a capacitor is chosen with reasonably low ESR, and if the capacitor is the right type of capacitor for the application (typically aluminum electrolytic or tantalum), then the capacitance will be sufficient.

ESR and printed circuit board layout have strong influence on RF interference levels. Special care must be taken to optimize PCB layout and component placement.

THE BENEFITS OF INPUT FILTERING

In practice, it may be that the peak current (calculated in Equation 4) flowing out of the battery and into the converter will cause a substantial input ripple voltage dropped across the resistance inside the battery. This becomes a more likely case for cold temperature (when battery series resistance is higher), higher load rating converters (whose inductors must draw higher peak currents), and when the battery is undersized for the peak current application.

While the simple analysis used a parameter “VIN” to represent the converter input voltage in the equations, one may not know what “VIN” value to use if it is delivered by a battery that allows high ripple to occur. For example, assume that the converter draws a peak current of 100 mA for a 1 V input, and assume that the input is powered by a partially discharged AAA battery which might have a series resistance of 2 Ohms at 0 °C. (Environmentally clean, so called “green” batteries tend to have higher source resistance than their “unclean” predecessors). If such partially discharged battery voltage is 1 V without load, the converter battery voltage will sag to about 0.8 V during the on-time. This can cause two problems: 1) with the effective input voltage to the converter reduced in this way, the converter output current capability will decrease, 2) if the

Page 12 January 1999 TOKO, Inc.

ADVANCED INFORMATION

SINGLE-CELL APPLICATION (CONT.)

TK652xx

same battery is powering the TK652xx at the VIN pin (i.e., the normal case), then the IC may become inoperable due to insufficient VIN. This is why the application test circuit features an RC filter into the VIN pin. The current draw is very small, so the voltage drop across this filter resistor is negligible. The filter serves to average out the input ripple caused by the battery resistance. Note that this filter is optional, and the net effect of its use is the extension of battery life by allowing the battery to be discharged more deeply.

A more power-efficient method comes at the price of a large capacitor. This can be placed in parallel with the battery to help channel the converter current pulses away from the battery. The capacitor must have low ESR compared to the battery resistance in order to accomplish this effectively.

Still another solution is to filter the DC input with an LC filter. However, it is more likely that the filter will be either too large or too lossy. It is of questionable benefit to smooth the input if the DC loss through the filter is large.

Assuming that input ripple voltage at the battery terminal and converter input is large, and that we filter the VIN pin of the IC as in the test circuit, then the parameter “VIN” in the previous equations is not usable, and we will need to use parameters to represent both the source voltage source resistance.

SWITCH ON-RESISTANCE, INDUCTOR WINDING RESISTANCE, AND CAPACITANCE ESR

The on-resistance of the TK652xx’s internal switch is about 1 Ohm maximum. Using the previously stated example of 100 mA peak current, the voltage drop across the switch would reach 100 mV during the on-time. This subtracts from the voltage which is impressed across the inductor to store energy during the on-time. As a result, less energy is delivered to the output during the off-time.

If the winding resistance of the inductor increases to 1 Ohm or greater, the voltage drop across the winding resistance also subtracts from the voltage used to store energy in the core. Thus, efficiency degradation occurs.

As the inductor delivers energy into the output capacitor during the off-time, its current decays at a rate proportional to the voltage drop across it. The idealized equations assume that the voltage at the switching node is clamped

and

the

at a diode drop above the output voltage. However, the ESR of the output capacitor can increase the voltage drop across the inductor by the additional voltage dropped across the ESR when the peak current flows in it. For example, the voltage across a capacitor with an ESR of 2 Ohms (not unusual at cold temperature) would jump by 200 mV when 100 mA peak current began to flow in it. This extra voltage drop would cause the inductor current to ramp down more quickly, thus, depleting the available output current. Possible choices for low ESR capacitors are: Panasonic TE series (surface mount); AVX TPS series (surface mount); Matsuo 267 series (surface mount); Sanyo OS-CON series.

LBI AND LOI FEATURES

In a properly designed system that has its maximum load current under the limit of the maximum output current of the converter, the Low Battery Indicator (LBI) signal is the first indicator that the battery is getting low. The latched Low

Output indicator (LOI) signal is the second and final indicator that the battery should be replaced.

The LBI output provides a warning signal to a system controller that the battery is near the end of its life and will need replacing soon. The LBI threshold voltage is between

1.05 and 1.15 V. There is no hysteresis in its on-off trigger levels.

The LOI output can provide a reset signal to a microprocessor or other external system controller. When the output voltage falls below the LOI threshold (during startup of the converter or under a current overload fault condition), the LOI signal is asserted low, indicating that the system controller )e.g., microprocessor) should be in a reset mode. This method of reset control can be used to prevent improper system operation which might occur at low supply voltage levels. The LOI threshold voltage is between 87% and 93% of the regulated output voltage value. The LOI threshold also has about 40 mV hysteresis between its on-off trigger levels.

When the output voltage dips below the LOI threshold and the LBI signal is asserted low, the LOI signal will be

latched low by the LBI signal. That is, the LOI signal will remain asserted low until the LBI signal goes high,

even if the output voltage regains its regulated value. Although the converter can start-up and work with the battery voltage even as low as 0.9 V, the battery should

be replaced when the LOI is latched low.

January 1999 TOKO, Inc. Page 13

TK652xx

ADVANCED INFORMATION

SINGLE-CELL APPLICATION (CONT.)

RIPPLE AND NOISE CONSIDERATIONS

The filtered test circuit of the TK652xx is shown below in Figure 2.

300 kΩ

V

R

N

1 K

C

N

10 µF

I

B

V

IN

L = 95 µF

IN

LBI GND

C

S

220 pF

R

1 K

FIGURE 2: FILTERED TEST CIRCUIT

Compared to the simple boost circuit, this Filtered Test Circuit adds the following circuitry: the RC filter into the V pin, the RC snubber, the RC filter at the converter output, and the pull-up resistor to the LOI pin.

The RC filter at the VIN pin is used only to prevent the ripple voltage at the battery terminals from prematurely causing undervoltage lockout of the IC. This is only needed when the inductor value is relatively small and the battery resistance is relatively high and the VIN range must extend as low as possible.

The snubber (optional) is composed of a series RC network from the switch pin to ground (or to the output or input if preferred). Its function is to dampen the resonant LC circuit which rings during the inductor current deadtime. When the current flowing in the inductor through the output diode decays to zero, the parasitic capacitance at the switch pin from the switch, the diode, and the inductor winding has energy which rings back into the inductor, flowing back into the battery. If there is no snubbing, it is feasible that the switch pin voltage could ring below ground. Although the IC is well protected against latch-up, this ringing may be undesirable due to radiated noise. To be effective, the snubber capacitor should be large (e.g., 5 - 20 times) in comparison to the parasitic capacitance. If it is unnecessarily large, it dissipates extra energy every time the converter switches. The resistor of the snubber should be chosen such that it drops a substantial voltage as the ringing parasitic capacitance attempts to pull the snubber capacitor along for the ride. If the resistor is too small (e.g., zero), the snubber capacitance just adds to the ringing energy. If the resistor is too large (e.g., infinite), it effectively disengages the snubber capacitor from fighting the ringing.

LOI

300 kΩ

V

OUTSW

I

C

10 µF

OUT

V

OUT

D

R

OF

15

++

D

S

C

10 µF

U

The RC filter at the converter output attenuates the conducted noise; the converter may not require this.

Finally, the pull-up resistors at the LOI pin are needed only if this output signals is used. Most of this circuitry which appears in the test circuit has been added to minimize ripple and noise effects. But when this is not critical, the circuit can be minimized.

When any DC-DC converter is used to convert power in RF circuits (e.g., pagers) the spectral noise generated by the converter, whether conducted or radiated, is of concern. The oscillator of the TK652xx has been trimmed and stabilized to 83 +/- 4 kHz with the intention of greatly minimizing interference at the common IF frequency of 455 kHz.

In comparison with conventional IC solutions, where the oscillator frequency is not controlled tightly, the TK652xx

IN

can achieve as much as 20 - 30 dB improvements in RF interference reduction by means of its accurately controlled oscillator frequency. This IF frequency is halfway between the fifth and sixth harmonics of the oscillator. The fifth harmonic of the maximum oscillator frequency and the sixth harmonic of the minimum oscillator frequency still leave a 39 kHz band centered around 455 kHz within which a fundamental harmonic of the oscillator will not fall. Since the TK652xx operates by pulse burst modulation (PBM), the switching pattern can be a subharmonic of the oscillator frequency. The simplest example, and the one to be avoided the most, is that of the converter causing every other oscillator pulse to be skipped. This means that the switching pattern would have a fundamental frequency of one-half the oscillator frequency, or 41.5 kHz. This is the eleventh harmonic, which lands at 456.5 kHz, right in the IF band. Fortunately, the energy is rather weak at the eleventh harmonic. Even more fortunate is the ease with which that regulation mode is avoided.

The internal regulator comparator has a finite hysteresis. When an additional output filter is used (e.g., the RC filter of the test circuit, or an LC filter), the ripple at the regulation node is minimized. This limits the rate at which the oscillator can be gated. In practice, this means that rather than exhibiting a switching pattern of skipping every other oscillator pulse, it would be more likely to exhibit a switching pattern of three or four pulses followed by the same number of pulses skipped. Although this also tends to increase the output ripple, it is low frequency and has low magnitude (e.g., 10 kHz and 10 mV) which tends to be of little consequence.

Page 14 January 1999 TOKO, Inc.

ADVANCED INFORMATION

SINGLE-CELL APPLICATION (CONT.)

HIGHER-ORDER DESIGN EQUATION

TK652xx

2

]

OUT(TGT)

F

L

OF S

BB

2

+ (V

+ VF)2 + (V

OUT

ƒCS [V

-

)

Targeted output current capability

Diode forward voltage

Inductor winding resistance Output filter resistance Snubber capacitance

2(V

OUT

+ VF)

+ VF - VBB)2 ]

(6)

D

(

D

2ƒ L

D

2ƒ L

capability

)

V

BB2

I

=

OUT

V

+ ROFI

OUT

The equation above was developed as a closed form approximation. In order to allow for a closed form, the design variable that required the least approximation was “I

The approximations made in the equation development have the primary consequence that error is introduced as resistive losses become relatively large. As it is normally a practical design goal to ensure that resistive losses will be relatively small, this seems acceptable. The variables used are:

V

OUT(TGT)

I

Output current

OUT

Output voltage V

OUT

V

Battery voltage, unloaded D Oscillating duty ratio of main switch

BB

f

Oscillator frequency L Inductance value

R

Source resistance (battery + filter) R

S

R

Switch on-state resistance R

SW

R

ESR of upstream output capacitor C

U

+ (VBBRU) + VF - V

D

1- (RS + RL + RSW)

[

2ƒ L

D

1 - (RS+ RL)

(

BB

OUT

2ƒ L

,” as opposed to “L.”

I

Deriving a design solution with this equation is necessarily an iterative process. Use worst-case tolerances as described for inductor selection, using different values for “L” to approximately achieve an “I fine tune the parasitic values as needed and, if necessary, readjust “L” again and reiterate the process.

January 1999 TOKO, Inc. Page 15

” equal to the targeted value. Then,

OUT

TK652xx

ADVANCED INFORMATION

STEP-DOWN CONVERTER APPLICATION

HOW TO MAKE A STEP-DOWN CONVERTER USING THE TK652xx AND AN IRF7524D1 “FETKY” PART

The TK652xx can be used as a controller in a step-down converter with the following two additional changes. See Fig 3.

V

BATT

R

10 k

R

3.9 k

1

2

R

1 k

U1

TK65221

V

U2

IRF7524D1

3

V

1

OUT

IN

LOI

5

GND

5,6, 7,8

1,24

R

4

150

SW

2

R

300 k

4

R

300 k

6

L1

V

120 µH

R

7

1 k

C

1

220 pFLBI

5

6

OUT

+

+ C2

47 µFC347 µF

Note: L = 120 µH Toko P/N: 646CY-121M D75C Coil

FIGURE 3: STEP-DOWN CONVERTER USING THE TK65221 SCHEMATIC

1) Change the main switch orientation for use in a step-down converter. An external P-channel power MOSFET is used as the main switch in a step-down converter configuration. The gate of FET is turned on through a resistor divider being pulled down to GND by the internal output transistor of the TK652xx. This application requires both a logic level P-channel MOSFET and a Schottky diode. An IRF7524D1 “FETKY” part contains both in a small micro 8 package.

2) Change the voltage seen at the V divider between the converter V than the V

pin, the TK652xx will not regulate the output, but will continue to pulse its output transistor.

OUT

IN

pin of the TK652xx to below the regulation voltage at the V

IN

and the chip VIN pin drops the voltage seen at the VIN pin. If the VIN pin is a higher voltage

pin. A resistor

OUT

WHERE TO USE THIS STEP-DOWN CONVERTER

The TK65221 is a Pulse Burst Modulation (PBM) controller with a fixed duty cycle of approximately 50%. Therefore, only if V can and does regulate in DCM (discontinuous current mode) for lighter output current loads with V voltage of V

is more than twice the voltage of V

BATT

. But DCM produces more peak current and more ripple current than CCM. Below is a table giving some

OUT

can the converter run in CCM (continuous current mode). The converter

OUT

less than twice the

IN

examples of where this type of step-down converter might be used.

Type Battery # of Cells V

Range V

BATT

OUT

Typ. Max I

Oper Mode Inductor

OUT

Li-ion 2 (Note 1) 5.4 to 8.4 V 2.1 V 500 mA CCM 120 µH NiMH 4 (Note 2) 4.4 to 5.2 V 2.1 V 500 mA CCM 120 µH NiMH 6 (Note 2) 6.6 to 7.8 V 2.1 V 500 mA CCM 120 µH

Note 1: Li-ion cell voltage range 2.7 V to 4.2 V Note 2: NiMH cell voltage range 1.1 V to 1.3 V

Page 16 January 1999 TOKO, Inc.

ADVANCED INFORMATION

TK652xx

STEP-DOWN CONVERTER APPLICATION (CONT.)

THE AMOUNT OF BOARD SPACE NEEDED TO IMPLEMENT THIS STEP-DOWN CONVERTER

An evaluation board for this converter has been made using a TOKO D73 or D75 series inductor, using only 0.96 sq. inches of board space. The artwork for the surface-mount circuit board is shown below in Figure 4.

Actual Size

VOUT

G

VIN

LOI

LBI

1.2 "

FIGURE 4: TK652xx STEP-DOWN CONVERTER EVALUATION BOARD ARTWORK

.8 "

SELECTING INPUT RESISTOR DIVIDER VALUES

Since the TK652xx draws so little quiescent current into VIN, relatively large resistance values can be used in the input resistor divider. The following shows the relationship equation between V

and VIN pin of the TK652xx, and

BATT

an example method of finding the resistor values to be used in the resistor divider.

V

IN

(R1 + R2)

BATT

=

2

- R1 x (IQ of VIN)

V

X R

therefore:

R1 + R

2

V

BATT

=

R

x [ V

2

and typical (IQ of VIN) = 15 µA and typical LBI trigger value of VIN = 1.1 V and letting R1 = 10 k solving for R

=

V

BATT

- 1.25 V

R

2

10 k x 1.25 V

then for the case of 4 NiMH cells picks a LBI trigger at V

+ R1 x (IQ of VIN)]

IN

= 4.4 V, then R

BATT

= 3.97 k

2

:

2

(1)

January 1999 TOKO, Inc. Page 17

TK652xx

ADVANCED INFORMATION

STEP-DOWN CONVERTER APPLICATION (CONT.)

2) Now, after selecting the R1 and R2 values, check that the V max V

is:

yes: 1.33 V < 2.1 V

Below are the resistor values for the three battery cases listed in “Where to use the TK652xx Step-Down Converter” section.

Type Battery # of Cells V Li-ion 2 5.4 to 8.4 V 2.1 V 10 K 3.01 K 1.79 V < 2.1 NiMH 4 4.4 to 5.2 V 2.1 V 10 K 3.97 K 1.33 V < 2.1 NiMH 6 6.6 to 7.8 V 2.1 V 10 K 2.34 K 1.33 V < 2.1

= 5.2 V and the TK65221 VIN > V

BATT

2.1V >

5.2 V x 3.97 k 10 k + 3.97 k

BATT

= 2.1 V, using the first equation.

REG

- 10 k x 15 µA

Range V

REG

is below V

IN

R1 VALUE ~R2 VALUE VIN CHECK

for the maximum V

REG

. For NiMH cells

BATT

Page 18 January 1999 TOKO, Inc.

ADVANCED INFORMATION

TK652xx

PULSED LOAD APPLICATION

Often in the world of power conversion, the current draw of the load circuit is not constant, but rather pulsed. It is common in power supply design to size the power path large enough, and make the feedback loop fast enough to support these pulsed maximum currents. For applications where the pulse width is long or unpredictable, this approach may be warranted. However, in applications where the pulse width and maximum frequency of occurrence is predictable, such as digital cell phones or two-way pagers, it may be easier and wiser to increase the energy storage in the output filter of the power supply and keep the power path small. This leads to the need for a very large value output capacitor. Panasonic makes a series AL gold cap “super cap” which is a low voltage, large value capacitor in the one farad range.

Before designing a low power DC-DC converter with a “super cap” in its output filter, it is necessary to know the loading profile (the waveform of the current going into the load from the output of the converter) of the application in which it is to be used. The converter can then be designed so that the “super cap” can be recharged in the time before the next big discharge current pulse comes along.

Figure 5 is an example “super cap” charge/discharge diagram showing that the charge into the cap needs to equal the charge leaving the cap during discharge. This diagram comes from the loading and unloading profile information. In reality, some extra charge needs to go into the cap to make up for the losses caused by ESR of the cap.

1A

60 ms

Note: Equal charge into and out of “supercap”

2 s (30 mA) = 60 ms (1A)

I

OUT

30 mA

time

2 s

FIGURE 5: “SUPER CAP” CHARGE/DISCHARGE DIAGRAM

Figure 6 is a schematic for this “super cap” example application.

R

N

1 k

C

N

10 µF

I

B

V

IN

L = 39 µF

V

IN

LBI GND

R 1 k

D

S

C

S

220 pF

V

LOI

OUTIND

+

"SUPERCAP"

GOLD CAP

1 F

Drawing not to scale

C

D

10 µF

+

I

OUT

V

OUT

FIGURE 6: PULSED LOAD “SUPER CAP” APPLICATION SCHEMATIC

January 1999 TOKO, Inc. Page 19

TK652xx

ADVANCED INFORMATION

PACKAGE OUTLINE

SOT-23L-6

123

e e

0.95 0.95

Marking Information

Marking

TK65218 18M TK65221 21M

+0.15

- 0.05

0.4

0.1

M

456

marking

+0.15

- 0.05

0.32

+0.3

- 0.1

3.5

3.4

5 PL

0.1

M

0.3

0.6

1.0

Recommended Mount Pad

ee

0.95 0.95

2.2

3.0 e1

TK65224 24M

0.1

Dimensions are shown in millimeters Tolerance: x.x = ± 0.2 mm (unless otherwise specified)

1.4 max

1.2

0 ~ 0.1

+0.15

- 0.05

0.15

+ 0.3

3.3

max 15

0.4

Toko America, Inc. Headquarters 1250 Feehanville Drive, Mount Prospect, Illinois 60056 Tel: (847) 297-0070 Fax: (847) 699-7864

TOKO AMERICA REGIONAL OFFICES

Midwest Regional Office Toko America, Inc. 1250 Feehanville Drive Mount Prospect, IL 60056 Tel: (847) 297-0070 Fax: (847) 699-7864

Western Regional Office Toko America, Inc. 2480 North First Street , Suite 260 San Jose, CA 95131 Tel: (408) 432-8281 Fax: (408) 943-9790

Eastern Regional Office Toko America, Inc. 107 Mill Plain Road Danbury, CT 06811 Tel: (203) 748-6871 Fax: (203) 797-1223

Semiconductor Technical Support Toko Design Center 4755 Forge Road Colorado Springs, CO 80907 Tel: (719) 528-2200 Fax: (719) 528-2375

Visit our Internet site at http://www.tokoam.com

The information furnished by TOKO, Inc. is believed to be accurate and reliable. However, TOKO reserves the right to make changes or improvements in the design, specification or manufacture of its products without further notice. TOKO does not assume any liability arising from the application or use of any product or circuit described herein, nor for any infringements of patents or other rights of third parties which may result from the use of its products. No license is granted by implication or otherwise under any patent or patent rights of TOKO, Inc.

Page 20 January 1999 TOKO, Inc.

IC-xxx-TK652xx

0798O0.0K