Datasheet SA5212AD, SA5212AFE, SA5212AN Datasheet (Philips)

INTEGRATED CIRCUITS

SA5212A

Transimpedance amplifier (140MHz)

Product specification
Replaces datasheet NE/SA/SE5212A of 1995 Apr 26
IC19 Data Handbook




1998 Oct 07

Philips Semiconductors Product specification

P

SA5212ATransimpedance amplifier (140MHz)

DESCRIPTION

The SA5212A is a 14kΩ transimpedance, wideband, low noise
differential output amplifier, particularly suitable for signal recovery in
fiber optic receivers and in any other applications where very low
signal levels obtained from high-impedance sources need to be
amplified.

FEA TURES

•Extremely low noise: 2.5pA/√Hz

•Single 5V supply

•Large bandwidth: 140MHz

PIN CONFIGURATION

N, FE, D Packages

1

I

IN

2

V

CC

3

GND

1

45

GND

1

Figure 1. Pin Configuration

8

7

6

SD00336

GND

OUT (–)

GND

2

OUT (+)

2

•Differential outputs

•Low input/output impedances

•14kΩ differential transresistance

•ESD hardened

•Wideband gain block

•Medical and scientific instrumentation

•Sensor preamplifiers

•Single-ended to differential conversion

APPLICA TIONS

•Fiber-optic receivers, analog and digital

•Low noise RF amplifiers

•RF signal processing

•Current-to-voltage converters

ORDERING INFORMATION

DESCRIPTION TEMPERATURE RANGE ORDER CODE DWG #

8-Pin Plastic Small Outline (SO) Package -40°C to +85°C SA5212AD SOT96-1
8-Pin Plastic Dual In-Line Package (DIP) -40°C to +85°C SA5212AN SOT97-1
8-Pin Ceramic Dual In-Line Package (DIP) -40°C to +85°C SA5212AFE 0580A

ABSOLUTE MAXIMUM RATINGS

SYMBOL PARAMETER SA5212A UNIT

V

CC

D MAX

I

IN MAX

T

A

T

J

T

STG

NOTES:

1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance:

2. The use of a pull-up resistor to V

8-Pin Plastic DIP: 110°C/W
8-Pin Plastic SO: 160°C/W
8-Pin Cerdip: 165°C/W

Power Supply 6 V
Power dissipation, TA=25°C (still air)

8-Pin Plastic DIP 1100 mW
8-Pin Plastic SO 750 mW

8-Pin Cerdip 750 mw
Maximum input current
Operating ambient temperature range -40 to 85 °C
Operating junction -55 to 150 °C
Storage temperature range -65 to 150 °C

CC

2

, for the PIN diode, is recommended

1

5 mA

1998 Oct 07 853-1266 20142

2

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

RECOMMENDED OPERATING CONDITIONS

SYMBOL PARAMETER RATING UNIT

V

CC

T

A

T

J

DC ELECTRICAL CHARACTERISTICS

Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V
and T

=25°C1.

A

SYMBOL

V

IN

V

±

O

V

OS

I

CC

I

OMAX

I

IN

I

N MAX

NOTES:

1. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.

Supply voltage range 4.5 to 5.5 V
Ambient temperature ranges -40 to +85 °C
Junction temperature ranges -40 to +105 °C

PARAMETER TEST CONDITIONS Min Typ Max UNIT

Input bias voltage 0.55 0.8 1.05 V
Output bias voltage 2.5 3.3 3.8 V
Output offset voltage 120 mV
Supply current 20 26 33 mA
Output sink/source current 3 4 mA
Maximum input current (2% linearity) Test Circuit 6, Procedure 2 ±40 ±80 µA
Maximum input current overload threshold T est Circuit 6, Procedure 4 ±60 ±120 µA

1998 Oct 07

3

Philips Semiconductors Product specification

g

T

SA5212ATransimpedance amplifier (140MHz)

AC ELECTRICAL CHARACTERISTICS

Minimum and Maximum limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V
and T

=25°C5.

A

SYMBOL

R
R

R
R

f

R
C

T

O
T
O

3dB

IN
IN

Transresistance (differential output)
Output resistance (differential output) DC tested 14 30 46 Ω

Transresistance (single-ended output) DC tested, R
Output resistance (single-ended output) DC tested 7 15 23 Ω

Bandwidth (-3dB) T

Input resistance 70 110 150 Ω
Input capacitance 10 18 pF

∆R/∆V Transresistance power supply sensitivity V
∆R/∆T

I

N

Transresistance ambient
temperature sensitivity

RMS noise current spectral density
(referred to input)

Integrated RMS noise current over the band­width (referred to input) C

I

T

C

= 1pF ∆f = 100MHz 32

S

PSRR Power supply rejection ratio

PSRR

V

O MAX

V

IN MAX

t

R

Power supply rejection ratio
(ECL configuration)

Maximum differential output voltage swing
Maximum input amplitude for output duty

cycle of 50 ±5%
Rise time for 50mV output signal

NOTES:

1. Package parasitic capacitance amounts to about 0.2pF.

2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in V

3. Guaranteed by linearity and over load tests.
defined as 20-80% rise time. It is guaranteed by -3dB bandwidth test.

4. t

R

5. As in all high frequency circuits, a supply bypass capacitor should be located as close to the part as possible.

PARAMETER TEST CONDITIONS Min Typ Max UNIT

DC tested, R

Test Circuit 6, Procedure 1

= ∞

L

= ∞ 4.5 7 9.5 kΩ

L

9.0 14 19 kΩ

Test Circuit 1

D package,

= 25°C 100 140 MHz

A

N, FE packages,

T

= 25°C 100 120

A

= 5 ±0.5V 9.6 %/V

CC

D package

= T

∆T

A

A MAX-TA MIN

Test Circuit 2

f = 10MHz T

T

= 25°C Test Circuit 2

A

= 25°C

A

∆f = 50MHz

1

= 0

S

∆f = 100MHz 27
∆f = 200MHz 40

0.05 %/°C

2.5 pA/√Hz

20

nA

∆f = 50MHz 22

∆f = 200MHz 52

Any package

2

DC tested

∆V

= 0.1V

CC

20 33 dB

Equivalent AC

Test Circuit 3

2

Any package

f = 0.1MHz

1

23 dB

Test Circuit 4

R

= ∞

Test Circuit 6, Procedure 3

L

3

4

Test Circuit 5 325 mV
Test Circuit 5 2.0 ns

1.7 3.2 V

CC

P-P

P-P

line.

1998 Oct 07

4

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TEST CIRCUITS

SINGLE-ENDED DIFFERENTIAL

Rt

V

OUT

2  S21  RR

V

IN

V

OUT



t

V

IN

4  S21  R

ZO = 50Ω

RO Z

PORT 1

0.1µF

50

1  S22





O

 33 RO 2Z

1  S22

NETWORK ANALYZER

S-PARAMETER TEST SET

R = 1k

IN DUT

GND

1

Test Circuit 1

1  S22





1  S22

RL = 50Ω

 66

SPECTRUM ANALYZER

V

CC

NC

IN DUT

GND

1

Test Circuit 2

OUT

OUT

GND

AV = 60DB

1µF

33

1µF

33

R

= 50

L

2

SD00337

O

PORT 2

V

CC

1µF

33

OUT

1µF

33

OUT

GND

2

Figure 2. Test Circuits 1 and 2

5V + ∆V

10µF

10µF

0.1µF

10µF

NC

16

GND

V

CC

IN

DUT

1

CURRENT PROBE

1mV/mA

1µF

GND

33

33

1µF

2

OUT

OUT

Test Circuit 3

Figure 3. Test Circuit 3

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1 PORT 2

CAL

50

100
BAL.

TRANSFORMER

NH0300HB

UNBAL.

TEST

SD00338

1998 Oct 07

5

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TEST CIRCUITS (Continued)

NETWORK ANALYZER

–5.2V + ∆V

10µF

10µF

0.1µF

0.1µF

NC

PULSE GEN.

GND

V

CC

16

1

OUT

IN

OUT

CURRENT PROBE

1mV/mA

GND

2

1µF

33

33

1µF

Test Circuit 4

Figure 4. Test Circuit 4

5V

S-PARAMETER TEST SET

PORT 1 PORT 2

CAL

50

100
BAL.

TRANSFORMER

NH0300HB

UNBAL.

TEST

SD00339

1998 Oct 07

50

0.1µF

GND

1

OUT

DUT

OUT

GND

IN

1k

Test Circuit 5

Figure 5. Test Circuit 5

1µF

33

33

1µF

2

A

Z

= 50Ω

O

OSCILLOSCOPE

B

ZO = 50Ω

Measurement done using

differential wave forms

SD00545

6

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TEST CIRCUITS (Continued)

Typical Differential Output Voltage

vs Current Input

5V

OUT +

IN

DUT

GND

1

OUT –

GND

2

CURRENT INPUT (µA)

IIN (µA)

2.00

1.60

1.20

0.80

0.40

0.00

–0.40

–0.80

DIFFERENTIAL OUTPUT VOLTAGE (V)

–1.20

–1.60

–2.00

–200 –160 –120 –80 –40 0 40 80 120 160 200

+

(V)

V

OUT

–

1998 Oct 07

NE5212A TEST CONDITIONS

Procedure 1

Procedure 2

Procedure 3

Procedure 4

Test Circuit 8

Figure 6. Test Circuit 8

measured at 30µA

R

T

= (VO1 – VO2)/(+30µA – (–30µA))

R

T

Where: V

Linearity = 1 – ABS((V
Where: V

V

OMAX

Where: VO7 Measured at IIN = +130µA

IIN Test Pass Conditions:
V

O7

Where: V

Measured at IIN = +30µA

O1

V

Measured at IIN = –30µA

O2

– VOB) / (VO3 – VO4))

OA

Measured at IIN = +60µA

O3

V

Measured at IIN = –60µA

O4

VOA RT ( 60A)  V

VOB RT ( 60A)  V

= VO7 – V

– VO5 > 20mV and V06 – VO5 > 20mV

O8

Measured at IIN = –130µA

V

O8

Measured at IIN = +800µA

O5

V

Measured at IIN = –80µA

O6

V

Measured at IIN = +130µA

O7

Measured at IIN = –130µA

V

O8

7

OB
OB

SD00340

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TYPICAL PERFORMANCE CHARACTERISTICS

NE5212A Supply Current

vs Temperature

30

VCC = 5.0V

29

28

27

SUPPLY CURRENT (mA)

26

25

–60 –40 –20 0 20 40 10060 12080 140

AMBIENT TEMPERATURE (°C)

NE5212A Differential Output Swing

vs Temperature

3.8
VCC = 5.0V

3.6
DC TESTED

RL = ∞

3.4

3.2

3.0

2.8

DIFFERENTIAL OUTPUT SWING (V)

2.6

2.4

–60 –40 –20 0 20 40 10060 12080 140

AMBIENT TEMPERATURE (°C)

NE5212A Input Bias Voltage

vs Temperature

950

900

850

800

750

700

INPUT BIAS VOLTAGE (mV)

650

600

–60 –40 –20 0 20 40 10060 12080 140

AMBIENT TEMPERATURE (°C)

VCC = 5.0V

NE5212A Output Offset Voltage

vs Temperature

80

VCC = 5.0V

60

= V

OUT5

– V

OUT7

V

OS

40

20

0

–20

–40

OUTPUT OFFSET VOLTAGE (mV)

–60

–60 –40 –20 0 20 40 10060 12080 140

AMBIENT TEMPERATURE (°C)

NE5212A Output Bias Voltage

vs Temperature

3.50
VCC = 5.0V

3.45

PIN 5

3.40

3.35

3.30

OUTPUT BIAS VOLTAGE (V)

3.25

–60 –40 –20 0 20 40 10060 12080 140

AMBIENT TEMPERATURE (°C)

PIN 7

NE5212A Differential Transresistance

vs Temperature

17.0

Ω

VCC = 5.0V

16.5
DC TESTED

R

= ∞

L

16.0

15.5

15.0

14.5

14.0

DIFFERENTIAL TRANSRESISTANCE (k )

–60 –40 –20 0 20 40 10060 12080

AMBIENT TEMPERATURE (°C)

140

NE5212A Power Supply Rejection Ratio

vs Temperature

40

VCC = 5.0V

39

38

37

36

35

34

33

POWER SUPPLY REJECTION RATIO (dB)

–60 –40 –20 0 20 40 10060 12080

= ±0.1V

∆V

CC

DC TESTED
OUTPUT REFERRED

AMBIENT TEMPERATURE (°C)

1998 Oct 07

NE5212A Output Resistance

vs Temperature

17

VCC = 5.0V

DC TESTED

PIN 7

PIN 5

140

16

Ω

15

14

13

12

11

OUTPUT RESISTANCE ( )

10

9

–60 –20 0 20 40 60 80 100 120–40

AMBIENT TEMPERATURE (°C)

Figure 7. Typical Performance Characteristics

8

140

NE5212A Typical

Bandwidth Distribution

(75 Parts from 3 Wafer Lots)

50

PIN 5
SINGLE-ENDED

= 50Ω

R

40

L

30

20

POPULATION (%)

10

0

112.5 122.5 132.5 142.5 152.5 162.5
FREQUENCY (MHz)

N, F PKG
V
T

CC

= 25°C

A

= 5.0V

SD00341

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TYPICAL PERFORMANCE CHARACTERISTICS (Continued)

Gain vs Frequency Gain vs Frequency

12
11
10

9
8
7

GAIN (dB)

6

N PKG
PIN 5

5

T

= 25°C

A

4
3

0.1 1 10 100
FREQUENCY (MHz)

5.5V

5.0V

4.5V

Output Resistance

vs Frequency

100

90

D PKG
TA = 25°C

80

VCC = 5V

70
60
50
40
30
20

OUTPUT RESISTANCE ( )Ω

10

0.1 1 10 100
FREQUENCY (MHz)

11
10

9
8
7

GAIN (dB)

6

N PKG

5

PIN 7

= 25°C

T

4

A

3

0.1 1 10 100
FREQUENCY (MHz)

5.0V

Gain vs Frequency

11
10

9
8
7
6

N PKG

GAIN (dB)

5

PIN 7

4

V

3

0.1 1 10 100

125°C

–55°C

= 5V

CC

FREQUENCY (MHz)

125°C

4.5V

5.5V

–55°C

Output Resistance

vs Frequency

80

N PKG

70

VCC = 5V

60

T

= 25°C

A

50
40
30
20
10

OUTPUT RESISTANCE ( )Ω

0.1 1 10 100
FREQUENCY (MHz)

Gain vs Frequency

11

10

9
8
7
6

GAIN (dB)

5
4
3

0.1 1 10 100

125°C

25°C

N PKG
PIN 5

= 5V

V

CC

FREQUENCY (MHz)

85°C

–55°C

PIN 5

PIN 7

–55°C

125°C

Gain and Phase Shift

vs Frequency

11
10

9
8
7

TA = 25°C

6

GAIN (dB)

5
4
3

0.1 1 10 100

φ

N PKG
PIN 5

= 5V

V

CC

FREQUENCY (MHz)

1998 Oct 07

Gain and Phase Shift

vs Frequency

–45

–135

–225

11

10

o

9
8

D PKG

7

PHASE ( )

PIN 7

6

V

= 5V

CC

5

GAIN (dB)

= 25°C

T

A

4
3

0.1 1 10 100
FREQUENCY (MHz)

φ

–180

–270

–360

Figure 8. Typical Performance Characteristics (cont.)

9

Gain and Phase Shift

vs Frequency

11

10

9
8

o

PHASE ( )

N PKG

7

PIN 7

6
5

GAIN (dB)

4
3

= 5V

V

CC

T

= 25°C

A

0.1 1 10 100
FREQUENCY (MHz)

φ

–180

–270

–360

SD00342

o

PHASE ( )

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

TYPICAL PERFORMANCE CHARACTERISTICS (Continued)

Gain and Phase Shift

vs Frequency

11
10

9
8
7

D PKG
PIN 5

6

GAIN (dB)

V

= 5V

CC

5

= 25°C

T

A

4
3

0.1 1 10 100
FREQUENCY (MHz)

Differential Output Voltage

vs Input Current

2.000

0

–55°C

25°C

85°C

125°C

0

–90

–180

4.5

o

PHASE ( )

2.0

DIFFERENTIAL OUTPUT VOLTAGE (V)

–150.0 150.00

10

8
6
4
2
0

DELAY (ns)

Output Voltage

vs Input Current

125°C

85°C

–55°C

25°C

INPUT CURRENT (µA)

Group Delay

vs Frequency

85°C

125°C

25°C–55°C

Differential Output Voltage

vs Input Current

2.0

0

5.5V

5.0V

DIFFERENTIAL OUTPUT VOLTAGE (V)

–2.0

4.5V

–150.0 150.00

INPUT CURRENT (µA)

5.0V

5.5V

4.5V

OUTPUT VOLTAGE (V)

–55°C

25°C

–2.000

125°C

–150.0 150.0

85°C

INPUT CURRENT (µA)

0.1 20 40 60 80 100 120 140 160
FREQUENCY (MHz)

Output Step Response

VCC = 5V
TA = 25°C

20mV/Div

0 2 4 6 8 10 12 14 16 18 20

(ns)

Figure 9. Typical Performance Characteristics (cont.)

SD00343

1998 Oct 07

10

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

THEORY OF OPERATION

Transimpedance amplifiers have been widely used as the
preamplifier in fiber-optic receivers. The SA5212A is a wide
bandwidth (typically 140MHz) transimpedance amplifier designed
primarily for input currents requiring a large dynamic range, such as
those produced by a laser diode. The maximum input current before
output stage clipping occurs at typically 240µA. The SA5212A is a
bipolar transimpedance amplifier which is current driven at the input
and generates a differential voltage signal at the outputs. The
forward transfer function is therefore a ratio of the differential output
voltage to a given input current with the dimensions of ohms. The
main feature of this amplifier is a wideband, low-noise input stage
which is desensitized to photodiode capacitance variations. When
connected to a photodiode of a few picoFarads, the frequency
response will not be degraded significantly. Except for the input
stage, the entire signal path is differential to provide improved
power-supply rejection and ease of interface to ECL type circuitry. A
block diagram of the circuit is shown in Figure 10. The input stage
(A1) employs shunt-series feedback to stabilize the current gain of
the amplifier. The transresistance of the amplifier from the current
source to the emitter of Q
feedback resistor, R

is approximately the value of the

3

=7kΩ. The gain from the second stage (A2)

F

and emitter followers (A3 and A4) is about two. Therefore, the
differential transresistance of the entire amplifier , R

V

(diff)

OUT



R

T

 2RF 2(7.2K)  14.4kW

I

IN

is

T

The single-ended transresistance of the amplifier is typically 7.2kΩ.
The simplified schematic in Figure 11 shows how an input current is

converted to a differential output voltage. The amplifier has a single
input for current which is referenced to Ground 1. An input current
from a laser diode, for example, will be converted into a voltage by
the feedback resistor R
open loop gain of the circuit, A
minimizes loading on Q1. The transistor Q4, resistor R7, and V

. The transistor Q1 provides most of the

F

≈70. The emitter follower Q

VOL

2

B1

provide level shifting and interface with the Q15– Q16 differential
pair of the second stage which is biased with an internal reference,
V

. The differential outputs are derived from emitter followers Q

B2

which are biased by constant current sources. The collectors of

Q

12

Q

– Q12 are bonded to an external pin, V

11

, in order to reduce

CC2

11

the feedback to the input stage. The output impedance is about 17Ω
single-ended. For ease of performance evaluation, a 33Ω resistor is
used in series with each output to match to a 50Ω test system.

BANDWIDTH CALCULATIONS

The input stage, shown in Figure 12, employs shunt-series feedback
to stabilize the current gain of the amplifier. A simplified analysis can
determine the performance of the amplifier. The equivalent input
capacitance, C

7.5pF, assuming that C
capacitance.

Since the input is driven by a current source the input must have a
low input resistance. The input resistance, R
incremental input voltage, V
and can be calculated as:

, in parallel with the source, IS, is approximately

IN

=0 where CS is the external source

S

, is the ratio of the

, to the corresponding input current, I

IN

IN

V

IN



R

IN

I

IN



R

1  A

F

VOL

7.2K



70

 103W

More exact calculations would yield a higher value of 110Ω.
Thus C

and RIN will form the dominant pole of the entire amplifier;

IN

f

3dB

1



2p R

INCIN

Assuming typical values for RF = 7.2kΩ, RIN = 110Ω, CIN = 10pF

f

3dB



2p (110) 10  10

1

12

 145MHz

The operating point of Q1, Figure 2, has been optimized for the
lowest current noise without introducing a second dominant pole in
the pass-band. All poles associated with subsequent stages have
been kept at sufficiently high enough frequencies to yield an overall
single pole response. Although wider bandwidths have been
achieved by using a cascade input stage configuration, the present
solution has the advantage of a very uniform, highly desensitized
frequency response because the Miller effect dominates over the
external photodiode and stray capacitances. For example, assuming
a source capacitance of 1pF, input stage voltage gain of 70, R
60Ω then the total input capacitance, C

= (1+7.5) pF which will

IN

=

IN

lead to only a 12% bandwidth reduction.

OUTPUT +

A3

INPUT

–

A1 A2

R

F

A4

OUTPUT –

SD00327

Figure 10. SA5212A – Block Diagram

NOISE

Most of the currently installed fiber-optic systems use non-coherent
transmission and detect incident optical power. Therefore, receiver
noise performance becomes very important. The input stage
achieves a low input referred noise current (spectral density) of

3.5pA/√Hz
external high value bias resistors often required for photodiode
biasing will not contribute to the total noise system noise. The
equivalent input
quiescent current of Q
bandwidth; however, it is not dependent upon the internal
Miller-capacitance. The measured wideband noise was 52nA RMS

IN

in a 200MHz bandwidth.

. The transresistance configuration assures that the

noise current is strongly determined by the

RMS

, the feedback resistor RF, and the

1

1998 Oct 07

11

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

V

CC1

R

1

R

3

R

12

R

13

V

CC2

Q

INPUT

GND

PHOTODIODE

2

Q

Q

1

1

3

R

2

R

5

Figure 11. Transimpedance Amplifier

V

CC

I

C1

R1

INPUT

I

IN

V

IN

I

B

Q1

I

F

R

F

Q2

R3

Q3

R2

V

EQ3

R4

SD00329

Figure 12. Shunt-Series Input Stage

DYNAMIC RANGE

The electrical dynamic range can be defined as the ratio of
maximum input current to the peak noise current:

Electrical dynamic range, D
I

= 120µA and a wideband noise of IEQ=52nA

INMAX

external source capacitance of C

(Max. input current)

D

+

E

(Peak noise current)

DE(dB) + 20log

DE(dB) + 20log

(120 @ 10

(120mA)

In order to calculate the optical dynamic range the incident optical
power must be considered.

For a given wavelength λ;
Energy of one Photon =

Where h=Planck’s Constant = 6.6 ×10
c = speed of light = 3 × 10
c / λ = optical frequency

, in a 200MHz bandwidth assuming

E

= 1pF.

S

*6

)

(2Ǹ52nA)

(73nA)

+ 64dB

hc

watt sec (Joule)

l

8

m/sec

-34

Joule sec.

RMS

for an

Q

4

+

Q

15

R

14

R

7

R

4

GND

Q

16

R

15

VB2

2

Q

11

Q

12

OUT–

+

OUT+

SD00328

No. of incident photons/sec= where P=optical incident power

P

No. of incident photons/sec =

hc

l

where P = optical incident power

P

No. of generated electrons/sec =

h @

hc

l

where η = quantum efficiency

no. of generated electron hole paris

+

no. of incident photons

P

NI + h @

where e = electron charge = 1.6 × 10
Responsivity R =

hc

@ e Amps (Coulombsńsec.)

l

h@e

Amp/watt

hc

-19

Coulombs

l

I + P @ R

Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the
noise parameter Z may be calculated as:

I

EQ

Z +

qB

+

(1.6 @ 10

where Z is the ratio of

*9

52 @ 10

*19

)(200 @ 106)

noise output to the peak response to a

RMS

1

+ 1625

ǒ

Amp
Amp

Ǔ

single hole-electron pair. Assuming 100% photodetector quantum
efficiency, half mark/half space digital transmission, 850nm
lightwave and using Gaussian approximation, the minimum required
optical power to achieve 10

+ 12

hc

l

P

avMIN

-9

BER is:

BZ+ 12 (2.3@ 10

*19

)

200 @ 1061625 + 897nW +*30.5dBm,

where h is Planck’s Constant, c is the speed of light, λ is the
wavelength. The minimum input current to the SA5212A, at this
input power is:

I

avMIN

+

+ qP

897 @ 10

avMIN

hc

*9

@ 1.6 @ 10

2.3 @ 10

l

*19

*19

= 624nA

1998 Oct 07

12

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

Choosing the maximum peak overload current of I

avMAX

=120µA, the

maximum mean optical power is:

V

R = 560

IN

IN

a. Non-inverting 20dB Amplifier

V

R = 560

IN

IN

b. Inverting 20dB Amplifier

V

R = 560

IN

IN

c. Differential 20dB Amplifier

NE5212A

NE5212A

NE5212A

OUT–

OUT+

OUT+

OUT–

OUT+

OUT–

SD00344

Figure 13. Variable Gain Circuit

P

avMAX

hcI

avMAX



lq



2.3  10

19

(120  106)

1.6  10

19

= 172µW or –7.6dBm
Thus the optical dynamic range, D
D

= P

O

avMAX

- P

= -30.5 -(-7.6) = 22.8dB.

avMIN

is:

O

This represents the maximum limit attainable with the SA5212A
operating at 200MHz bandwidth, with a half mark/half space digital
transmission at 820nm wavelength.

APPLICATION INFORMATION

Package parasitics, particularly ground lead inductances and
parasitic capacitances, can significantly degrade the frequency
response. Since the SA5212A has differential outputs which can
feed back signals to the input by parasitic package or board layout
capacitances, both peaking and attenuating type frequency
response shaping is possible. Constructing the board layout so that
Ground 1 and Ground 2 have very low impedance paths has
produced the best results. This was accomplished by adding a
ground-plane stripe underneath the device connecting Ground 1,
Pins 8–11, and Ground 2, Pins 1 and 2 on opposite ends of the
SO14 package. This ground-plane stripe also provides isolation
between the output return currents flowing to either V
2 and the input photodiode currents to flowing to Ground 1. Without
this ground-plane stripe and with large lead inductances on the
board, the part may be unstable and oscillate near 800MHz. The
easiest way to realize that the part is not functioning normally is to
measure the DC voltages at the outputs. If they are not close to their
quiescent values of 3.3V (for a 5V supply), then the circuit may be
oscillating. Input pin layout necessitates that the photodiode be
physically very close to the input and Ground 1. Connecting Pins 3
and 5 to Ground 1 will tend to shield the input but it will also tend to
increase the capacitance on the input and slightly reduce the
bandwidth.

or Ground

CC2

As with any high-frequency device, some precautions must be
observed in order to enjoy reliable performance. The first of these is
the use of a well-regulated power supply. The supply must be
capable of providing varying amounts of current without significantly
changing the voltage level. Proper supply bypassing requires that a
good quality 0.1µF high-frequency capacitor be inserted between
V

CC1

and V

, preferably a chip capacitor, as close to the package

CC2

pins as possible. Also, the parallel combination of 0.1µF capacitors
with 10µF tantalum capacitors from each supply, V

CC1

and V

CC2

, to
the ground plane should provide adequate decoupling. Some
applications may require an RF choke in series with the power
supply line. Separate analog and digital ground leads must be
maintained and printed circuit board ground plane should be
employed whenever possible.

BASIC CONFIGURATION

A trans resistance amplifier is a current-to-voltage converter. The
forward transfer function then is defined as voltage out divided by
current in, and is stated in ohms. The lower the source resistance,
the higher the gain. The SA5212A has a differential transresistance
of 14kΩ typically and a single-ended transresistance of 7kΩ
typically. The device has two outputs: inverting and non-inverting.
The output
voltage in the differential output mode is twice that of the output
voltage in the single-ended mode. Although the device can be used
without coupling capacitors, more care is required to avoid upsetting
the internal bias nodes of the device. Figure 13 shows some basic
configurations.

VARIABLE GAIN

Figure 14 shows a variable gain circuit using the SA5212A and the
SA5230 low voltage op amp. This op amp is configured in a
non-inverting gain of five. The output drives the gate of the SD210
DMOS FET. The series resistance of the FET changes with this
output voltage which in turn changes the gain of the SA5212A. This
circuit has a distortion of less than 1% and a 25dB range, from

-42.2dBm to -15.9dBm at 50MHz, and a 45dB range, from -60dBm
to -14.9dBm at 10MHz with 0 to 1V of control voltage at V

OUT+

RF

OUT–

SD00345

CC

IN

0–1V

2.4k

0.1µF

51

+5V

10k

SD210

IN

0–5V

NE5212A

RF

V

Figure 14. Variable Gain Circuit

OUT

CC

.

1998 Oct 07

13

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

16MHZ CRYSTAL OSCILLATOR

Figure 15 shows a 16MHz crystal oscillator operating in the series
resonant mode using the SA5212A. The non-inverting input is fed
back to the input of the SA5212A in series with a 2pF capacitor. The
output is taken from the inverting output.

+5V

OUT+

NE5212A

IN

Figure 15. 16MHz Crystal Oscillator

V

EE

V

CC

–15V

V

+5.0

BPF31

EE

0.01µF

1

4.7

NE5212A

2

3

1.0µF

0.01µF

6

4

OUT–

SD00346

0.1µF

7
5

8

–5.2V

0.1µF

4.7µF 4.7µF0.1µF

1

V

BB

1k

1k

2.7µH

DIGIT AL FIBER OPTIC RECEIVER

Figures 16 and 17 show a fiber optic receiver using off-the-shelf
components.

The receiver shown in Figure 16 uses the SA5212A, the Philips
Semiconductors 101 16 ECL line receiver, and Philips/Amperex
BPF31 PIN diode. The circuit is a capacitor-coupled receiver and
utilizes positive feedback in the last stage to provide the hysteresis.
The amount of hysteresis can be tailored to the individual application
by changing the values of the feedback resistors to maintain the
desired balance between noise immunity and sensitivity. At room
temperature, the circuit operates at 50Mbaud with a BER of 10E-10
and over the automotive temperature range at 40Mbaud with a BER
of 10E-9. Higher speed experimental diodes have been used to
operate this circuit at 220Mbaud with a BER of 10E-10.

Figure 17 depicts a TTL receiver using the SA5212A and the
SA5214 fast amplifier system along with the Philips/Amperex PIN
diode. The system shown is optimized for 50 Mb/s Non Return to
Zero (NRZ) data. A link status indication is provided along with a
jamming function when the input level is below a
user-programmable threshold level.

1

V

BB

510

1

16

9

7

1/3
10116

10

6

11

0.01µF

1

V

BB

510

0.1µF

V

EE

5

4

1/3
10116

8

2

3

0.01µF

510

100pF

100pF

510

1k

1k

13

15

1/3
10116

12

14

1k

1k

1

V

BB

510

ECL

ECL

510

NOTE:

1. Tie all V

1998 Oct 07

points together.

BB

SD00347

Figure 16. ECL Fiber Optic Receiver

14

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

+V

CC

47µF

C1

C2

.01µF

GND

L1

10µH

R1
100

10µH

L3

10µH

L2

C10

10µF

C12

10µF

R2

220

C9

R3
47k

C11

.01µF

.01µF

D1

LED

100pF

C13

V

LED

1

C

PKDET

2

THRESH

3

GND

A

4

FLAG

5

JAM

6

V

CCD

7

V

CCA

8

GND

D

9

TTL

OUT

10 11

(TTL)

OUT

Figure 17. A 50Mb/s TTL Digital Fiber Optic Receiver

NE5214

R

IN

IN

C

C

OUT

IN

OUT

IN

R

HYST

PKDET

AZP

AZN

C3

10µF

20

1B

19

1A

18

17

16

2B

15

8B

14

2A

13

8A

C7

100pF

C8

0.1µF

OUT+

GND

6

2

OUT–

7

GND

2

8

GND

GND

NE5212A

1

45

3

1

V

CC

2

I

IN

1

C4

.01µF

1.0µF

.01µF

BPF31

OPTICAL

INPUT

C5

C6

12

R4

5.1k

SD00348

1998 Oct 07

15

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

GND2

I

IN

1

8

2

3

4

ECN No.: 99918
1990 Jul 5

V

GND1

GND1

CC

Figure 18. SA5212A Bonding Diagram

Die Sales Disclaimer

Due to the limitations in testing high frequency and other parameters
at the die level, and the fact that die electrical characteristics may
shift after packaging, die electrical parameters are not specified and
die are not guaranteed to meet electrical characteristics (including
temperature range) as noted in this data sheet which is intended
only to specify electrical characteristics for a packaged device.

All die are 100% functional with various parametrics tested at the
wafer level, at room temperature only (25°C), and are guaranteed to
be 100% functional as a result of electrical testing to the point of
wafer sawing only. Although the most modern processes are
utilized for wafer sawing and die pick and place into waffle pack

7

6

5

OUT–

GND2

OUT+

SD00489

carriers, it is impossible to guarantee 100% functionality through this
process. There is no post waffle pack testing performed on
individual die.

Since Philips Semiconductors has no control of third party
procedures in the handling or packaging of die, Philips
Semiconductors assumes no liability for device functionality or
performance of the die or systems on any die sales.

Although Philips Semiconductors typically realizes a yield of 85%
after assembling die into their respective packages, with care
customers should achieve a similar yield. However, for the reasons
stated above, Philips Semiconductors cannot guarantee this or any
other yield on any die sales.

1998 Oct 07

16

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

DIP8: plastic dual in-line package; 8 leads (300 mil) SOT97-1

1998 Oct 07

17

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

SO8: plastic small outline package; 8 leads; body width 3.9mm SOT96-1

1998 Oct 07

18

1998 Oct 07

853–0580A 006688

0580A 8-PIN (300 mils wide) CERAMIC DUAL IN-LINE (F) PACKAGE

Philips Semiconductors Product specification

19

0.055 (1.40)

0.030 (0.76)

– E –

PIN # 1

– D –

– T –

SEATING

PLANE

0.408 (10.36)

0.376 (9.55)

0.070 (1.78)

0.050 (1.27)

0.023 (0.58)

0.015 (0.38)

0.055 (1.40)

0.030 (0.76)

0.303 (7.70)

0.245 (6.22)

0.100 (2.54) BSC

0.200 (5.08)

0.165 (4.19)

0.165 (4.19)

0.125 (3.18)

0.010 (0.254)TED

NOTES:

1. Controlling dimension: Inches. Millimeters are
shown in parentheses.

2. Dimension and tolerancing per ANSI Y14. 5M-1982.

3. “T”, “D”, and “E” are reference datums on the body
and include allowance for glass overrun and meniscus
on the seal line, and lid to base mismatch.

4. These dimensions measured with the leads
constrained to be perpendicular to plane T.

5. Pin numbers start with Pin #1 and continue
counterclockwise to Pin #8 when viewed
from the top.

0.320 (8.13)

0.290 (7.37)
(NOTE 4)

0.175 (4.45)

0.145 (3.68)

0.035 (0.89)

0.020 (0.51)

BSC

0.300 (7.62)
(NOTE 4)

0.015 (0.38)

0.010 (0.25)

0.395 (10.03)

0.300 (7.62)

SA5212ATransimpedance amplifier (140MHz)

Philips Semiconductors Product specification

SA5212ATransimpedance amplifier (140MHz)

Data sheet status

Data sheet
status

Objective
specification

Preliminary
specification

Product
specification

Product
status

Development

Qualification

Production

Definition

This data sheet contains the design target or goal specifications for product development.
Specification may change in any manner without notice.

This data sheet contains preliminary data, and supplementary data will be published at a later date.
Philips Semiconductors reserves the right to make chages at any time without notice in order to
improve design and supply the best possible product.

This data sheet contains final specifications. Philips Semiconductors reserves the right to make
changes at any time without notice in order to improve design and supply the best possible product.

[1]

[1] Please consult the most recently issued datasheet before initiating or completing a design.

Definitions

Short-form specification — The data in a short-form specification is extracted from a full data sheet with the same type number and title. For

detailed information see the relevant data sheet or data handbook.
Limiting values definition — Limiting values given are in accordance with the Absolute Maximum Rating System (IEC 134). Stress above one

or more of the limiting values may cause permanent damage to the device. These are stress ratings only and operation of the device at these or
at any other conditions above those given in the Characteristics sections of the specification is not implied. Exposure to limiting values for extended
periods may affect device reliability.

Application information — Applications that are described herein for any of these products are for illustrative purposes only. Philips
Semiconductors make no representation or warranty that such applications will be suitable for the specified use without further testing or
modification.

Disclaimers

Life support — These products are not designed for use in life support appliances, devices or systems where malfunction of these products can

reasonably be expected to result in personal injury . Philips Semiconductors customers using or selling these products for use in such applications
do so at their own risk and agree to fully indemnify Philips Semiconductors for any damages resulting from such application.

Right to make changes — Philips Semiconductors reserves the right to make changes, without notice, in the products, including circuits, standard
cells, and/or software, described or contained herein in order to improve design and/or performance. Philips Semiconductors assumes no
responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright, or mask work right to these
products, and makes no representations or warranties that these products are free from patent, copyright, or mask work right infringement, unless
otherwise specified.

Philips Semiconductors
811 East Arques Avenue
P.O. Box 3409
Sunnyvale, California 94088–3409
Telephone [phone]

 Copyright Philips Electronics North America Corporation 1998

All rights reserved. Printed in U.S.A.

Date of release: 10-98

Document order number: 9397 750 04625




1998 Oct 07

20