Datasheet RT8202-DS8202AB-05 Datasheet

Page 1

Single Synchronous Buck Controller

RT8202/A/B

General Description

The constant on-time PWM control sche me handles wide input/output voltage ratios with ea se and provides 100ns “instant-on” response to load tran sients while maintaining a relatively constant switching frequency .

The RT8202/A/B achieves high efficiency at a reduced cost by eliminating the current sense resistor found in traditional current mode PWMs. Efficiency is further enhanced by its ability to drive very large synchronous rectifier MOSFET s. The buck conversion allows this device to directly step down high voltage batteries for the highest possible efficiency . The RT8202/A/B is intended f or CPU core, chipset, DRAM, or other low voltage supplies as low as 0.75V. RT8202 is available in WQFN-16L 4x4, RT8202A is available in WQF N-16L 3x3 and R T8202B is available in WQF N-14L 3.5x3.5 pack ages.

Ordering Information

RT8202/A/B

Package Type QW : WQFN-16L 4x4 (W-Type) (RT8202) QW : WQFN-16L 3x3 (W-Type) (RT8202A) QW : WQFN-14L 3.5x3.5 (W-Type) (RT8202B)

Lead Plating System P : Pb Free G : Green (Halogen Free and Pb Free)

Note : Richtek products are :

` RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

` Suitable for use in SnPb or Pb-free soldering processes.

Marking Information

For marking information, contact our sales re presentative directly or through a Richtek distributor located in your area.

Features

Ultra-High Efficiency

z Resistor Programmable Current Limit by Low Side

Sense (Lossless Limit) or Sense Resistor

DS(ON)

(High Accuracy)

z Quick Load Step Response within 100ns

z 1% V

z Adjustable 0.75V to 3.3V Output Range

z 4.5V to 26V Battery Input Range

z Resistor Programmable Frequency

z Over/Under Voltage Protection

z 2 Steps Current Limit During Soft-Start

z Drives Large Synchronous-Rectifier FET s

z Power Good Indicator

z RoHS Compliant and 100% Lead (Pb)-Free

Accuracy over Line and Load

OUT

Applications

z Notebook Computers z CPU Core Supply z Chipset/RAM Supply a s Low as 0.75V

Pin Configurations

(TOP VIEW)

TON

EN/DEM

BOOT

13141516

UGATE

GND

8765

PGND

141

12 11 10

LGATE

BOOT

13 12 11

PGND

PHASE OC VDDP

UGATE PHASE OC VDDP

VOUT

1 2

VDD

PGOOD

GND

3 4

WQFN-16L 4x4/WQFN-16L 3x3

EN/DEM

TON

VOUT

VDD

PGOOD LGATE

GND

4 5

GND

WQFN-14L 3.5x3.5

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RT8202/A/B

Typical Application Circuit

T V

N N

Functional Pin Description

Pi n No.

RT8202/A RT8202B

1 3 VOU T

2 4 VDD

3 5 FB

4 6 PGO OD

5, 14 -- NC No In ternal Connection.

Exposed Pad

(17)

Exposed Pad

(15)

7 8 PGN D Power Groun d.

8 9 LGATE

9 10 VDD P

10 11 OC

11 12 PHASE

12 13 UGATE

13 14 BOOT

15 1 EN/DEM

16 2 TON

Pin Name Pin Function

VOUT Sense Input. Conn ect to the output of PWM converter. VOUT is an input of the PWM con trol ler . Analog Supply Voltage Input for the internal analog integrated circuit. B y pa s s to GN D with a 1 μF ceramic capacito r. VOUT Feedback Input. Connect FB to a resistor voltage divider from VOUT to GND to adjust the output from 0.75V to 3.3 V. Power Good Signal Open-Drain Output of PWM Converter. This pin will be pulled high when the output voltage is within the target range.

GND

Ground for Analog Circuitry. The expos ed pad must be sol der ed t o a large

PCB and connected to GND for maximum power dissipation.

Low side N-MOSFET Gate-Drive Output for PWM. This pin swings betw een G ND and VDDP .

VDDP is th e gate driver supply for the external MO SFETs. Bypass to GND with a 1μF ceramic capacitor .

PWM Current Limit Set ting and sense. Con nect a resistor betwe en OC to PHASE for c urre n t li mit sett ing . Inductor Connection. This pin is not only the zero-current-sense input for the PWM converter, but also the UGATE high side gate driver return. High Side N-MOSFET Floating Gate-Driver Output for the PWM con ver te r. This pi n swi ngs betw een PHASE and BOOT. Boost Capacitor Connection for PWM Converter. Connect an external ceramic capacitor to PHAS E and an ext er nal diode to VDDP. PWM Enable and Operation Mode Selection Input. Connect to VDD for diode-emulation mode, connect to GND for shutdown mode and floating the pi n for CCM mode. VIN Sense Input. Connect to VIN through a resistor. TON is an input of the P WM controller.

C 0

C 2

8 k

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Function Block Diagram

TRIG

VOUT

TON

70% V

0.75V V

REF

SS Timer

SS(Internal)

115% V

VDD

GND

On-time

Compute

1-SHOT

90% V

Latch

S1 Q

Latch

S1 Q

REF

Thermal

Shutdown

Comp

Min. T

OFF

QTRIG

1-SHOT

Emulation

Diode

RT8202/A/B

BOOT

DRV

20uA

UGATE PHASE

VDDP LGATE PGND

EN/DEM

PGOOD

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RT8202/A/B

Absolute Maximum Ratings (Note 1)

z Input V oltage, TON to GN D ---------------------------------------------------------------------------------------------- −0.3V to 32V z BOOT to GND -------------------------------------------------------------------------------------------------------------- −0.3V to 38V z PHASE to BOOT ---------------------------------------------------------------------------------------------------------- −6V to 0.3V z V DD, V DDP, VOUT, EN/DEM, FB, PGOOD to GN D -------------------------------------------------------------- −0.3V to 6V z UGA TE to PHASE -------------------------------------------------------------------------------------------------------- −0.3V to 6V z OC to GND ------------------------------------------------------------------------------------------------------------------ −0.3V to 32V z LGA TE to GN D ------------------------------------------------------------------------------------------------------------- −0.3V to 6V z PGND to GND -------------------------------------------------------------------------------------------------------------- −0.3V to 0.3V z Power Dissipation, P

WQFN-16L 4x4 ------------------------------------------------------------------------------------------------------------ 1.852W WQFN-16L 3x3 ------------------------------------------------------------------------------------------------------------ 1.471W WQFN-14L 3.5x3.5 ------------------------------------------------------------------------------------------------------- 1.667W

z Package Thermal Re sistance (Note 2)

WQF N-16L 4x4, θJA------------------------------------------------------------------------------------------------------- 54°C/W WQFN-16L 4x4, θJC------------------------------------------------------------------------------------------------------ 7°C/W WQF N-16L 3x3, θJA------------------------------------------------------------------------------------------------------- 68°C/W WQFN-16L 3x3, θJC------------------------------------------------------------------------------------------------------ 7.5°C/W WQF N-14L 3.5x3.5, θJA-------------------------------------------------------------------------------------------------- 60°C/W WQFN-14L 3.5x3.5, θJC------------------------------------------------------------------------------------------------- 7°C/W

z Lead Te mperature (Soldering, 10 sec.)------------------------------------------------------------------------------- 260°C z Junction T emperature ---------------------------------------------------------------------------------------------------- 150°C z Storage T emperature Range -------------------------------------------------------------------------------------------- −65°C to 150°C z ESD Susceptibility (Note 3)

HBM (Human Body Mode) ---------------------------------------------------------------------------------------------- 2kV MM (Ma chine Mode)------------------------------------------------------------------------------------------------------ 200V

@ TA = 25°C

Recommended Operating Conditions (Note 4)

z Input Voltage, V z Supply Voltage, V z Junction T emperature Range-------------------------------------------------------------------------------------------- z Ambient T emperature Range--------------------------------------------------------------------------------------------

---------------------------------------------------------------------------------------------------------- 4.5V to 26V

, V

---------------------------------------------------------------------------------------------- 4.5V to 5.5V

DDP

−40°C to 125°C

−40°C to 85°C

Electrical Characteristics

= V

PWM Controller

Quiescent Supply Current VDD + VDDP, FB = 0.8V -- -- 1250 μA TON Operating Current R

Shutdown Current I

DDP

= 5V, V

= 15V, V

= 1.25V, EN/DEM = VDD, R

OUT

= 1MΩ, T

TON

= 25°C, unless otherwise specified)

Parameter Symbol Test Conditions Min Typ Max Unit

= 1M -- 15 -- μA

TON

VD D + VDDP - - 1 10 μA TON -- 1 5 μA

SHDN

EN/DEM = 0V −10 −1 -- μA

To be continued

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RT8202/A/B

Parameter Symbol Test Conditions Min Typ Max Unit

FB Reference Voltage VFB V

= 4.5 to 5.5V 0.742 0.75 0.758 V

FB Input Bias Curr ent FB = 0.75V −1 0.1 1 μA Output Voltage Range V

On-Time VIN = 15V, V

0.75 -- 3.3 V

OUT

= 1.25V, R

OUT

= 1M 267 334 401 ns

TON

Minimum Off-Time 250 400 550 ns V

Shutdown Discharge

OUT

Resistance

EN/DEM = GND -- 2 0 -- Ω

Current Sensing

ILI M Source Cur ren t LGATE = High 18 20 22 μA Current Comparator Offset GND − OC −10 -- 10 mV

Cur ren t Limit Se tting Range R

2.5 -- 10 kΩ

ILIM

Zero Cro ss ing Thre shold GND − PHASE, EN/DEM = 5V −10 -- 5 mV

Fa ult Pr ot ec t i on

Cur ren t Limit Se nse Voltage

RILIM

GND − PH ASE, R GND − PH ASE, R

= 2.5k 35 50 65 mV

ILIM

= 10k 170 200 230 mV

ILIM

Output UV Threshold 60 70 80 % OVP Threshold

W ith re spec t to erro r com para t or threshold

10 15 20 % OV Fault Delay FB forced above OV threshol d - - 20 -- μs VDD UVLO Th re s hold

Soft-Start Ramp Time

Rising edge, Hysteresis = 20mV, PWM disabled below this level

From EN high to internal V

REF

reac h

0.71V (0Æ95%)

4.1 4.3 4.5 V

-- 1.35 -- ms

UV Blank Time From EN signal going high -- 3.1 -- ms Thermal Shutdown -- 155 -- °C

Thermal Shutdown Hysteresis

-- 10 -- °C

Driver O n-Resi st ance

UG ATE Dr iv e r Pu ll Up BOOT − PHAS E = 5V -- 1.5 5 Ω UG ATE Dr iv e r Si nk R

UGATEsk

BOOT − PH ASE = 5V -- 1. 5 5 Ω LGATE Dri ver Pull Up LGATE, High St at e (S our ce) -- 1 .5 5 Ω LGATE Dri ver Pull Down LGA TE , Low S tate (Sink) -- 0.6 2.5 Ω

UGATE Driver Source/Sink Current

LGATE Dri ver Sour ce Current

LGATE forced to 2.5V -- 1 -- A

UGATE − PH ASE = 2.5V, BOOT − PHAS E = 5V

-- 1 -- A

LGATE Dri ver Sink Curr ent LGA TE forc ed to 2.5V -- 3 -- A

LGATE Rising (PHASE = 1.5V) -- 30 - -

Dead Time

UGATE Ris ing -- 30 --

To be continued

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RT8202/A/B

Parameter Symbol Test Conditions Min Typ Max Unit

Logic I/O

EN/DE M Logic Low V oltage -- -- 0.8 V EN/DE M Logic Hig h Voltage 2.9 -- -- V EN/DE M Float ing V oltage

Logic In put Cur rent

EN/D EM Open -- 2 - - V EN/DEM = VDD -- 1 5

μA

EN/DEM = 0 −5 −1 --

PGO OD (u pper side thresh ol d de ci de by OV th reshold)

Measured at FB, with respect to

Trip Thr eshold (Falli ng)

reference, no load.

−13 −10 −7 %

Hysteresis = 3%

Fault Propagation Delay Output Low Voltage I

Falling edge, FB forced below PGOOD trip threshold

= 1mA -- -- 0.4 V

SINK

-- 2.5 -- μs

Leakage Current High state, forced to 5.0V -- -- 1 μA

Note 1. Stresses listed as the above “Absolute Maximum Ratings” may cause permanent damage to the device. These are for

stress ratings. Functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may remain possibility to affect device reliability.

Note 2. θ

Note 3. Devices are ESD sensitive. Handling precaution is recommended. Note 4. The device is not guaranteed to function outside its operating conditions.

is measured in the natural convection at TA = 25°C on a high effective four layers thermal conductivity test board of

JEDEC 51-7 thermal measurement standard. The case point of θ

is on the expose pad of the package.

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Typical Operating Characteristics

RT8202/A/B

Efficiency vs. Output Current

100

90 80 70 60 50 40

Eff iciency ( % )

30 20 10

0.001 0.01 0.1 1 10

DEM

PWM

Output Current (A)

Efficiency vs. Output Current

100

90 80 70 60 50 40

Efficiency (%)

30 20 10

0.001 0.01 0.1 1 10

DEM

PWM

VIN = 12V

Output Current (A)

VIN = 8V

Switching Frequency vs. Output Current

300 275 250 225 200 175 150 125 100

75 50

Swit ching Frequency (kHz)

0.001 0.01 0.1 1 10

PWM

DEM

Output Current (A)

Switching Frequency vs. Output Current

300 275 250 225 200 175 150 125 100

75 50

Swit ching Frequency (kHz)

0.001 0.01 0.1 1 10

PWM

DEM

Output Current (A)

VIN = 8V

VIN = 12V

Efficiency vs. Output Current

100

90 80 70 60 50 40

Efficiency (%)

30 20 10

0.001 0.01 0.1 1 10

DEM

PWM

VIN = 24V

Output Current (A)

Switching Frequency vs. Output Current

300 275 250 225 200 175 150 125 100

75 50

Switching Frequency (kHz)

0.001 0.01 0.1 1 10

PWM

DEM

Output Current (A)

VIN = 24V

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RT8202/A/B

400 390 380 370 360 350 340 330 320 310

Standby Cur rent (uA)

300 290 280

OUT

(1V/Div)

Standby Current vs. Input Voltage

EN = 5V, No Loa d

7 9 11 13 15 17 19 21 23 25

Input Volt age (V)

Power On from EN

PWM-Mode

3.0

2.5

2.0

1.5

1.0

Shutdown Current (uA)

0.5

0.0

OUT

(1V/Div)

Shutdown Current vs. Input Voltage

EN = GND, No Load

7 9 11 13 15 17 19 21 23 25

Inpu t Voltage (V)

Power On from EN

DEM-Mode

PHASE

(10V/Div)

EN/DEM

(2V/Div) PGOOD

(2V/Div)

OUT

(1V/Div)

EN/DEM

(2V/Div)

UGATE

(20V/Div)

LGATE

(5V/Div)

= 12V, EN = Floating, No Load

Time (800μs/Div)

Power Off from EN

= 12V, EN = Floating, No Load

PHASE

(10V/Div)

EN/DEM

(5V/Div) PGOOD

(2V/Div)

OUT_ac

(100mV/Div)

LOAD

(5A/Div)

UGATE

(20V/Div)

LGATE

(5V/Div)

= 12V, EN = 5V, No Load

Time (800μs/Div)

Load Transient Response

OUT

= 12V, EN = Floating, I

= 0A to 6A

OUT

Time (4ms/Div)

Time (10μs/Div)

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RT8202/A/B

OUT

(1V/Div)

UGATE

(10V/Div)

LGATE

(5V/Div)

OUT

(1V/Div)

LOAD

(10A/Div)

OVP

= 12V, EN = 5V, No Load

Time (40μs/Div)

Power On in Short Condition

OUT

(1V/Div)

LOAD

(20A/Div)

UGATE

(20V/Div)

LGATE

(5V/Div)

UVP

= 12V, EN = Floating, No Load

Time (20μs/Div)

UGATE

(20V/Div)

LGATE

(5V/Div)

= 12V, EN = Floating, V

Time (800μs/Div)

OUT

Short

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RT8202/A/B

Application Information

The RT8202/A/B PWM controller provides high efficiency , excellent transient response, a nd high DC output a ccuracy needed for stepping down high voltage batteries to generate low voltage CPU core, I/O, and chipset RAM supplies in notebook computers. Richtek Mach ResponseTM technology is specifically designed for providing 100ns “instant-on” response to load steps while maintaining a relatively constant operating frequency a nd inductor operating point over a wide range of input voltages. The topology circumvents the poor load transient timing problems of fixed-frequency current mode PWMs while avoiding the problems caused by widely varying switching frequencies in conventional constant-on-ti me and constantoff-time PWM schemes. The DRVTM mode PWM modulator is specifically designed to have better noise immunity for such a single output application.

PWM Operation

The Mach Response

DRVTM mode controller relies on

the output filter capacitor's effective series resistance (ESR) to act as a current sense resistor, so the output ripple voltage provides the PWM ra mp signal. Refer to the function diagra ms of RT8202/A/B, the synchronous high side MOSFET is turned on at the beginning of each cycle. After the internal one-shot timer expires, the MOSFET is turned off. The pulse width of this one shot is determined by the converter's input and output voltages to keep the frequency fairly constant over the input voltage range. Another one-shot sets a minimum off-time (400ns typ.).

On-Time Control (TON)

The on-time one-shot comparator has two inputs. One input monitors the output voltage, while the other input samples the input voltage and converts it to a current. This input voltage proportional current is used to charge an internal on-time capacitor. The on-time is the time required for the voltage on this capacitor to charge from zero volts to V

, thereby making the on-time of the high

OUT

side switch directly proportional to output voltage and inversely proportional to input voltage. The implementation results in a nearly constant switching frequency without the need a clock generator.

And then the switching frequency is : Frequency = V R

is a resistor connected from the input supply (VIN)

TON

/ (VIN x TON)

OUT

to TON pin.

Mode Selection (EN/DEM) Operation

The EN/DEM pin enables the supply. When EN/DEM is tied to VDD, the controller is enabled and operates in diode-emulation mode. When the EN/DEM pin is floating, the RT8202/A/B will operate in forced-CCM mode.

Diode-Emulation Mode (EN/DEM = High)

In diode-emulation mode, RT8202/A/B automatically reduces switching frequency at light-load conditions to maintain high efficiency. This reduction of frequency is achieved smoothly a nd without increasing V

ripple or

OUT

load regulation. As the output current decreases from heavy-load condition, the inductor current is also reduced, and eventually comes to the point that its valley touches zero current, which is the boundary between continuous conduction and discontinuous conduction modes. By emulation the behavior of diodes, the low-side MOSFET allows only partial of negative current when the inductor freewheeling current reach negative. As the loa d current is further decrea sed, it takes longer and longer to discharge the output capacitor to the level than requires the next “ON” cycle. The on-time is kept the same as that in the heavy-load condition. In reverse, when the output current increases from light load to heavy load, the switching frequency increases to the preset value as the inductor current reaches the continuous condition. The tra n sition load point to the light load operation ca n be calculated as follows (Figure 1) :

(V V )

−

LOAD ON

IN OUT

≈×

where TON is On-time.

TON = 3.85p x R

TON

x V

OUT

/ (V

− 0.5)

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Page 11

Slope = (VIN -V

OUT

) / L

L, peak

Load

= i

L, peak

/ 2

RT8202/A/B

L, peak

Load

LIM

Figure 1. Boundary condition of CCM/DEM

The switching waveforms may appear noisy and asynchronous when light loa ding causes diode-emulation operation, but this is a normal operating condition that results in high light-load efficiency . T rade-offs in DEM noise vs. light-load efficiency are made by varying the inductor value. Generally, low inductor values produce a broader efficiency vs. load curve, while higher values result in higher full-load efficiency (assuming that the coil resistance remains fixed) and less output voltage ripple. The disadvantages for using higher inductor values include larger physical size and degrades loa d-transient response (especially at low input voltage levels).

Forced-CCM Mode (EN/DEM = floating)

The low noise, forced-CCM mode (EN/DEM = floating) disables the zero-crossing comparator, which controls the low-side switch on-time. This causes the low side gatedrive waveform to become the complement of the high side gate-drive waveform. This in turn causes the inductor current to reverse at light loads as the PWM loop to maintain a duty ratio V

OUT/VIN

. The benefit of forced-CCM mode is to keep the switching frequency fairly constant, but it comes at a cost : The no-load battery current can be up to 10mA to 40mA, depending on the external MOSFETs.

Current Limit Setting (OCP)

RT8202/A/B has cycle-by-cycle current limiting control. The current limit circuit employs a unique “valley” current sensing algorithm. If the magnitude of the current-sense signal at OC is above the current limit threshold, the PWM is not allowed to initiate a new cycle (Figure 2).

Figure 2. V alley Current-Limit

Current sensing of the RT8202/A/B can be a ccomplished in two ways. Users can either use a current sense resistor or the on-state of the low side MOSFET (R

DS(ON)

). For resistor sensing, a sense resistor is pla ced between the source of low-side MOSFET and PGND (Figure 3(a)). R

sensing is more efficient a nd less expensive (Figure

DS(ON)

3(b)). There is a compromise between current-limit accura cy a nd sense re sistor power dissipation.

PHASE

LGATE

ILIM

LGATE

ILIM

(a) (b)

Figure 3. Current-Sense Methods

In both case s, the R

resistor between the OC pin and

ILIM

PHASE pin sets the over current threshold. This resistor R

is connected to a 20μA current source within the

ILIM

RT8202/A/B which is turned on when the low side MOSFET turns on. When the voltage drop across the sense resistor or low side MOSFET equals the voltage across the R

resistor, positive current li mit will activate.

ILIM

The high side MOSFET will not be turned on until the voltage drop across the sense element (resistor or MOSFET) falls below the voltage across the R

resistor.

ILIM

Choose a current limit resistor by following Equation : R

= I

ILIM

x R

LIMIT

SENSE

/ 20μA

Carefully observe the PC board layout guidelines to ensure that noise and DC errors do not corrupt the current-sense signal seen by OC and PGND. Mount the IC close to the low-side MOSFET and sense resistor with short, direct

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RT8202/A/B

traces, making a Kelvin sense connection to the sense resistor.

MOSFET Gate Driver (UGATE, LGA TE)

The high side driver is designed to drive high current, low R

N-MOSFET(s). When configured as a floating

DS(ON)

driver, 5V bia s voltage is delivered from V DDP supply . The average drive current is proportional to the gate charge at VGS = 5V times switching frequency. The instantaneous drive current is supplied by the flying capacitor between BOOT and PHASE pins.

A dead time to prevent shoot through is internally generated between high side MOSFET off to low side MOSFET on, and low side MOSFET off to high side MOSFET on.

The low side driver is designed to drive high current, low R

N-MOSFET(s). The internal pull-down tra nsistor

DS(ON)

that drives LGATE low is robust, with a 0.6Ω typical on- resistance. A 5V bias voltage is delivered form VDDP supply . The instanta neous drive current is supplied by the flying capa citor between V DDP and PGND.

For high current application s, some combinations of high and low side MOSFETs might be encountered that will cause excessive gate-drain coupling, which can lead to efficiency killing, EMI producing shoot through currents. This is often remedied by adding a resistor in series with BOOT, which increa ses the turn-on ti me of the high side MOSFET without degrading the turn-off time (Figure 4).

+5V

BOOT

UGATE

PHASE

Figure 4. Reducing the UGA TE Rise T ime

Power Good Output (PGOOD)

The power good output is an open-drain output and requires a pull up resistor. When the output voltage is 15% a bove or 10% below its set voltage, PGOOD gets pulled low. It is held low until the output voltage returns to within these tolerances once more. In soft start, PGOOD is actively

held low and is allowed to tra nsition high until soft start is over and the output rea ches 93% of its set voltage. There is a 2.5μs delay built into PGOOD circuitry to prevent false transition.

POR, UVLO and Soft-Start

Power on reset (POR) occurs when VDD rises a bove to approximately 4.3V, the RT8202/A/B will reset the fault latch and preparing the PWM for operation. Below

4.1V

, the V DD under voltage-lockout (UVLO) circuitry

(MIN)

inhibits switching by keeping UGA TE and LGATE low . A built-in soft-start is used to prevent surge current from

power supply input after EN/DEM is enabled. It clamps the ramping of intern al reference voltage which is compared with FB signal. The typical soft-start duration is 1.35ms.

Furthermore, the maximum allowed current limit is segment in 2 steps during 1.35ms period.

Output Over Voltage Protection (OVP)

The output voltage can be continuously monitored for over voltage protection. When the output voltage exceeds 15% of the its set voltage threshold, over voltage protection is triggered and the low side MOSFET is latched on. This activates the low side MOSFET to discharge the output ca p acitor.

RT8202/A/B is latched once OVP is triggered and can only be relea sed by V DD or EN/DEM power on reset. There is 20us delay built into the over voltage protection circuit to prevent false transitions.

Output Under Voltage Protection (UVP)

The output voltage can be continuously monitored for under voltage protection. When the output voltage is less than 70% of its set voltage threshold, under voltage protection is triggered and then both UGA TE and LGA TE gate drivers are forced low . In order to remove the residual charge on the output capacitor during the under voltage period, if PHASE is greater than 1V, the LGA TE is forced high until PHASE is lower than 1V. There is 2.5us delay built into the under voltage protection circuit to prevent false transitions. During soft-start, the UVP will be blanked around 3.1ms.

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Page 13

)

Output V oltage Setting (FB)

RT8202/A/B

The output voltage can be adjusted from 0.75V to 3.3V by setting the feedback resistor R1 a nd R2 (Figure 5). Choose R2 to be approxi mately 10kΩ, and solve f or R1 using the equation:

⎡⎤

V = V 1

OUT FB

⎛⎞

×+

⎜⎟

⎢⎥

⎝⎠

⎣⎦

where VFB is 0.75V. Note that in order for the device to regulate in a controlled

manner , the ripple content at the feedba ck pin, VFB, should be approxi mately 15mV at minimum V no smaller than 10mV. If V

at minimum V

ripple

, and worst ca se

BAT

is less than 15mV, the above component values should be revisited in order to improve this. Quite often a small cap acitor , C1, is required in parallel with the top feedbac k resistor, R1, in order to ensure that VFB is large enough. The value of C1 can be calculated a s follows, where R2 is the bottom feedback resistor .

Firstly calculating the value of Z1 required :

Z1 = V 0.015

×−Ω

()

0.015

ripple_VBAT(MIN)

OUT

UGATE PHASE

BOOT VOUT

GND

Figure 5. Setting The Output Voltage

For application that output voltage is higher than 3.3V, user can also use a voltage divider to keep VOUT pin voltage within 0.75V to 2.8V as shown in Figure 6. For this case, T

If R < 2M then T = 3.85p

TON ON

If R 2M then T = 3.55p

TON ON

Where R

can be determined as below :

Ω×

≥Ω ×

is TON set resistor and the V

TON

TON OUT_FB

V0.5

R V

TON OUT_FB

V0.4

−

OUT_FB

is the output signal of resistor divider. Since the switching frequency is

Secondly calculating the value of C1 required to achieve

F =

this :

−

(

C1 = F

Z1 R1

××

SW_VBAT(MIN)

Finally using the equation a s follows to verify the value of V

V = V

FB_VBAT(MIN) ripple_VBAT(MIN)

For a given switching frequency , we can obtain the R a s below

If R < 2M then

TON

R =

TON

If R 2M then

TON

R =

TON

⎡⎤ ⎢⎥ ⎢⎥

⎢⎥

R2+

⎢⎥

⎢⎥ ⎣⎦

where V minimum V

sw_VBAT(MIN)

FB_VBAT(MIN)

BAT

+×× ×

ripple_VBAT(MIN)

;

BAT

is the switching frequency in minimum V

is the ripple voltage into FB pin in minimum

2f C1

SW_VBAT(MIN)

is the output ripple voltage in

BAT

;

OUT

VT×

IN ON

−

V0.5V

OUT OUT

V V F 3.85p

IN OUT_FB S

≥Ω

−

V0.4V

OUT OUT

V V F 3.55p

IN OUT_FB S

VIN

UGATE PHASE

BOOT

VOUT

GND

TON

××

OUT_FB

TON

OUT

Figure 6. Output Voltage Setting for V

OUT

> 3.3V

Application

DS8202/A/B-05 April 2011 www.richtek.com

Page 14

RT8202/A/B

Output Inductor Selection

The switching frequency (on-time) and operating point (% ripple or LIR) determine the inductor value as f ollows :

T(V - V)

ON IN OUT

L =

IR LOAD(MAX)

Find a low pass inductor having the lowest possible DC resistance that fits in the allowed dimen sions. Ferrite cores are often the best choice, although powdered iron is inexpensive and ca n work well at 200kHz. The core must be large enough and not to saturate at the pea k inductor current (I

= I

PEAK

) :

PEAK

LOAD(MAX)

+ [(LIR / 2) x I

LOAD(MAX)

]

Output Capacitor Selection

The output filter ca pacitor must have ESR low enough to meet output ripple and loa d transient requirement, yet have high enough ESR to satisfy stability requirements. Also, the cap acitance value must be high enough to a bsorb the inductor energy going from a full load to no load condition without tripping the OVP circuit.

Do not put high value ceramic capacitors dire ctly across the outputs without taking precautions to ensure sta bility . Large ceramic capacitors can have a high ESR zero frequency and cause erratic and unstable operation. However, it is easy to add sufficient series resistance by placing the ca pacitors a couple of inche s downstream from the inductor and connecting V

or FB divider close to

OUT

the inductor. There are two related but distinct ways including double

pulsing and feedback loop instability to identify the unstable operation.

Double-pulsing occurs due to noise on the output or because the ESR is too low that there is not enough voltage ramp in the output voltage sign al. The “fools” the error comparator into triggering a new cycle immediately after 400ns minimum off-time period ha s expired. Doublepulsing is more annoying tha n harmful, resulting in nothing worse than increased output ripple. However, it may indicate the possible presence of loop instability, which is caused by insufficient ESR.

For CPU core voltage converters and other applications where the output is subject to violent load transient, the output capacitor's size depends on how much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag due to f inite cap acita nce :

ESR

≤

P-P

LOAD(MAX)

In non-CPU applications, the output capacitor's size depends on how much ESR is needed to maintain at an accepta ble level of output voltage ripple :

ESR

≤

P-P

IR LOAD(MAX)

Organic semiconductor ca pa citor(s) or specially polymer cap acitor(s) are recommended.

Output Capacitor Stability

Stability is determined by the value of the ESR zero relative to the switching frequency . The point of instability is given by the following equation :

f =

ESR

2 ESR C 4

×× ×

OUT

≤

Loop instability ca n result in oscillation at the output after line or load perturbations that can trip the over voltage protection latch or cause the output voltage to fall below the tolerance limit.

The easiest method for stability checking is to apply a very zero-to-max load tran sient and carefully observe the output-voltage-ripple envelope for overshoot a nd ringing. It helps to simultaneously monitor the inductor current with AC probe. Do not allow more tha n one ringing cycle after the initial step-response under- or over-shoot.

Thermal Considerations

For continuous operation, do not exceed absolute maximum operation junction temperature.

The maximum power dissipation depends on the thermal resistance of IC package, PCB layout, the rate of surroundings airflow and temperature difference between junction to ambient. The maximum power dissipation ca n be calculated by following formula :

P Where T

temperature 125°C, T

= ( T

D(MAX)

J(MAX)

is the junction to ambient thermal resistance.

- TA ) / θ

J(MAX)

is the maximum operation junction

is the ambient temperature a nd the

DS8202/A/B-05 April 2011www.richtek.com

Page 15

RT8202/A/B

For recommended operating conditions specification of RT8202/A/B, where T

is the maximum junction

J(MAX)

temperature of the die (125°C) and TA is the maximum ambient temperature. The junction to ambient thermal resistance θ

is layout dependent. For WQFN-16L 3x3

packages, the thermal resistance θJA is 68°C/W on the standard JEDEC 51-7 four layers thermal test board. For WQFN-14L 3.5x3.5 pack age, the thermal resistance θ

is 60°C/W on the standard JEDEC 51-7 f our layers thermal test board. The maximum power dissipation at TA = 25°C can be calculated by following f ormula :

= (125°C − 2°C) / (68°C/W) = 1.471W for

D(MAX)

WQF N-16L 3x3 pa ckages P

= (125°C − 25°C) / (54°C/W) = 1.852W for

D(MAX)

WQF N-16L 4x4 pa ckages P

= (125°C − 25°C) / (60°C/W) = 1.667W for

D(MAX)

WQF N-14L 3.5x3.5 packages The maximum power dissipation depends on operating

ambient temperature for fixed T

and thermal

J(MAX)

resistance θJA. For RT8202/A/B packages, the Figure 7 of derating curves allows the designer to see the effect of rising ambient temperature on the maximum power allowed.

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

Maximum Power Dissipation (W)

0.0 0 25 50 75 100 125

WQFN -16L 3x3

WQFN -16L 4x4

Ambient Temperature (°C)

Four Layer PCB

WQFN -14L 3.5x3.5

Layout Considerations

Layout is very important in high frequency switching converter design. If designed improperly , the PCB could radiate excessive noise and contribute to the converter instability. Certain points must be considered before starting a layout for RT8202/A/B.

` Connect RC low pa ss filter from V DDP to V DD, 1uF a nd

10Ω are recommended. Place the filter ca pa citor close to the IC.

` Keep current limit setting network a s close a s possible

to the IC. Routing of the network should avoid coupling to high voltage switching node.

` Connections from the drivers to the respective gate of

the high side or the low side MOSFET should be as short as possible to reduce stray inductance.

` All sensitive analog traces and components such as

VOUT, FB, GND, EN/DEM, PGOOD, OC, VDD, and TON should be placed away from high voltage switching nodes such as PHASE, LGATE, UGATE, or BOOT nodes to avoid coupling. Use internal layer(s) a s ground plane(s) and shield the feedba ck trace from power tra ces and components.

` Current sense connections must always be made using

Kelvin connections to ensure an accurate signal, with the current limit resistor located at the device.

` Power sections should connect directly to ground

plane(s) using multiple vias as required for current handling (including the chip power ground connection s). Power components should be placed to minimize loops and reduce losses.

Figure 7. Derating Curves f or RT8202/A/B Pack ages

DS8202/A/B-05 April 2011 www.richtek.com

Page 16

RT8202/A/B

Outline Dimension

SEE DETAIL A

1 2

DETAIL A

Pin #1 ID a nd T ie Bar Mark Option s

Note : The configuration of the Pin #1 identifier is optional, but must be located within the zone indicated.

Dimensions In Millimeters Dimensions In Inches

Symbol

Min Max Min Max

A 0.700 0.800 0.028 0.031 A1 0.000 0.050 0.000 0.002 A3 0.175 0.250 0.007 0.010

b 0.180 0.300 0.007 0.012

D 2.950 3.050 0.116 0.120

D2 1.300 1.750 0.051 0.069

E 2.950 3.050 0.116 0.120 E2 1.300 1.750 0.051 0.069

e 0.500 0.020

L 0.350 0.450

0.014 0.018

W-Type 16L QFN 3x3 Package

DS8202/A/B-05 April 2011www.richtek.com

Page 17

RT8202/A/B

SEE DETAIL A

1 2

DETAIL A

Pin #1 ID a nd T ie Bar Mark Option s

Note : The configuration of the Pin #1 identifier is optional, but must be located within the zone indicated.

Dimensions In Millimeters Dimensions In Inches

Symbol

Min Max Min Max

A 0.700 0.800 0.028 0.031 A1 0.000 0.050 0.000 0.002 A3 0.175 0.250 0.007 0.010

b 0.250 0.380 0.010 0.015

D 3.950 4.050 0.156 0.159

D2 2.000 2.450 0.079 0.096

E 3.950 4.050 0.156 0.159 E2 2.000 2.450 0.079 0.096

e 0.650 0.026

L 0.500 0.600

0.020 0.024

W-Type 16L QFN 4x4 Package

DS8202/A/B-05 April 2011 www.richtek.com

Page 18

RT8202/A/B

DETAIL A

Pin #1 ID a nd T ie Bar Mark Option s

Note : The configuration of the Pin #1 identifier is optional, but must be located within the zone indicated.

Dimensions In Millimeters Dimensions In Inches

Symbol

Min Max Min Max

A 0.700 0.800 0.028 0.031 A1 0.000 0.050 0.000 0.002 A3 0.175 0.250 0.007 0.010

b 0.180 0.300 0.007 0.012

D 3.400 3.600 0.134 0.142

D2 1.950 2.150 0.077 0.085

E 3.400 3.600 0.134 0.142 E2 1.950 2.150 0.077 0.085

e 0.500 0.020

e1 1.500 0.060

L 0.300 0.500

0.012 0.020

W-Type 14L QFN 3.5x3.5 Package

Richtek Technology Corporation

Headquarter 5F, No. 20, Taiyuen Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Fax: (8863)5526611

Information that is provided by Richtek Technology Corporation is believed to be accurate and reliable. Richtek reserves the right to make any change in circuit design, specification or other related things if necessary without notice at any time. No third party intellectual property infringement of the applications should be guaranteed by users when integrating Richtek products into any application. No legal responsibility for any said applications is assumed by Richtek.

Richtek Technology Corporation

Taipei Office (Marketing) 5F, No. 95, Minchiuan Road, Hsintien City Taipei County, Taiwan, R.O.C. Tel: (8862)86672399 Fax: (8862)86672377 Email: [email protected]

DS8202/A/B-05 April 2011www.richtek.com