Datasheet RF2516, RF2516PCBA Datasheet (RF Micro Devices)

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11

TRANSCEIVERS

Preliminary

Product Description

Ordering Information

Typical Applications

Features

Functional Block Diagram

RF Micro Devices, Inc.
7625 Thorndike Road
Greensboro,NC 27409, USA

Tel [phone]

Fax [phone]

http://www.rfmd.com

Optimum Technology Matching® Applied

Si BJT GaAs MESFETGaAs HBT
Si Bi-CMOS

ü

SiGe HBT

Si CMOS

Prescaler

32/64

Phase

Detector &

Charge Pump

Lock

Detect

LOOP FLT

14

16

DIV CTRL

DC

Bias

5TX OUT

2

OSC E

1

OSC B

15

LD FLT

MOD IN

8

RESNTR+13RESNTR-

12

3PD

RF2516

VHF/UHF TRANSMITTER

• 315/433MHz Band Systems

• Local Oscillator Source

• Part 15.231 Applications

• Remote Keyless Entry

• Wireless Security Systems

• AM/ASK/OOK Transmitter

The RF2516 is a monolithic integrated circuit intended for
use as a low-cost AM/ASK transmitter. The device is pro­vided in a 16-pin QSOP-16 package and is designed to
provide a phased locked frequency source for use in local
oscillator or transmitter applications. The chip can be
used in applicationsin the North American and European
VHF/UHF bands. The integrated VCO, phase detector,
prescaler, and reference oscillator transistor require only
the addition of an external crystal to provide a complete
phase-locked loop. In addition to the standard power­down mode, the chip also includes an automatic lock­detect feature that disables the transmitter output when
the PLL is out-of-lock.

• Fully Integrated PLL Circuit

• Integrated VCO and Reference Oscillator

• 2.0V to 3.6V Supply Voltage

• Low Current and Power Down Capability

• 100MHz to 500MHz Frequency Range

• Out-of-Lock Inhibit Circuit

RF2516 VHF/UHF Transmitter
RF2516 PCBA Fully Assembled EvaluationBoard

11

Rev A10 010613

0.157

0.150

0.196

0.189

0.2440

0.2284

0.0688

0.0532

0.050

0.016

0.0098

0.0075

8° MAX

0°MIN

NOTES:

1. Shaded lead is Pin 1.

2. All dimensions are excluding mold flash.

3. Lead coplanarity - 0.005 with respect to datum "A".

0.012

0.008

0.025

-A-

0.0098

0.0040

Package Style: QSOP-16

Preliminary

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Absolute Maximum Ratings

Parameter Rating Unit

Supply Voltage -0.5 to +3.6 V

DC

Power Down Voltage(VPD) -0.5toV

CC

V

MOD IN -0.5 to 1.1 V
Operating Ambient Temperature -40 to +85 °C
Storage Temperature -40 to +150 °C

Parameter

Specification

Unit Condition

Min. Typ. Max.

Overall

T=25°C, VCC=2.8V, Freq=433MHz,
R

MODIN

=3kΩ

Frequency Range 100 to 500 MHz
Modulation AM/ASK
Modulation Frequency 1 MHz
Incidental FM 15 kHz p-p
Output Power +8.5 +10 dBm 50Ω load
ON/OFF Ratio 75 dB

PLL and Prescaler

Prescaler Divide Ratio 32/64
VCO Gain, K

VCO

20 MHz/V Frequency and board layout dependent.

PLL Phase Noise -97 dBc/Hz 10kHz Offset, 50kHz loop bandwidth

-102 dBc/Hz 100kHz Offset, 50kHz loop bandwidth
Harmonics -60 dBc With output tuning.
Reference Frequency 17 MHz
Crystal Frequency Spurs -50 dBc 50kHz PLL loop bandwidth
Max CrystalR

S

TBD 35 50 Ω For a typ. 1ms turn-on time.

Max Crystal Motional Inductance 60 mH For a typ. 1 ms turn-on time.
Charge Pump Current 100 µAK

PD

=100µA/2π=0.0159mA/rad

Power Down Control

Power Down “ON” VCC-0.3V V Voltage supplied to the input; device is “ON”
Power Down “OFF” +0.3 V Voltage supplied to the input; device is “OFF”

Control Input Impe dance 100k Ω
Turn On Time 1 2 ms Crystal start-up, 13.57734MHz crystal.
Turn Off Time 1 2 ms

Power Supply

Voltage 2.8 V Specifications

2.0 3.6 V Operating limits

Current Consumption (Avg.) 6 10.5 mA 50% Duty Cycle 10kHz Data applied to the

MOD IN input. R

MODIN

(R10)=3kΩ. Output

power/DC current consumption externally
adjustable by modulation input resistor (see
applicable Application Schematic).

Power Down Current 0 1 uA PD

=0V, MOD IN=0V,DIV CTRL=0V

Caution! ESD sensitive device.

RF Micro Devices believes the furnished information is correct and accurate
at the time of this printing. However, RF Micro Devices reserves the right to
make changes to its products without notice. RF Micro Devices does not
assume responsibility for the use of the described product(s).

Preliminary

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Pin Function Description Interface Schematic

1OSCB

This pin is connected directly to the reference oscillator transistor base.
The intended reference oscillator configuration is a modified Colpitts. A
68pF capacitor should be conne cted between pin 1 and pin 2. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.

2OSCE

This pin is connected directly to the emitter of the reference oscillator
transistor. A 33pF capaci tor should be connected from this pin to
ground. Diodes shown in the interface schematic provide 3kV electro­static discharge (ESD) protection using the human body model.

See pin 1.

3PD

Power Down control for all circuitry.When this pin is a logic “low” all c ir­cuits are turned off. When this pin is a logic “high”, all circuits are oper­ating normally. A “high” is V

CC

. Diodes shown in the interface

schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.

4GND

Ground connection for the TX OUT amp. Keep traces physically short
and connect immediately to ground plane for best performance. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.

5TXOUT

Transmitter output. This output is an open collector and requires a pull­up inductor for bias/matching and a tapped capacitor for matching.

6GND1

Ground connection for the TX output buffer amplifier. Diodes shown in
the interface schematic provide 3kV electrostatic discharge (ESD) pro­tection using the human body model.

See pin 4.

7VCC1

This pin is used to supply bias to the TX buffer amplifier. Diodes shown
in the interface schematic provide 3kV electrostatic discharge (ESD)
protection using the human body model.

8MODIN

AM analog or digital modulation can be imparted to the carrier by an
input to this pin. An external resistor is used to bias the output amplifi­ers through this pin. The voltage at this pin must not exceed 1.1V.
Higher voltages may damage the device. Diodes shown in the interface
schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.

9VCC2

This pin is used to supply DC bias to the VCO, crystal oscillator, pre­scaler, phase detector, and charge pump. An IF bypass capacitor
should be connected directly to this pin and returned to ground. Diodes
shown in the interface schematic provide 3kV electrostatic discharge
(ESD) protection using the human body model.

See pin 7.

10 GND2

Digital PLL ground connection. Diodes shown in the interface sche­matic provide 3kV electrostatic discharge (ESD) protection using the
human body mo del.

See pin 4.

OSC E

V

CC

OSC B

V

CC

PD

GND

V

CC

TX OUT

MOD IN

RF IN

VCC1

V

CC

1k

Ω

MOD IN

TX OUT

V

CC

Preliminary

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Pin Function Description Interface Schematic

11 VREF P

Bias voltage reference pin for bypassing. The bypass capacitor should
be of appropriate size to provide filtering of the reference crystal fre­quency and be connected directly to this pin. Diodes shown in the inter­face schematic provide 3kV electrostatic discharge (ESD) protection
using the h uman body model.

12 RESNTR-

The RESNTR pins are used to supply DC voltage to the VCO, as well
as to tune the center frequency of the VCO. Equal value inductors
should be connected to this pin and pin 13.

13 RESNTR+

See pin 12.

14 LOOP FLT

Output of th e charge pump. An RC network from this pin to ground is
used to establish the PLL bandwidth. Diodes s h own in the interface
schematic provide 3kV electrostatic discharge (ESD) protection using
the human body model.

15 LD FLT

This pin is used to set the threshold of the lock-detect circuit. A shunt
capacitor should be used to set an RC time constant with the on-chip
series 1k resistor. This signal is used to clamp (enable or disable) the
MOD IN circuitry. The time constant should be approximately 10 times
the reference period. Diodes shown in the interface schematic provide
3kV electrostatic discharge (ESD) protection using the human body
model.

16 DIV CTRL

Logic “High” input selects divide-by-64 prescaler. Logic “Low” input
selects divide-by-32 prescaler. Diodes shown in the interface sche­matic provide 3kV electrostatic discharge (ESD) protection using the
human body model.

VREF P

V

CC

RESNTR-RESNTR+

LOOP FLT

4k

Ω

LOOP FLT

V

CC

V

CC

LD FLT

1k

Ω

V

CC

DIV CTRL

Preliminary

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RF2516 Theory of Operation

Introduction

Short range radio devices are becoming commonplace
in today’s environment. The most common examples
are the remote keyless entr y systems popular on many
new cars and trucks, and the ubiquitous garage door
opener. Other applications are emerging with the
growth in home security, automation and the advent of
various remote control applications. Typically these
deviceshave been simplex, or one-way, links.They are
also typically built using surface acoustic wave (SAW)
devices as the frequency control elements. This
approach has been attractive because the SAW
devices have been readily available and a transmitter,
for example, could be built with only a few additional
components. Recently however, RF Micro Devices,
Inc. (RFMD), has introduced several new components
that enable a new class of short-range radio devices
based on the use of crystals and phase-locked loops
for frequency control. These devices are superior in
performance and comparable in cost to the traditional
SAW-based designs. The RF2516 is an example of
such a device. The RF2516 is targeted for applications
such as 315MHz and 433MHz band remote keyless
entry systems and wireless security systems, as well
as other remote control applications.

The RF2516 Transmitter

The RF2516 is a low-cost AM/ASK VHF/UHF transmit­ter designed for applications operating within the fre­quency range of 100MHz to 500MHz. In particular, it is
intended for 315MHz to 433MHz band systems,
remote keyless entry systems, and FCC Part 15.231
periodic transmitters. It can also be used as a local
oscillator signal source. The integrated VCO, phase
detector, prescaler, and reference oscillator require
only the addition of an external crystal to provide a
complete phase-locked loop. In addition to the stan­dard power-down mode, the chip also includes an
automatic lock-detect feature that disables the trans­mitter output when the PLL is out-of-lock.

Thedeviceismanufacturedona25GHzSiliconBipo­lar-CMOS process and packaged in an industry stan­dard SSOP-16 plastic package. This, combined with
the low external parts count, enables the designer to
achieve small-footprint, high-performance, low-cost
designs.

The RF2516 is designed to operate from a supply volt­age ranging from 2.0V to 3.6V, accommodating
designs using three NiCd battery cells, two AAA flash­light cells, or a lithium button battery. The device is

capable of providing up to +10dBm output power into a
50 Ω load, and is intended to comply with FCC require­ments for unlicensed remote control transmitters. ESD
protection is provided on all pins except VCO and TX
OUT.

While this device is intended for OOK operation, it is
possible to use narrowband FM. This is accomplished
by modulating the r eference oscillator rather than
applying the data to the MOD IN input pin. The MOD
INpinshouldbetiedhightocausethedevicetotrans­mit. The deviation will be set by pulling limits of the
crystal. Deviation sufficient for the transmission of
voice and other low data rate signals can therefore be
accomplished. Refer to the Application Schematic in
the data sheet for details.

The RF2516 Functional Blocks

A PLL consists of a reference oscillator, a phase detec­tor, a loop filter, a voltage controlled oscillator (VCO),
and a programmable divider in the feedback path. The
RF2516 includes all of these internally, except for the
loop filter and the reference oscillator’s crystal and two
feedback capacitors.

The reference oscillator is a Colpitts type oscillator.
Pins 1 (OSC B) and 2 (OSC E) provide connections to
a transistor that is used as the reference oscillator. The
Colpitts configuration is a low parts count topology with
reliable performance and reasonable phase noise.
Alternatively, an external signal could be injected into
the base of the transistor. The drive level should, in
either case, be around 500mV

PP

. This level prevents

overdriving the device and keeps the phase noise and
reference spurs to a minimum.

The prescaler divides the VCO frequency by either 64
or 32, using a series of flip-flops, depending upon the
logic level present at the DIV CTRL pin. A high logic
level will select the 64 divisor. A low logic level will
select the 32 divisor. This divided signal is then fed into
the phase detector where it is compared with the refer­ence frequency.

The RF2516 contains an onboard phase detector and
charge pump. The phase detector compares the
phase of the reference oscillator to the phase of the
VCO. The phase detector is implemented using flip­flops in a topology referred to as either “digital phase/
frequency detector” or “digital tri-state comparator”.
The circuit consists of two D flip-flops whose outputs
are combined with a NAND gate which is then tied to

Preliminary

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11

TRANSCEIVERS

the reset on each flip-flop. The outputs of the flip-flops
are also connected to the charge pump. Each flip-flop
output signal is a series of pulses whose frequency is
related to the flip-flop input frequency.

When both inputs of the flip-flops are identical, the sig­nals are both frequency- and phase-locked. If they are
different, they will provide signals to the charge pump
which will either charge or discharge the loop filter, or
enter into a high impedance state. The name “tri-state
comparator” comes from this.

The main benefit of this type of detector is the ability to
correct for errors in both phase and frequency. When
locked, the detector uses phase error for correction.
When unlocked, it uses frequency error for correction.
This type of detector will lock under all conditions.

The charge pump consists of two transistors, one for
charging the loop filter and the other for discharging
the loop filter. Its inputs are the outputs of the phase
detector flip-flops. Since there are two flip-flops, there
are four possible states. If both amplifier inputs are low,
then the amplifier pair goes into a high impedance
state, maintaining the charge on the loop filter. The
state where both inputs are high will not occur. The
other states are either charging or discharging the loop
filter. The loop filter integrates the pulses coming from
the charge pump to create a control voltage for the
voltage controlled oscillator.

The VCO is a tuned differential amplifier with the bases
and collectors cross-coupled to provide positive feed­back and a 360° phase shift. The tuned circuit is
located in the collectors, and is comprised of internal
varactors and external inductors. The designer selects
the inductors for the desired frequency of operation.
These inductors also provide DC bias for the VCO.

The output of the VCO is buffered and applied to the
prescaler circuit, where it is divided by either 32 or 64,
as selected by the designer, and compared to the ref­erence oscillator frequency.

The transmit amplifier is a two-stage amplifier con-
sisting of a driver and an open collector final stage.It is
capable of providing 10dBm of output power into a
50 Ω load while operating from a 3.6V power supply.

The lock-detect circuitry connects to the output of the
phasedetectorcircuitryandisusedtodisablethe
transmitter when the VCO is not phase-locked to the
reference oscillator. This is necessary to avoid
unwanted out-of-band transmission and to provide
compliance with regulatory limits during an unlocked
condition.

There are many possible reasons for the PLL not to be
locked. For instance, there is a shor t period during the
start of any VCO in which the VCO begins oscillating
and the reference oscillator builds up to full amplitude.
During this period, the frequency will likely be outside
the authorized band. Typically, the VCO starts much
faster than the reference oscillator. Once both VCO
and reference oscillators are running, the phase detec­tor can start slewing the VCO to the correct frequency,
slowly sliding across 200MHz of occupied spectrum. In
competitive devices, the VCO radiates at full power
under all of these conditions.

The lock protection circuit in the RF2516 is intended to
stabilize quickly after power is applied to the chip, and
to disable the base drive to the transmit amplifier. This
attenuates the output to levels that will be generally
acceptable to regulatory boards as spurious emis­sions. Once the phase detector has locked the oscilla­tors, then the lock circuit enables the MOD IN pin for
transmission of the desired data. There is no need for
an external microprocessor to monitor the lock status,
although that can be done with a low current A/D con­verter in a system micro, if needed. The lock-detect cir­cuitry contains an internal resistor which, combined
with a designer-chosen capacitor for a particular RC
time constant, filters the lock-detect signal. This signal
is then passed through an internal Schmitt tr igger and
used to enable or disable the transmit amplifier.

If the oscillator unlocks, even momentarily, the protec­tion circuit quickly disables the output until the lock is
stable. These unlocks can be caused by low battery
voltage, poor power supply regulation, severe shock of
the crystal or VCO, antenna loading, component fail­ure, or a myriad of unexpectedsingle-point failures.

The RF2516 contains onboard band gap reference
voltage circuitry which provides a stable DC bias over
varying temperature and supply voltages. Additionally,
the device features a power-down mode, eliminating
battery disconnect switches.

Preliminary

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TRANSCEIVERS

Designing With the RF2516

The reference oscillator is built around the onboard
transistor at pins 1 and 2. The intended topology is that
of a Colpitts oscillator. The Colpitts oscillator is quite
common and requires few external components, mak­ing it ideal for low-cost solutions. The topology of this
type of oscillator is as seen in the following figure.

This type of oscillator is a parallel resonant circuit for a
fundamental mode crystal. The transistor amplifier is
an emitter follower and the voltage gain is developed
by the tapped capacitor impedance transformer. The
series combination of C

1

and C2act in parallel with the

input capacitance of the transistor to capacitively load
the crystal.

The nominal capacitor values can be calculated with
the following equations

6

:

and

The load capacitance is usually 32pF. The variable freq
is the oscillator frequency in MHz. The frequency can
be adjusted by either changing C

2

or by placing a vari-

able capacitor in series with the crystal. As an exam­ple, assume a desired frequency of 14MHz and a load
capacitance of 32pF. C

1

=137.1 pF and C2= 41.7pF.

These capacitor values provide a starting point. The
drive level of the oscillator should be checked by look­ing at the signal at pin 2 (OSC E). It has been found
that the level at this pin should generally be around
500 mV

PP

or less. This will reduce the reference spur

levels and reduce noise from distor tion. If this level is
higher than 500mV

PP

then decrease the value of C1.

The values of these capacitors are usually tweaked
during design to meet performance goals, such as
minimizing the start-up time.

Additionally, by placing a variable capacitor in series
with the crystal, one is able to adjust the frequency.
This will also alter the drive level, so it should be
checked again.

An impor tant part of the overall design is the voltage
controlled oscillator.TheVCOisconfiguredasadif­ferential amplifier.The VCO is tuned via internal varac­tors. The varactors are tuned by the loop filter output
voltage through a 4 kΩ resistor.

As mentioned earlier, the inductors and the varactors
aretuningadifferentialamplifier.TotunetheVCOthe
designer only needs to calculate the value of the induc­tors connected to pins 12 and 13 (RESNTR- and
RESNTR+). The inductor value is determined by the
equation:

In this equation, f is the desired operating frequency
and L is the value of the inductor required. The value C
is the amount of capacitance presented by the varac­tors and parasitics. For calculation purposes 1.5pF
should be used. The factor of one-half is due to the
inductors being in each leg. As an example, assume
an operating frequency of 433MHz. The calculated
value of each inductor is 45nH. A 47nH inductor would
be appropriate as the closest available value.

X1 C2

C1

V

CC

C

1

60 C

load

⋅

freq

MHz

----------------------- -

= C

2

1

1

C

load

------------ -

1

C

1

------

–

--------------------------

=

4k

Ω

LOOP FLT

L L

RESNTR+ RESNTR-

L

1

2

π f⋅⋅

----------------

èø

æö

2

1

C

--- -

1
2

-- -

⋅⋅

=

Preliminary

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TRANSCEIVERS

ThesetupoftheVCOcanbesummarizedasfollows.
First, open the loop. Next, get the VCO to run on the
desired frequency by selecting the proper inductor and
capacitor values. The capacitor value will need to
include the varactor and circuit parasitics.

After the VCO is running at the desired frequency, set
the VCO sensitivity. The sensitivity is determined by
connecting the control voltage input point to ground
and noting the frequency.

Connect the same point to the supply, and again note
the frequency. The difference between these two fre­quencies divided bythe supply voltageis the VCO sen­sitivity expressedin Hz/V. Increasing the inductor value
while decreasing the capacitor value will increase the
sensitivity. Decreasing the inductor value while
increasing the capacitor value will lowerthe sensitivity.

When increasing or decreasing component values,
make sure that the center frequency remains constant.
Finally, close the loop.

External to the part, the designer needs to implement a
loop filter to completethe PLL. The loop filter converts
the output of the charge pump into a voltage that is
used to control the VCO. Internally, the VCO is con­nected to the charge pump output through a 4k Ω resis­tor.Theloopfilteristhenconnectedinparalleltothis
point at pin 14 (LOOP FLT). This limits the loop filter
topology to a second order filter usually consisting of a
shunt capacitor and a shunt series RC. A passive filter
is most common, as it is a low-cost and low-noise
design. An additional pole could be used for reducing
the reference spurs, however there is not a way to add
the series resistor. However, this should not be a rea­son for concern.

The schematic of the loop filter is:

The transfer function is:

where the time constants are defined as:

and

The frequency at which unity gain occurs is given by:

This is also the loop bandwidth.
If the phase margin (PM) and the loop bandwidth

(ω

LBW

) are known, it is possible to calculate the time

constants. These are found using the equations

4

:

and

With these known, it is then possible to determine the
values of the filter components.

4

As an example, consider a loop bandwidth of 50 kHz, a
phase margin of 45°, a divide ratio of 64, a K

VCO

of

20 MHz/V, and a KPD of 0.01592mA/2πrad. Time con­stant τ1 is 1.31848µs, time constant τ2 is 7.68468µs,
C

1

is 131.15pF, C2is 633.26pF, and R2is 12.14kΩ.

In order to perform these calculations, one will need to
know the value of two constants, K

VCO

and KPD.KPDis

calculated by dividing the charge pump current by 2π.
For the RF2516, the charge pump current is 100µA.
K

VCO

is best found empirically as it will change with

frequency and board parasitics. By briefly connecting
pin 14 (LOOP FLT) to VCC and then to ground, the fre­quency tuning range of the VCO can be seen. Dividing
the difference between these two frequencies by the
difference in the voltage gives K

VCO

in MHz/V.

V

CC

R2

C2

C1

VCO

Charge Pump

Loop Filter

Fs() R

2

s τ21+⋅

s τ2s τ11)+⋅(⋅⋅

-------------------------------------------

⋅

=

τ

2

R2C

2

⋅= τ

1

R

2

C1C

2

⋅

C1C2+

------------------ -

ø

ö

è

æ

⋅

=

ω

LBW

1

τ1τ2⋅

------------------ -

=

τ

1

PM()sec PM()tan–

ω

LBW

--------------------------------------------------

= τ

2

1

ω

LBW

2

τ1⋅

----------------------- -

=

C

1

τ

1

τ

2

---- -

KPDK

VCO

⋅

ω

LBW

2

N⋅

-----------------------------

1 ω

LBWτ2

⋅()

2

+

1

ω

LBWτ1

⋅()

2

+

----------------------------------------⋅⋅=

C

2

C

1

τ

2

τ

1

---- -

1–

èø

æö

⋅

=

R

2

τ

2

C

2

------

=

Preliminary

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TRANSCEIVERS

The control lines provide an interface for connecting
the device to a microcontroller or other signal generat­ing mechanism. The designer can treat pin 8 (MOD
IN), pin 16 (DIV CTRL), and pin 3 (PD) as control pins
whose voltage levelcan be set. The lock-detect voltage
at pin 15 (LD FLT) is an output that can be monitored
by the microcontroller.

Pin 15 (LD FLT) is used to set the threshold of the
lock-detect circuit. A shunt capacitor is used to set an
RC time constant with an on-chip series 1kΩ resistor.
The time constant should be approximately 10 times
the reference period.

General RF bypassing techniques must be observed
to get the best performance. Choose capacitors such
that they are series resonant near the frequency of
operation.

Board layout is always an area in which great care
must be taken. The board material and thickness are
used in calculating the RF line widths. The use of vias
for connection to the ground plane allows one to con­nect to ground as close as possible to ground pins.
When laying out the traces around the VCO, it is desir­able to keep the parasitics equal between the two legs.
This will allow equal valued inductors to be used.

Pre-compliance testing should be performed during
the design process. This can be done with a GTEM cell
or at a compliance testing laboratory. It is recom­mended that pre-compliance testing be performed so
that there are no surprises during final compliance
testing. This will help keep the product development
and release on schedule.

Working with a laboratory offers the benefit of years of
compliance testing experience and familiarity with the
regulatory issues. Also, the laboratory can often pro­vide feedback that will help the designer make the
product compliant.

On the other hand, having a GTEM cell or an open air
test site locally offers the designer the ability to rapidly
determine whether or not design changes impact the
product's compliance. Set-up of an open air test site
and the associated calibration is not trivial. An alterna­tive is to use a GTEM test c ell.

After the design has been completed and passes com­pliance testing, application will need to be made with
the respective regulatory bodies for the geographic
region in which the product will be operated to obtain
final certifications.

RF2516 Typical Applications

FCC Part 15.231 Periodic Transmitter - 315MHz Auto­motive Keyless Entry Transmitter

The following information is taken or paraphrased from
the Code of FederalRegulations Title 47, Part 15, Sec­tion 231 (47 CFR 15.231). Part 15 discusses radio fre­quency devices and section 231 discusses periodic
transmissions. Please refer to the regulation itself as
the final authority. Additional information may be found
on the Internet at www.fcc.gov.

To highlight the main guidelines outlined by this sec­tion, there are five main limitations: operating fre­quency, transmission content, transmission duration,
emission bandwidth, and spur ious emissions.

Part 15.231 allows operation in two bands: 40.66MHz
to 40.70MHz and above 70MHz. Transmission is lim­ited to control signals such as alarm systems, door
openers, remote switches, etc. Radio control of toys is
not permitted, nor is continuous transmission such as
voice or video. Data transmission other than a recogni­tion code is not permitted. Transmission time is limited
to 5 seconds (paragraph a) or for 1 second with greater
than ten seconds off (paragraph e).

Emission bandwidth between 70 MHz and 900MHz
can not be more than 0.25% of the center frequency.
Above 900MHz, the emission bandwidth cannot be
greater than 0.50% of the center frequency. The emis­sion bandwidth is determined from the points that are
20dB down from the modulated carrier. This corre­sponds to an occupied bandwidth of 4.5 MHz at a cen­ter frequency of 902MHz, 1.1MHz at 433 M Hz, and
788 kHz at 315MHz.

Spurious emissions limits are listed in tabular form for
various frequency ranges in the Section 231. Above
470 MHz with a manually activated transmitter, the fun­damental field strength at a distance of 3 meters shall
not exceed 12,500microvolts/meter. The spurious
emissions shall not exceed 1,250microvolt/meter at a
distance of 3meters above 470MHz. Refer to Appen­dix A for a method of convertingfield strength to power.

In the frequency range of 260MHz to 470MHz, one
needs to linearly interpolate the maximum emissions
levelfor both the fundamental and spurious emissions.
The equation for this line is given by:

E

µV

m

------ -

41

2
3

-- -

Freq

MHz

7083

1
3

-- -

–⋅=

Preliminary

11-44

RF2516

Rev A10 010613

11

TRANSCEIVERS

This equation is derived from the endpoints of the fre­quency range and their respective field strengths. Note
that the field strength is in microvolts per meter and the
frequency is in megahertz. To determine the spurious
level, divide the level calculated above for the spurious
frequency by ten.

As an example, assume the fundamental is 315MHz
and the reference frequency is 9.8MHz. The field
strengths of the fundamental, the r eference spurs, and
the harmonics of the fundamental up through the tenth
harmonic are calculated in the following table The
occupied bandwidth limit is 787.5kHz. As shown in
Table A, the fifth, seventh,and ninth harmonics fall into
restricted bands as called out in section 15.205. The
limits for these restricted bands are called out in sec­tion 15.209. The power level in the last column is the
level if the output is connected directly to a spectrum
analyzer. Refer to Appendix A as to how this column
was calculated.

Local Oscillator Source

Since the RF2516 has a phase-locked VCO, it can be
used as a signal source. The device is an ASK/OOK
transmitter, with the data provided at the MOD IN pin.
WhentheMODINisahighlogiclevel,thecarrieris
transmitted.When MOD IN is a low logic level, then the
carrier is not transmitted. Therefore, to use the RF2516
as signal source, simply tie the MOD IN pin to the sup­ply voltage, through a suitable series resistor (mini­mum 3kΩ).

Conclusions

The RF2516 is an AM/OOK VHF/UHF transmitter that
features a phase-locked output. This device is suitable
for use in a CFR Part 15.231 compliant product as well
as a local oscillator signal source. Two examplesshow­ing these applicationswere discussed.

The RF2516 is packaged in a low-cost plastic package
and requires few external parts, thus making it suitable
for low-cost designs.

Table A

Frequency

(MHz)

15.205 Limits
(µV/m@3m)

15.231 Limits
(µV/m @3m)

Final FCC Mask

(µV/m@3m)

Final FCC Mask

(µV/m @3m)

Power Level
(dBm, 50Ω)

Ref Spur 305.2 - 604.17 604.17 55.62 -39.61

1 315.0 - 6041.67 6041.67 75.62 -19.61

Ref Spur 324.8 - 604.17 604.17 55.62 -39.61

2 630.0 - 604.17 604.17 55.62 -39.61
3 945.0 - 604.17 604.17 55.62 -39.61
4 1260.0 - 604.17 604.17 55.62 -39.61
5 1575.0 500 - 500.00 53.98 -41.25
6 1890.0 - 604.17 604.17 55.62 -39.61
7 2205.0 500 - 500.00 53.98 -41.25
8 2520.0 - 604.17 604.17 55.62 -39.61
9 2835.0 500 - 500.00 53.98 -41.25

10 3150.0 - 604.17 604.17 55.62 -39.61

Preliminary

11-45

RF2516

Rev A10 010613

11

TRANSCEIVERS

Pin Out

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

DIV CTRL

LD FLT

LOOP FLT

RESNTR+

RESNTR-

VREFP

GND2

VCC2

OSC B

OSC E

PD

GND

TX OUT

GND1

VCC1

MOD IN

Preliminary

11-46

RF2516

Rev A10 010613

11

TRANSCEIVERS

Applicatio n Schematic

315MHz

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

68 pF

9.83 MH z

33pF

*Not populated on standard Evaluation Board.

OSC B

OSC E

GND

TX O UT

GND1

VCC1

MOD IN

DIV CTRL

LD F LT

LOO P F LT

RESNTR+

RESNTR -

VREFP

GND2

VCC2

50Ωµstrip

4pF

50Ωµstrip

56 nH

16 k

Ω

J1

TX OUT

220 pF

10

Ω

TX VC C

100 pF

MOD IN

S1

CAS-120B

10

Ω

82 nH

82 nH

2k

Ω

10 nF

4.3 k

Ω

2.2 nF

1nF

V

C

C

V

C

C

V

C

C

PD

Preliminary

11-47

RF2516

Rev A10 010613

11

TRANSCEIVERS

Application Schematic

315MHz

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

68 pF

9.83 MH z

33pF

OSC B

OSC E

GND

TX OUT

GND1

VCC1

MOD IN

DIV CTRL

LD F LT

LOO P FLT

RESNTR+

RESNTR-

VREFP

GND2

VCC2

50Ωµstrip

4pF

50Ωµstrip

56 nH

16 k

Ω

J1

TX OUT

220 pF

10

Ω

TX VCC

100 pF

S1

CAS-120B

10

Ω

82 nH

82 nH

2k

Ω

10 nF

4.3 k

Ω

2.2 nF

1nF

PD

D1

SMV1249-011

150 k

Ω

AUDIO

V

CC

V

CC

RF2516 Audio Transmitter

V

CC

V

CC

Preliminary

11-48

RF2516

Rev A10 010613

11

TRANSCEIVERS

Applicatio n Schematic

433MHz

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

68 pF

13.57734 MHz

PWR DWN

4.3 k

Ω

220 pF

220 pF 10 nF

220 pF

DIV CTRL

10 nF

TX OUT

33pF

1nF

2.2 nF

39 nH

2k

Ω

10

Ω

10 nF

*Not populated on standard Evaluation Board.

*

2.8

VCC(V) Mod.in Res. Value

(R5)

I

CC

(mA)

P

OUT

(dBm)

1k
3k
5k
7k

9k
11k
13k
15k
17k
19k
21k

17.38

10.51

8.68

7.82

7.18

6.75

6.45

6.18

5.99

5.80

5.66

7.45

8.78

7.23

6.00

4.73

3.81

2.98

2.30

1.63

1.00

0.35

2.0

VCC(V) Mod.in Res. Value

(R5)

1k
3k
5k
7k

9k
11k
13k
15k
17k
19k
21k

I

CC

(mA)

11.08

10.83

4.61

4.00

3.63

3.42

3.26

3.15

3.07

3.01

2.95

P

OUT

(dBm)

-6.23

-4.40

-5.61

-6.66

-8.08

-8.93

-10.04

-10.71

-11.58

-12.32

-13.10

2.4

VCC(V) Mod.in Res. Value

(R5)

1k
3k
5k
7k

9k
11k
13k
15k
17k
19k
21k

I

CC

(mA)

14.05

9.00

7.48

6.73

6.16

5.79

5.53

5.29

5.13

4.98

4.86

P

OUT

(dBm)

7.94

7.63

5.95

4.64

3.35

2.40

1.47

0.75

0.05

-0.60

-1.26

3.2

VCC(V) Mod.in Res. Value

(R5)

1k
3k
5k
7k

9k
11k
13k
15k
17k
19k
21k

P

OUT

(dBm)

6.77

9.70

8.30

7.11

5.91

5.02

4.16

3.51

2.89

2.26

1.66

I

CC

(mA)

20.90

12.12

9.66

8.95

8.23

7.75

7.42

7.10

6.89

6.68

6.52

3.6

VCC(V) Mod.in Res. Value

(R5)

1k
3k
5k
7k

9k
11k
13k
15k
17k
19k
21k

P

OUT

(dBm)

5.78

10.42

9.18

8.08

6.88

6.02

5.19

4.52

3.93

3.35

2.72

I

CC

(mA)

24.68

13.88

10.94

10.14

9.34

8.81

8.44

8.09

7.86

7.63

7.44

OSC B

OSC E

GND

TX OUT

GND1

VCC1

MOD IN

DIV CTRL

LD FLT

LOOP FLT

RESNTR+

RESNTR-

VREFP

GND2

VCC2

50Ωµstrip

4pF

50Ωµstrip

68 nH

2pF

50Ωµstrip

15 pF 10 nH 15 pF10 nH

22 nH

220 pF10 nF

220 pF10 nF

10

Ω

3kΩ10 nF

MOD IN

39 nH

V

CC

V

CC

V

CC

PD

Preliminary

11-49

RF2516

Rev A10 010613

11

TRANSCEIVERS

Evaluation Board Schematic

315MHz

(Download Bill of Materials from www.rfmd.com.)

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

C8

68 pF

Y1

9.83 MHz

C7

33pF

*Not populated on standard Evaluation Board.

OSC B

OSC E

GND

TX OUT

GND1

VCC1

MOD IN

DIV CTRL

LD FLT

LOOP FLT

RESNTR+

RESNTR-

VREFP

GND2

VCC2

50Ωµstrip

C6

4pF

50Ωµstrip

L3

56 nH

R4

16 k

Ω

J1

TX OUT

C5

220 pF

TX VCC

C4

100 pF

MOD IN

S1

CAS-120B

R2

10

Ω

L2

82 nH

L1

82 nH

R1

2k

Ω

C1

1

µ

F

R3

4.3 k

Ω

C2

2.2 nF

C3

1nF

VCC

VCC

VCC

GND

P1-1 VCC1

P1-3 MOD IN

P1

1
2
3

CON3

B1

LITH BATT

VCC

+

-

2516400, rev A

PD

R5

10

Ω

Preliminary

11-50

RF2516

Rev A10 010613

11

TRANSCEIVERS

Evaluation Board Schema t ic

433MHz

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

C2

68 pF

X1

13.57734 MHz

PWR DWN

R2

4.3k

Ω

C7

220 pF

C9

220 pF

C10

10 nF

VCC

C12

220 pF

DIV CTRL

C11

10 nF

C1

33pF

C6

1nF

C8

2.2 nF

L2

39 nH

R4

2k

Ω

R3

10

Ω

C13

10 nF

*Not populated on standard Evaluation Board.

C17*

OSC B

OSC E

GND

TX OUT

GND1

VCC1

MOD IN

DIV CTRL

LD FLT

LOOP FLT

RESNTR+

RESNTR-

VREFP

GND2

VCC2

50Ωµstrip

C3

4pF

50Ωµstrip

L4

68 nH

C15
2pF

50Ωµstrip

C14

15 pFL310 nH

C16

15 pF

L5

10 nH

L6

22 nH

C18

220 pF

C19

10 nF

VCC

C5

220 pF

C4

10 nF

R1

10

Ω

VCC

R5

3k

Ω

C20

10 nF

L1

39 nH

J1

TX OUT

J2

MOD IN

P1

1
2
3

CON3

VCC

NC
GND

P2

1
2
3

CON3

DIV CTRL

GND

PWR DWN

PD

Preliminary

11-51

RF2516

Rev A10 010613

11

TRANSCEIVERS

Evaluation Board Layout (315MHz)

Board Size 1.285” x 1.018”

Board Thickness 0.062”, Board Material FR-4

Preliminary

11-52

RF2516

Rev A10 010613

11

TRANSCEIVERS

Evaluation Board Layout (433MHz)

Board Size 1.392” x 1.392”

Board Thickness 0.031”, Board Material FR-4

Preliminary

11-53

RF2516

Rev A10 010613

11

TRANSCEIVERS

433MHz Phase Noise

0

1.0

1.0-1.0

10.0

1

0

.

0

-

1

0

.

0

5.0

5

.

0

-

5

.

0

2.0

2

.

0

-

2

.

0

3.0

3

.

0

-

3

.

0

4.0

4

.

0

-

4

.

0

0.2

0

.

2

-

0

.

2

0.4

0

.

4

-

0

.

4

0.6

0

.

6

-

0

.

6

0.8

0

.

8

-

0

.

8

RF2516 Output Z

Swp Max

1GHz

Swp Min

0.1GHz

VCC = 3 V
VCC = 2 V
VCC = 3.3 V

1.0 GHz 0.1 GHz

Preliminary

11-54

RF2516

Rev A10 010613

11

TRANSCEIVERS