Page 1
Preliminary
RF2514
11
Typical Applications
• 868MHz/915MHz ISM Band Systems
• Local Oscillator Source
• Remote Keyless Entry
Product Description
The RF2514 is a monolithic integrated circuit intended for
use as a low-cost AM/ASK transmitter. The device is provided in a4mmx4 mm, 16-pin leadless chip carrier and is
designed to provide a phased locked frequency source
for use in local oscillator or transmitter applications. The
chip can be used in applications in the North American
and European VHF/UHF ISM bands. The integrated
VCO, phase detector, reference divider, and reference
oscillator transistor require only the addition of an external crystal to provide a complete phase-locked oscillator.
In addition to the standard power-down mode, the chip
also includes an automatic lock detect feature that disables the transmitter output when the PLL is out-of-lock.
VHF/UHF TRANSMITTER
• AM/ASK/OOK Transmitter
• Wireless Security Systems
3.75
Dimensions in mm.
INDEX AREA
3.75
12°
3
0.75
0.65
0.05
0.00
0.75
0.50
1.00
0.90
NOTES:
Shaded Pin is Lead 1.1
Dimension applies to plated terminal and is measured between
2
0.10 mm and 0.25 mm from terminal tip.
The terminal #1 identifier and terminal numbering convention
3
shall conform to JESD 95-1 SPP-012. Details of terminal #1
identifier are optional, but must be located within the zone
indicated. The identifier may be either a mold or marked
feature.
4
Pins 1 and 9 are fused.
5 Package Warpage: 0.05 max.
2
0.45
0.28
1.50 SQ
3.20
4.00
0.80
TYP
+
1
1
4.00
1.60
Optimum Technology Matching® Applied
Si BJT GaAs MESFETGaAs HBT
Si Bi-CMOS
ü
SiGe HBT
RESNTR+11RESNTR-
10
3TX OUT
Lock
Detect
5 13
LD FLT
MOD IN
Detector &
Charge Pump
Prescaler
12
Phase
32/64
14
LOOP FLT
DIV CTRL
Si CMOS
OSC E
16
Bias
OSC B
15
DC
Functional Block Diagram
2PD
Package Style: LCC, 16-Pin, 4x4
Features
• Fully Integrated PLL Circuit
• Integrated VCO and Reference Oscillator
• 2.2V to 3.6V Supply Voltage
• Low Current and Power Down Capability
• 100MHz to 1000MHz Frequency Range
• Out-of-Lock Inhibit Circuit
Ordering Information
RF2514 VHF/UHF Transmitter
RF2514 PCBA Fully Assembled Evaluation Board
RF Micro Devices, Inc.
7625 Thorndike Road
Greensboro,NC 27409, USA
Tel (336)664 1233
Fax (336)664 0454
http://www.rfmd.com
11
TRANSCEIVERS
Rev A2 010215
11-27
Page 2
RF2514
Absolute Maximum Ratings
Parameter Rating Unit
Supply Voltage -0.5 to +3.6 V
Power Down Voltage (VPD) -0.5toV
Operating Ambient Temperature -40 to +85 °C
Storage Temperature -40 to +150 °C
CC
DC
V
Preliminary
Caution! ESD sensitive device.
RF Micro Devices believes the furnished information is correct and accurate
at the time of this printing. However, RF Micro Devices reserves the right to
make changes to its products without notice. RF Micro Devices does not
assume responsibility for the use of the described product(s).
11
Parameter
Overall
Frequency Range 100 868/915 1000 MHz
Modulation AM/ASK
Modulation Frequency 4 20 kHz Square wave, 50% duty cycle, 300kHz loop
Incidental FM 15 kHz
Output Power 1 dBm 50Ω load, CW
ON/OFF Ratio 52 dB
Specification
Min. Typ. Max.
Unit Condition
T=25°C, VCC=3.0V, Freq=916MHz,
R
=10kΩ
MODIN
bandwidth
P-P
PLL and Prescaler
Prescaler Divide Ratio 32/64
VCO Gain, K
PLL Phase Noise -90 dBc/Hz 10kHz Offset, 300kHz loop bandwidth
Harmonics -25 dBc With matched output and no additional filter-
Reference Frequency 14.318 17 MHz
Crystal Frequency Spurs -52 dBc 300kHz PLL loop bandwidth
Max Crystal R
Max Crystal Motional Inductance 10 mH For a typ. 2ms turn-on time.
Charge Pump Current 100 µA KPD=100µA/2π=0.0159µA/rad
VCO
S
40 MHz/V Frequency and board layout dependent
-95 dBc/Hz 100kHz Offset, 300kHz loop bandwidth
ing.
10 50 Ω For a typ. 2ms turn-on time.
Power Down Control
Power Down (VIL) 0 0.3 V Voltage supplied to the input; device is “OFF”
Power Down (V
Control Input Impedance 100 kΩ
Tu rn On Time 2 ms Crystal start-up, 14.318MHz crystal.
Turn Off Time 2 ms
)V
IH
-0.3 V
CC
CC
V Voltage supplied to the inp ut; device is “ON”
Power Supply
Voltage 2.2 3.0 3.6 V Specifications
TRANSCEIVERS
Current Consumption
Average 8 mA 50% Duty Cycle 4kHz Data applied to the
Sleep Mode 1 µAPD
Operating limits
MODINinput.R
put power/DC current consumption exter-
nally adjustable by modulation input resistor
(see applicable Application Schematic).
=0
(R7+R8)=10 kΩ.Out-
MODIN
11-28
Rev A2 010215
Page 3
Preliminary
RF2514
Pin Function Description Interface Schematic
1GND1
2PD
Ground connection for the analog circuits, including TX buffer and output amplifier. Internally connected to die flag. For best perfor mance,
keep traces physically short and connect immediately to ground plane.
Power Down control for all circuitry. When this pin is a logic “ low” all circuits are turned off. W hen this pin is a logic “high”, all circuits are operating normally. See electrical parameters for “high” and “low”
thresholds.
V
CC
PD
3TXOUT
4VCC1
5MODIN
6VCC2
7GND2
8VREFP
9GND3
10 RESNTR-
Transmitter output. This output is an open collector and requires a pullup inductor for bias/matching and a tapped capacitor for matching.
This pin is used to supply bias to the TX buffer amplifier.
AM analog or digital modulation can be imparted to the c a rrier by an
input to this pin. An external resistor is used to bias the output amplifiers through this pin. The voltage at this pin must not exceed 1.1V.
Higher voltages may damage the device.
This pin is used to supply DC bias to the VCO, crystal o scillator, prescaler, phase detector, and charge pump. An IF bypass capacitor
should be connected directly to this pin and returned to ground.
Digital PLL ground connection.
Bias voltage reference pin for bypassing the prescaler and phase
detector. The bypass capacitor should be of appropriate size to provide
filtering of the reference crystal frequency and be connected directly to
this pin.
See pin 1.
The RESNTR pi ns are used to supply DC voltage to the VCO, as well
as to tune the center frequency of the VCO. Equal value inductors
should be connected to this pin and pin 11.
RF IN
See pin 3.
VREFP
LOOP FLT
4k
TX OUT
MOD IN
V
CC
11
RESNTR-RESNTR+
Ω
11 RESNTR+
12 LOOP FLT
Rev A2 010215
See pin 10.
Output of the charge pump. An RC network from this pin to ground is
used to establish the PLL bandwidth.
TRANSCEIVERS
V
CC
LOOP FLT
11-29
Page 4
RF2514
Preliminary
Pin Function Description Interface Schematic
13 LD FLT
This pin is used to set the threshold of the lock detect circuit. A shunt
capacitor should be used to set an RC time constant with the on-chip
series 1k resistor. The time constant should be approximately 10 times
the reference period.
LD FLT
V
CC
11
14 DIV CTRL
15 OSC B
16 OSC E
Die
GND
Flag
ESD
Logic “High” input selects divide-by-64 prescaler. Logic “Low” input
selects divide-by-32 prescaler.s
This pin is connected directly to the reference oscillator transistor base.
The intended reference oscillator configuration is a modified Colpitts. A
68pF capacitor should be connected between pin 15 and pin 16.
This pin is connected directly to the emitter of the reference oscillator
transistor. A 33pF capacitor should be connected from this pin to
ground.
Exposed die flag is centered and measures 1.5mmx1.5mm
(0.059in.x0.059in.). For best results, provide a solder pad for the flag
and connect immediately to ground plane (see evaluation board layout). Internally connected to pins 1 and 9.
This diode structure is used to provide electrostatic discharge protection to 3kV using the Human body model. The following pins are protected: 1, 2, 4-9, 12-14. The die flag is not protected.
DIV CTRL
OSC B
See pin 15.
V
CC
V
CC
OSC E
V
CC
TRANSCEIVERS
11-30
Rev A2 010215
Page 5
Preliminary
RF2514
RF2514 Theory of Operation
Introduction
Short range radio devices are becoming commonplace
in today's environment. The most common examples
are the remote keyless entry systems popular on many
new cars and trucks and the ubiquitous garage door
opener. Other applications are emerging along with the
growth in home security and automation and the
advent of various remote control applications. Typically
these devices have been simplex, or one way, links.
They are also typically built using surface acoustic
wave (SAW) devices as the frequency control elements. This approach has been attractive because the
SAW devices have been readily available and a transmitter, for example, could be built with only a few additional components. Recently, however, RF Micro
Devices has introduced several new components that
enable a new class of short range radio devices based
on the use of crystals and phase locked loops for frequency control. These devices are superior in performance and comparable in cost to the traditional SAW
based designs. The RF2514 is an example of such a
device. The RF2514 is targeted for applications such
as 315, 433, 868 and 915MHz band remote keyless
entry systems, wireless security systems, and other
remote control applications.
The RF2514 Transmitter
The RF2514 is a low cost AM/ASK VHF/UHF transmitter designed for applications operating within the frequency range of 100MHz to 1000 MHz. In particular, it
is intended for 868 and 915MHz band systems (ETS
300 220 applications and FCC Parts 15.231 and
15.249 transmitters) and remote keyless entry systems. It can also be used as a local oscillator signal
source. The integrated VCO, phase detector, prescaler, and reference oscillator require only the addition
of an external crystal to provide a complete phaselocked loop. In addition to the standard power down
mode, the chip also includes an automatic lock detect
feature that disables the transmitter output when the
PLL is out-of-lock.
The device is manufactured on a 25GHz silicon bipolar-CMOS process and packaged in a n industry s tandard MLF16 plastic package. This, combined with the
low external parts count, enables the designer to
achieve small-footprint, high-performance, low-cost
designs.
The RF2514 is designed to operate from a supply voltage ranging from 2.2V to 3.6V, accommodating
designs using three NiCd batter y cells, two AAA flash-
light cells, or a lithium button battery. The device is
capable of providing up to +5dBm output power into a
50Ω load and is intended to comply with FCC and
ETSI requirements for unlicensed remote control transmitters. ESD protection is provided on all pins except
for OSCB, O SCE, RESNTR-, RESNTR+, TXOUT, and
the two analog ground pins (1 and 9).
While this device is intended for OOK operation, it is
possible to use narrowband FM. This is accomplished
by modulating the r eference oscillator rather than
applying the data to the MOD IN input pin. The MOD
INpinshouldbetiedhightocausethedevicetotransmit. The deviation will be set by pulling limits of the
crystal. Deviation sufficient for the transmission of
voice and other low data rate signals can therefore be
accomplished. Refer to the Application Schematic in
the data sheet for details.
RF2514 Functional Blocks
A PLL consists of a reference oscillator, a phase detector, a loop filter, a voltage controlled oscillator (VCO),
and a programmable divider in the feedback path. The
RF2514 includes all of these internally except for the
loop filter and the reference oscillator's cr y stal and two
feedback capacitors.
The reference oscillator is a Colpitts type oscillator.
Pins OSC B and OSC E provide c onnections to a transistor that is used as the reference oscillator. The Colpitts c onfiguration is a low par ts count topology with
reliable perfor mance and reasonable phase noise.
Alternatively, an external signal could be injected into
the base of the transistor. The dri ve level should, in
either case, be around 500mV
overdriving the device and keeps the phase noise and
reference spurs to a minimum.
The p rescaler uses a series of flip-flops to divide the
VCO frequency by either 64 or 32, depending upon the
logic level present at the DIV CTRL pin. A high logic
level will select the 64 divisor. A low logic level will
select the 32 divisor. This divided signal is then fed into
the phase detector where it is compared with the reference frequency.
The RF2514 contains an onboard phase detector and
charge pump. The phase detector c ompares the phase
of the reference oscillator to the phase of the prescaler
output. The phase detector is implemented using flipflops in a topology referred to as either "digital phase/
frequency detector" or "digital tri-state comparator".
. This level prevents
PP
11
TRANSCEIVERS
Rev A2 010215
11-31
Page 6
RF2514
Preliminary
11
The circuit consists of two D flip-flops whose outputs
are combined with a NAND gate which is then tied to
the reset on each flip-flop. The outputs of the flip-flops
are also connected to the charge pump inputs. Each
flip-flop output signal is a series of pulses whose frequency is related to the flip-flop input frequency. When
both inputs of the flip-flops are identical, the signals are
both frequency and phase locked. If they are different,
they will provide signals to the charge pump which will
either charge or discharge the loop filter or place the
charge pump in a high impedance state, maintaining
thechargeontheloopfilter.Thename"tri-statecomparator" comes from this. The main benefit of this type
of detector is the ability to correct for errors in both
phase and frequency. When locked, the detector uses
phase error for correction. When unlocked, it will use
the frequency error for correction. This type of detector
will lock under all conditions.
The charge pump consists of two transistors, one for
charging the loop filter and the other for discharging
the loop filter. The charge pump inputs are the outputs
of the phase detector flip-flops. If both amplifier inputs
are low, then the amplifier pair goes into a high impedance state, maintaining the charge on the loop filter. In
the charge and discharge states, the loop filter integrates the pulses coming from the charge pump to create a control voltage for the voltage controlled
oscillator.
The VCO is a tuned-differential a mplifier with the
bases and collectors cross-coupled to provide positive
feedback and a 360° phase shift. The tuned circuit is
located in the collectors and is comprised of internal
varactors and external inductance, which also provides
DC bias for the VCO. The varactor diodes are internally configured for negative tuning. That is, a higher
control voltage results in a lower VCO frequency by
reducing the varactor reverse bias which correspond-
TRANSCEIVERS
ingly increases the capacitance. The inductance is
selected by the designer for the desired frequency of
operation. Two inductor configurations are possible.
In the first configuration, two inductors are connected
in series between RESNTR- and RESNTR+. A resistor
is then used to provide the DC bias to the balanced
inductance node formed by the series connection of
the inductors. Ideally, the two inductors should be
equal in value, but a slight imbalance is acceptable if
necessary for VCO centering.
In the second configuration, a single inductor is placed
across RESNTR- and RESNTR+ and one resistor is
used to provide bias to the differential amplifier. The
resistor is connected in series from VCC to either
RESNTR- or RESNTR+. The inductor provides the DC
bias path for the other resonator pin. This configuration
has the advantage of lower cost and parts count, as
only one inductor is required; the disadvantage is
potentially suboptimal VCO centering due to limited
standard inductor values. For example, 20nH may be
the optimal inductance to center the VCO at the
desired operating frequency, but only 18nH and 22nH
inductors are available as standard values. However,
for the two-inductor configuration, both inductors can
be 10nH, thus giving the optimal 20nH of inductance.
Of course, the problem of optimization can also be
resolved by increasing (or decreasing) the inductance
of the traces running to the inductor in the single-inductor configuration.
The output of the VCO is buffered and applied to the
prescaler circuit, where it is divided by either 32 or 64,
as selected by the designer, and compared to the reference oscillator frequency.
The transmit amplifier is a two-stage amplifier consisting of a driver and an open collector final stage. It is
capable of providing 5dBm of output power into a 50Ω
load while operating from a 3.6V power supply.
The lock-detect circuitry connects to the output of the
phasedetectorcircuitryandisusedtodisablethe
transmitter when the VCO is not phase-locked to the
reference oscillator. This is necessary to avoid
unwanted out-of-band transmission and to provide
compliance with regulatory limits during an unlocked
condition.
There are many possible reasons that the PLL could
be unlocked. For instance, there is a shor t period during the start of any VCO in which the VCO star ts oscillating and the reference oscillator builds up to full
amplitude. During this period, the frequency will likely
be outside the authorized band. Typically the VCO
starts much faster than the reference oscillator. Once
both VCO and reference oscillators are running, the
phase detector can start slewing the VCO to the correct frequency, sliding across 200MHz of occupied
spectrum. In some competitive devices, the transmitter
output operates at full power under all of these conditions.
The lock protection circuit in the RF2514 is intended to
stabilize quickly after power is applied to the chip and
to disable the base drive to the transmit amplifier. This
attenuates the output to levels that will be generally
acceptable to regulatory boards as spurious emissions. Once the phase detector has locked the oscillators, then the lock circuit enables the MOD IN pin for
11-32
Rev A2 010215
Page 7
Preliminary
RF2514
transmission of the desired data. There is no need for
an external microprocessor to monitor the lock status,
although that can be done with a low current A/D converter in a system micro, if needed. The lock detect circuitry contains an internal 1kΩ resistor which,
combined with a designer-chosen capacitor for a particular RC time constant, filters the lock detect signal.
This signal is then passed through an internal Schmitt
trigger and used to enable or disable the transmit
amplifier.
If the oscillator unlocks, even momentarily, the protection circuit quickly disables the output until lock is
achieved. These unlocks can be caused by low battery
voltage, poor power supply regulation, severe shock of
the crystal or VCO, antenna loading, component failure, or a myriad of unexpected single-point failures.
The RF2514 contains onboard band gap reference
voltage circuitry which provides a stable DC bias over
varying temperature and supply voltages. Additionally,
the device features a power-down mode, eliminating
battery disconnect switches.
Designing with the RF2514
The reference oscillator is built around the onboard
transistor at pins 15 and 16. The intended topology is
that of a Colpitts oscillator. The Colpitts oscillator is
quite common and requires few external components,
making it ideal for low cost solutions. The topology of
this type of oscillator is as seen in the following figure.
V
CC
X1 C2
C1
This type of oscillator is a parallel resonant circuit for a
fundamental mode crystal. The transistor amplifier is
an emitter follower and the voltage gain is developed
by the tapped capacitor impedance transformer. The
series combination of C
input capacitance of the transistor to capacitively load
the crystal.
and C2act in parallel with the
1
The nominal capacitor values can be calculated with
the following equations
60 C
⋅
load
----------------------- -
C
= C
1
freq
The load capacitance, C
crystal used; freq
MHz. The frequency can be adjusted by either chang-
ing C2 or by placing a variable capacitor in series with
the crystal. As an example, assume a desired oscillator
frequency of 14MHz and a load capacitance of 32 pF.
C
= 137.1pF and C2=41.7pF.
1
These capacitor values provide a starting point. The
drive level of the oscillator should be checked by looking at the signal at the OSC E pin. It has been found
that the level at this pin should generally be around
500 mV
levels and reduce noise produced by distortion. If this
level is higher than 500 mV
of C
adjusted dur ing design to meet performance goals,
such as minimizing the start-up time.
An important part of the overall design is the voltage
controlled oscillator. The VCO is configured as a differential amplifier. The VCO range is set by the external
inductor(s) and is fine-tuned via internal varactor
diodes.The varactorsare tuned by the loop filter output
voltage through a 4kΩ resistor. (Refer to the internal
schematic for RESNTR- in the pin description table.)
To tune the VCO the designer only needs to calculate
the value of the inductor(s) connected to RESNTRand RESNTR+. The inductor value is determined by
the equation:
or less. This will reduce the reference spur
PP
. The values of these capacitors are usually
1
L
In this equation, f is the desired operating frequency
and L is the value of the inductor required. In the case
of a two-inductor resonator configuration,the value of L
is halved due to the inductors being in each leg. The
value C is the amount of capacitance presented by the
varactors and parasitics. For calculation purposes,
1.5 pF should be used. As an example, assume an
operating frequency of 868MHz. The calculated inductor value is 22.4nH. A 22nH inductor (two 10nH inductors for the two-inductor configuration) would be
appropriate as the closest available value. Be aware
and
MHz
load
is the oscillator frequency in
MHz
1
æö
----------------
=
èø
2
=
2
, is a characteristic of the
then decrease the value
PP
2
⋅
π f⋅⋅
1
--------------------------
1
------------ -
C
load
1
--- -
C
1
------
–
C
1
11
TRANSCEIVERS
Rev A2 010215
11-33
Page 8
RF2514
Preliminary
11
that any inductance in the traces connecting the inductor(s) to the VCO pins will contribute to the overall resonator inductance and should be subtracted from the
calculated value of L.
A parameter of the VCO that is necessary for calculating the loop filter values is the VCO sensitivity, K
(sometimes r eferred to as VCO gain). To determine the
VCO sensitivity, first connect the control voltage input
point (LOOP FLT pin) to ground and note the frequency. (The frequency can be observed at the output
if the LD FLT pin is connected to VCC.) Then connect
the same point to the supply and again note the frequency. The difference between these two frequencies
divided by the supply voltage is the VCO sensitivity
expressed in Hz/V. There is little that the designer can
do to increase the VCO sensitivity since it is largely
determined by the tuning capacitance of the on-chip
varactors. While increasing the inductor value will
increase the tuning sensitivity, it will also lower the center frequency of the VCO's tuning range. A very small
capacitance (1pF or less) may be added across the
VCO pins, which will have the effect of lowering the
VCO center frequency and decreasing VCO sensitivity,
but this is likely to be neither necessary nor desirablein
most applications.
Should adequate centering of the VCO range be
unachievable with standard inductor values, two
options are available for proper centering. First, a twoinductor resonator may be used with one inductor
being one standard value higher than the other. Second, the tuning range of the VCO m ay be extended at
the upper limit of the control voltage by increasing the
VCO bias resistor(s). This allows the internal varactor
diodes to be slightly forward biased, further increasing
the resonator capacitance and thereby extending the
lower frequency operation. Care should be taken not to
reduce the VCO bias so much that the circuit ceases
TRANSCEIVERS
operation at the minimum required supply voltage.
External to the part, the designer needs to implement a
loop filter to complete the PLL. The loop filter converts
the output of the charge pump into a voltage that is
used to control the VCO. Internally, the VCO is connected to the charge pump output through a 4kΩ resistor.The loop filter is then connected in parallel with this
point at pin 12 (LOOP FLT). This limits the loop filter
topology to a second order filter usually consisting of a
shunt capacitor and a shunt series RC, as shown in the
following schematic.
VCO
Charge Pump
V
CC
The transfer function is
Loop Filter
R2
C2
VCO
C1
s τ21+⋅
-------------------------------------------
Fs() R
where the time constants are defined as
τ
R2C
2
The frequency at which unity gain occurs is given by
This is defined as the loop bandwidth.
Once the desired phase m argin (PM) and loop band-
width (ω
time constants. These are found using the equations
τ
1
The phase detector gain, K
the charge pump current by 2π. For the RF2514, the
charge pump current is 100µA.
With these known, it is then possible to determine the
values of the filter components.
C
C
) are chosen, it is possible to calculate the
LBW
PM()sec PM()tan–
--------------------------------------------------
= τ
τ
1
---- -
1
τ
2
C
=
2
1
⋅
=
2
s τ2s τ11)+⋅(⋅⋅
and
⋅= τ
2
ω
ω
LBW
K
PDKVCO
-----------------------------
ω
τ
æö
---- -
⋅
èø
τ
LBW
2
LBW
2
1
=
⋅
N⋅
1–
C1C
⋅
2
R
=
1
1
------------------ -
⋅
2
æ
------------------ -
è
C1C2+
ö
ø
τ1τ2⋅
and
, is calculated by dividing
PD
1 ω
+
----------------------------------------⋅⋅=
1
+
=
2
LBWτ2
ω
LBWτ1
----------------------- -
ω
2
LBW
⋅()
⋅()
1
τ
------
R
=
2
C
τ1⋅
2
2
2
2
11-34
Rev A2 010215
Page 9
Preliminary
RF2514
As an example, consider a loop bandwidth of 300kHz,
a phase margin of 60°, a divide ratio of 64, a K
33 M Hz/V, and a K
stant τ
3.9pF, C
is 142.15ns, time constant τ2is 1.98ms, C1is
1
is 50.3pF, and R2is 39.4kΩ.
2
of 0.01592mA/2πrad. Time con-
PD
VCO
The control lines provide an interface for connecting
the device to a microcontroller or other signal generating mechanism. The designer can treat pin 5 (MOD
IN), pin 14 (DIV CTRL), and pin 2 (PD) as control pins
whose voltage level can be set. The lock detect voltage
at pin 13 (LD FLT) is an output that can be monitored
by the microcontroller.
Pin 5 (MOD IN) is the data input to the modulator and
must have a series resistor (R
the raw data source. The value of R
) between it and
MOD_IN
MOD_IN
and the
voltage at its input determine the output power level,
with maximum power obtained for R
MOD_IN
=3kΩ,the
minimum allowable resistance. A three-element filter
structure (series R, shunt C, series R) has been found
to be effective in reducing the out-of-band spectral content by filtering the higher frequency components of the
baseband data. For this filter, R
MOD_IN
is the sum of
the two series resistors. The filter values will vary
according to the par ticular data rate of a given application and are best determined experimentally. When the
input to R
MOD_IN
is a high logic level, the carrier is
transmitted; when the input is a low logic level, the carrier is not transmitted. For use as a local oscillator (LO)
source, simply tie the MOD IN pin to the supply voltage
through a suitable series resistor.
Pin 13 (LD FLT) is used to set the threshold of the lock
detect circuit. A shunt capacitor is used to set an RC
time constant with an on-chip series 1kΩ resistor. The
time constant should be approximately 10 times the
reference period.
General RF bypassing techniques must be observed to
get the best performance. Choose capacitors such that
they are s eries resonant near the frequency of operation.
Board layout is always an area in which great care
must be taken. The board material and thickness are
used in calculating the RF line widths. The use of vias
allows IC and component ground pins to be connected
closely to the ground plane, minimizing ground inductance. When laying out the traces around the VCO, it is
desirable to keep the parasitics equal between the two
legs. This will allow equal valued inductors to be used.
It is recommended that pre-compliance testing be per-
of
formed during the design process to avoid surprises
during final compliance testing, helping to keep the
product development and release on s chedule. Precompliance testing can be done with a GTEM cell, an
open area test site, or at a compliance testing laboratory.
After the design has been completed and passes compliance testing, then application will need to be made
to obtain final certifications with the respective regulatory bodies for the geographic region in which the
product will be operated.
TROUBLESHOOTING GUIDE
The following measurements were obtained from a
915MHz Evaluation Board.
Test conditions are: V
V
MOD_IN=VCC
Pin
Number
1GND10.00 0
2 PD 3.00 2.7M
3 TX OUT 3.00 1.6M
4 VCC1 3.00 1.6M
5 MOD IN 0.90 1.1M
6 VCC2 2.96 1.6M
7GND20.00 0
8 VREF P 0.91 1.1M
9GND30.00 0
10 RESNTR- 2.63 1.6M
11 RESNTR+ 2.63 1.6M
12 LOOP FLT 2.52* 1.9M
13 LD FLT 2.77 234 k
14 DIV CTL 3.00 1.6M
15 OSC B 2.83 1.7M
16 OSC E 2.00 Open
.
Pin
Name
=3.00V, R
CC
Typical DC
Voltage
MOD_IN
Ω to GND
(Power Off)
=10kΩ,
11
* Dependent on frequency of operation, board layout,
and component variations.
TRANSCEIVERS
Bibliography
1. Keese, William O., An Analysis and Performance
Evaluation of a PassiveFilter Design Technique for
Charge Pump Phase-Locked Loops: Application
Note 1001, National Semiconductor Cor p., M ay
1996.
2. Rhea, Randall W., Oscillator Design and Computer
Simulation, 2nd Ed., Atlanta: Noble Publishing,
1995.
Rev A2 010215
11-35
Page 10
RF2514
TX OUT
VCC1
Pin Out
GND1
OSCE
1 16
2PD
3
4
6
5
OSCB15DIV CTRL14LD FLT
13
12
11
10
8
7
Preliminary
LOOP FLT
RESNTR+
RESNTR-
9
11
VCC2
MOD IN
TRANSCEIVERS
GND2
GND3
VREFP
11-36
Rev A2 010215
Page 11
Preliminary
P1
1
GND, DGND
P1-2 PD
P1-3 VCC
2
3
CON3
Evaluation Board Schemati c
868MHz
C1
3 - 10 pF
X1
13.577 MHz
C2
33 pF
C3
68 pF
C4
1nF
RF2514
VCC
R1
0 Ω
R2
0 Ω
PD
C17*
J1
TX OUT
VCC
J2
MOD IN
*Components not populated on PCB.
50 Ωµstrip
C16
4pF
L3
10 nH
50 Ωµstrip
C15
5pF
C14
1.5 pF
C13*
R7
3.9 kΩ
C11
10 nF
L2
18 nH
0.1 uF
C12
1 16 15 14 13
2
3
4
R8
6.2 kΩ
C10
10 nF
R6
10 Ω
C9
10 nF
C5
7pF
12
L1 is placed 130 mils from the edge of U1 so an 18 nH standard
inductor can be used.
11
10
98765
18 nH
FLAG
2514400-
R5
L1
1.5 kΩ
C7
0.1 uF
R3
22 kΩ
C8*
C6
100 pF
VCC
VCC
11
Rev A2 010215
TRANSCEIVERS
11-37
Page 12
RF2514
P1
1
GND, DGND
P1-2 PD
P1-3 VCC
2
3
CON3
Evaluation Board Schema t ic
915MHz
C1
3-10pF
X1
14.318 MHz
C2
33 pF
C3
68 pF
C4
1nF
Preliminary
VCC
R1
0 Ω
R2
0 Ω
11
PD
C17*
J1
TX OUT
VCC
J2
MOD IN
*Components not populated on PCB.
50 Ωµstrip
C16
4pF
L3
10 nH
50 Ωµstrip
C15
5pF
C14
1.5 pF
C13*
R7
3.9 kΩ
C11
10 nF
L2
15 nH
0.1 uF
C12
1 16 15 14 13
2
3
4
R8
6.2 kΩ
C10
10 nF
R6
10 Ω
C9
10 nF
C5
7pF
12
11
10
98765
18 nH
FLAG
2514401-
R4
L1
1.5 kΩ
R5*
1.5 kΩ
R3
22 kΩ
C7
0.1 uF
C6
100 pF
VCC
C8*
VCC
TRANSCEIVERS
11-38
Rev A2 010215
Page 13
Preliminary
RF2514
Evaluation Board Layout (868MHz)
Board Size 1.242” x 1.242”
Board Thickness 0.031”, Board Material FR-4
Evaluation Board Layout (915MHz)
Board Size 1.242” x 1.242”
Board Thickness 0.031”, Board Material FR-4
Rev A2 010215
11
TRANSCEIVERS
11-39
Page 14
RF2514
Preliminary
11
TRANSCEIVERS
11-40
Rev A2 010215
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