Datasheet OPA632N-250, OPA632U, OPA632N-3K, OPA632U-2K5, OPA631U-2K5 Datasheet (Burr Brown)

®

OPA631
OPA632

© 1999 Burr-Brown Corporation PDS-1377A Printed in U.S.A. June, 1999

International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: [phone]

Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: [phone] • Immediate Product Info: [phone]

OPA632

V

IN

750Ω562Ω

2.26kΩ

374Ω

22pF

+3V

100Ω

Pwrdn

DIS

Disable

+3V

ADS901

10-Bit

20Msps

For most current data sheet and other product

information, visit www.burr-brown.com

SINGLES DUALS

Medium Speed, No Disable OPA631 OPA2631

With Disable OPA632 —

High Speed, No Disable OPA634 OPA2634

With Disable OPA635 —

RELATED PRODUCTS

Low Power, Single-Supply

OPERATIONAL AMPLIFIERS

TM

DESCRIPTION

The OPA631 and OPA632 are low power, high-speed,
voltage-feedback amplifiers designed to operate on a
single +3V or +5V supply. Operation on ±5V or +10V
supplies is also supported. The input range extends
below ground and to within 1V of the positive supply.
Using complementary common-emitter outputs pro­vides an output swing to within 30mV of ground and
130mV of the positive supply. The high output drive
current and low differential gain and phase errors also
make them ideal for single-supply consumer video
products.

Low distortion operation is ensured by the high gain
bandwidth (68MHz) and slew rate (100V/µs), making
the OPA631 and OPA632 ideal input buffer stages to
3V and 5V CMOS converters. Unlike other low power,
single-supply amplifiers, distortion performance im­proves as the signal swing is decreased. A low 6nV
input voltage noise supports wide dynamic range op­eration. Channel multiplexing or system power reduc­tion can be achieved using the high speed disable line.
Power dissipation can be reduced to zero by taking the
disable line High.

The OPA631 and OPA632 are available in an industry­standard SO-8 package. The OPA631 is also available
in an ultra-small SOT23-5 package, while the OPA632
is available in the SOT23-6. Where higher full-power
bandwidth and lower distortion are required in a single­supply operational amplifier, consider the OPA634
and OPA635.

APPLICATIONS

● SINGLE SUPPLY ADC INPUT BUFFER

● SINGLE SUPPLY VIDEO LINE DRIVER

● CCD IMAGING CHANNELS

● LOW POWER ULTRASOUND

● PLL INTEGRATORS

● PORTABLE CONSUMER ELECTRONICS

FEATURES

● HIGH BANDWIDTH: 75MHz (G = +2)

● LOW SUPPLY CURRENT: 6mA

● ZERO POWER DISABLE (OPA632)

● +3V AND +5V OPERATION

● INPUT RANGE INCLUDES GROUND

● 4.8V OUTPUT SWING ON +5V SUPPLY

● HIGH SLEW RATE: 100V/µs

● LOW INPUT VOLTAGE NOISE: 6nV/√HZ

● AVAILABLE IN SOT23 PACKAGE

OPA631

2

®

OPA631, OPA632

OPA631U, N
OPA632U, N

TYP GUARANTEED

0°C to –40°C to

MIN/

TEST

PARAMETER CONDITIONS +25°C +25°C70°C +85°C UNITS MAX

LEVEL

(1)

SPECIFICATIONS: VS = +5V

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1).

AC PERFORMANCE (Figure 1)

Small-Signal Bandwidth G = +2, V

O

≤ 0.5Vp-p 75 50 40 32 MHz min B

G = +5, V

O

≤ 0.5Vp-p 16 12 10 8.5 MHz min B

G = +10, V

O

≤ 0.5Vp-p 7.6 5.6 4.2 3.7 MHz min B
Gain Bandwidth Product G ≥ [phone] MHz min B
Peaking at a Gain of +1 V

O

≤ 0.5Vp-p 5 — — — dB typ C
Slew Rate G = +2, 2V Step 100 64 52 47 V/µs min B
Rise Time 0.5V Step 5.3 8.0 11 12.8 ns max B
Fall Time 0.5V Step 5.4 7.5 10 11.6 ns max B
Settling Time to 0.1% G = +2, 1V Step 17 28 38 42 ns max B
Spurious Free Dynamic Range V

O

= 2Vp-p, f = 5MHz 42 40 38 35 dBc min B
Input Voltage Noise f > 1MHz 6.0 6.8 7.6 7.9 nV/√Hz max B
Input Current Noise f > 1MHz 1.9 2.6 2.9 3.6 pA/√Hz max B
NTSC Differential Gain 0.5 — — — % typ C
NTSC Differential Phase 1.2 — — — degrees typ C

DC PERFORMANCE

Open-Loop Voltage Gain 62 56 50 46 dB min A
Input Offset Voltage 2.5 6 8 11 mV max A
Average Offset Voltage Drift — — — 50 µV/°C max B
Input Bias Current V

CM

= 2.0V 11 21 27 40 µA max B

Input Offset Current V

CM

= 2.0V 0.3 1 1.3 2 µA max B

Input Offset Current Drift — — — 7 nA/°C max B

INPUT

Least Positive Input Voltage –0.5 –0.1 –0.1 –0.1 V max B
Most Positive Input Voltage 4.0 3.7 3.7 3.5 V min A
Common-Mode Rejection Ratio (CMRR)

Input Referred 74 70 68 60 dB min A

Input Impedance

Differential-Mode 10 || 2.1 — — — kΩ || pF typ C
Common-Mode 400 || 1.2 — — — kΩ || pF typ C

OUTPUT

Least Positive Output Voltage R

L

= 1kΩ to 2.5V 0.03 0.06 0.09 0.12 V max B

R

L

= 150Ω to 2.5V 0.16 0.17 0.20 1.7 V max A

Most Positive Output Voltage R

L

= 1kΩ to 2.5V 4.87 4.8 4.7 4.6 V min B

R

L

= 150Ω to 2.5V 4.60 4.4 4.4 3.1 V min A
Current Output, Sourcing 80 25 20 5 mA min A
Current Output, Sinking 90 38 24 10 mA min A
Short-Circuit Current (output shorted to either supply) 100 — — — mA typ C
Closed-Loop Output Impedance G = +2, f ≤ 100kHz 0.2 — — — Ω typ C

DISABLE (OPA632 only)

On Voltage

(device enabled Low) 1.0 1.0 1.0 1.0 V min A

Off Voltage

(device disabled High) 3.7 3.8 4.0 4.2 V max A
On Disable Current (DIS pin) 70 110 120 120 µA max A
Off Disable Current (DIS pin) 0 — — — µA typ C
Disabled Quiescent Current 0 20 25 30 µA max A
Disable Time 100 — — — ns typ C
Enable Time 60———nstypC
Off Isolation f = 5MHz, Input to Output 70 — — — dB typ C

POWER SUPPLY

Minimum Operating Voltage — 2.7 2.7 2.7 V min A
Maximum Operating Voltage — 10.5 10.5 10.5 V max A
Maximum Quiescent Current V

S

= +5V 6 6.4 6.7 6.9 mA max A

Minimum Quiescent Current V

S

= +5V 6 5.8 5.5 4.8 mA min A

Power Supply Rejection Ratio (PSRR) Input Referred 59 52 49 48 dB min A

THERMAL CHARACTERISTICS

Specification: U, N

–40 to +85

°C typ C

Thermal Resistance

U SO-8 125 °C/W typ C
N SOT23-5, SOT23-6 150 °C/W typ C

NOTE: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information.

3

®

OPA631, OPA632

OPA631U, N
OPA632U, N

TYP GUARANTEED

0°C to

MIN/ TEST

PARAMETER CONDITIONS +25

°C +25°C70°C UNITS MAX

LEVEL

(1)

SPECIFICATIONS: VS = +3V

At TA = 25°C, G = +2 and RL = 150Ω to VS/2, unless otherwise noted (see Figure 2).

The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no
responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice.
No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product
for use in life support devices and/or systems.

AC PERFORMANCE (Figure 2)

Small-Signal Bandwidth G = +2, V

O

≤ 0.5Vp-p 61 45 35 MHz min B

G = +5, V

O

≤ 0.5Vp-p 15 11 9 MHz min B

G = +10, V

O

≤ 0.5Vp-p 7.7 4.6 4.0 MHz min B
Gain Bandwidth Product G ≥ [phone] MHz min B
Peaking at a Gain of +1 V

O

≤ 0.5Vp-p 5 — — dB typ C
Slew Rate 1V Step 95 52 46 V/µs min B
Rise Time 0.5V Step 5.6 9 11.3 ns max B
Fall Time 0.5V Step 5.6 9 11.3 ns max B
Settling Time to 0.1% 1V Step 40 63 85 ns max B
Spurious Free Dynamic Range V

O

= 1Vp-p, f = 5MHz 44 37 34 dBc min B
Input Voltage Noise f > 1MHz 6.2 7.0 7.8 nV/√Hz max B
Input Current Noise f > 1MHz 2.0 2.6 2.9 pA/√Hz max B

DC PERFORMANCE

Open-Loop Voltage Gain 60 54 50 dB min A
Input Offset Voltage 0.5 3.5 4 mV max A
Average Offset Voltage Drift ——45µV/°C max B
Input Bias Current V

CM

= 1.0V 12 21 26 µA max B

Input Offset Current V

CM

= 1.0V 0.3 1 1.3 µA max B

Input Offset Current Drift — — 2 nA/°C max B

INPUT

Least Positive Input Voltage –0.5 –0.3 –0.1 V max B
Most Positive Input Voltage 2 1.75 1.3 V min A
Common-Mode Rejection Ratio (CMRR) Input Referred 72 66 65 dB min A
Input Impedance

Differential-Mode 10 || 2.1 — — kΩ || p typ C
Common-Mode 400 || 1.2 — — kΩ || p typ C

OUTPUT

Least Positive Output Voltage R

L

= 1kΩ to 1.5V 0.03 0.05 0.05 V max A

R

L

= 150Ω to 1.5V 0.05 0.15 0.16 V max A

Most Positive Output Voltage R

L

= 1kΩ to 1.5V 2.95 2.85 2.84 V min A

R

L

= 150Ω to 1.5V 2.85 2.66 2.60 V min A
Current Output, Sourcing 55 21 14 mA min A
Current Output, Sinking 55 21 14 mA min A
Short Circuit Current (output shorted to either supply) 80 — — mA typ C
Closed-Loop Output Impedance Figure 2, f < 100kHz 0.2 — — Ω typ C

DISABLE (OPA632 only)

On Voltage (device enabled Low)

1.0 1.0 1.0 V min A

Off Voltage

(device disabled High)

1.7 1.8 1.8 V max A
On Disable Current (DIS pin) 66 100 110 µA max A
Off Disable Current (DIS pin) 0——µA typ C
Disabled Quiescent Current 0 20 25 µA max A
Disable Time 100 — — ns typ C
Enable Time 60 — — ns typ C
Off Isolation f = 5MHz, Input to Output 70 — — dB typ C

POWER SUPPLY

Minimum Operating Voltage — 2.7 2.7 V min A
Maximum Operating Voltage — 10.5 10.5 V max A
Maximum Quiescent Current V

S

= +3V 5.3 5.7 6.2 mA max A

Minimum Quiescent Current V

S

= +3V 5.3 5.0 4.8 mA min A

Power Supply Rejection Ratio (PSRR) Input Referred 57 50 48 dB min A

THERMAL CHARACTERISTICS

Specification: U, N

–40 to +85

°C typ C

Thermal Resistance

U SO-8 125 °C/W typ C
N SOT23-5, SOT23-6 150 °C/W typ C

NOTE: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.
(C) Typical value only for information.

4

®

OPA631, OPA632

1

2

3

4

8

7

6

5

NC

Inverting Input

Non-Inverting Input

GND

DIS (OPA632 only)

+V

S

Output

NC

PIN CONFIGURATIONS

Top View—OPA631, OPA632 SO-8

ABSOLUTE MAXIMUM RATINGS

Power Supply ................................................................................ +11V

DC

Internal Power Dissipation ....................................See Thermal Analysis

Differential Input Voltage .................................................................. ±1.2V

Input Voltage Range .................................................... –0.5 to +V

S

+ 0.3V

Storage Temperature Range: P, U, N ........................... –40°C to +125°C

Lead Temperature (soldering, 10s) .............................................. +300°C

Junction Temperature (T

J

) ........................................................... +175°C

ELECTROSTATIC
DISCHARGE SENSITIVITY

Electrostatic discharge can cause damage ranging from perfor­mance degradation to complete device failure. Burr-Brown Corpo­ration recommends that all integrated circuits be handled and stored
using appropriate ESD protection methods.

ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes
could cause the device not to meet published specifications.

1

2

3

6

5

4

Output

GND

Non-Inverting Input

+V

S

DIS

Inverting Input

A32

1

2

3

6

5

4

Pin Orientation/Package Marking

Top View—OPA632 SOT23-6

PACKAGE SPECIFIED
DRAWING TEMPERATURE PACKAGE ORDERING TRANSPORT

PRODUCT PACKAGE NUMBER

(1)

RANGE MARKING NUMBER

(2)

MEDIA

OPA631U SO-8 Surface-Mount 182 –40°C to +85°C OPA631U OPA631U Rails

"""""OPA631U/2K5 Tape and Reel

OPA631N 5-Lead SOT23-5 331 –40°C to +85°C A31 OPA631N/250 Tape and Reel

"""""OPA631N/3K Tape and Reel

OPA632U SO-8 Surface-Mount 182 –40°C to +85°C OPA632U OPA632U Rails

"""""OPA632U/2K5 Tape and Reel

OPA632N 6-Lead SOT23-6 332 –40°C to +85°C A32 OPA632N/250 Tape and Reel

"""""OPA632N/3K Tape and Reel

NOTES: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) Models with a slash (/ ) are
available only in Tape and Reel in the quantities indicated (e.g., /3K indicates 3000 devices per reel). Ordering 3000 pieces of “OPA632N/3K” will get a single 3000­piece Tape and Reel. For detailed Tape and Reel mechanical information, refer to Appendix B of Burr-Brown IC Data Book.

PACKAGE/ORDERING INFORMATION

1

2

3

6

4

Output

GND

Non-Inverting Input

+V

S

Inverting Input

A31

1

2

3

6

4

Pin Orientation/Package Marking

Top View—OPA631 SOT23-5

5

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +5V

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1).

6
3

0
–3
–6
–9

–12
–15
–18
–21
–24

SMALL-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Normalized Gain (dB)

1 10 100 300

VO = 200mVp-p

G = +10

G = +5

G = +2

12

9
6
3

0
–3
–6
–9

–12
–15
–18

LARGE-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Gain (dB)

1 10 100 300

VO = 0.2Vp-p

VO = 4Vp-p

VO = 2Vp-p

VO = 1Vp-p

SMALL-SIGNAL PULSE RESPONSE

Time (10ns/div)

Input and Output Voltage (50mV/div)

VO = 200mVp-p

V

O

V

IN

LARGE-SIGNAL DISABLE/ENABLE RESPONSE

Time (10ns/div)

Input and Output Voltage (500mV/div)

VO = 4Vp-p

V

O

V

IN

LARGE-SIGNAL DISABLE/ENABLE RESPONSE

Time (50ns/div)

Disable Voltage (1V/div)

Output Voltage (250mV/div)

VIN = 0.5V

OPA632 Only

V

O

V

DIS

–35
–40
–45
–50
–55
–60
–65
–70
–75
–80
–85

DISABLE FEEDTHROUGH vs FREQUENCY

Frequency (MHz)

1 10 100 1000

Feedthrough (dB)

OPA632 Only

V

DIS

= +5V

6

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.)

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1).

–30

–40

–50

–60

–70

–80

–90

1MHz 2nd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

2nd Harmonic Distortion (dBc)

RL = 150Ω

RL = 250Ω

–30

–40

–50

–60

–70

–80

–90

1MHz 3rd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

3rd Harmonic Distortion (dBc)

RL = 250Ω

RL = 150Ω

RL = 500Ω

–30

–40

–50

–60

–70

–80

–90

5MHz 2nd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

2nd Harmonic Distortion (dBc)

RL = 150Ω

RL = 250Ω

RL = 500Ω

–30

–40

–50

–60

–70

–80

–90

5MHz 3rd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

3rd Harmonic Distortion (dBc)

RL = 250Ω

RL = 150Ω

RL = 500Ω

–30

–40

–50

–60

–70

–80

–90

10MHz 2nd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

2nd Harmonic Distortion (dBc)

RL = 500Ω

RL = 150Ω

RL = 250Ω

–30

–40

–50

–60

–70

–80

–90

10MHz 3rd HARMONIC DISTORTION

vs OUTPUT VOLTAGE

Output Voltage (Vp-p)

10.1 4

3rd Harmonic Distortion (dBc)

RL = 500Ω

RL = 150Ω

RL = 250Ω

7

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.)

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1).

–30

–40

–50

–60

–70

–80

–90

2nd HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

10.1 10

2nd Harmonic Distortion (dBc)

VO = 2Vp-p

R

L

= 150Ω

G = +2

G = +5

G = +10

–30

–40

–50

–60

–70

–80

–90

3rd HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

10.1 10

3rd Harmonic Distortion (dBc)

VO = 2Vp-p

R

L

= 150Ω

G = +2

G = +5

G = +10

–30

–40

–50

–60

–70

–80

–90

HARMONIC DISTORTION vs LOAD RESISTANCE

R

L

(Ω)

100 200 300 400 500

Harmonic Distortion (dBc)

VO = 2Vp-p

f

O

= 5MHz

3rd Harmonic Distortion

2nd Harmonic Distortion

80
75
70
65
60
55
50
45
40
35
30

CMRR AND PSRR vs FREQUENCY

Frequency (Hz)

100 1k 10k 100k 1M 10M

Rejection Ratio, Input Referred (dB)

CMRR

PSRR

100

10

1

INPUT NOISE DENSITY vs FREQUENCY

Frequency (Hz)

100 1k 10k 100k 1M 10M

Voltage Noise (nV/√Hz)

Current Noise (pA/√Hz)

Voltage Noise, eni = 6.0nV/√Hz

Current Noise, ini = 1.9pA/√Hz

–30

–40

–50

–60

–70

–80

–90

TWO-TONE, 3rd-ORDER

INTERMODULATION SPURIOUS

Single-Tone Load Power (dBm)

–16 –14 –12 –10 –8 –6 –4 –2 0

3rd-Order Spurious Level (dBc)

Load Power at

Matched 50Ω Load

fO = 1MHz

fO = 5MHz

fO = 10MHz

8

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.)

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1).

1000

100

10

1

R

S

vs CAPACITIVE LOAD

Capacitive Load (pF)

1 10 100 1000

R

S

(Ω)

2
1

0
–1
–2
–3
–4
–5
–6
–7
–8

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (MHz)

1 10 100 300

Normalized Gain (dB)

R

S

OPA63x

V

O

1kΩC

L

+VS/2

CL = 100pF

CL = 1000pF

CL = 10pF

100

90
80
70
60
50
40
30
20
10

0
–10
–20

OPEN-LOOP GAIN AND PHASE

Frequency (Hz)

1k 10k 100k 1M 10M 100M 1G

Open-Loop Gain (dB)

0
–30
–60
–90
–120
–150
–180
–210
–240
–270
–300
–330
–360

Open-Loop Phase (°)

Open-Loop Phase

Open-Loop Gain

100

10

1

0.1

CLOSED-LOOP OUTPUT IMPEDANCE

vs FREQUENCY

Frequency (Hz)

1k 10k 100k 1M 10M 100M

Output Impedance (Ω)

G = +1

R

F

= 25Ω

5.0

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

0.0

INPUT DC ERRORS vs TEMPERATURE

Temperature (°C)

–40 –20 0 20 40 60 80 100

Input Offset Voltage (mV)

20
18
16
14
12
10
8
6
4
2
0

Input Bias Current (µA)

10x Input Offset Current (µA)

Input Offset Voltage

Input Bias Current

10X Input Offset Current

12

10

8

6

4

2

0

POWER SUPPLY AND OUTPUT CURRENT

vs TEMPERATURE

Temperature (°C)

–40 –20 0 20 40 60 80 100

Quiescent Supply Current (mA)

120

100

80

60

40

20

0

Output Current (mA)

Sinking Output Current

Sourcing Output Current

Quiescent Supply Current

9

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +3V

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 2).

6
3

0
–3
–6
–9

–12
–15
–18
–21
–24

SMALL-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Normalized Gain (dB)

1 10 100 300

VO = 200mVp-p

G = +10

G = +2

G = +5

12

9
6
3

0
–3
–6
–9

–12
–15
–18

LARGE-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Gain (dB)

1 10 100 300

VO = 2Vp-p

VO = 200mVp-p

VO = 1Vp-p

–30

–40

–50

–60

–70

–80

–90

2nd HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

10.1 10

2nd Harmonic Distortion (dBc)

VO = 1Vp-p

R

L

= 150Ω

G = +2

G = +5

G = +10

–30

–40

–50

–60

–70

–80

–90

3rd HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

10.1 10

3rd Harmonic Distortion (dBc)

VO = 1Vp-p

R

L

= 150Ω

G = +2

G = +5

G = +10

–30

–40

–50

–60

–70

–80

–90

HARMONIC DISTORTION vs LOAD RESISTANCE

R

L

(Ω)

100 200 300 400 500

Harmonic Distortion (dBc)

VO = 1Vp-p

f

O

= 5MHz

3rd Harmonic Distortion

2nd Harmonic Distortion

–30

–40

–50

–60

–70

–80

–90

TWO-TONE, 3rd-ORDER

INTERMODULATION SPURIOUS

Single-Tone Load Power (dBm)

–16 –14 –12 –10 –8 –6 –4

3rd-Order Spurious Level (dBc)

Load Power at

Matched 50Ω Load

fO = 10MHz

fO = 1MHz

fO = 5MHz

10

®

OPA631, OPA632

TYPICAL PERFORMANCE CURVES: VS = +3V (Cont.)

At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 2).

1000

100

10

1

R

S

vs CAPACITIVE LOAD

Capacitive Load (pF)

1 10 100 1000

R

S

(Ω)

3.0

2.9

2.8

2.7

2.6

2.5

2.4

2.3

2.2

2.1

2.0

OUTPUT SWING vs LOAD RESISTANCE

R

L

(Ω)

50 100 1000

Maximum Output Voltage (V)

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1

0.0

Minimum Output Voltage (V)

Maximum V

O

Minimum V

O

6
3

0
–3
–6
–9

–12
–15
–18
–21
–24

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (MHz)

1 10 100 300

Normalized Gain (dB)

VO = 0.2Vp-p

R

S

OPA63x

V

O

1kΩC

L

+VS/2

CL = 1000pF

CL = 10pF

CL = 100pF

10

9
8
7
6
5
4
3
2
1
0

SUPPLY AND OUTPUT CURRENTS

vs SUPPLY VOLTAGE

Supply Voltage (V)

345678910

Quiescent Supply Current (mA)

200
180
160
140
120
100
80
60
40
20
0

Output Current (mA)

Quiescent Supply Current

Output Current, Sourcing

Output Current, Sinking

120

100

80

60

40

20

0

SLEW RATE AND GAIN BANDWIDTH PRODUCT

vs SUPPLY VOLTAGE

Supply Voltage (V)

345678910

Slew Rate (V/µs)

120

100

80

60

40

20

0

Gain Bandwidth Product (MHz)

Slew Rate

Gain Bandwidth Product

11

®

OPA631, OPA632

APPLICATIONS INFORMATION

WIDEBAND VOLTAGE-FEEDBACK OPERATION

The OPA631 and OPA632 are unity-gain stable, very high­speed, voltage-feedback op amps designed for single-supply
operation (+3V to +5V). The input stage supports input
voltages below ground, and within 1.0V of the positive
supply. The complementary common-emitter output stage
provides an output swing to within 30mV of ground and
130mV of the positive supply. They are compensated to
provide stable operation with a wide range of resistive loads.
The OPA632’s internal disable circuitry is intended to mini­mize system power when disabled.

Figure 1 shows the AC-coupled, gain of +2 configuration
used for the +5V Specifications and Typical Performance
Curves. For test purposes, the input impedance is set to 50Ω
with a resistor to ground. Voltage swings reported in the
Specifications are taken directly at the input and output pins.
For the circuit of Figure 1, the total effective load on the
output at high frequencies is 150Ω || 1500Ω. The disable pin
(OPA635 only) needs to be driven by a low impedance
source, such as a CMOS inverter. The 1.50kΩ resistors at
the non-inverting input provide the common-mode bias
voltage. Their parallel combination equals the DC resistance
at the inverting input, minimizing the output DC offset.

SINGLE-SUPPLY ADC CONVERTER INTERFACE

The front page shows a DC-coupled, single-supply ADC
driver circuit. Many systems are now requiring +3V supply
capability of both the ADC and its driver. The OPA632
provides excellent performance in this demanding applica­tion. Its large input and output voltage ranges, and low
distortion, support converters such as the ADS901 shown in
this figure. The input level-shifting circuitry was designed
so that VIN can be between 0V and 0.5V, while delivering an
output voltage of 1V to 2V for the ADS901. Both the
OPA632 and ADS901 have power reduction pins with the
same polarity for those systems that need to conserve power.

DC LEVEL SHIFTING

Figure 3 shows a DC-coupled non-inverting amplifier that
level-shifts the input up to accommodate the desired output
voltage range. Given the desired signal gain (G), and the
amount V

OUT

needs to be shifted up (∆V

OUT

) when VIN is at
the center of its range, the following equations give the
resistor values that produce the best DC offset.

NG = G + ∆V

OUT/VS

R1 = R4/G
R2 = R4/(NG – G)
R3 = R4/(NG –1)

where:

NG = 1 + R4/R3 (Noise Gain)
V

OUT

= (G)VIN + (NG – G)V

S

Make sure that VIN and V

OUT

stay within the specified input

and output voltage ranges.

Figure 2 shows the DC-coupled, gain of +2 configuration
used for the +3V Specifications and Typical Performance
Curves. For test purposes, the input impedance is set to 50Ω
with a resistor to ground. Though not strictly a “rail-to-rail”
design, these parts come very close, while maintaining
excellent performance. They will deliver ≈ 2.9Vp-p on a
single +3V supply with 61MHz bandwidth. The 374Ω and

2.26kΩ resistors at the input level-shift VIN so that V

OUT

is
within the allowed output voltage range when VIN = 0. See
the Typical Performance Curves for information on driving
capacitive loads.

FIGURE 1. AC-Coupled Signal—Resistive Load to Supply

Midpoint.

FIGURE 2. DC-Coupled Signal—Resistive Load to Supply

Midpoint.

OPA63x

+VS = 5V

DIS (OPA632 only)

V

OUT

53.6Ω

V

IN

750Ω

1.50kΩ

1.50kΩ

R

L

150Ω

+V

S

2

750Ω

6.8µF
+

0.1µF

0.1µF

0.1µF
+

OPA63x

+VS = 3V

DIS (OPA632 only)

V

OUT

57.6Ω

V

IN

562Ω

374Ω

2.26kΩ

R

L

150Ω

+V

S

2

750Ω

6.8µF
+

0.1µF
+

12

®

OPA631, OPA632

The front page circuit is a good example of this type of
application. It was designed to take VIN between 0V and

0.5V, and produce V

OUT

between 1V and 2V, when using a

+3V supply. This means G = 2.00, and ∆V

OUT

= 1.50V – G

• 0.25V = 1.00V. Plugging into the above equations gives:
NG = 2.33, R1 = 375Ω, R2 = 2.25kΩ, and R3 = 563Ω. The
resistors were changed to the nearest standard values.

NON-INVERTING AMPLIFIER WITH
REDUCED PEAKING

Figure 4 shows a non-inverting amplifier that reduces peak­ing at low gains. The resistor RC compensates the OPA631
or OPA632 to have higher Noise Gain (NG), which reduces
the AC response peaking (typically 5dB at G = +1 without
RC) without changing the DC gain. VIN needs to be a low
impedance source, such as an op amp. The resistor values
are low to reduce noise. Using both RT and RF helps
minimize the impact of parasitic impedances.

A unity gain buffer can be designed by selecting RT = RF =

20.0Ω and RC = 40.2Ω (do not use RG). This gives a Noise
Gain of 2, so its response will be similar to the Characteris­tics Plots with G = +2 which typically gives a flat frequency
response, but with less bandwidth.

DESIGN-IN TOOLS

DEMONSTRATION BOARDS

Two PC boards are available to assist in the initial evaluation
of circuit performance using the OPA631 and OPA632 in
their three package styles. These are available free as an
unpopulated PC board delivered with descriptive documen­tation. The summary information for these boards is shown
below:

FIGURE 3. DC Level-Shifting Circuit.

OPA63x

V

OUT

V

IN

R

G

R

T

R

F

R

C

FIGURE 4. Compensated Non-Inverting Amplifier.

The Noise Gain can be calculated as follows:

Contact the Burr-Brown Applications support line to request
any of these boards.

OPERATING SUGGESTIONS

OPTIMIZING RESISTOR VALUES

Since the OPA631 and OPA632 are voltage feedback op
amps, a wide range of resistor values may be used for the
feedback and gain setting resistors. The primary limits on
these values are set by dynamic range (noise and distortion)
and parasitic capacitance considerations. For a non-inverting
unity gain follower application, the feedback connection
should be made with a 25Ω resistor, not a direct short (see
Figure 4). This will isolate the inverting input capacitance
from the output pin and improve the frequency response
flatness. Usually, for G > 1 application, the feedback resistor
value should be between 200Ω and 1.5kΩ. Below 200Ω, the
feedback network will present additional output loading
which can degrade the harmonic distortion performance.
Above 1.5kΩ, the typical parasitic capacitance (approxi­mately 0.2pF) across the feedback resistor may cause unin­tentional band-limiting in the amplifier response.

A good rule of thumb is to target the parallel combination of
RF and RG (Figure 1) to be less than approximately 400Ω.
The combined impedance RF || RG interacts with the invert­ing input capacitance, placing an additional pole in the
feedback network and thus, a zero in the forward response.
Assuming a 3pF total parasitic on the inverting node, hold­ing RF || RG <400Ω will keep this pole above 130MHz. By
itself, this constraint implies that the feedback resistor R

F

can increase to several kΩ at high gains. This is acceptable
as long as the pole formed by RF and any parasitic capaci­tance appearing in parallel is kept out of the frequency range
of interest.

BOARD LITERATURE

PART REQUEST

PRODUCT PACKAGE NUMBER NUMBER

OPA63xU 8-Pin SO-8 DEM-OPA68xU MKT-351
OPA63xN 5-Pin SOT23-5 DEM-OPA6xxN MKT-348

6-Pin SOT23-6

OPA63x

+V

S

V

OUT

V

IN

R

3

R

2

R

1

R

4

G

R
R

G

RRG

R

NG G G

F
G

TF

C

1

2

1

12

11=+

=+

+=/

13

®

OPA631, OPA632

BANDWIDTH VS GAIN: NON-INVERTING OPERATION

Voltage feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90°, as it
does in high gain configurations. At low gains (increased
feedback factors), most amplifiers will exhibit a more com­plex response with lower phase margin. The OPA631 and
OPA632 are compensated to give a slightly peaked response
in a non-inverting gain of 2 (Figure 1). This results in a
typical gain of +2 bandwidth of 75MHz, far exceeding that
predicted by dividing the 68MHz GBP by 2. Increasing the
gain will cause the phase margin to approach 90° and the
bandwidth to more closely approach the predicted value of
(GBP/NG). At a gain of +10, the 7.6MHz bandwidth shown
in the Typical Specifications is close to that predicted using
the simple formula and the typical GBP.

The OPA631 and OPA632 exhibit minimal bandwidth re­duction going to +3V single supply operation as compared
with +5V supply. This is because the internal bias control
circuitry retains nearly constant quiescent current as the total
supply voltage between the supply pins is changed.

INVERTING AMPLIFIER OPERATION

Since the OPA631 and OPA632 are general purpose,
wideband voltage feedback op amps, all of the familiar op
amp application circuits are available to the designer. Figure
5 shows a typical inverting configuration where the I/O
impedances and signal gain from Figure 1 are retained in an
inverting circuit configuration. Inverting operation is one of
the more common requirements and offers several perfor­mance benefits. The inverting configuration shows improved
slew rate and distortion. It also allows the input to be biased
at VS/2 without any headroom issues. The output voltage can
be independently moved to be within the output voltage
range with coupling capacitors, or bias adjustment resistors.

In the inverting configuration, three key design consider­ation must be noted. The first is that the gain resistor (RG)
becomes part of the signal channel input impedance. If input
impedance matching is desired (which is beneficial when­ever the signal is coupled through a cable, twisted pair, long
PC board trace or other transmission line conductor), R

G

may be set equal to the required termination value and R

F

adjusted to give the desired gain. This is the simplest
approach and results in optimum bandwidth and noise per­formance. However, at low inverting gains, the resultant
feedback resistor value can present a significant load to the
amplifier output. For an inverting gain of 2, setting RG to
50Ω for input matching eliminates the need for RM but
requires a 100Ω feedback resistor. This has the interesting
advantage that the noise gain becomes equal to 2 for a 50Ω
source impedance—the same as the non-inverting circuits
considered above. However, the amplifier output will now
see the 100Ω feedback resistor in parallel with the external

load. In general, the feedback resistor should be limited to
the 200Ω to 1.5kΩ range. In this case, it is preferable to
increase both the RF and RG values as shown in Figure 5, and
then achieve the input matching impedance with a third
resistor (RM) to ground. The total input impedance becomes
the parallel combination of RG and RM.

The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes
part of the noise gain equation and hence influences the
bandwidth. For the example in Figure 5, the RM value
combines in parallel with the external 50Ω source imped­ance, yielding an effective driving impedance of 50Ω ||
576Ω = 26.8Ω. This impedance is added in series with R

G

for calculating the noise gain. The resultant is 2.87 for
Figure 5, as opposed to only 2 if RM could be eliminated as
discussed above. The bandwidth will therefore be lower for
the gain of –2 circuit of Figure 5 (NG = +3) than for the
gain of +2 circuit of Figure 1.

The third important consideration in inverting amplifier
design is setting the bias current cancellation resistors on
the non-inverting input (a parallel combination of RT =
263Ω). If this resistor is set equal to the total DC resistance
looking out of the inverting node, the output DC error, due
to the input bias currents, will be reduced to (Input Offset
Current) • RF. If the 50Ω source impedance is DC-coupled
in Figure 5, the total resistance to ground between the
inverting input and the source will be 401Ω. Combining
this in parallel with the feedback resistor gives the RT =
263Ω used in this example. To reduce the additional high
frequency noise introduced by this resistor, and power
supply feedthrough, RT is bypassed with a capacitor. As
long as RT < 400Ω, its noise contribution will be minimal.
As a minimum, the OPA631 and OPA632 require an R

T

FIGURE 5. Gain of –2 Example Circuit.

OPA63x

50Ω

R

F

750Ω

R

G

374Ω

2R

T

523Ω

R

M

57.6Ω

Source

DIS

+5V

2R

T

523Ω

R

O

50Ω

0.1µF 6.8µF

+

0.1µF
50Ω Load

14

®

OPA631, OPA632

value of 50Ω to damp out parasitic-induced peaking—a
direct short to ground on the non-inverting input runs the
risk of a very high frequency instability in the input stage.

OUTPUT CURRENT AND VOLTAGE

The OPA631 and OPA632 provide outstanding output volt­age capability. Under no-load conditions at +25°C, the
output voltage typically swings closer than 130mV to either
supply rail; the guaranteed swing limit is within 400mV of
either rail (VS = +5V).

The minimum specified output voltage and current specifi­cations over temperature are set by worst-case simulations at
the cold temperature extreme. Only at cold start-up will the
output current and voltage decrease to the numbers shown in
the guaranteed tables. As the output transistors deliver power,
their junction temperatures will increase, decreasing their
VBE’s (increasing the available output voltage swing) and
increasing their current gains (increasing the available out­put current). In steady-state operation, the available output
voltage and current will always be greater than that shown
in the over-temperature specifications since the output stage
junction temperatures will be higher than the minimum
specified operating ambient.

To maintain maximum output stage linearity, no output
short-circuit protection is provided. This will not normally
be a problem since most applications include a series match­ing resistor at the output that will limit the internal power
dissipation if the output side of this resistor is shorted to
ground.

DRIVING CAPACITIVE LOADS

One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an A/D converter—including
additional external capacitance which may be recommended
to improve A/D linearity. A high speed, high open-loop gain
amplifier like the OPA631 and OPA632 can be very suscep­tible to decreased stability and closed-loop response peaking
when a capacitive load is placed directly on the output pin.
When the primary considerations are frequency response
flatness, pulse response fidelity and/or distortion, the sim­plest and most effective solution is to isolate the capacitive
load from the feedback loop by inserting a series isolation
resistor between the amplifier output and the capacitive
load.

The Typical Performance Curves show the recommended
RS versus capacitive load and the resulting frequency re­sponse at the load. Parasitic capacitive loads greater than
2pF can begin to degrade the performance of the OPA631
and OPA632. Long PC board traces, unmatched cables, and
connections to multiple devices can easily exceed this value.
Always consider this effect carefully, and add the recom­mended series resistor as close as possible to the output pin
(see Board Layout Guidelines).

The criterion for setting this RS resistor is a maximum
bandwidth, flat frequency response at the load. For a gain of
+2, the frequency response at the output pin is already
slightly peaked without the capacitive load, requiring rela­tively high values of RS to flatten the response at the load.
Increasing the noise gain will also reduce the peaking (see
Figure 4).

DISTORTION PERFORMANCE

The OPA631 and OPA632 provide good distortion perfor­mance into a 150Ω load. Relative to alternative solutions, it
provides exceptional performance into lighter loads and/or
operating on a single +3V supply. Generally, the 3rd har­monic will dominate the distortion. Focusing then on the 3rd
harmonic, increasing the load impedance improves distor­tion directly. Remember that the total load includes
the feedback network; in the non-inverting configuration
(Figure 1) this is sum of RF + RG, while in the inverting
configuration, it is just RF.

NOISE PERFORMANCE

High slew rate, unity gain stable, voltage feedback op amps
usually achieve their slew rate at the expense of a higher
input noise voltage. The 6.0nV/√Hz input voltage noise for
the OPA631 and OPA632 is, however, much lower than
comparable amplifiers. The input-referred voltage noise,
and the two input-referred current noise terms (1.9pA/√Hz),
combine to give low output noise under a wide variety of
operating conditions. Figure 6 shows the op amp noise
analysis model with all the noise terms included. In this
model, all noise terms are taken to be noise voltage or
current density terms in either nV/√Hz or pA/√Hz.

The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 1 shows the general form for the
output noise voltage using the terms shown in Figure 6.

FIGURE 6. Noise Analysis Model.

4kT

R

G

R

G

R

F

R

S

OPA63x

I

BI

E

O

I

BN

4kT = 1.6 • 10

–20

J

at 290°K

E

RS

E

NI

4kTR

S

√

4kTRF√

15

®

OPA631, OPA632

Equation 1:

Dividing this expression by the noise gain (NG = (1+RF/RG))
will give the equivalent input-referred spot noise voltage at
the non-inverting input, as shown in Equation 2.

Equation 2:

Evaluating these two equations for the circuit and compo­nent values shown in Figure 1 will give a total output spot
noise voltage of 13.1nV/√Hz and a total equivalent input
spot noise voltage of 6.6nV/√Hz. This is including the noise
added by the resistors. This total input-referred spot noise
voltage is not much higher than the 6.0nV/√Hz specification
for the op amp voltage noise alone. This will be the case as
long as the impedances appearing at each op amp input are
limited to the previously recommend maximum value of
400Ω, and the input attenuation is low. Since the resistor­induced noise is relatively negligible, additional capacitive
decoupling across the bias current cancellation resistor (RT)
for the inverting op amp configuration of Figure 5 is not
required.

DC ACCURACY AND OFFSET CONTROL

The balanced input stage of a wideband voltage feedback op
amp allows good output DC accuracy in a wide variety of
applications. The power supply current trim for the OPA631
and OPA632 gives even tighter control than comparable
products. Although the high-speed input stage does require
relatively high input bias current (typically 11µA out of each
input terminal), the close matching between them may be
used to reduce the output DC error caused by this current.
This is done by matching the DC source resistances appear­ing at the two inputs. Evaluating the configuration of Figure
1 (which has matched DC input resistances), using worst­case +25°C input offset voltage and current specifications,
gives a worst-case output offset voltage equal to: (NG = non­inverting signal gain at DC)

±(NG • V

OS(MAX)

) ± (RF • I

OS(MAX)

)

= ±(1 • 6.0mV) ± (750Ω • 2.0µA)
= ±6.8mV = Output Offset Range for Figure 1

A fine scale output offset null, or DC operating point
adjustment, is often required. Numerous techniques are
available for introducing DC offset control into an op amp
circuit. Most of these techniques are based on adding a DC
current through the feedback resistor. In selecting an offset
trim method, one key consideration is the impact on the
desired signal path frequency response. If the signal path is
intended to be non-inverting, the offset control is best
applied as an inverting summing signal to avoid interaction
with the signal source. If the signal path is intended to be

inverting, applying the offset control to the non-inverting
input may be considered. Bring the DC offsetting current
into the inverting input node through resistor values that are
much larger than the signal path resistors. This will insure
that the adjustment circuit has minimal effect on the loop
gain and hence the frequency response.

DISABLE OPERATION

The OPA632 provides an optional disable feature that may
be used either to reduce system power or to implement a
simple channel multiplexing operation. To disable, the con­trol pin must be asserted HIGH. Figure 7 shows a simplified
internal circuit for the disable control feature.

In normal operation, base current to Q1 is provided through
the DIS pin and the 50kΩ resistor.

EN= E

NI

2

+ IBNR

S

()

2

+4kTRS+

I

BIRF

NG







2

+

4kTR

F

NG

EO= E

NI

2

+ IBNR

S

()

2

+4kTR

S

()

NG2+ IBIR

F

()

2

+4kTRFNG

50kΩ

IS Control

V

DIS

+V

S

Q1

FIGURE 7. Simplified Disable Control Circuit (OPA632).

One key parameter in disable operation is the output glitch
when switching in and out of the disabled mode.

The transition edge rate (dv/dt) of the DIS control line will
influence this glitch. Adding a simple RC filter into the DIS
pin from a higher speed logic line will reduce the glitch. If
extremely fast transition logic is used, a 1kΩ series resistor
will provide adequate band limiting using just the parasitic
input capacitance on the DIS pin while still ensuring ad­equate logic level swing.

THERMAL ANALYSIS

Maximum desired junction temperature will set the maxi­mum allowed internal power dissipation as described below.
In no case should the maximum junction temperature be
allowed to exceed 175°C.

Operating junction temperature (TJ) is given by TA + PD•

θ

JA

.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is

16

®

OPA631, OPA632

simply the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for resistive load
connected to mid-supply (VS/2), be at a maximum when the
output is fixed at a voltage equal to VS/4 or 3VS/4. Under this
condition, PDL = V

S

2

/(16 • RL), where RL includes feedback

network loading.
Note that it is the power in the output stage and not into the

load that determines internal power dissipation.
As a worst-case example, compute the maximum TJ using an

OPA632 (SOT23-6 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85°C and driving a 150Ω load at mid-supply.

PD = 10V • 6.9mA + 52/(16 • (150Ω || 1500Ω)) = 80mW
Maximum TJ = +85°C + (0.08W • 150°C/W) = 97°C.

Although this is still well below the specified maximum
junction temperature, system reliability considerations may
require lower guaranteed junction temperatures. The highest
possible internal dissipation will occur if the load requires
current to be forced into the output at high output voltages
or sourced from the output at low output voltages. This puts
a high current through a large internal voltage drop in the
output transistors.

BOARD LAYOUT GUIDELINES

Achieving optimum performance with a high frequency
amplifier like the OPA631 and OPA632 requires careful
attention to board layout parasitics and external component
types. Recommendations that will optimize performance
include:

a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the output
and inverting input pins can cause instability: on the non­inverting input, it can react with the source impedance to
cause unintentional bandlimiting. To reduce unwanted ca­pacitance, a window around the signal I/O pins should be
opened in all of the ground and power planes around those
pins. Otherwise, ground and power planes should be unbro­ken elsewhere on the board.

b) Minimize the distance (<0.25") from the power supply
pins to high frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power supply
connections should always be decoupled with these capaci­tors. An optional supply decoupling capacitor (0.1µF) across
the two power supplies (for bipolar operation) will improve
2nd harmonic distortion performance. Larger (2.2µF to

6.8µF) decoupling capacitors, effective at lower frequency,
should also be used on the main supply pins. These may be
placed somewhat farther from the device and may be shared
among several devices in the same area of the PC board.

c) Careful selection and placement of external compo­nents will preserve the high frequency performance.

Resistors should be a very low reactance type. Surface­mount resistors work best and allow a tighter overall layout.
Metal film or carbon composition axially-leaded resistors
can also provide good high frequency performance. Again,
keep their leads and PC board traces as short as possible.
Never use wirewound type resistors in a high frequency
application. Since the output pin and inverting input pin are
the most sensitive to parasitic capacitance, always position
the feedback and series output resistor, if any, as close as
possible to the output pin. Other network components, such
as non-inverting input termination resistors, should also be
placed close to the package. Where double-side component
mounting is allowed, place the feedback resistor directly
under the package on the other side of the board between the
output and inverting input pins. Even with a low parasitic
capacitance shunting the external resistors, excessively high
resistor values can create significant time constants that can
degrade performance. Good axial metal film or surface­mount resistors have approximately 0.2pF in shunt with the
resistor. For resistor values > 1.5kΩ, this parasitic capaci­tance can add a pole and/or zero below 500MHz that can
effect circuit operation. Keep resistor values as low as
possible consistent with load driving considerations. The
750Ω feedback used in the typical performance specifica­tions is a good starting point for design. See Figure 4 for the
unity gain follower application.

d) Connections to other wideband devices on the board
may be made with short direct traces or through on-board
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set RS from the
plot of Recommended RS vs Capacitive Load. Low parasitic
capacitive loads (< 5pF) may not need an RS since the
OPA631 and OPA632 are nominally compensated to oper­ate with a 2pF parasitic load. Higher parasitic capacitive
loads without an RS are allowed as the signal gain increases
(increasing the unloaded phase margin) If a long trace is
required, and the 6dB signal loss intrinsic to a doubly
terminated transmission line is acceptable, implement a
matched impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook for
microstrip and stripline layout techniques). A 50Ω environ­ment is normally not necessary on board, and in fact, a
higher impedance environment will improve distortion as
shown in the distortion versus load plots. With a character­istic board trace impedance defined (based on board mate­rial and trace dimensions), a matching series resistor into the
trace from the output of the OPA631 and OPA632 is used
as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating
impedance will be the parallel combination of the shunt
resistor and the input impedance of the destination device;
this total effective impedance should be set to match the
trace impedance. If the 6dB attenuation of a doubly termi-

17

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OPA631, OPA632

nated transmission line is unacceptable, a long trace can be
series-terminated at the source end only. Treat the trace as a
capacitive load in this case and set the series resistor value
as shown in the plot of Recommended RS vs Capacitive
Load. This will not preserve signal integrity as well as a
doubly terminated line. If the input impedance of the desti­nation device is low, there will be some signal attenuation
due to the voltage divider formed by the series output into
the terminating impedance.

e) Socketing a high speed part is not recommended. The
additional lead length and pin-to-pin capacitance introduced
by the socket can create an extremely troublesome parasitic
network which can make it almost impossible to achieve a
smooth, stable frequency response. Best results are obtained
by soldering the OPA631 and OPA632 onto the board.

INPUT AND ESD PROTECTION

The OPA631 and OPA632 are is built using a very high
speed complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very small
geometry devices. These breakdowns are reflected in the
Absolute Maximum Ratings table. All device pins are pro­tected with internal ESD protection diodes to the power
supplies as shown in Figure 8.

These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with ±15V supply
parts driving into the OPA631 and OPA632), current-limit­ing series resistors should be added into the two inputs. Keep
these resistor values as low as possible since high values
degrade both noise performance and frequency response.

External

Pin

+V

CC

–V

CC

Internal
Circuitry

FIGURE 8. Internal ESD Protection.