Datasheet OP295GP, OP495GP, OP295GS Datasheet (Analog Devices)

OUT A

–IN A

+IN A

V–

V+

OUT B

–IN B

+IN B

1

2

3

4

5

6

7

8

OP295

OUT A

–IN A

+IN A

V–

V+

OUT B

–IN B

+IN B

1

2

3

4

5

6

7

8

OP295

C

D

Dual/Quad Rail-to-Rail

a

FEATURES
Rail-to-Rail Output Swing
Single-Supply Operation: 3 V to 36 V
Low Offset Voltage: 300 ␮V
Gain Bandwidth Product: 75 kHz
High Open-Loop Gain: 1,000 V/mV
Unity-Gain Stable
Low Supply Current/Per Amplifier: 150 ␮A max

APPLICATIONS
Battery-Operated Instrumentation
Servo Amplifiers
Actuator Drives
Sensor Conditioners
Power Supply Control

GENERAL DESCRIPTION

Rail-to-rail output swing combined with dc accuracy are the key
features of the OP495 quad and OP295 dual CBCMOS operational
amplifiers. By using a bipolar front end, lower noise and higher
accuracy than that of CMOS designs has been achieved. Both input
and output ranges include the negative supply, providing the user
“zero-in/zero-out” capability. For users of 3.3 V systems such as
lithium batteries, the OP295/OP495 is specified for 3 V operation.

Maximum offset voltage is specified at 300 mV for 5 V operation,
and the open-loop gain is a minimum of 1000 V/mV. This yields
performance that can be used to implement high accuracy systems,
even in single-supply designs.

The ability to swing rail-to-rail and supply 15 mA to the load makes
the OP295/OP495 an ideal driver for power transistors and “H”
bridges. This allows designs to achieve higher efficiencies and to
transfer more power to the load than previously possible without
the use of discrete components. For applications that require

Operational Amplifiers

OP295/OP495

PIN CONNECTIONS

8-Lead Narrow-Body SO 8-Lead Epoxy DIP

(S Suffix) (P Suffix)

14-Lead Epoxy DIP 16-Lead SO (300 Mil)

(P Suffix) (S Suffix)

1

OUT A

2

–IN A

3

+IN A

4

V+

OP495

5

+IN B

6

–IN B

OUT B

7

driving inductive loads, such as transformers, increases in efficiency
are also possible. Stability while driving capacitive loads is another
benefit of this design over CMOS rail-to-rail amplifiers. This is
useful for driving coax cable or large FET transistors. The
OP295/OP495 is stable with loads in excess of 300 pF.

The OP295 and OP495 are specified over the extended industrial
(–40∞C to +125∞C) temperature range. OP295s are available in
8-lead plastic DIP plus SO-8 surface-mount packages. OP495s are
available in 14-lead plastic and SO-16 surface-mount packages.
Contact your local sales office for MIL-STD-883 data sheet.

14

OUT

13

–IN D

12

+IN D

11

V–

10

+IN C

9

–IN C

8

OUT

1

OUT A

–IN A

2

3

+IN A

4

V+

5

+IN B

6

–IN B

7

OUT B

8

NC

NC = NO CONNECT

OP495

TOP VIEW

(Not to Scale)

16

15

14

13

12

11

10

9

OUT D

–IN D

+IN D

V–

+IN C

–IN C

OUT C

NC

REV. C

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
that may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] © Analog Devices, Inc., 2002

OP295/OP495–SPECIFICATIONS

ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Unit

INPUT CHARACTERISTICS

Offset Voltage V

Input Bias Current I

Input Offset Current I

Input Voltage Range V
Common-Mode Rejection Ratio CMRR 0 V £ VCM £ 4.0 V, –40∞C £ TA £ +125∞C90110 dB
Large Signal Voltage Gain A

Offset Voltage Drift DVOS/DT 15 mV/∞C

OUTPUT CHARACTERISTICS

Output Voltage Swing High V

Output Voltage Swing Low V

Output Current I

POWER SUPPLY

Power Supply Rejection Ratio PSRR ± 1.5 V £ VS £ ± 15 V 90 110 dB

Supply Current Per Amplifier I

DYNAMIC PERFORMANCE

Skew Rate SR RL = 10 kW 0.03 V/ms
Gain Bandwidth Product GBP 75 kHz
Phase Margin q

NOISE PERFORMANCE

Voltage Noise en p-p 0.1 Hz to 10 Hz 1.5 mV p-p
Voltage Noise Density e
Current Noise Density i

Specifications subject to change without notice.

OS

B

OS

CM

VO

OH

OL

OUT

SY

O

n

n

(@ VS = 5.0 V, VCM = 2.5 V, TA = 25ⴗC unless otherwise noted.)

–40∞C £ TA £ +125∞C 800 mV

–40∞C £ TA £ +125∞C30nA

–40∞C £ TA £ +125∞C ±5nA

0 4.0 V

RL = 10 kW, 0.005 £ V
RL = 10 kW, –40∞C £ TA £ +125∞C 500 V/mV

£ 4.0 V 1,000 10,000 V/mV

OUT

RL = 100 kW to GND 4.98 5.0 V
RL = 10 kW to GND 4.90 4.94 V
I

= 1 mA, –40∞C £ TA £ +125∞C 4.7 V

OUT

RL = 100 kW to GND 0.7 2 mV
RL = 10 kW to GND 0.7 2 mV
I

= 1 mA, –40∞C £ TA £ +125∞C90mV

OUT

±11 ±18 mA

±1.5 V £ VS £ ± 15 V,

–40∞C £ TA £ +125∞C85dB
V

= 2.5 V, RL = •, –40∞C £ TA £ +125∞C 150 mA

OUT

f = 1 kHz 51 nV/÷Hz
f = 1 kHz <0.1 pA/÷Hz

30 300 mV

820 nA

±1 ±3nA

86 Degrees

ELECTRICAL CHARACTERISTICS

(@ VS = 3.0 V, VCM = 1.5 V, TA = 25ⴗC unless otherwise noted.)

Parameter Symbol Conditions Min Typ Max Unit

INPUT CHARACTERISTICS

Offset Voltage V
Input Bias Current I
Input Offset Current I
Input Voltage Range V
Common-Mode Rejection Ratio CMRR 0 V £ VCM £ 2.0 V, –40∞C £ TA £ +125∞C90110 dB
Large Voltage Gain A
Offset Voltage Drift DVOS/DT 1 mV/∞C

OS

B

OS

CM

VO

RL = 10 kW 750 V/mV

0 2.0 V

30 500 mV
820 nA
±1 ±3nA

OUTPUT CHARACTERISTICS

Output Voltage Swing High V
Output Voltage Swing Low V

OH

OL

RL = 10 kW to GND 2.9 V
RL = 10 kW to GND 0.7 2 mV

POWER SUPPLY

Power Supply Rejection Ratio PSRR ± 1.5 V £ VS £ ± 15 V 90 110 dB

±1.5 V £ VS £ ± 15 V,
–40∞C £ TA £ +125∞C85dB

Supply Current Per Amplifier I

SY

V

= 1.5 V, RL = •, –40∞C £ TA £ +125∞C 150 mA

OUT

DYNAMIC PERFORMANCE

Slew Rate SR RL = 10 kW 0.03 V/ms
Gain Bandwidth Product GBP 75 kHz
Phase Margin q

O

85 Degrees

NOISE PERFORMANCE

Voltage Noise en p-p 0.1 Hz to 10 Hz 1.6 mV p-p
Voltage Noise Density e
Current Noise Density i

Specifications subject to change without notice.

n

n

f = 1 kHz 53 nV/÷Hz
f = 1 kHz <0.1 pA/÷Hz

–2–

REV. C

OP295/OP495

ELECTRICAL CHARACTERISTICS

(@ VS = ±15.0 V, TA = 25ⴗC unless otherwise noted.)

Parameter Symbol Conditions Min Typ Max Unit

INPUT CHARACTERISTICS

Offset Voltage V

Input Bias Current I

OS

–40∞C £ T

B

VCM = 0 V 7 20 nA

£ +125∞C 800 mV

A

30 300 mV

VCM = 0 V, –40∞C £ TA £ +125∞C30nA

Input Offset Current I

Input Voltage Range V

OS

CM

Common-Mode Rejection Ratio CMRR –15.0 V £ V
Large Signal Voltage Gain A

VO

VCM = 0 V ±1 ± 3nA

= 0 V, –40∞C £ TA £ +125∞C ±5nA

V

CM

–15 13.5 V

£ +13.5 V, –40∞C £ TA £ +125∞C90 110 dB

CM

RL = 10 kW 1,000 4,000 V/mV

Offset Voltage Drift DVOS/DT 1 mV/∞C

OUTPUT CHARACTERISTICS

Output Voltage Swing High V

OH

RL = 100 kW to GND 14.95 V
RL = 10 kW to GND 14.80 V

Output Voltage Swing Low V

Output Current I

OL

OUT

RL = 100 kW to GND –14.95 V
R

= 10 kW to GND –14.85 V

L

±15 ±25 mA

POWER SUPPLY

Power Supply Rejection Ratio PSRR VS = ± 1.5 V to ± 15 V 90 110 dB

VS = ±1.5 V to ±15 V, –40∞C £ TA £ +125∞C85 dB

Supply Current I

SY

VO = 0 V, RL = •, VS = ±18 V,
–40∞C £ TA £ +125∞C 175 mA

Supply Voltage Range V

S

3 (±1.5) 36 (± 18) V

DYNAMIC PERFORMANCE

Slew Rate SR RL = 10 kW 0.03 V/ms
Gain Bandwidth Product GBP 85 kHz
Phase Margin q

O

83 Degrees

NOISE PERFORMANCE

Voltage Noise en p-p 0.1 Hz to 10 Hz 1.25 mV p-p
Voltage Noise Density e
Current Noise Density i

Specifications subject to change without notice.

n

n

f =1 kHz 45 nV/÷Hz
f = 1 kHz <0.1 pA/÷Hz

REV. C

–3–

OP295/OP495

WARNING!

ESD SENSITIVE DEVICE

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .±18 V

Input Voltage
Differential Input Voltage

2

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V

2

. . . . . . . . . . . . . . . . . . . . . . . . . 36 V

1

Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite

Storage Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65∞C to +150∞C

Operating Temperature Range

OP295G, OP495G . . . . . . . . . . . . . . . . . . . –40∞C to +125∞C

Junction Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65∞C to +150∞C

Lead Temperature Range (Soldering, 60 Sec) . . . . . . . . 300∞C

NOTES

1

Absolute maximum ratings apply to packaged parts, unless otherwise noted.

2

For supply voltages less than ± 18 V, the absolute maximum input voltage is
equal to the supply voltage.

ORDERING GUIDE

Temperature Package Package

Model Range Description Option

OP295GP –40∞C to +125∞C 8-Lead Plastic DIP N-8
OP295GS –40∞C to +125∞C 8-Lead SOIC SO-8
OP495GP –40∞C to +125∞C 14-Lead Plastic DIP N-14
OP495GS –40∞C to +125∞C 16-Lead SOL R-16

Package Type ␪JA* ␪

JC

Unit

8-Lead Plastic DIP (P) 103 43 ∞C/W
8-Lead SOIC (S) 158 43 ∞C/W
14-Lead Plastic DIP (P) 83 39 ∞C/W
16-Lead SO (S) 98 30 ∞C/W

*qJA is specified for the worst case conditions, i.e., qJA is specified for device in

socket for cerdip, P-DIP, and LCC packages; qJA is specified for device soldered
in circuit board for SOIC package.

Typical Performance Characteristics

140

120

100

80

60

SUPPLY CURRENT – ␮A

40

20

–50

–25

TEMPERATURE – ⴗC

VS = 36V

VS = 5V

V

= 3V

S

7550250

100

TPC 1. Supply Current Per Amplifier vs. Temperature

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the OP295/OP495 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

15.2

15.0

14.8

14.6

14.4

14.2

–14.4

–14.6

–14.8

–15.0

–15.2

– OUTPUT SWING – V + OUTPUT SWING – V

–50

–25

TEMPERATURE –

TPC 2. Output Voltage Swing vs. Temperature

VS = 15V

C

R

= 100k⍀

L

RL = 10k⍀

R

= 2k⍀

L

RL = 2k⍀

RL = 10k⍀

RL = 100k⍀

7550250

100

–4–

REV. C

OP295/OP495

500

0

300

150

50

–50

100

–100

300

200

250

350

400

450

250200150100500

INPUT OFFSET VOLTAGE – ␮V

UNITS

VS = 5V
T

A

= 25ⴗC

BASED ON 1200 OP AMPS

500

0

3.2

150

50

0.4

100

0

300

200

250

350

400

450

2.82.42.01.61.20.8

T

C

– VOS – ␮V/ⴗC

UNITS

VS = 5V

–40ⴗ

TA +85ⴗC

BASED ON 1200 OP AMPS

3.10

VS = 3V

3.00

2.90

2.80

2.70

OUTPUT VOLTAGE SWING – V

2.60

2.50
–50

–25

TEMPERATURE – ⴗC

RL = 100k⍀

RL = 10k⍀

RL = 2k⍀

7550250

TPC 3. Output Voltage Swing vs. Temperature

200

BASED ON 600 OP AMPS

175

150

125

100

UNITS

75

VS = 5V

T

= 25ⴗC

A

100

5.10

VS = 5V

5.00

4.90

4.80

4.70

OUTPUT VOLTAGE SWING – V

4.60

4.50
–50

–25

TEMPERATURE – ⴗC

RL = 100k⍀

RL = 10k⍀

RL = 2k⍀

7550250

100

TPC 6. Output Voltage Swing vs. Temperature

50

25

0

–200–250

TPC 4. OP295 Input Offset (VOS) Distribution

250

BASED ON 600 OP AMPS

225

200

175

150

125

UNITS

100

75

50

25

0

0

TPC 5. OP295 TC–VOS Distribution

REV. C

INPUT OFFSET VOLTAGE – ␮V

0.4

T

– VOS – ␮V/ⴗC

C

VS = 5V

–40ⴗ

200150100500–50–100–150

TA +85ⴗC

2.82.42.01.61.20.8

250

TPC 7. OP495 Input Offset (VOS) Distribution

3.2

TPC 8. OP495 TC–VOS Distribution

–5–

OP295/OP495

20

VS = 5V

16

12

8

INPUT BIAS CURRENT – nA

4

0

–50

–25

TEMPERATURE – ⴗC

7550250

TPC 9. Input Bias Current vs. Temperature

40

35

30

25

20

15

OUTPUT CURRENT – mA

10

5

SINK

SOURCE

VS = 15V

SOURCE

SINK

VS = 5V

100

100

VS = 15V
V

= 10V

O

50

RL = 100k⍀

RL = 10k⍀

RL = 2k⍀

75

10

OPEN-LOOP GAIN – V/␮V

1

–50 25

–25

0

TEMPERATURE – ⴗC

TPC 11. Open-Loop Gain vs. Temperature

12

VS = 5V
VO = 4V

10

8

RL = 100k⍀

6

4

OPEN-LOOP GAIN – V/␮V

2

RL = 10k⍀

RL = 2k⍀

100

0

–25–50

TEMPERATURE – ⴗC

7550250

TPC 10. Output Current vs. Temperature

VS = 5V

= 25ⴗC

T

A

1V

100mV

10mV

OUTPUT VOLTAGE TO RAIL

1mV

100␮V

1␮A10␮A

TPC 13. Output Voltage to Supply Rail vs. Load Current

100

SOURCE

SINK

100␮A

LOAD CURRENT

0

–50

–25

TEMPERATURE – ⴗC

7550250

TPC 12. Open-Loop Gain vs. Temperature

10mA1mA

100

REV. C–6–

OP295/OP495

APPLICATIONS
Rail-to-Rail Application Information

The OP295/OP495 has a wide common-mode input range extending
from ground to within about 800 mV of the positive supply.
There is a tendency to use the OP295/OP495 in buffer applications
where the input voltage could exceed the common-mode input
range. This may initially appear to work because of the high input
range and rail-to-rail output range. But above the common-mode
input range the amplifier is, of course, highly nonlinear. For this
reason it is always required that there be some minimal amount
of gain when rail-to-rail output swing is desired. Based on the
input common-mode range, this gain should be at least 1.2.

Low Drop-Out Reference

The OP295/OP495 can be used to gain up a 2.5 V or other low
voltage reference to 4.5 V for use with high resolution A/D convert­ers that operate from 5 V only supplies. The circuit in Figure 1 will
supply up to 10 mA. Its no-load drop-out voltage is only 20 mV.
This circuit will supply over 3.5 mA with a 5 V supply.

16k⍀

5V

2

REF43

4

20k⍀

6

0.001␮F

1/2

OP295/

OP495

5V

10⍀

V = 4.5V

OUT

1␮F TO

10 ␮F

Figure 1. 4.5 V, Low Drop-Out Reference

Low Noise, Single-Supply Preamplifier

Most single-supply op amps are designed to draw low supply
current, at the expense of having higher voltage noise. This
tradeoff may be necessary because the system must be powered
by a battery. However, this condition is worsened because all
circuit resistances tend to be higher; as a result, in addition to
the op amp’s voltage noise, Johnson noise (resistor thermal noise)
is also a significant contributor to the total noise of the system.

The choice of monolithic op amps that combine the characteris­tics of low noise and single-supply operation is rather limited.
Most single-supply op amps have noise on the order of 30 nV/÷
to 60 nV/÷
5 nV/÷

Hz and single-supply amplifiers with noise below

Hz do not exist.

Hz

In order to achieve both low noise and low supply voltage opera­tion, discrete designs may provide the best solution. The circuit
on Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a
matched PNP transistor pair—the MAT03—to achieve zero-in/
zero-out single-supply operation with an input voltage noise of

3.1 nV/÷

Hz at 100 Hz. R5 and R6 set the gain of 1,000, making

this circuit ideal for maximizing dynamic range when amplifying
low level signals in single-supply applications. The OP295/OP495
provides rail-to-rail output swings, allowing this circuit to oper­ate with 0 V to 5 V outputs. Only half of the OP295/OP495 is
used, leaving the other uncommitted op amp for use elsewhere.

0.1␮F

R1LED

Q2

2N3906

35

V

IN

2

R2
27k⍀

Q1 Q2

MAT-03

R7

510⍀

C1

R3

1500pF

R8

100⍀

6

71

2

3

R4

10␮F

R5

10k⍀

8

1

4

OP295/OP495

10⍀

C2
10␮F

R6

V

OUT

Figure 2. Low Noise Single-Supply Preamplifier

The input noise is controlled by the MAT03 transistor pair and
the collector current level. Increasing the collector current reduces
the voltage noise. This particular circuit was tested with 1.85 mA
and 0.5 mA of current. Under these two cases, the input voltage
noise was 3.1 nV/÷

Hz and 10 nV/÷Hz, respectively. The high

collector currents do lead to a tradeoff in supply current, bias
current, and current noise. All of these parameters will increase
with increasing collector current. For example, typically the
MAT03 has an h

= 165. This leads to bias currents of 11 mA

FE

and 3 mA, respectively. Based on the high bias currents, this circuit
is best suited for applications with low source impedance such
as magnetic pickups or low impedance strain gages. Furthermore,
a high source impedance will degrade the noise performance.
For example, a 1 kW resistor generates 4 nV/÷

Hz of broadband

noise, which is already greater than the noise of the preamp.

The collector current is set by R1 in combination with the LED
and Q2. The LED is a 1.6 V Zener diode that has a temperature
coefficient close to that of Q2’s base-emitter junction, which
provides a constant 1.0 V drop across R1. With R1 equal to 270 W,
the tail current is 3.7 mA and the collector current is half that, or

1.85 mA. The value of R1 can be altered to adjust the collector cur­rent. Whenever R1 is changed, R3 and R4 should also be adjusted.
To maintain a common-mode input range that includes ground,
the collectors of the Q1 and Q2 should not go above 0.5 V—otherwise
they could saturate. Thus, R3 and R4 must be small enough to
prevent this condition. Their values and the overall performance
for two different values of R1 are summarized in Table I. Lastly,
the potentiometer, R8, is needed to adjust the offset voltage to
null it to zero. Similar performance can be obtained using an OP90
as the output amplifier with a savings of about 185 mA of supply
current. However, the output swing will not include the positive
rail, and the bandwidth will reduce to approximately 250 Hz.

REV. C

–7–

OP295/OP495

Table I. Single-Supply Low Noise Preamp Performance

IC = 1.85 mA IC = 0.5 mA

R1 270 W 1.0 kW
R3, R4 200 W 910 W

@ 100 Hz 3.15 nV/÷Hz 8.6 nV/÷Hz

e

n

e

@ 10 Hz 4.2 nV/÷Hz 10.2 nV/÷Hz

n

I

SY

I

B

4.0 mA 1.3 mA
11 mA3 mA

Bandwidth 1 kHz 1 kHz
Closed-Loop Gain 1,000 1,000

Driving Heavy Loads

The OP295/OP495 is well suited to drive loads by using a
power transistor, Darlington or FET to increase the current to
the load. The ability to swing to either rail can assure that the
device is turned on hard. This results in more power to the load
and an increase in efficiency over using standard op amps with
their limited output swing. Driving power FETs is also possible
with the OP295/OP495 because of its ability to drive capacitive
loads of several hundred picofarads without oscillating.

Without the addition of external transistors, the OP295/OP495
can drive loads in excess of ±15 mA with ± 15 V or +30 V
supplies. This drive capability is somewhat decreased at lower
supply voltages. At ±5 V supplies, the drive current is ±11 mA.

Driving motors or actuators in two directions in a single-supply
application is often accomplished using an “H” bridge. The prin­ciple is demonstrated in Figure 3a. From a single 5 V supply this
driver is capable of driving loads from 0.8 V to 4.2 V in both direc­tions. Figure 3b shows the voltages at the inverting and noninverting
outputs of the driver. There is a small crossover glitch that is frequency
dependent and would not cause problems unless this was a low
distortion application such as audio. If this is used to drive induc­tive loads, be sure to add diode clamps to protect the bridge from
inductive kickback.

5V

2N2222

VIN 2.5V

0

5k⍀

10k⍀

2N2222

OUTPUTS

Direct Access Arrangement

OP295/OP495 can be used in a single-supply Direct Access
Arrangement (DAA) as is shown in Figure 4. This figure shows
a portion of a typical DM capable of operating from a single 5 V
supply and it may also work on 3 V supplies with minor modifi­cations. Amplifiers A2 and A3 are configured so that the transmit
signal TXA is inverted by A2 and is not inverted by A3. This arrange­ment drives the transformer differentially so that the drive to the
transformer is effectively doubled over a single amplifier arrange­ment. This application takes advantage of the OP295/OP495’s
ability to drive capacitive loads, and to save power in single-supply
applications.

390pF

37.4k⍀

0.1␮F

RXA

0.0047␮F

OP295/

A1

3.3k⍀

A2

OP295/
OP495

20k⍀

20k⍀

475⍀

OP495

22.1k⍀

0.1␮F

TXA

2.5V REF

20k⍀

OP295/
OP495

750pF

20k⍀

20k⍀

A3

0.033␮F

1:1

Figure 4. Direct Access Arrangement

A Single-Supply Instrumentation Amplifier

The OP295/OP495 can be configured as a single-supply instru­mentation amplifier as in Figure 5. For our example, V

V +

equal to

common-mode voltage range includes ground and the output

and VO is measured with respect to V

2

REF

is set

REF

. The input

swings to both rails.

1.67V

10k⍀ 10k⍀

2N2907

2N2907

Figure 3a. “H” Bridge

100

90

10

0%

2V

2V

1ms

Figure 3b. “H” Bridge Outputs

1/2

V+

OP295/

5

8

V

IN

R1

100k⍀

V

REF

1/2

OP295/

OP495

3

2

1

R2

20k⍀ 20k⍀ 100k⍀

VO = 5 +

R

G

200k⍀

R

G

R3

VIN + V

4

6

R4

REF

OP495

V

7

O

Figure 5. Single-Supply Instrumentation Amplifier

Resistor RG sets the gain of the instrumentation amplifier. Mini­mum gain is 6 (with no R

). All resistors should be matched in

G

absolute value as well as temperature coefficient to maximize

–8–

REV. C

OP295/OP495

common-mode rejection performance and minimize drift. This
instrumentation amplifier can operate from a supply voltage as
low as 3 V.

A Single-Supply RTD Thermometer Amplifier

This RTD amplifier takes advantage of the rail-to-rail swing of
the OP295/OP495 to achieve a high bridge voltage in spite of a
low 5 V supply. The OP295/OP495 amplifier servos a constant
200 mA current to the bridge. The return current drops across
the parallel resistors 6.19 kW and the 2.55 MW, developing a
voltage that is servoed to 1.235 V, which is established by the
AD589 bandgap reference. The 3-wire RTD provides an equal
line resistance drop in both 100 W legs of the bridge, thus improving
the accuracy.

The AMP04 amplifies the differential bridge signal and converts it
to a single-ended output. The gain is set by the series resistance of
the 332 W resistor plus the 50 W potentiometer. The gain scales the
output to produce a 4.5 V full scale. The 0.22 mF capacitor to the
output provides a 7 Hz low-pass filter to keep noise at a minimum.

ZERO ADJ

10-TURNS

26.7k⍀

2.55M⍀
1%

0.5%

100⍀
RTD

200⍀

100⍀

0.5%

6.19k⍀
1%

26.7k⍀

0.5%

2

AD589

1

3

7

3

2

1/2

OP295/

OP495

1.235

37.4k⍀

5V

1

AMP04

4

8

5

5V

50⍀

332⍀

0.22␮F

6

4.5V = 450ⴗC
0V = 0ⴗC

V

O

To calibrate, immerse the thermocouple measuring junction in a
0∞C ice bath, adjust the 500 W Zero Adjust pot to zero volts out.
Then immerse the thermocouple in a 250∞C temperature bath
or oven and adjust the Scale Adjust pot for an output voltage of

2.50 V, which is equivalent to 250∞C. Within this temperature
range, the K-type thermocouple is quite accurate and produces
a fairly linear transfer characteristic. Accuracy of ±3∞C is achiev­able without linearization.

Even if the battery voltage is allowed to decay to as low as 7 V, the
rail-to-rail swing allows temperature measurements to 700∞C.
However, linearization may be necessary for temperatures above
250∞C where the thermocouple becomes rather nonlinear. The
circuit draws just under 500 mA supply current from a 9 V battery.

A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V

Figure 8 shows a complete voltage output DAC with wide output
voltage swing operating off a single 5 V supply. The serial input
12-bit D/A converter is configured as a voltage output device with
the 1.235 V reference feeding the current output pin (I
DAC. The V

which is normally the input now becomes the output.

REF

OUT

) of the

The output voltage from the DAC is the binary weighted voltage
of the reference, which is gained up by the output amplifier such
that the DAC has a 1 mV per bit transfer function.

5V 5V

1.23V

R1

17.8k⍀

3

AD589

I

OUT

V

DAC8043

GND CLK SRI

8

DD

4765

2

R

FB

1

V

REF

LD

5V

D

V

= (4.096V)

O

8

3

2

4

OP295/

4096

1

OP495

Figure 6. Low Power RTD Amplifier

A Cold Junction Compensated, Battery-Powered
Thermocouple Amplifier

The OP295/OP495’s 150 mA quiescent current per amplifier
consumption makes it useful for battery-powered temperature
measuring instruments. The K-type thermocouple terminates
into an isothermal block where the terminated junctions’ ambi­ent temperatures can be continuously monitored and corrected
by summing an equal but opposite thermal EMF to the ampli­fier, thereby canceling the error introduced by the cold junctions.

1.235V

24.9k⍀

ISOTHERMAL

ALUMEL

AL

CR

CHROMEL

K-TYPE
THERMOCOUPLE

40.7␮V/ ⴗC

BLOCK

1N914

COLD
JUNCTIONS

AD589

7.15k⍀
1%

1.5M⍀1%24.9k⍀
1%

475⍀

1%

24.3k⍀
1%

4.99k⍀
1%

500⍀
10-TURN

ZERO
ADJUST

2.1k⍀
1%

9V

2

3

1.33M⍀

8

4

SCALE

ADJUST

20k⍀

1

OP295/
OP495

V

O

5V = 500ⴗC
0V = 0ⴗC

Figure 7. Battery-Powered, Cold-Junction Compensated
Thermocouple Amplifier

DIGITAL

CONTROL

TOTA L POWER DISSIPATION = 1.6mW

R2

41.2k⍀

R3
5k⍀

R4

100k⍀

Figure 8. A 5 V 12-Bit DAC with 0 V to 4.095 Output Swing

4 mA to 20 mA Current Loop Transmitter

Figure 9 shows a self powered 4 mA to 20 mA current loop
transmitter. The entire circuit floats up from the single-supply
(12 V to 36 V) return. The supply current carries the signal within
the 4 mA to 20 mA range. Thus the 4 mA establishes the baseline
current budget with which the circuit must operate. This circuit
consumes only 1.4 mA maximum quiescent current, making 2.6 mA
of current available to power additional signal conditioning circuitry
or to power a bridge circuit.

SPAN ADJ

V

IN

0 + 3V

10k⍀

10-TURN

182k⍀

1%

100k⍀

10-TURN

1.21M
1%

HP
5082-2800

NULL ADJ

3

2

220pF

100k⍀

1%

62

REF02

GND

4

5V

8

4

1/2

100⍀

220⍀

1

2N1711

OP295/

OP495

100⍀

1%

4mA

TO

20mA

12V

TO

36V

R

L

100⍀

REV. C

Figure 9. 4 mA to 20 mA Current Loop Transmitter

–9–

OP295/OP495

O

A 3 V Low-Dropout Linear Voltage Regulator

Figure 10 shows a simple 3 V voltage regulator design. The
regulator can deliver 50 mA load current while allowing a 0.2 V
dropout voltage. The OP295/OP495’s rail-to-rail output swing
handily drives the MJE350 pass transistor without requiring special
drive circuitry. At no load, its output can swing less than the pass
transistor’s base-emitter voltage, turning the device nearly off. At
full load, and at low emitter-collector voltages, the transistor beta
tends to decrease. The additional base current is easily handled
by the OP295/OP495 output.

The amplifier servos the output to a constant voltage, which
feeds a portion of the signal to the error amplifier.

Higher output current, to 100 mA, is achievable at a higher
dropout voltage of 3.8 V.

I

< 50mA

1.235V

L

44.2k⍀
1%

30.9k⍀
1%

OP295/

OP495

V

100␮F

1/2

V

5V TO 3.2V

MJE 350

IN

43k⍀

1

1000pF

8

4

AD589

3

2

Figure 10. 3 V Low Dropout Voltage Regulator

Figure 11 shows the regulator’s recovery characteristic when its
output underwent a 20 mA to 50 mA step current change.

If the output current greater than 1 amp persists, the current limit
loop forces a reduction of current to the load, which causes a
corresponding drop in output voltage. As the output voltage
drops, the current limit threshold also drops fractionally, resulting
in a decreasing output current as the output voltage decreases,
to the limit of less than 0.2 A at 1 V output. This “fold-back”
effect reduces the power dissipation considerably during a short
circuit condition, thus making the power supply far more forgiving
in terms of the thermal design requirements. Small heat sinking
on the power MOSFET can be tolerated.

The OP295’s rail-to-rail swing exacts higher gate drive to the
power MOSFET, providing a fuller enhancement to the transistor.
The regulator exhibits 0.2 V dropout at 500 mA of load current.
At 1 amp output, the dropout voltage is typically 5.6 V.

R

5

6

3

124k⍀

2

0.1⍀
1/4W

210k⍀
1%

45.3k⍀
1%

1%

IRF9531

SD

6V

G

1N4148

8

7

A2

1/2

OP295/

5%

OP495

0.01␮F

1/2

OP295/

A1

1

4

100k⍀

OP495

2

REF43

4

6

2.500V

SENSE

(NORM) = 0.5A

I

O

(MAX) = 1A

I

O

205k⍀
1%

45.3k⍀
1%

124k⍀

1%

5V V

O

2V

100

50mA

20mA

OUTPUT

90

10

0%

20mV

1ms

STEP

CURRENT

CONTROL

WAVEFORM

Figure 11. Output Step Load Current Recovery

Low-Dropout, 500 mA Voltage Regulator with Fold-Back
Current Limiting

Adding a second amplifier in the regulation loop as shown in
Figure 12 provides an output current monitor as well as fold­back current limiting protection.

Amplifier A1 provides error amplification for the normal voltage
regulation loop. As long as the output current is less than 1 A,
amplifier A2’s output swings to ground, reverse biasing the di­ode and effectively taking itself out of the circuit. However, as
the output current exceeds 1 amp, the voltage that develops
across the 0.1 W sense resistor forces the amplifier A2’s output
to go high, forward-biasing the diode, which in turn closes the
current limit loop. At this point A2’s lower output resistance
dominates the drive to the power MOSFET transistor, thereby
effectively removing the A1 voltage regulation loop from the circuit.

Figure 12. Low Dropout, 500 mA Voltage Regulator with
Fold-Back Current Limiting

Square Wave Oscillator

The circuit in Figure 13 is a square wave oscillator (note the
positive feedback). The rail-to-rail swing of the OP295/OP495
helps maintain a constant oscillation frequency even if the supply
voltage varies considerably. Consider a battery powered system
where the voltages are not regulated and drop over time. The
rail-to-rail swing ensures that the noninverting input sees the full
V+/2, rather than only a fraction of it.

The constant frequency comes from the fact that the 58.7 kW
feedback sets up Schmitt Trigger threshold levels that are directly
proportional to the supply voltage, as are the RC charge voltage
levels. As a result, the RC charge time, and therefore, the frequency,
remains constant independent of supply voltage. The slew rate
of the amplifier limits oscillation frequency to a maximum of
about 800 Hz at a 5 V supply.

Single-Supply Differential Speaker Driver

Connected as a differential speaker driver, the OP295/OP495
can deliver a minimum of 10 mA to the load. With a 600 W
load, the OP295/OP495 can swing close to 5 V peak-to-peak
across the load.

–10–

REV. C

OP295/OP495

V+

100k⍀

100k⍀

58.7k⍀

C

8

3

1

1/2

4

2

OP295/
OP495

R

FREQ OUT

F

=

OSC

1

< 350Hz @ V+ = 5V

RC

Figure 13. Square Wave Oscillator Has Stable Frequency
Regardless of Supply Changes

90.9k⍀90.9k⍀

V

IN

20k⍀ 20k⍀

V+

10k⍀

2.2␮F

10k⍀

1/4
OP295/
OP495

100k⍀

V+

1/4
OP295/
OP495

1/4
OP295/
OP495

SPEAKER

Figure 14. Single-Supply Differential Speaker Driver

High Accuracy, Single-Supply, Low Power Comparator

The OP295/OP495 makes an accurate open-loop comparator.
With a single 5 V supply, the offset error is less than 300 mV. Figure
15 shows the OP295/OP495’s response time when operating
open-loop with 4 mV overdrive. It exhibits a 4 ms response time at
the rising edge and a 1.5 ms response time at the falling edge.

1V

100

90

INPUT

(5mV OVERDRIVE

@ OP-295 INPUT)

OUTPUT

10

0%

2V

5ms

Figure 15. Open-Loop Comparator Response Time with
5 mV Overdrive

OP295/OP495 SPICE MODEL Macro-Model

* Node Assignments
* Noninverting Input
* Inverting Input
* Positive Supply
* Negative Supply
* Output
*
*
.SUBCKT OP295 1 2 99 50 20
*
* INPUT STAGE
*
I1 99 4 2E-6
R1 1 6 5E3

R2 2 5 5E3
CIN 1 2 2E-12
IOS 1 2 0.5E-9
D1 5 3 DZ
D2 6 3 DZ
EOS 7 6 POLY (1) (31,39) 30E-6 0.024
Q1 8 5 4 QP
Q2 9 74QP
R3 8 50 25.8E3
R4 9 50 25.8E3
*
* GAIN STAGE
*
R7 10 98 270E6
G1 98 10 POLY (1) (9,8) –4.26712E-9 27.8E-6
EREF 98 0 (39, 0) 1
R5 99 39 100E3
R6 39 50 100E3
*
* COMMON MODE STAGE
*
ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5
R12 30 31 1E6
R13 31 98 100
*
* OUTPUT STAGE
*
I2 18 50 1.59E-6
V2 99 12 DC 2.2763
Q4 10 14 50 QNA 1.0
R11 14 50 33
M3 15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10
M4 13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11
D8 10 22 DX
V3 22 50 DC 6
M2 20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
Q5 17 17 99 QPA 1.0
Q6 18 17 99 QPA 4.0
R8 18 99 2.2E6
Q7 18 19 99 QPA 1.0
R9 99 19 8
C2 18 99 20E-12
M6 15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12
M1 20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9
D4 21 18 DX
V4 99 21 DC 6
R10 10 11 6E3
C3 11 20 50E-12
.MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3
+ ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4
+ ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4
RC=209
+ CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534
MJC=0.5
+ CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30)
.MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4
+ ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5
+ ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31
RC=354.4
+ CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762
MJC=0.4
+ CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30)
.MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3
+ TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5
+ XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-4
PB=0.837
+ MJ=0.407 CJSW=0.5E-9 MJSW=0.33)
.MODEL MP PMOS (LEVEL=3 VTO=–1.1 RS=0.7 RD=0.7
+ TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5
+ XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=–1 CJ=1.55E-4
PB=0.56
+ MJ=0.442 CJSW=0.4E-9 MJSW=0.33)
.MODEL DX D(IS=1E-15)
.MODEL DZ D (IS=1E-15, BV=7)
.MODEL QP PNP (BF=125)

.ENDS

REV. C

–11–

OP295/OP495

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm)

PIN 1

0.210

(5.33)

MAX

0.160 (4.06)

0.115 (2.93)

PIN 1

0.210 (5.33)
MAX

0.160 (4.06)

0.115 (2.93)

8-Lead Plastic DIP

0.430 (10.92)

0.348 (8.84)

8

14

0.100 (2.54)
BSC

0.022 (0.558)

0.014 (0.356)

0.070 (1.77)

0.045 (1.15)

14-Lead Plastic DIP

0.795 (20.19)

0.725 (18.42)

14

17

0.100 (2.54)
BSC

0.022 (0.558)

0.014 (0.356)

(P Suffix)

5

0.280 (7.11)

0.240 (6.10)

0.060 (1.52)

0.015 (0.38)

(P Suffix)

8

0.280 (7.11)

0.240 (6.10)

0.060 (1.52)

0.015 (0.38)

0.070 (1.77)

0.045 (1.15)

0.130
(3.30)
MIN

SEATING
PLANE

0.130
(3.30)
MIN

SEATING
PLANE

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

0.195 (4.95)

0.115 (2.93)

0.195 (4.95)

0.115 (2.93)

0.1574 (4.00)

0.1497 (3.80)

PIN 1

0.0098 (0.25)

0.0040 (0.10)

SEATING

16

1

PIN 1

0.1968 (5.00)

0.1890 (4.80)

85

0.0500 (1.27)

PLANE

0.4133 (10.50)

0.3977 (10.00)

0.050 (1.27)
BSC

8-Lead Narrow-Body SO

(S Suffix)

0.2440 (6.20)

0.2284 (5.80)

41

BSC

0.0192 (0.49)

0.0138 (0.35)

0.0688 (1.75)

0.0532 (1.35)

0.0098 (0.25)

0.0075 (0.19)

16-Lead Wide-Body SO

(S Suffix)

9

0.2992 (7.60)

0.2914 (7.40)

0.4193 (10.65)

8

0.3937 (10.00)

0.1043 (2.65)

0.0926 (2.35)

0.0196 (0.50)

0.0099 (0.25)

8ⴗ

0.0500 (1.27)

0ⴗ

0.0160 (0.41)

0.0291 (0.74)

0.0098 (0.25)

ⴛ 45ⴗ

C00331–0–4/02(C)

ⴛ 45ⴗ

0.0118 (0.30)

0.0040 (0.10)

0.0192 (0.49)

0.0138 (0.35)

SEATING
PLANE

0.0125 (0.32)

0.0091 (0.23)

8ⴗ
0ⴗ

0.0500 (1.27)

0.0157 (0.40)

Revision History

Location Page

03/02—Data Sheet changed from REV. B to REV. C.

Figure changes to PIN CONNECTIONS ..................................................................................................................................... 1

Deletion of OP295GBC and OP495GBC from ORDERING GUIDE .......................................................................................... 3

Deletion of WAFER TEST LIMITS table .................................................................................................................................... 3

Changes to ABSOLUTE MAXIMUM RATINGS........................................................................................................................ 4

Deletion of DICE CHARACTERISTICS .................................................................................................................................... 4

PRINTED IN U.S.A.

–12–

REV. C