Datasheet OP295, OP495 Datasheet (Analog Devices)

Page 1

Dual/Quad Rail-to-Rail

FEATURES Rail-to-Rail Output Swing Single-Supply Operation: +3 V to +36 V Low Offset Voltage: 300 mV Gain Bandwidth Product: 75 kHz High Open-Loop Gain: 1000 V/mV Unity-Gain Stable Low Supply Current/Per Amplifier: 150 mA max

APPLICATIONS Battery Operated Instrumentation Servo Amplifiers Actuator Drives Sensor Conditioners Power Supply Control

GENERAL DESCRIPTION

Rail-to-rail output swing combined with dc accuracy are the key features of the OP495 quad and OP295 dual CBCMOS operational amplifiers. By using a bipolar front end, lower noise and higher accuracy than that of CMOS designs has been achieved. Both input and output ranges include the negative supply, providing the user “zero-in/zero-out” capability. For users of 3.3 volt systems such as lithium batteries, the OP295/OP495 is specified for three volt operation.

Maximum offset voltage is specified at 300 µV for +5 volt opera- tion, and the open-loop gain is a minimum of 1000 V/mV. This yields performance that can be used to implement high accuracy systems, even in single supply designs.

The ability to swing rail-to-rail and supply +15 mA to the load makes the OP295/OP495 an ideal driver for power transistors and “H” bridges. This allows designs to achieve higher efficiencies and to transfer more power to the load than previously possible without the use of discrete components. For applications

Operational Amplifiers

OP295/OP495

PIN CONNECTIONS

8-Lead Narrow-Body SO 8-Lead Epoxy DIP

(S Suffix) (P Suffix)

OP495

V+

OUT B –IN B

+IN B

OUT D

–IN D

+IN D

V–

+IN C

–IN C

OUT C

OUT A

–IN A

+IN A

V–

OUT A

–IN A +IN A

V+ +IN B –IN B

OUT B

1 2

OP295

1 2 3 413

OP495

5 6 7 89

NC = NO CONNECT

V+ OUT B

–IN B

+IN B

OUT D

–IN D

15 14

+IN D

V–

+IN C

–IN C

OUT C NC

OUT A

1 2

–IN A +IN A

OP295

3 45

V–

14-Lead Epoxy DIP 16-Lead SO (300 Mil)

(P Suffix) (S Suffix)

OUT A

1 2

–IN A +IN A

V+

+IN B

–IN B

OUT B

that require driving inductive loads, such as transformers, increases in efficiency are also possible. Stability while driving capacitive loads is another benefit of this design over CMOS rail-to-rail amplifiers. This is useful for driving coax cable or large FET transistors. The OP295/OP495 is stable with loads in excess of 300 pF.

The OP295 and OP495 are specified over the extended industrial (–40°C to +125°C) temperature range. OP295s are available in 8-pin plastic and ceramic DIP plus SO-8 surface mount packages. OP495s are available in 14-pin plastic and SO-16 surface mount packages. Contact your local sales office for MIL-STD-883 data sheet.

REV. B

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703

Page 2

OP295/OP495–SPECIFICA TIONS

ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Units

INPUT CHARACTERISTICS

Offset Voltage V Input Bias Current I Input Offset Current I Input Voltage Range V

Common-Mode Rejection Ratio CMRR 0 V ≤ VCM ≤ 4.0 V, –40°C ≤ TA ≤ +125°C 90 110 dB Large Signal Voltage Gain A

Offset Voltage Drift ∆VOS/∆T 15 µV/°C

OUTPUT CHARACTERISTICS

Output Voltage Swing High V

Output Voltage Swing Low V

Output Current I

POWER SUPPLY

Power Supply Rejection Ratio PSRR ±1.5 V ≤ VS ≤ ±15 V 90 110 dB

Supply Current Per Amplifier I

DYNAMIC PERFORMANCE

Skew Rate SR RL = 10 kΩ 0.03 V/µs Gain Bandwidth Product GBP 75 kHz Phase Margin θ

NOISE PERFORMANCE

Voltage Noise en Voltage Noise Density e Current Noise Density i

Specifications subject to change without notice.

OUT

p-p

(@ VS = +5.0 V, VCM = +2.5 V, TA = +258C unless otherwise noted)

–40°C ≤ TA ≤ +125°C 800 µV –40°C ≤ TA ≤ +125°C30nA –40°C ≤ TA ≤ +125°C ±5nA

0 +4.0 V

RL = 10 kΩ, 0.005 ≤ V RL = 10 kΩ, –40°C ≤ TA ≤ +125°C 500 V/mV

≤ 4.0 V 1000 10,000 V/mV

OUT

RL = 100 kΩ to GND 4.98 5.0 V RL = 10 kΩ to GND 4.90 4.94 V I

= 1 mA, –40°C ≤ TA ≤ +125°C 4.7 V

OUT

RL = 100 kΩ to GND 0.7 2 mV RL = 10 kΩ to GND 0.7 2 mV I

= 1 mA, –40°C ≤ TA ≤ +125°C90mV

OUT

± 11 ± 18 mA

±1.5 V ≤ VS ≤ ±15 V,

–40°C ≤ TA ≤ +125°C85dB V

= 2.5 V, RL = ∞, –40°C ≤ TA ≤ +125°C 150 µA

OUT

0.1 Hz to 10 Hz 1.5 µV p-p f = 1 kHz 51 nV/√Hz f = 1 kHz <0.1 pA/√Hz

30 300 µV 820 nA ±1 ±3nA

86 Degrees

ELECTRICAL CHARACTERISTICS

(@ VS = +3.0 V, VCM = +1.5 V, TA = +258C unless otherwise noted)

Parameter Symbol Conditions Min Typ Max Units

INPUT CHARACTERISTICS

Offset Voltage V Input Bias Current I Input Offset Current I Input Voltage Range V Common-Mode Rejection Ratio CMRR 0 V ≤ VCM ≤ 2.0 V, –40°C ≤ TA ≤ +125°C 90 110 dB Large Voltage Gain A Offset Voltage Drift ∆VOS/∆T 1 µV/°C

OS B OS

RL = 10 kΩ 750 V/mV

0 +2.0 V

30 500 µV 820 nA ±1 ±3nA

OUTPUT CHARACTERISTICS

Output Voltage Swing High V Output Voltage Swing Low V

RL = 10 kΩ to GND 2.9 V RL = 10 kΩ to GND 0.7 2 mV

POWER SUPPLY

Power Supply Rejection Ratio PSRR ±1.5 V ≤ VS ≤ ±15 V 90 110 dB

±1.5 V ≤ VS ≤ ±15 V, –40°C ≤ TA ≤ +125°C85dB

Supply Current Per Amplifier I

= 1.5 V, RL = ∞, –40°C ≤ TA ≤ +125°C 150 µA

OUT

DYNAMIC PERFORMANCE

Slew Rate SR RL = 10 kΩ 0.03 V/µs Gain Bandwidth Product GBP 75 kHz Phase Margin θ

85 Degrees

NOISE PERFORMANCE

Voltage Noise en Voltage Noise Density e Current Noise Density i

Specifications subject to change without notice.

p-p

0.1 Hz to 10 Hz 1.6 µV p-p f = 1 kHz 53 nV/√Hz f = 1 kHz <0.1 pA/√Hz

–2–

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Page 3

OP295/OP495

ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Units

INPUT CHARACTERISTICS

Offset Voltage V Input Bias Current I Input Offset Current I Input Voltage Range V

Common-Mode Rejection Ratio CMRR –15.0 V ≤ VCM ≤ +13.5 V, –40°C ≤ TA ≤ +125°C 90 110 dB Large Signal Voltage Gain A Offset Voltage Drift ∆VOS/∆T 1 µV/°C

OUTPUT CHARACTERISTICS

Output Voltage Swing High V Output Voltage Swing Low V Output Current I

POWER SUPPLY

Power Supply Rejection Ratio PSRR VS = ±1.5 V to ±15 V 90 110 dB Supply Current I Supply Voltage Range V

DYNAMIC PERFORMANCE

Slew Rate SR RL = 10 kΩ 0.03 V/µs Gain Bandwidth Product GBP 85 kHz Phase Margin θ

NOISE PERFORMANCE

Voltage Noise e Voltage Noise Density e Current Noise Density i

Specifications subject to change without notice.

OUT

n p-p n

(@ VS = ±15.0 V, TA = +258C unless otherwise noted)

–40°C ≤ TA ≤ +125°C 800 µV VCM = 0 V 7 20 nA VCM = 0 V, –40°C ≤ TA ≤ +125°C30nA VCM = 0 V ±1 ±3nA V

= 0 V, –40°C ≤ TA ≤ +125°C ±5nA

RL = 10 kΩ 1000 4000 V/mV

RL = 100 kΩ to GND 14.95 V RL = 10 kΩ to GND 14.80 V RL = 100 kΩ to GND –14.95 V RL = 10 kΩ to GND –14.85 V

VS = ±1.5 V to ±15 V, –40°C ≤ TA ≤ +125°C85 dB VO = 0 V, RL = ∞, VS = ±18 V, –40°C ≤ TA ≤ +125°C 175 µA

0.1 Hz to 10 Hz 1.25 µV p-p f =1 kHz 45 nV/√Hz f = 1 kHz <0.1 pA/√Hz

–15 +13.5 V

± 15 ±25 mA

+3 (±1.5) +36 (±18) V

30 300 µV

83 Degrees

W AFER TEST LIMITS

(@ VS = +5.0 V, VCM = 2.5 V, TA = +258C unless otherwise noted)

Parameter Symbol Conditions Limit Units

Offset Voltage Vos 300 µV max Input Bias Current I Input Offset Current I Input Voltage Range

Common-Mode Rejection Ratio CMRR 0 V ≤ V Power Supply Rejection Ratio PSRR ±1.5 V ≤ V Large Signal Voltage Gain A Output Voltage Swing High V Supply Current Per Amplifier I

NOTES Electrical tests and wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard product dice. Consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing.

Guaranteed by CMRR test.

B OS

VO OH

RL = 10 kΩ 1000 V/mV min RL = 10 kΩ 4.9 V min V

OUT

≤ 4 V 90 dB min

≤ ±15 V 90 µV/V

= 2.5 V, RL = ∞ 150 µA max

20 nA max ±2 nA max 0 to +4 V min

ORDERING GUIDE

Temperature Package Package

Model Range Description Option

OP295GP –40°C to +125°C 8-Pin Plastic DIP N-8 OP295GS –40°C to +125°C 8-Pin SOIC SO-8 OP295GBC +25°C DICE

Model Range Description Option

OP495GP –40°C to +125°C 14-Pin Plastic DIP N-14 OP495GS –40°C to +125°C 16-Pin SOL R-16 OP495GBC +25°C DICE

Temperature Package Package

REV. B

–3–

Page 4

OP295/OP495

15.2

–15.2

100

–14.6

–15.0

–25

–14.8

–50

14.2

–14.4

14.4

14.6

14.8

15.0

7550250

TEMPERATURE – °C

– OUTPUT SWING – Volts

+ OUTPUT SWING – Volts

VS = ±15V

RL = 100k

RL = 2k

RL = 100k

RL = 10k

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .±18 V

Input Voltage Differential Input Voltage

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .±18 V

. . . . . . . . . . . . . . . . . . . . . . . +36 V

Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite

Storage Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . .–65°C to +150°C

Operating Temperature Range

OP295G, OP495G . . . . . . . . . . . . . . . . . . .–40°C to +125°C

Junction Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . .–65°C to +150°C

Lead Temperature Range (Soldering, 60 Sec) . . . . . . . +300°C

Package Type u

Unit

8-Pin Plastic DIP (P) 103 43 °C/W 8-Pin SOIC (S) 158 43 °C/W 14-Pin Plastic DIP (P) 83 39 °C/W 16-Pin SO (S) 98 30 °C/W

NOTES

Absolute maximum ratings apply to both DICE and packaged parts, unless otherwise noted.

For supply voltages less than ±18 V, the absolute maximum input voltage is equal to the supply voltage.

θJA is specified for the worst case conditions, i.e., θJA is specified for device in socket

for cerdip, P-DIP, and LCC packages; θJA is specified for device soldered in circuit board for SOIC package.

DICE CHARACTERISTICS

OP295 Die Size 0.066 × 0.080 inch, 5,280 sq. mils. Substrate (Die Backside) Is Connected to V+. Transistor Count, 74.

Typical Characteristics

140

120

100

SUPPLY CURRENT PER AMPLIFIER – µA

–50

–25

Supply Current Per Amplifier vs. Temperature Output Voltage Swing vs. Temperature

TEMPERATURE – °C

VS = +36V

VS = +5V

VS = +3V

7550250

100

–4–

OP495 Die Size 0.113 × 0.083 inch, 9,380 sq. mils. Substrate (Die Backside) Is Connected to V+. Transistor Count, 196.

REV. B

Page 5

Typical Characteristics–OP295/OP495

500

3.2

150

0.4

100

300

200

250

350

400

450

2.82.42.01.61.20.8

TC – VOS – µV/°C

UNITS

VS = +5V –40° ≤ TA ≤ +85°C

BASED ON 1200 OP AMPS

3.10

3.00

2.90

2.80

2.70

OUTPUT VOLTAGE SWING – Volts

2.60

2.50 –50

–25

TEMPERATURE – °C

Output Voltage Swing vs. Temperature

200

BASED ON 600 OP AMPS

175

150

125

100

UNITS

–200–250

INPUT OFFSET VOLTAGE – µV

RL = 100k

RL = 10k

7550250

VS = +5V

= +25°C

VS = +3V

RL = 2k

200150100500–50–100–150

250

100

5.10

5.00

4.90

4.80

4.70

OUTPUT VOLTAGE SWING – Volts

4.60

4.50 –50

–25

TEMPERATURE – °C

Output Voltage Swing vs. Temperature

500

BASED ON 1200 OP AMPS

450

400

350

300

250

UNITS

200

150

100

–50

–100

INPUT OFFSET VOLTAGE – µV

VS = +5V

RL = 100k

RL = 10k

RL = 2k

7550250

VS = +5V

= +25°C

250200150100500

100

300

REV. B

OP295 Input Offset (VOS) Distribution

250

BASED ON 600 OP AMPS

225

200

175

150

125

UNITS

100

0.4 – VOS – µV/°C

OP295 TC–VOS Distribution

VS = +5V

–40° ≤ T

≤ +85°C

OP495 Input Offset (VOS) Distribution

3.2

2.82.42.01.61.20.8

OP495 TC–VOS Distribution

–5–

Page 6

OP295/OP495–Typical Characteristics

INPUT BIAS CURRENT – nA

–50

–25

TEMPERATURE – °C

Input Bias Current vs. Temperature

OUTPUT CURRENT – mA

SOURCE

SINK

SOURCE

SINK

VS = ±15V

VS = +5V

7550250

100

OPEN-LOOP GAIN – V/µV

–50 25

–25

TEMPERATURE – °C

Open-Loop Gain vs. Temperature

OPEN-LOOP GAIN – V/µV

VS = ±15V VO = ±10V

RL = 100k

RL = 10k

RL = 2k

VS = +5V VO = +4V

RL = 100k

RL = 10k

RL = 2k

1000

–25–50

TEMPERATURE – °C

Output Current vs. Temperature

100

7550250

–50

–25

Open-Loop Gain vs. Temperature

100mV

10mV

OUTPUT VOLTAGE ∆ TO RAIL

1mV

100µV

1µA10µA 10mA1mA100µA

SOURCE

SINK

LOAD CURRENT

VS = +5V

= +25°C

Output Voltage to Supply Rail vs. Load Current

TEMPERATURE – °C

100

7550250

REV. B–6–

Page 7

OP295/OP495

MAT- 03

10Ω

OUT

LED

10kΩ

C2 10µF

510Ω

1500pF

100Ω

R2 27kΩ

0.1µF

10µF

2N3906

OP295/

OP495

APPLICATIONS Rail-to-Rail Applications Information

The OP295/OP495 has a wide common-mode input range extending from ground to within about 800 mV of the positive supply. There is a tendency to use the OP295/OP495 in buffer applications where the input voltage could exceed the commonmode input range. This may initially appear to work because of the high input range and rail-to-rail output range. But above the common-mode input range the amplifier is, of course, highly nonlinear. For this reason it is always required that there be some minimal amount of gain when rail-to-rail output swing is desired. Based on the input common-mode range this gain should be at least 1.2.

Low Drop-Out Reference

The OP295/OP495 can be used to gain up a 2.5 V or other low voltage reference to 4.5 volts for use with high resolution A/D converters that operate from +5 volt only supplies. The circuit in Figure 1 will supply up to 10 mA. Its no-load drop-out voltage is only 20 mV. This circuit will supply over 3.5 mA with a +5 volt supply.

16k

+5V

REF43

20k

1/2 OP295/ OP495

0.001µF

+5V

10Ω

1 TO 10µF

= 4.5V

OUT

Figure 1. 4.5 Volt, Low Drop-Out Reference

Low Noise, Single Supply Preamplifier

Most single supply op amps are designed to draw low supply current, at the expense of having higher voltage noise. This tradeoff may be necessary because the system must be powered by a battery. However, this condition is worsened because all circuit resistances tend to be higher; as a result, in addition to the op amp’s voltage noise, Johnson noise (resistor thermal noise) is also a significant contributor to the total noise of the system.

The choice of monolithic op amps that combine the characteristics of low noise and single supply operation is rather limited. Most single supply op amps have noise on the order of 30 nV/√ to 60 nV/√ 5 nV/√

In order to achieve both low noise and low supply voltage operation, discrete designs may provide the best solution. The circuit on Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a matched PNP transistor pair—the MAT03—to achieve zero-in/ zero-out single supply operation with an input voltage noise of

3.1 nV/√ this circuit ideal for maximizing dynamic range when amplifying low level signals in single supply applications. The OP295/OP495 provides rail-to-rail output swings, allowing this circuit to operate with 0 to 5 volt outputs. Only half of the OP295/OP495 is used, leaving the other uncommitted op amp for use elsewhere.

Hz and single supply amplifiers with noise below

Hz do not exist.

Hz at 100 Hz. R5 and R6 set the gain of 1000, making

Figure 2. Low Noise Single Supply Preamplifier

The input noise is controlled by the MAT03 transistor pair and the collector current level. Increasing the collector current reduces the voltage noise. This particular circuit was tested with

1.85 mA and 0.5 mA of current. Under these two cases, the input voltage noise was 3.1 nV/√

Hz and 10 nV/√Hz, respectively. The high collector currents do lead to a tradeoff in supply current, bias current, and current noise. All of these parameters will increase with increasing collector current. For example, typically the MAT03 has an h

= 165. This leads to bias currents of

11 µA and 3 µA, respectively. Based on the high bias currents, this circuit is best suited for applications with low source impedance such as magnetic pickups or low impedance strain gages. Furthermore, a high source impedance will degrade the noise performance. For example, a 1 kΩ resistor generates 4 nV/√

of broadband noise, which is already greater than the noise of the preamp.

The collector current is set by R1 in combination with the LED and Q2. The LED is a 1.6 V “Zener” that has a temperature coefficient close to that of Q2’s base-emitter junction, which provides a constant 1.0 V drop across R1. With R1 equal to 270 Ω, the tail current is 3.7 mA and the collector current is half that, or 1.85 mA. The value of R1 can be altered to adjust the collector current. Whenever R1 is changed, R3 and R4 should also be adjusted. To maintain a common-mode input range that includes ground, the collectors of the Q1 and Q2 should not go above 0.5 V—otherwise they could saturate. Thus, R3 and R4 have to be small enough to prevent this condition. Their values and the overall performance for two different values of R1 are summarized in Table I. Lastly, the potentiometer, R8, is needed to adjust the offset voltage to null it to zero. Similar performance can be obtained using an OP90 as the output amplifier with a savings of about 185 µA of supply current. However, the output swing will not include the positive rail, and the bandwidth will reduce to approximately 250 Hz.

REV. B

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Page 8

OP295/OP495

V+

REF

100k

20k

100k

VIN + V

REF

VO = 5 +

200k

(

)

1/2

OP295/

OP495

1/2

OP295/

OP495

Table I. Single Supply Low Noise Preamp Performance

IC = 1.85 mA IC = 0.5 mA

R1 270 Ω 1.0 kΩ R3, R4 200 Ω 910 Ω e

@ 100 Hz 3.15 nV/√Hz 8.6 nV/√Hz

@ 10 Hz 4.2 nV/√Hz 10.2 nV/√Hz

4.0 mA 1.3 mA

11 µA3 µA Bandwidth 1 kHz 1 kHz Closed-Loop Gain 1000 1000

Driving Heavy Loads

The OP295/OP495 is well suited to drive loads by using a power transistor, Darlington or FET to increase the current to the load. The ability to swing to either rail can assure that the device is turned on hard. This results in more power to the load and an increase in efficiency over using standard op amps with their limited output swing. Driving power FETs is also possible with the OP295/OP495 because of its ability to drive capacitive loads of several hundred picofarads without oscillating.

Without the addition of external transistors the OP295/OP495 can drive loads in excess of ± 15 mA with ±15 or +30 volt supplies. This drive capability is somewhat decreased at lower supply voltages. At ±5 volt supplies the drive current is ±11 mA.

Driving motors or actuators in two directions in a single supply application is often accomplished using an “H” bridge. The principle is demonstrated in Figure 3a. From a single +5 volt supply this driver is capable of driving loads from 0.8 V to 4.2 V in both directions. Figure 3b shows the voltages at the inverting and noninverting outputs of the driver. There is a small crossover glitch that is frequency dependent and would not cause problems

+5V

2N2222 2N2222

0 ≤ V

≤ 2.5V

1.67V

10k

2N2907

OUTPUTS

2N2907

unless this was a low distortion application such as audio. If this is used to drive inductive loads, be sure to add diode clamps to protect the bridge from inductive kickback.

Direct Access Arrangement

OP295/OP495 can be used in a single supply Direct Access Arrangement (DAA) as is shown an in Figure 4. This figure shows a portion of a typical DM capable of operating from a single +5 volt supply and it may also work on +3 volt supplies with minor modifications. Amplifiers A2 and A3 are configured so that the transmit signal TXA is inverted by A2 and is not inverted by A3. This arrangement drives the transformer differentially so that the drive to the transformer is effectively doubled over a single amplifier arrangement. This application takes advantage of the OP295/OP495’s ability to drive capacitive loads, and to save power in single supply applications.

390pF

37.4kΩ

RXA

TXA

2.5V REF

0.1µF

0.0047µF

OP295/ OP495

3.3kΩ

22.1kΩ

750pF20kΩ

20kΩ 20kΩ

OP295/ OP495

20kΩ

475Ω

0.033µF

1:1

Figure 4. Direct Access Arrangement

A Single Supply Instrumentation Amplifier

The OP295/OP495 can be configured as a single supply instrumentation amplifier as in Figure 5. For our example, V

equal to put common-mode voltage range includes ground and the out-

V+

and VO is measured with respect to V

REF

. The in-

REF

is set

put swings to both rails.

Figure 3a. “H” Bridge

100

10 0%

Figure 3b. “H” Bridge Outputs

2V 1ms

Figure 5. Single Supply Instrumentation Amplifier

Resistor RG sets the gain of the instrumentation amplifier. Minimum gain is 6 (with no R

). All resistors should be matched in

absolute value as well as temperature coefficient to maximize

–8–

REV. B

Page 9

OP295/OP495

+5V

41.2k R3

5kΩ

100kΩ

DD R

REF

SRICLKGND

OUT

DAC8043

+5V

DIGITAL

CONTROL

+5V

AD589

17.8kΩ

+1.23V

VO = (4.096V)

4096

TOTAL POWER DISSIPATION = 1.6mW

OP295/

OP495

220pF

220Ω

REF02

GND

100Ω

2N1711

100k

HP 5082-2800

100Ω

0 + 3V

100kΩ

10-TURN

1.21M 1%

NULL ADJ

SPAN ADJ

182k

10kΩ

10-TURN

+12V

+36V

100Ω

4–20mA

1/2

OP295/

OP495

common-mode rejection performance and minimize drift. This instrumentation amplifier can operate from a supply voltage as low as 3 volts.

A Single Supply RTD Thermometer Amplifier

This RTD amplifier takes advantage of the rail-to-rail swing of the OP295/OP495 to achieve a high bridge voltage in spite of a low 5 V supply. The OP295/OP495 amplifier servos a constant 200 µA current to the bridge. The return current drops across the parallel resistors 6.19 kΩ and the 2.55 MΩ, developing a voltage that is servoed to 1.235 V, which is established by the AD589 bandgap reference. The 3-wire RTD provides an equal line resistance drop in both 100 Ω legs of the bridge, thus improving the accuracy.

The AMP04 amplifies the differential bridge signal and converts it to a single-ended output. The gain is set by the series resistance of the 332 Ω resistor plus the 50 Ω potentiometer. The gain scales the output to produce a 4.5 V full scale. The

0.22 µF capacitor to the output provides a 7 Hz low-pass filter to keep noise at a minimum.

ZERO ADJ

10-TURNS

26.7k

100Ω RTD

2.55M 1%

0.5%

200Ω

100Ω

0.5%

6.19k 1%

26.7k

0.5%

AD589

2 3

1/2

OP295/

OP495

1.235

37.4k

+5V

AMP04

+5V

50Ω 332Ω

0.22µF

4.5V = 450°C 0V = 0°C

To calibrate, immerse the thermocouple measuring junction in a 0°C ice bath, adjust the 500 Ω Zero Adjust pot to zero volts out. Then immerse the thermocouple in a 250°C temperature bath or oven and adjust the Scale Adjust pot for an output voltage of

2.50 V, which is equivalent to 250°C. Within this temperature range, the K-type thermocouple is quite accurate and produces a fairly linear transfer characteristic. Accuracy of ±3°C is achievable without linearization.

Even if the battery voltage is allowed to decay to as low as 7 volts, the rail-to-rail swing allows temperature measurements to 700°C. However, linearization may be necessary for temperatures above 250°C where the thermocouple becomes rather nonlinear. The circuit draws just under 500 µA supply current from a 9 V battery.

A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V

Figure 8 shows a complete voltage output DAC with wide output voltage swing operating off a single +5 V supply. The serial input 12-bit D/A converter is configured as a voltage output device with the 1.235 V reference feeding the current output pin (I

) of the DAC. The V

OUT

which is normally the input now

REF

becomes the output. The output voltage from the DAC is the binary weighted volt-

age of the reference, which is gained up by the output amplifier such that the DAC has a 1 mV per bit transfer function.

A Cold Junction Compensated, Battery Powered Thermocouple Amplifier

The OP295/OP495’s 150 µA quiescent current per amplifier consumption makes it useful for battery powered temperature measuring instruments. The K-type thermocouple terminates into an isothermal block where the terminated junctions’ ambient temperatures can be continuously monitored and corrected by summing an equal but opposite thermal EMF to the amplifier, thereby canceling the error introduced by the cold junctions.

ISOTHERMAL

ALUMEL

CHROMEL

K-TYPE THERMOCOUPLE

40.7µV/°C

Figure 7. Battery Powered, Cold-Junction Compensated Thermocouple Amplifier

REV. B

Figure 6. Low Power RTD Amplifier

24.9k

24.3k 1%

4.99k

500Ω 10-TURN

ZERO ADJUST

2.1k 1%

OP295/

OP495

BLOCK

1N914

AD589

1.5M 1%

COLD JUNCTIONS

1.235V

7.15k 1%

24.9k 1%

475Ω

1.33MΩ

SCALE

ADJUST

20k

0V = 0°C

5V = 500°C

Figure 8. A 5 Volt 12-Bit DAC with 0 V to +4.095 Output Swing

4–20 mA Current Loop Transmitter

Figure 9 shows a self powered 4–20 mA current loop transmitter. The entire circuit floats up from the single supply (12 V to 36 V) return. The supply current carries the signal within the 4 to 20 mA range. Thus the 4 mA establishes the baseline

Figure 9. 4–20 mA Current Loop Transmitter

–9–

Page 10

OP295/OP495

current budget with which the circuit must operate. This circuit consumes only 1.4 mA maximum quiescent current, making 2.6 mA of current available to power additional signal conditioning circuitry or to power a bridge circuit.

A 3 Volt Low-Dropout Linear Voltage Regulator

Figure 10 shows a simple 3 V voltage regulator design. The regulator can deliver 50 mA load current while allowing a 0.2 V dropout voltage. The OP295/OP495’s rail-to-rail output swing handily drives the MJE350 pass transistor without requiring special drive circuitry. At no load, its output can swing less than the pass transistor’s base-emitter voltage, turning the device nearly off. At full load, and at low emitter-collector voltages, the transistor beta tends to decrease. The additional base current is easily handled by the OP295/OP495 output.

The amplifier servos the output to a constant voltage, which feeds a portion of the signal to the error amplifier.

Higher output current, to 100 mA, is achievable at a higher dropout voltage of 3.8 V.

< 50mA

1.235V

AD589

44.2k 1%

30.9k 1%

100µF

5V TO 3.2V

MJE 350

43k

1000pF

1/2

OP295/

OP495

Figure 10. 3 V Low Dropout Voltage Regulator

Figure 11 shows the regulator’s recovery characteristic when its output underwent a 20 mA to 50 mA step current change.

current limit loop. At this point A2’s lower output resistance dominates the drive to the power MOSFET transistor, thereby effectively removing the A1 voltage regulation loop from the circuit.

If the output current greater than 1 amp persists, the current limit loop forces a reduction of current to the load, which causes a corresponding drop in output voltage. As the output voltage drops, the current limit threshold also drops fractionally, resulting in a decreasing output current as the output voltage decreases, to the limit of less than 0.2 A at 1 V output. This “fold-back” effect reduces the power dissipation considerably during a short circuit condition, thus making the power supply far more forgiving in terms of the thermal design requirements. Small heat sinking on the power MOSFET can be tolerated.

The OP295’s rail-to-rail swing exacts higher gate drive to the power MOSFET, providing a fuller enhancement to the transistor. The regulator exhibits 0.2 V dropout at 500 mA of load current. At 1 amp output, the dropout voltage is typically

5.6 volts.

SENSE

124k

0.1Ω 1/4W

210k1%205k

45.3k 1%

2.500V

IRF9531

100k

1N4148

OP295/

OP495

REF43

OP295/

OP495

0.01µF

1/2

IO (NORM) = 0.5A IO (MAX) = 1A

45.3k 1%

124k

+5V V

100

50mA

20mA

OUTPUT

20mV

1ms

STEP

CURRENT

CONTROL

WAVEFORM

Figure 11. Output Step Load Current Recovery

Low-Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting

Adding a second amplifier in the regulation loop as shown in Figure 12 provides an output current monitor as well as foldback current limiting protection.

Amplifier A1 provides error amplification for the normal voltage regulation loop. As long as the output current is less than 1 ampere, amplifier A2’s output swings to ground, reverse biasing the diode and effectively taking itself out of the circuit. However, as the output current exceeds 1 amp, the voltage that develops across the 0.1 Ω sense resistor forces the amplifier A2’s output to go high, forward-biasing the diode, which in turn closes the

Figure 12. Low Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting

Square Wave Oscillator

The circuit in Figure 13 is a square wave oscillator (note the positive feedback). The rail-to-rail swing of the OP295/OP495 helps maintain a constant oscillation frequency even if the supply voltage varies considerably. Consider a battery powered system where the voltages are not regulated and drop over time. The rail-to-rail swing ensures that the noninverting input sees the full V+/2, rather than only a fraction of it.

The constant frequency comes from the fact that the 58.7 kΩ feedback sets up Schmitt Trigger threshold levels that are directly proportional to the supply voltage, as are the RC charge voltage levels. As a result, the RC charge time, and therefore the frequency, remains constant independent of supply voltage. The slew rate of the amplifier limits the oscillation frequency to a maximum of about 800 Hz at a +5 V supply.

Single Supply Differential Speaker Driver

Connected as a differential speaker driver, the OP295/OP495 can deliver a minimum of 10 mA to the load. With a 600 Ω load, the OP295/OP495 can swing close to 5 volts peak-to-peak across the load.

–10–

REV. B

Page 11

OP295/OP495

V+

100k

58.7k

1/2

OP295/

OP495

FREQ OUT

= < 350Hz @ V+ = +5V

OSC

Figure 13. Square Wave Oscillator Has Stable Frequency Regardless of Supply Changes

90.9k

V+

1/4

OP295/

OP495

10k

2.2µF

10k 100k

20k20k

V+

1/4

OP295/

OP495

1/4

OP295/

OP495

SPEAKER

Figure 14. Single Supply Differential Speaker Driver

High Accuracy, Single-Supply, Low Power Comparator

The OP295/OP495 makes an accurate open-loop comparator. With a single +5 V supply, the offset error is less than 300 µV. Fig- ure 15 shows the OP295/OP495’s response time when operating open-loop with 4 mV overdrive. It exhibits a 4 ms response time at the rising edge and a 1.5 ms response time at the falling edge.

100

INPUT

(5mV OVERDRIVE

@ OP295 INPUT)

OUTPUT

10 0%

5ms2V

Figure 15. Open-Loop Comparator Response Time with 5 mV Overdrive

OP295/OP495 SPICE MODEL Macro-Model

* Node Assignments * Noninverting Input * Inverting Input * Positive Supply * Negative Supply * Output * * .SUBCKT OP295 1 2 99 50 20 * * INPUT STAGE * I1 99 4 2E-6 R1 1 6 5E3

R2 2 5 5E3 CIN 1 2 2E-12 IOS 1 2 0.5E-9 D1 5 3 DZ D2 6 3 DZ EOS 7 6 POLY (1) (31,39) 30E-6 0.024 Q1 8 5 4 QP Q2974QP R3 8 50 25.8E3 R4 9 50 25.8E3 * * GAIN STAGE * R7 10 98 270E6 G1 98 10 POLY (1)(9,8) –4.26712E-9 27.8E-6 EREF 98 0 (39, 0) 1 R5 99 39 100E3 R6 39 50 100E3 * * COMMON MODE STAGE * ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5 R12 30 31 1E6 R13 31 98 100 * * OUTPUT STAGE * I2 18 50 1.59E-6 V2 99 12 DC 2.2763 Q4 10 14 50 QNA 1.0 R11 14 50 33 M3 15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10 M4 13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11 D8 10 22 DX V3 22 50 DC 6 M2 20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 Q5 17 17 99 QPA 1.0 Q6 18 17 99 QPA 4.0 R8 18 99 2.2E6 Q7 18 19 99 QPA 1.0 R9 99 19 8 C2 18 99 20E-12 M6 15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405 E-12 M1 20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 D4 21 18 DX V4 99 21 DC 6 R10 10 11 6E3 C3 11 20 50E-12 .MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3 + ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4 + ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4 RC=209 + CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534 MJC=0.5 + CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30) .MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4 + ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5 + ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31 RC=354.4 + CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762 MJC=0.4 + CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30) .MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3 + TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5 + XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-4 PB=0.837 + MJ=0.407 CJSW=0.5E-9 MJSW=0.33) .MODEL MP PMOS (LEVEL=3 VTO=–1.1 RS=0.7 RD=0.7 + TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5 + XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=–1 CJ=1.55E-4 PB=0.56 + MJ=0.442 CJSW=0.4E-9 MJSW=0.33) .MODEL DX D(IS=1E-15) .MODEL DZ D (IS=1E-15, BV=7) .MODEL QP PNP (BF=125)

.ENDS

REV. B

–11–

Page 12

OP295/OP495

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm)

0.210 (5.33)

MAX

0.160 (4.06)

0.115 (2.93)

PIN 1

0.210 (5.33)

MAX

0.160 (4.06)

0.115 (2.92)

8 Lead Plastic DIP (P Suffix)

0.022 (0.558)

0.014 (0.356)

0.430 (10.92)

0.348 (8.84)

0.100 (2.54)

BSC

0.280 (7.11)

0.240 (6.10)

0.070 (1.77)

0.045 (1.15)

0.015 (0.381) TYP

SEATING PLANE

0.130 (3.30) MIN

0°- 15°

14-Lead Plastic DIP (P Suffix)

0.022 (0.558)

0.014 (0.36)

0.795 (20.19)

0.725 (18.42)

0.100

(2.54)

BSC

0.070 (1.77)

0.045 (1.15)

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

0.280 (7.11)

0.240 (6.10)

0.015 (0.381) MIN

0.130 (3.30) MIN

SEATING PLANE

0.325 (8.25)

0.300 (7.62)

0.015 (0.38)

0.008 (0.20)

15°

0°

0.195 (4.95)

0.115 (2.93)

0.0098 (0.25)

0.0040 (0.10)

PIN 1

0.0118 (0.30)

0.0040 (0.10)

8-Lead Narrow-Body SO (S Suffix)

0.1968 (5.00)

0.1890 (4.80)

0.0192 (0.49)

0.0138 (0.35)

0.0688 (1.75)

0.0532 (1.35)

0.1574 (4.00)

0.1497 (3.80)

0.2440 (6.20)

0.2284 (5.80)

0.0098 (0.25)

0.0075 (0.19)

PIN 1

0.0500 (1.27)

BSC

16-Lead Wide-Body SO (S Suffix)

0.2992 (7.60)

0.2914 (7.40)

0.4193 (10.65)

0.3937 (10.00)

0.0192 (0.49)

0.0138 (0.35)

0.1043 (2.65)

0.0926 (2.35)

0.0125 (0.32)

0.0091 (0.23)

0.4133 (10.50)

0.3977 (10.10)

0.0500 (1.27) BSC

0.0196 (0.50)

0.0099 (0.25)

0.0291 (0.74)

0.0098 (0.25)

x 45

0.0500 (1.27)

0.0160 (0.41)

x 45

0.0500 (1.27)

0.0157 (0.40)

C1806a–10–7/95

–12–

PRINTED IN U.S.A.

REV. B