Datasheet OP295 Datasheet (Analog Devices)

Page 1
OP295/OP495
FEATURES Rail-to-Rail Output Swing Single-Supply Operation: 3 V to 36 V Low Offset Voltage: 300 mV Gain Bandwidth Product: 75 kHz High Open-Loop Gain: 1,000 V/mV Unity-Gain Stable Low Supply Current/Per Amplifier: 150 A max
APPLICATIONS Battery-Operated Instrumentation Servo Amplifiers Actuator Drives Sensor Conditioners Power Supply Control

GENERAL DESCRIPTION

Rail-to-rail output swing combined with dc accuracy are the key features of the OP495 quad and OP295 dual CBCMOS operational amplifiers. By using a bipolar front end, lower noise and higher accuracy than that of CMOS designs has been achieved. Both input and output ranges include the negative supply, providing the user zero-in/zero-out capabil­ity. For users of 3.3 V systems such as lithium batteries, the OP295/OP495 is specified for 3 V operation.
Maximum offset voltage is specified at 300 µV for 5 V operation, and the open-loop gain is a minimum of 1000 V/mV. This yields performance that can be used to implement high accuracy systems, even in single-supply designs.
The ability to swing rail-to-rail and supply 15 mA to the load makes the OP295/OP495 an ideal driver for power transistors and “H” bridges. This allows designs to achieve higher efficien­cies and to transfer more power to the load than previously possible without the use of discrete components. For applica­tions such as transformers that require driving inductive loads,

PIN CONNECTIONS

8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
1
2
OP295
3
V–
4
8
7
6
5
V+
OUT B
–IN B
+IN B
14-Lead PDIP
(P Suffix)
1
OUT A
–IN A
+IN A
+IN B
–IN B
OUT B
2
3
4
V+
OP495
5
6
7
14
13
12
11
10
9
8
OUT D
–IN D
+IN D
V–
+IN C
–IN C
OUT C
8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
1
OP295
2
3
V–
4
8
7
6
5
V+
OUT B
–IN B
+IN B
14-Lead PDIP
(P Suffix)
OUT D
1
OUT A
–IN A
2
3
+IN A
4
V+
5
+IN B
6
–IN B
7
OUT B
8
NC
NC = NO CONNECT
OP495
TOP VIEW
(Not to Scale)
16
15
14
13
12
11
10
9
–IN D
+IN D
V–
+IN C
–IN C
OUT C
NC
increases in efficiency are also possible. Stability while driving capacitive loads is another benefit of this design over CMOS rail-to-rail amplifiers. This is useful for driving coax cable or large FET transistors. The OP295/OP495 is stable with loads in excess of 300 pF.
The OP295 and OP495 are specified over the extended industrial (–40°C to +125°C) temperature range. OP295s are available in 8-lead plastic DIP plus SOIC-8 surface-mount packages. OP495s are available in 14-lead plastic and SOIC-16 surface­mount packages.
REV. D
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 www.analog.com Fax: 781/326-8703 © 2004 Analog Devices, Inc. All rights reserved.
Page 2
OP295/OP495–SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
(@ VS = 5.0 V, VCM = 2.5 V, TA = 25C, unless otherwise noted.)
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
Input Bias Current I
Input Offset Current I
Input Voltage Range V Common-Mode Rejection Ratio CMRR 0 V ≤ VCM 4.0 V, –40°C TA +125°C90110 dB Large Signal Voltage Gain A
OS
B
OS
CM
VO
–40°C TA +125°C 800 µV
–40°C TA +125°C30nA
–40°C TA +125°C ±5nA
0 4.0 V
RL = 10 k, 0.005 V RL = 10 k, –40°C TA +125°C 500 V/mV
4.0 V 1,000 10,000 V/mV
OUT
30 300 µV
820 nA
±1 ±3nA
Offset Voltage Drift ∆VOS/T 15 µV/°C
OUTPUT CHARACTERISTICS
Output Voltage Swing High V
Output Voltage Swing Low V
Output Current I
OH
OL
OUT
RL = 100 k to GND 4.98 5.0 V RL = 10 k to GND 4.90 4.94 V I
= 1 mA, –40°C TA +125°C 4.7 V
OUT
RL = 100 k to GND 0.7 2 mV RL = 10 k to GND 0.7 2 mV I
= 1 mA, –40°C TA +125°C90mV
OUT
±11 ±18 mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR ± 1.5 V ≤ VS ± 15 V 90 110 dB
±1.5 V VS ± 15 V, –40°C TA +125°C85dB
Supply Current Per Amplifier I
SY
V
= 2.5 V, RL = , –40°C TA +125°C 150 µA
OUT
DYNAMIC PERFORMANCE
Skew Rate SR RL = 10 k 0.03 V/µs Gain Bandwidth Product GBP 75 kHz Phase Margin θ
O
86 Degrees
NOISE PERFORMANCE
Voltage Noise en p-p 0.1 Hz to 10 Hz 1.5 µV p-p Voltage Noise Density e Current Noise Density i
Specifications subject to change without notice.
n
n
f = 1 kHz 51 nV/Hz f = 1 kHz <0.1 pA/Hz
ELECTRICAL CHARACTERISTICS
(@ VS = 3.0 V, VCM = 1.5 V, TA = 25C, unless otherwise noted.)
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V Input Bias Current I Input Offset Current I Input Voltage Range V Common-Mode Rejection Ratio CMRR 0 V ≤ VCM 2.0 V, –40°C TA +125°C90110 dB Large Voltage Gain A Offset Voltage Drift ∆VOS/T 1 µV/°C
OS
B
OS
CM
VO
RL = 10 k 750 V/mV
0 2.0 V
100 500 µV 820 nA ±1 ±3nA
OUTPUT CHARACTERISTICS
Output Voltage Swing High V Output Voltage Swing Low V
OH
OL
RL = 10 k to GND 2.9 V RL = 10 k to GND 0.7 2 mV
POWER SUPPLY
Power Supply Rejection Ratio PSRR ± 1.5 V ≤ VS ± 15 V 90 110 dB
±1.5 V VS ± 15 V, –40°C TA +125°C85dB
Supply Current Per Amplifier I
SY
V
= 1.5 V, RL = , –40°C TA +125°C 150 µA
OUT
DYNAMIC PERFORMANCE
Slew Rate SR RL = 10 k 0.03 V/µs Gain Bandwidth Product GBP 75 kHz Phase Margin θ
O
85 Degrees
NOISE PERFORMANCE
Voltage Noise en p-p 0.1 Hz to 10 Hz 1.6 µV p-p Voltage Noise Density e Current Noise Density i
Specifications subject to change without notice.
n
n
f = 1 kHz 53 nV/Hz f = 1 kHz <0.1 pA/Hz
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Page 3
OP295/OP495
ELECTRICAL CHARACTERISTICS
(@ VS = ±15.0 V, TA = 25C, unless otherwise noted.)
Parameter Symbol Conditions Min Typ Max Unit
INPUT CHARACTERISTICS
Offset Voltage V
Input Bias Current I
OS
–40°C T
B
VCM = 0 V 7 20 nA
+125°C 800 µV
A
300 500 µV
VCM = 0 V, –40°C TA +125°C30nA
Input Offset Current I
Input Voltage Range V
OS
CM
Common-Mode Rejection Ratio CMRR –15.0 V ≤ V Large Signal Voltage Gain A
VO
VCM = 0 V ±1 ±3nA V
= 0 V, –40°C TA +125°C ±5nA
CM
–15 13.5 V
+13.5 V, –40°C TA +125°C90 110 dB
CM
RL = 10 k 1,000 4,000 V/mV
Offset Voltage Drift ∆VOS/T 1 µV/°C
OUTPUT CHARACTERISTICS
Output Voltage Swing High V
OH
RL = 100 k to GND 14.95 V RL = 10 kΩ to GND 14.80 V
Output Voltage Swing Low V
Output Current I
OL
OUT
RL = 100 k to GND –14.95 V R
= 10 k to GND –14.85 V
L
±15 ±25 mA
POWER SUPPLY
Power Supply Rejection Ratio PSRR VS = ± 1.5 V to ± 15 V 90 110 dB
VS = ±1.5 V to ±15 V, –40°C TA +125°C85 dB
Supply Current I
SY
VO = 0 V, RL = , VS = ±18 V, –40°C TA +125°C 175 µA
Supply Voltage Range V
S
3 (±1.5) 36 (±18) V
DYNAMIC PERFORMANCE
Slew Rate SR RL = 10 k 0.03 V/µs Gain Bandwidth Product GBP 85 kHz Phase Margin θ
O
83 Degrees
NOISE PERFORMANCE
Voltage Noise en p-p 0.1 Hz to 10 Hz 1.25 µV p-p Voltage Noise Density e
Current Noise Density i
Specifications subject to change without notice.
n
n
f =1 kHz 45 nV/Hz f = 1 kHz <0.1 pA/Hz
REV. D
–3–
Page 4
OP295/OP495

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
Input Voltage Differential Input Voltage
2
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
3
. . . . . . . . . . . . . . . . . . . . . . . . . 36 V
1, 2
Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Operating Temperature Range
OP295G, OP495G . . . . . . . . . . . . . . . . . . .–40°C to +125°C
Junction Temperature Range
P, S Package . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300°C

ORDERING GUIDE

Temperature Package Package
Model Range Description Option
OP295GP –40°C to +125°C 8-Lead Plastic DIP P-8 (N-8) OP295GS –40°C to +125°C 8-Lead SOIC S-8 (R-8) OP295GS-REEL –40°C to +125°C 8-Lead SOIC S-8 (R-8) OP295GS-REEL7 –40°C to +125°C 8-Lead SOIC S-8 (R-8) OP495GP –40°C to +125°C 14-Lead Plastic DIP P-14 (N-14) OP495GS –40°C to +125°C 16-Lead SOIC S-16 (RW-16) OP495GS-REEL –40°C to +125°C 16-Lead SOIC S-16 (RW-16) OP495GSZ* –40°C to +125°C 16-Lead SOIC S-16 (RW-16) OP495GSZ-REEL7* –40°C to +125°C 16-Lead SOIC S-16 (RW-16)
*Z = Pb-free part.
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma­nent damage to the device. This is a stress rating only; and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
2
Absolute maximum ratings apply to packaged parts, unless otherwise noted.
3
For supply voltages less than ± 18 V, the absolute maximum input voltage is equal to the supply voltage.
Package Type JA*
JC
Unit
8-Lead Plastic DIP (P) 103 43 °C/W 8-Lead SOIC (S) 158 43 °C/W 14-Lead Plastic DIP (P) 83 39 °C/W 16-Lead SOIC (S) 98 30 °C/W
*JA is specified for worst case mounting conditions, i.e., JA is specified for device
in socket for CERDIP, PDIP, and LCC packages; JA is specified for device soldered to printed circuit board for SOIC package.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the OP295/OP495 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.

Typical Performance Characteristics

140
120
100
80
60
SUPPLY CURRENT – ␮A
40
20
–50
–25
TEMPERATURE – ⴗC
VS = 36V
VS = 5V
V
= 3V
S
7550250
100
TPC 1. Supply Current Per Amplifier vs. Temperature
15.2
15.0
14.8
14.6
14.4
14.2
–14.4
–14.6
–14.8
–15.0
–15.2
– OUTPUT SWING – V + OUTPUT SWING – V
–50
–25
TEMPERATURE –
TPC 2. Output Voltage Swing vs. Temperature
VS = 15V
C
R
= 100k
L
RL = 10k
= 2k
R
L
RL = 2k
RL = 10k
RL = 100k
7550250
100
REV. D–4–
Page 5
OP295/OP495
500
0
300
150
50
–50
100
–100
300
200
250
350
400
450
250200150100500
INPUT OFFSET VOLTAGE – ␮V
UNITS
VS = 5V T
A
= 25ⴗC
BASED ON 1200 OP AMPS
500
0
3.2
150
50
0.4
100
0
300
200
250
350
400
450
2.82.42.01.61.20.8
T
C
– VOS – V/ⴗC
UNITS
VS = 5V –40
TA +85ⴗC
BASED ON 1200 OP AMPS
3.10
VS = 3V
3.00
2.90
2.80
2.70
OUTPUT VOLTAGE SWING – V
2.60
2.50 –50
–25
TEMPERATURE – ⴗC
RL = 100k
RL = 10k
RL = 2k
7550250
100
TPC 3. Output Voltage Swing vs. Temperature
200
BASED ON 600 OP AMPS
175
150
125
100
UNITS
75
VS = 5V
T
= 25ⴗC
A
5.10
VS = 5V
5.00
4.90
4.80
4.70
OUTPUT VOLTAGE SWING – V
4.60
4.50 –50
–25
TEMPERATURE – ⴗC
RL = 100k
RL = 10k
RL = 2k
7550250
100
TPC 6. Output Voltage Swing vs. Temperature
50
25
0
–200–250
TPC 4. OP295 Input Offset (VOS) Distribution
250
BASED ON 600 OP AMPS
225
200
175
150
125
UNITS
100
75
50
25
0
0
TPC 5. OP295 TC–VOS Distribution
REV. D
INPUT OFFSET VOLTAGE – ␮V
0.4
T
– VOS – V/ⴗC
C
VS = 5V
–40
200150100500–50–100–150
TA +85ⴗC
2.82.42.01.61.20.8
250
TPC 7. OP495 Input Offset (VOS) Distribution
3.2
TPC 8. OP495 TC–VOS Distribution
–5–
Page 6
OP295/OP495
20
VS = 5V
16
12
8
INPUT BIAS CURRENT – nA
4
0
–50
–25
TEMPERATURE – ⴗC
7550250
TPC 9. Input Bias Current vs. Temperature
40
35
30
25
20
15
OUTPUT CURRENT – mA
10
5
SINK
SOURCE
VS = 15V
SOURCE
SINK
VS = 5V
100
12
VS = 5V VO = 4V
10
8
RL = 100k
6
4
OPEN-LOOP GAIN – V/␮V
2
0
–50
–25
TEMPERATURE – ⴗC
RL = 10k
RL = 2k
7550250
TPC 12. Open-Loop Gain vs. Temperature
VS = 5V
= 25ⴗC
T
A
1V
100mV
OUTPUT VOLTAGE TO RAIL
10mV
1mV
SOURCE
SINK
100
0
–25–50
TEMPERATURE – ⴗC
7550250
TPC 10. Output Current vs. Temperature
100
VS = 15V V
= 10V
O
50
RL = 100k
RL = 10k
RL = 2k
75
10
OPEN-LOOP GAIN – V/␮V
1
–50 25
–25
0
TEMPERATURE – ⴗC
TPC 11. Open-Loop Gain vs. Temperature
100
100
100V
1A10␮A
100A
LOAD CURRENT
TPC 13. Output Voltage to Supply Rail vs.
Load Current
10mA1mA
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OP295/OP495
APPLICATIONS Rail-to-Rail Application Information
The OP295/OP495 has a wide common-mode input range extending from ground to within about 800 mV of the positive supply. There is a tendency to use the OP295/OP495 in buffer applications where the input voltage could exceed the common­mode input range. This may initially appear to work because of the high input range and rail-to-rail output range. But above the common-mode input range, the amplifier is, of course, highly nonlinear. For this reason, it is always required that there be some minimal amount of gain when rail-to-rail output swing is desired. Based on the input common-mode range, this gain should be at least 1.2.

Low Drop-Out Reference

The OP295/OP495 can be used to gain up a 2.5 V or other low voltage reference to 4.5 V for use with high resolution ADCs that operate from 5 V only supplies. The circuit in Figure 1 will supply up to 10 mA. Its no-load drop-out voltage is only 20 mV. This circuit will supply over 3.5 mA with a 5 V supply.
16k
5V
2
REF43
4
20k
6
OP295/OP495
0.001␮F
1/2
5V
10
V = 4.5V
OUT
1F TO
10 F
Figure 1. 4.5 V, Low Drop-Out Reference

Low Noise, Single-Supply Preamplifier

Most single-supply op amps are designed to draw low supply current at the expense of having higher voltage noise. This tradeoff may be necessary because the system must be powered by a battery. However, this condition is worsened because all circuit resistances tend to be higher; as a result, in addition to the op amp’s voltage noise, Johnson noise (resistor thermal noise) is also a significant contributor to the total noise of the system.
The choice of monolithic op amps that combine the character­istics of low noise and single-supply operation is rather limited. Most single-supply op amps have noise on the order of 30 nV/Hz to 60 nV/Hz and single-supply amplifiers with noise below 5 nV/Hz do not exist.
In order to achieve both low noise and low supply voltage opera­tion, discrete designs may provide the best solution. The circuit in Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a matched PNP transistor pair—the MAT03—to achieve zero-in/ zero-out single-supply operation with an input voltage noise of
3.1 nV/Hz at 100 Hz. R5 and R6 set the gain of 1,000, making this circuit ideal for maximizing dynamic range when amplifying low level signals in single-supply applications. The OP295/OP495 provide rail-to-rail output swings, allowing this circuit to operate with 0 V to 5 V outputs. Only half of the OP295/OP495 is used, leaving the other uncommitted op amp for use elsewhere.
0.1␮F
R1LED
Q2
2N3906
35
V
IN
2
R2 27k
Q1 Q2
MAT-03
R7
510
C1
R3
1500pF
R8
100
6
71
2
3
R4
10F
R5
10k
8
1
4
OP295/OP495
10
C2 10F
R6
V
OUT
Figure 2. Low Noise Single-Supply Preamplifier
The input noise is controlled by the MAT03 transistor pair and the collector current level. Increasing the collector current reduces the voltage noise. This particular circuit was tested with 1.85 mA and 0.5 mA of current. Under these two cases, the input voltage noise was 3.1 nV/Hz and 10 nV/Hz, respec­tively. The high collector currents do lead to a tradeoff in supply current, bias current, and current noise. All of these parameters increase with increasing collector current. For example, typi­cally the MAT03 has an h
= 165. This leads to bias currents
FE
of 11 µA and 3 µA, respectively. Based on the high bias cur- rents, this circuit is best suited for applications with low source impedance such as magnetic pickups or low impedance strain gages. Furthermore, a high source impedance degrades the noise performance. For example, a 1 kresistor generates 4 nV/Hz of broadband noise, which is already greater than the noise of the preamp.
The collector current is set by R1 in combination with the LED and Q2. The LED is a 1.6 V Zener diode that has a temperature coefficient close to that of Q2’s base-emitter junction, which provides a constant 1.0 V drop across R1. With R1 equal to 270 Ω, the tail current is 3.7 mA and the collector current is half that, or 1.85 mA. The value of R1 can be altered to adjust the collector current. Whenever R1 is changed, R3 and R4 should also be adjusted. To maintain a common-mode input range that includes ground, the collectors of the Q1 and Q2 should not go above
0.5 V—otherwise they could saturate. Thus, R3 and R4 must be small enough to prevent this condition. Their values and the overall performance for two different values of R1 are summa­rized in Table I. Lastly, the potentiometer, R8, is needed to adjust the offset voltage to null it to zero. Similar performance can be obtained using an OP90 as the output amplifier with a savings of about 185 µA of supply current. However, the output swing will not include the positive rail, and the bandwidth will reduce to approximately 250 Hz.
REV. D
–7–
Page 8
OP295/OP495
Table I. Single-Supply Low Noise Preamp Performance
IC = 1.85 mA IC = 0.5 mA
R1 270 1.0 k R3, R4 200 910
@ 100 Hz 3.15 nV/Hz 8.6 nV/Hz
e
n
en @ 10 Hz 4.2 nV/Hz 10.2 nV/Hz I
SY
I
B
4.0 mA 1.3 mA 11 µA3 µA
Bandwidth 1 kHz 1 kHz Closed-Loop Gain 1,000 1,000

Driving Heavy Loads

The OP295/OP495 is well suited to drive loads by using a power transistor, Darlington, or FET to increase the current to the load. The ability to swing to either rail can assure that the device is turned on hard. This results in more power to the load and an increase in efficiency over using standard op amps with their limited output swing. Driving power FETs is also possible with the OP295/OP495 because of its ability to drive capacitive loads of several hundred picofarads without oscillating.
Without the addition of external transistors, the OP295/OP495 can drive loads in excess of ±15 mA with ±15 V or +30 V supplies. This drive capability is somewhat decreased at lower supply voltages. At ±5 V supplies, the drive current is ±11 mA.
Driving motors or actuators in two directions in a single-supply application is often accomplished using an “H” bridge. The principle is demonstrated in Figure 3a. From a single 5 V sup­ply, this driver is capable of driving loads from 0.8 V to 4.2 V in both directions. Figure 3b shows the voltages at the inverting and non- inverting outputs of the driver. There is a small cross­over glitch that is frequency dependent and would not cause problems unless this was a low distortion application such as audio. If this is used to drive inductive loads, be sure to add diode clamps to protect the bridge from inductive kickback.
5V
2N2222
2N2907
0
VIN 2.5V
5k
1.67V
10k10k
10k
2N2222
OUTPUTS
2N2907
Figure 3a. “H” Bridge
100
90

Direct Access Arrangement

OP295/OP495 can be used in a single-supply direct access arrangement (DAA) as is shown in Figure 4. This figure shows a portion of a typical DM capable of operating from a single 5 V supply and it may also work on 3 V supplies with minor modifica­tions. Amplifiers A2 and A3 are configured so that the transmit signal TxA is inverted by A2 and is not inverted by A3. This arrangement drives the transformer differentially so that the drive to the transformer is effectively doubled over a single amplifier arrangement. This application takes advantage of the OP295/ OP495’s ability to drive capacitive loads, and to save power in single-supply applications.
390pF
37.4k
0.1␮F
RXA
0.0047␮F
OP295/
A1
3.3k
A2
OP295/ OP495
20k
20k
475
OP495
22.1k
0.1␮F
TXA
2.5V REF
20k
OP295/ OP495
750pF
20k
20k
A3
0.033␮F
1:1
Figure 4. Direct Access Arrangement

A Single-Supply Instrumentation Amplifier

The OP295/OP495 can be configured as a single-supply instru­mentation amplifier as in Figure 5. For our example, V equal to V+/2 and V
is measured with respect to V
O
REF
is set
REF
. The input common-mode voltage range includes ground and the output swings to both rails.
1/2
V+
OP295/
5
8
OP495
V
IN
R1
100k
V
REF
1/2
OP295/
OP495
3
2
1
R2
20k 20k 100k
VO = 5 +
R
G
200k
R
G
R3
VIN + V
4
6
R4
REF
V
7
O
10
0%
2V
2V
1ms
Figure 3b. “H” Bridge Outputs
Figure 5. Single-Supply Instrumentation Amplifier
Resistor RG sets the gain of the instrumentation amplifier. Mini­mum gain is 6 (with no R
). All resistors should be matched in
G
absolute value as well as temperature coefficient to maximize common-mode rejection performance and minimize drift. This instrumentation amplifier can operate from a supply voltage as low as 3 V.
REV. D–8–
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OP295/OP495
5V 5V
R1
17.8k
1.23V
AD589
3
4 765
GND CLK SRI
LD
V
DD
V
REF
R
FB
2
1
3
2
4
1
8
DIGITAL
CONTROL
5V
OP295/ OP495
R3 5k
R2
41.2k
R4
100k
D
4096
V
O
= (4.096V)
TOTA L POWER DISSIPATION = 1.6mW
I
OUT
8
DAC8043

A Single-Supply RTD Thermometer Amplifier

This RTD amplifier takes advantage of the rail-to-rail swing of the OP295/OP495 to achieve a high bridge voltage in spite of a low 5 V supply. The OP295/OP495 amplifier servos a constant 200 µA current to the bridge. The return current drops across the parallel resistors 6.19 kand the 2.55 M, developing a voltage that is servoed to 1.235 V, which is established by the AD589 band gap reference. The 3-wire RTD provides an equal line resistance drop in both 100 legs of the bridge, thus improving the accuracy.
The AMP04 amplifies the differential bridge signal and converts it to a single-ended output. The gain is set by the series resis­tance of the 332 resistor plus the 50 potentiometer. The gain scales the output to produce a 4.5 V full scale. The 0.22 µF capacitor to the output provides a 7 Hz low-pass filter to keep noise at a minimum.
ZERO ADJ
10-TURNS
26.7k
2.55M 1%
0.5%
100 RTD
200
100
0.5%
6.19k 1%
26.7k
0.5%
2
AD589
1
OP295/
OP495
3
1.235
5V
7
3
2
1/2
37.4k
1
AMP04
4
8
5
5V
50
332
0.22␮F
6
4.5V = 450ⴗC 0V = 0ⴗC
V
O
bath or oven and adjust the scale-adjust potentiometer for an output voltage of 2.50 V, which is equivalent to 250°C. Within this temperature range, the K-type thermocouple is quite accu­rate and produces a fairly linear transfer characteristic. Accuracy of ±3°C is achievable without linearization.
Even if the battery voltage is allowed to decay to as low as 7 V, the rail-to-rail swing allows temperature measurements to 700°C. However, linearization may be necessary for temperatures above 250°C where the thermocouple becomes rather nonlinear. The circuit draws just under 500 µA supply current from a 9 V battery.

A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V

Figure 8 shows a complete voltage output DAC with wide out­put voltage swing operating off a single 5 V supply. The serial input 12-bit DAC is configured as a voltage output device with the 1.235 V reference feeding the current output pin (I the DAC. The V
, which is normally the input now becomes
REF
OUT
) of
the output.
The output voltage from the DAC is the binary weighted volt­age of the reference, which is gained up by the output amplifier such that the DAC has a 1 mV per bit transfer function.

A Cold Junction Compensated, Battery-Powered Thermocouple Amplifier

The OP295/OP495’s 150 µA quiescent current per amplifier consumption makes it useful for battery-powered temperature measuring instruments. The K-type thermocouple terminates into an isothermal block where the terminated junctions’ ambi­ent temperatures can be continuously monitored and corrected by summing an equal but opposite thermal EMF to the amplifier, thereby canceling the error introduced by the cold junctions.
ISOTHERMAL
BLOCK
ALUMEL
AL
CR
CHROMEL
K-TYPE THERMOCOUPLE
40.7V/ ⴗC
Figure 7. Battery-Powered, Cold-Junction Compensated Thermocouple Amplifier
To calibrate, immerse the thermocouple measuring junction in a 0°C ice bath, adjust the 500 zero-adjust potentiometer to 0 V out. Then immerse the thermocouple in a 250°C temperature
REV. D
Figure 6. Low Power RTD Amplifier
1.235V
24.9k
1N914
COLD JUNCTIONS
AD589
7.15k
1.5M1%24.9k
475
1%
1%
1%
24.3k 1%
4.99k 1%
500 10-TURN
ZERO ADJUST
2.1k 1%
9V
1.33M
8
2
3
4
SCALE
ADJUST
20k
1
OP295/ OP495
V
O
5V = 500ⴗC 0V = 0ⴗC
Figure 8. A 5 V 12-Bit DAC with 0 V to 4.095 Output Swing

4 mA to 20 mA Current Loop Transmitter

Figure 9 shows a self-powered 4 to 20 mA current loop trans­mitter. The entire circuit floats up from the single-supply (12 V to 36 V) return. The supply current carries the signal within the 4 to 20 mA range. Thus the 4 mA establishes the baseline current budget with which the circuit must operate. This circuit consumes only 1.4 mA maximum quiescent current, making 2.6 mA of current available to power additional signal conditioning circuitry or to power a bridge circuit.
SPAN ADJ
V
IN
0 + 3V
10k
10-TURN
10-TURN
182k
1%
100k
1.21M 1%
HP 5082-2800
NULL ADJ
3
2
220pF
100k
1%
62
REF02
GND
4
5V
8
4
1/2
100
220
1
2N1711
OP295/
OP495
100
1%
4 TO
20mA
12V
TO
36V
Figure 9. 4 to 20 mA Current Loop Transmitter
–9–
R
L
100
Page 10
OP295/OP495

A 3 V Low-Dropout Linear Voltage Regulator

Figure 10 shows a simple 3 V voltage regulator design. The regulator can deliver 50 mA load current while allowing a 0.2 V dropout voltage. The OP295/OP495’s rail-to-rail output swing handily drives the MJE350 pass transistor without requiring special drive circuitry. At no load, its output can swing less than the pass transistor’s base-emitter voltage, turning the device nearly off. At full load, and at low emitter-collector voltages, the transistor beta tends to decrease. The additional base current is easily handled by the OP295/OP495 output.
The amplifier servos the output to a constant voltage, which feeds a portion of the signal to the error amplifier.
Higher output current, to 100 mA, is achievable at a higher dropout voltage of 3.8 V.
< 50mA
I
1.235V
L
44.2k 1%
30.9k 1%
OP295/
OP495
1/2
100F
V
O
V
5V TO 3.2V
MJE 350
IN
43k
1
1000pF
8
4
AD589
3
2
Figure 10. 3 V Low Dropout Voltage Regulator
Figure 11 shows the regulator’s recovery characteristic when its output underwent a 20 mA to 50 mA step current change.
2V
100
50mA
20mA
OUTPUT
90
10
0%
20mV
1ms
STEP CURRENT CONTROL
WAVEFORM
Figure 11. Output Step Load Current Recovery

Low-Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting

Adding a second amplifier in the regulation loop, as shown in Figure 12, provides an output current monitor as well as fold­back current limiting protection.
Amplifier A1 provides error amplification for the normal voltage regulation loop. As long as the output current is less than 1 A, amplifier A2’s output swings to ground, reverse biasing the diode and effectively taking itself out of the circuit. However, as the output current exceeds 1 A, the voltage that develops across the 0.1 sense resistor forces the amplifier A2’s output to go high, forward-biasing the diode, which in turn closes the current limit loop. At this point A2’s lower output resistance dominates
the drive to the power MOSFET transistor, thereby effectively removing the A1 voltage regulation loop from the circuit.
If the output current greater than 1 A persists, the current limit loop forces a reduction of current to the load, which causes a corresponding drop in output voltage. As the output voltage drops, the current limit threshold also drops fractionally, resulting in a decreasing output current as the output voltage decreases, to the limit of less than 0.2 A at 1 V output. This “fold-back” effect reduces the power dissipation considerably during a short circuit condition, thus making the power supply far more forgiving in terms of the thermal design requirements. Small heat sinking on the power MOSFET can be tolerated.
The OP295’s rail-to-rail swing exacts higher gate drive to the power MOSFET, providing a fuller enhancement to the tran­sistor. The regulator exhibits 0.2 V dropout at 500 mA of load current. At 1 A output, the dropout voltage is typically 5.6 V.
I
SENSE
0.1 1/4W
(NORM) = 0.5A
O
(MAX) = 1A
I
O
205k 1%
45.3k 1%
124k
1%
5V V
O
IRF9531
SD
6V
G
1N4148
7
1/2
OP295/
5%
OP495
0.01␮F
1
1/2
100k
OP295/
R
210k 1%
8
5
A2
6
45.3k 1%
3
124k
A1
1%
4
2
OP495
REF43
2
6
4
2.500V
Figure 12. Low Dropout, 500 mA Voltage Regulator with Fold-Back Current Limiting

Square Wave Oscillator

The circuit in Figure 13 is a square wave oscillator (note the positive feedback). The rail-to-rail swing of the OP295/OP495 helps maintain a constant oscillation frequency even if the sup­ply voltage varies considerably. Consider a battery-powered system where the voltages are not regulated and drop over time. The rail-to-rail swing ensures that the noninverting input sees the full V+/2, rather than only a fraction of it.
The constant frequency comes from the fact that the 58.7 k feedback sets up Schmitt trigger threshold levels that are directly proportional to the supply voltage, as are the RC charge voltage levels. As a result, the RC charge time, and therefore, the fre­quency, remains constant independent of supply voltage. The slew rate of the amplifier limits oscillation frequency to a maxi­mum of about 800 Hz at a 5 V supply.

Single-Supply Differential Speaker Driver

Connected as a differential speaker driver, the OP295/OP495 can deliver a minimum of 10 mA to the load. With a 600 load, the OP295/OP495 can swing close to 5 V p-p across the load.
REV. D–10–
Page 11
OP295/OP495
V+
100k
100k
58.7k
C
8
3
1
1/2
4
2
OP295/ OP495
R
FREQ OUT
F
OSC
1
=
< 350Hz @ V+ = 5V
RC
Figure 13. Square Wave Oscillator Has Stable Frequency Regardless of Supply Changes
90.9k90.9k
V
IN
20k 20k
V+
10k
2.2␮F
10k
1/4 OP295/ OP495
100k
V+
1/4 OP295/ OP495
1/4 OP295/ OP495
SPEAKER
Figure 14. Single-Supply Differential Speaker Driver

High Accuracy, Single-Supply, Low Power Comparator

The OP295/OP495 makes an accurate open-loop comparator. With a single 5 V supply, the offset error is less than 300 µV. Figure 15 shows the OP295/OP495’s response time when operating open-loop with 4 mV overdrive. It exhibits a 4 ms response time at the rising edge and a 1.5 ms response time at the falling edge.
1V
100
90
INPUT
(5mV OVERDRIVE
@ OP-295 INPUT)
OUTPUT
10
0%
2V
5ms
Figure 15. Open-Loop Comparator Response Time with 5 mV Overdrive

OP295/OP495 SPICE MODEL Macro-Model

* Node Assignments * Noninverting Input * Inverting Input * Positive Supply * Negative Supply * Output * * .SUBCKT OP295 1 2 99 50 20 * * INPUT STAGE *
REV. D
–11–
I1 99 4 2E-6 R1 1 6 5E3 R2 2 5 5E3 CIN 1 2 2E-12 IOS 1 2 0.5E-9 D1 5 3 DZ D2 6 3 DZ EOS 7 6 POLY (1) (31,39) 30E-6 0.024 Q1 8 5 4 QP Q2 9 7 4QP R3 8 50 25.8E3 R4 9 50 25.8E3 * * GAIN STAGE * R7 10 98 270E6 G1 98 10 POLY (1) (9,8) –4.26712E-9 27.8E-6 EREF 98 0 (39, 0) 1 R5 99 39 100E3 R6 39 50 100E3 * * COMMON MODE STAGE * ECM 30 98 POLY(2) (1,39) (2,39) 0 0.5 0.5 R12 30 31 1E6 R13 31 98 100 * * OUTPUT STAGE * I2 18 50 1.59E-6 V2 99 12 DC 2.2763 Q4 10 14 50 QNA 1.0 R11 14 50 33 M3 15 10 13 13 MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10 M4 13 10 50 50 MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11 D8 10 22 DX V3 22 50 DC 6 M2 20 10 14 14 MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 Q5 17 17 99 QPA 1.0 Q6 18 17 99 QPA 4.0 R8 18 99 2.2E6 Q7 18 19 99 QPA 1.0 R9 99 19 8 C2 18 99 20E-12 M6 15 12 17 99 MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12 M1 20 18 19 99 MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9 D4 21 18 DX V4 99 21 DC 6 R10 10 11 6E3 C3 11 20 50E-12 .MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3 + ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4 + ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4 RC=209 + CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534 MJC=0.5 + CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30) .MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4 + ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5 + ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31 RC=354.4 + CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762 MJC=0.4 + CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30) .MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3 + TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5 + XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E­4 PB=0.837 + MJ=0.407 CJSW=0.5E-9 MJSW=0.33) .MODEL MP PMOS (LEVEL=3 VTO=–1.1 RS=0.7 RD=0.7 + TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5 + XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=–1 CJ=1.55E-4
PB=0.56 + MJ=0.442 CJSW=0.4E-9 MJSW=0.33) .MODEL DX D(IS=1E-15) .MODEL DZ D (IS=1E-15, BV=7) .MODEL QP PNP (BF=125)
.ENDS
Page 12
OP295/OP495

OUTLINE DIMENSIONS

8-Lead Plastic Dual In-Line Package [PDIP]
(N-8)
P-Suffix
Dimensions shown in inches and (millimeters)
0.375 (9.53)
0.365 (9.27)
0.355 (9.02)
8
1
0.100 (2.54)
0.180 (4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79)
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-095AA
BSC
5
4
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.015 (0.38) MIN
SEATING PLANE
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
8-Lead Standard Small Outline Package [SOIC]
Narrow Body
(R-8)
S-Suffix
Dimensions shown in millimeters and (inches)
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
0.25 (0.0098)
0.10 (0.0040)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
85
1.27 (0.0500)
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MS-012AA
BSC
6.20 (0.2440)
5.80 (0.2284)
41
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
0.31 (0.0122)
0.25 (0.0098)
0.17 (0.0067)
0.50 (0.0196)
0.25 (0.0099)
8
1.27 (0.0500)
0
0.40 (0.0157)
45
14-Lead Plastic Dual In-Line Package [PDIP]
(N-14)
P-Suffix
Dimensions shown in inches and (millimeters)
0.685 (17.40)
0.665 (16.89)
0.645 (16.38)
14
1
0.100 (2.54) BSC
0.015 (0.38)
0.180 (4.57) MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
COMPLIANT TO JEDEC STANDARDS MO-095-AB
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
8
7
MIN
0.295 (7.49)
0.285 (7.24)
0.275 (6.99)
SEATING PLANE
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
16-Lead Standard Small Outline Package [SOIC]
Wide Body
(RW-16) S-Suffix
Dimensions shown in millimeters and (inches)
10.50 (0.4134)
10.10 (0.3976)
16
1
1.27 (0.0500) BSC
0.30 (0.0118)
0.10 (0.0039)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
0.51 (0.0201)
0.31 (0.0122)
COMPLIANT TO JEDEC STANDARDS MS-013AA
9
7.60 (0.2992)
7.40 (0.2913)
8
2.65 (0.1043)
2.35 (0.0925)
SEATING PLANE
10.65 (0.4193)
10.00 (0.3937)
0.33 (0.0130)
0.20 (0.0079)
8 0
0.75 (0.0295)
0.25 (0.0098)
1.27 (0.0500)
0.40 (0.0157)
45
REV. D–12–
Page 13
OP295/OP495

Revision History

Location Page
2/04—Data Sheet changed from REV. C to REV. D.
Changes to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Changes to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
Changes to Absolute Maximum Ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
3/02—Data Sheet changed from REV. B to REV. C.
Figure changes to PIN CONNECTIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Deletion of OP295GBC and OP495GBC from ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Deletion of WAFER TEST LIMITS table . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Changes to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Deletion of DICE CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
REV. D
–13–
Page 14
–14–
Page 15
–15–
Page 16
C00331–0–2/04(D)
–16–
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