FEATURES
Rail-to-Rail Output Swing
Single-Supply Operation: 3 V to 36 V
Low Offset Voltage: 300 mV
Gain Bandwidth Product: 75 kHz
High Open-Loop Gain: 1,000 V/mV
Unity-Gain Stable
Low Supply Current/Per Amplifier: 150 A max
APPLICATIONS
Battery-Operated Instrumentation
Servo Amplifiers
Actuator Drives
Sensor Conditioners
Power Supply Control
GENERAL DESCRIPTION
Rail-to-rail output swing combined with dc accuracy are the
key features of the OP495 quad and OP295 dual CBCMOS
operational amplifiers. By using a bipolar front end, lower
noise and higher accuracy than that of CMOS designs has
been achieved. Both input and output ranges include the
negative supply, providing the user zero-in/zero-out capability. For users of 3.3 V systems such as lithium batteries, the
OP295/OP495 is specified for 3 V operation.
Maximum offset voltage is specified at 300 µV for 5 V operation,
and the open-loop gain is a minimum of 1000 V/mV. This yields
performance that can be used to implement high accuracy systems,
even in single-supply designs.
The ability to swing rail-to-rail and supply 15 mA to the load
makes the OP295/OP495 an ideal driver for power transistors
and “H” bridges. This allows designs to achieve higher efficiencies and to transfer more power to the load than previously
possible without the use of discrete components. For applications such as transformers that require driving inductive loads,
PIN CONNECTIONS
8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
1
2
OP295
3
V–
4
8
7
6
5
V+
OUT B
–IN B
+IN B
14-Lead PDIP
(P Suffix)
1
OUT A
–IN A
+IN A
+IN B
–IN B
OUT B
2
3
4
V+
OP495
5
6
7
14
13
12
11
10
9
8
OUT D
–IN D
+IN D
V–
+IN C
–IN C
OUT C
8-Lead Narrow-Body SOIC
(S Suffix)
OUT A
–IN A
+IN A
1
OP295
2
3
V–
4
8
7
6
5
V+
OUT B
–IN B
+IN B
14-Lead PDIP
(P Suffix)
OUT D
1
OUT A
–IN A
2
3
+IN A
4
V+
5
+IN B
6
–IN B
7
OUT B
8
NC
NC = NO CONNECT
OP495
TOP VIEW
(Not to Scale)
16
15
14
13
12
11
10
9
–IN D
+IN D
V–
+IN C
–IN C
OUT C
NC
increases in efficiency are also possible. Stability while driving
capacitive loads is another benefit of this design over CMOS
rail-to-rail amplifiers. This is useful for driving coax cable or
large FET transistors. The OP295/OP495 is stable with loads in
excess of 300 pF.
The OP295 and OP495 are specified over the extended industrial
(–40°C to +125°C) temperature range. OP295s are available
in 8-lead plastic DIP plus SOIC-8 surface-mount packages.
OP495s are available in 14-lead plastic and SOIC-16 surfacemount packages.
REV. D
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
Lead Temperature Range (Soldering, 60 sec) . . . . . . . . 300°C
ORDERING GUIDE
TemperaturePackagePackage
ModelRangeDescriptionOption
OP295GP–40°C to +125°C8-Lead Plastic DIPP-8 (N-8)
OP295GS–40°C to +125°C8-Lead SOICS-8 (R-8)
OP295GS-REEL–40°C to +125°C8-Lead SOICS-8 (R-8)
OP295GS-REEL7–40°C to +125°C8-Lead SOICS-8 (R-8)
OP495GP–40°C to +125°C14-Lead Plastic DIPP-14 (N-14)
OP495GS–40°C to +125°C16-Lead SOICS-16 (RW-16)
OP495GS-REEL–40°C to +125°C16-Lead SOICS-16 (RW-16)
OP495GSZ*–40°C to +125°C16-Lead SOICS-16 (RW-16)
OP495GSZ-REEL7* –40°C to +125°C16-Lead SOICS-16 (RW-16)
*Z = Pb-free part.
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; and functional operation
of the device at these or any other conditions above those indicated in the
operational section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
2
Absolute maximum ratings apply to packaged parts, unless otherwise noted.
3
For supply voltages less than ± 18 V, the absolute maximum input voltage is equal
to the supply voltage.
*JA is specified for worst case mounting conditions, i.e., JA is specified for device
in socket for CERDIP, PDIP, and LCC packages; JA is specified for device
soldered to printed circuit board for SOIC package.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
OP295/OP495 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
Typical Performance Characteristics
140
120
100
80
60
SUPPLY CURRENT – A
40
20
–50
–25
TEMPERATURE – ⴗC
VS = 36V
VS = 5V
V
= 3V
S
7550250
100
TPC 1. Supply Current Per Amplifier vs. Temperature
15.2
15.0
14.8
14.6
14.4
14.2
–14.4
–14.6
–14.8
–15.0
–15.2
– OUTPUT SWING – V+ OUTPUT SWING – V
–50
–25
TEMPERATURE –
TPC 2. Output Voltage Swing vs. Temperature
VS = 15V
C
R
= 100k⍀
L
RL = 10k⍀
= 2k⍀
R
L
RL = 2k⍀
RL = 10k⍀
RL = 100k⍀
7550250
100
REV. D–4–
Page 5
OP295/OP495
500
0
300
150
50
–50
100
–100
300
200
250
350
400
450
250200150100500
INPUT OFFSET VOLTAGE – V
UNITS
VS = 5V
T
A
= 25ⴗC
BASED ON 1200 OP AMPS
500
0
3.2
150
50
0.4
100
0
300
200
250
350
400
450
2.82.42.01.61.20.8
T
C
– VOS – V/ⴗC
UNITS
VS = 5V
–40ⴗ
TA +85ⴗC
BASED ON 1200 OP AMPS
3.10
VS = 3V
3.00
2.90
2.80
2.70
OUTPUT VOLTAGE SWING – V
2.60
2.50
–50
–25
TEMPERATURE – ⴗC
RL = 100k⍀
RL = 10k⍀
RL = 2k⍀
7550250
100
TPC 3. Output Voltage Swing vs. Temperature
200
BASED ON 600 OP AMPS
175
150
125
100
UNITS
75
VS = 5V
T
= 25ⴗC
A
5.10
VS = 5V
5.00
4.90
4.80
4.70
OUTPUT VOLTAGE SWING – V
4.60
4.50
–50
–25
TEMPERATURE – ⴗC
RL = 100k⍀
RL = 10k⍀
RL = 2k⍀
7550250
100
TPC 6. Output Voltage Swing vs. Temperature
50
25
0
–200–250
TPC 4. OP295 Input Offset (VOS) Distribution
250
BASED ON 600 OP AMPS
225
200
175
150
125
UNITS
100
75
50
25
0
0
TPC 5. OP295 TC–VOS Distribution
REV. D
INPUT OFFSET VOLTAGE – V
0.4
T
– VOS – V/ⴗC
C
VS = 5V
–40ⴗ
200150100500–50–100–150
TA +85ⴗC
2.82.42.01.61.20.8
250
TPC 7. OP495 Input Offset (VOS) Distribution
3.2
TPC 8. OP495 TC–VOS Distribution
–5–
Page 6
OP295/OP495
20
VS = 5V
16
12
8
INPUT BIAS CURRENT – nA
4
0
–50
–25
TEMPERATURE – ⴗC
7550250
TPC 9. Input Bias Current vs. Temperature
40
35
30
25
20
15
OUTPUT CURRENT – mA
10
5
SINK
SOURCE
VS = 15V
SOURCE
SINK
VS = 5V
100
12
VS = 5V
VO = 4V
10
8
RL = 100k⍀
6
4
OPEN-LOOP GAIN – V/V
2
0
–50
–25
TEMPERATURE – ⴗC
RL = 10k⍀
RL = 2k⍀
7550250
TPC 12. Open-Loop Gain vs. Temperature
VS = 5V
= 25ⴗC
T
A
1V
100mV
OUTPUT VOLTAGE TO RAIL
10mV
1mV
SOURCE
SINK
100
0
–25–50
TEMPERATURE – ⴗC
7550250
TPC 10. Output Current vs. Temperature
100
VS = 15V
V
= 10V
O
50
RL = 100k⍀
RL = 10k⍀
RL = 2k⍀
75
10
OPEN-LOOP GAIN – V/V
1
–5025
–25
0
TEMPERATURE – ⴗC
TPC 11. Open-Loop Gain vs. Temperature
100
100
100V
1A10A
100A
LOAD CURRENT
TPC 13. Output Voltage to Supply Rail vs.
Load Current
10mA1mA
REV. D–6–
Page 7
OP295/OP495
APPLICATIONS
Rail-to-Rail Application Information
The OP295/OP495 has a wide common-mode input range
extending from ground to within about 800 mV of the positive
supply. There is a tendency to use the OP295/OP495 in buffer
applications where the input voltage could exceed the commonmode input range. This may initially appear to work because of
the high input range and rail-to-rail output range. But above the
common-mode input range, the amplifier is, of course, highly
nonlinear. For this reason, it is always required that there be
some minimal amount of gain when rail-to-rail output swing is
desired. Based on the input common-mode range, this gain
should be at least 1.2.
Low Drop-Out Reference
The OP295/OP495 can be used to gain up a 2.5 V or other low
voltage reference to 4.5 V for use with high resolution ADCs
that operate from 5 V only supplies. The circuit in Figure 1 will
supply up to 10 mA. Its no-load drop-out voltage is only 20 mV.
This circuit will supply over 3.5 mA with a 5 V supply.
16k⍀
5V
2
REF43
4
20k⍀
6
OP295/OP495
0.001F
1/2
5V
10⍀
V = 4.5V
OUT
1F TO
10 F
Figure 1. 4.5 V, Low Drop-Out Reference
Low Noise, Single-Supply Preamplifier
Most single-supply op amps are designed to draw low supply
current at the expense of having higher voltage noise. This
tradeoff may be necessary because the system must be powered
by a battery. However, this condition is worsened because all
circuit resistances tend to be higher; as a result, in addition to the
op amp’s voltage noise, Johnson noise (resistor thermal noise) is
also a significant contributor to the total noise of the system.
The choice of monolithic op amps that combine the characteristics of low noise and single-supply operation is rather limited.
Most single-supply op amps have noise on the order of 30 nV/√Hz
to 60 nV/√Hzand single-supply amplifiers with noise below
5 nV/√Hzdo not exist.
In order to achieve both low noise and low supply voltage operation, discrete designs may provide the best solution. The circuit
in Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a
matched PNP transistor pair—the MAT03—to achieve zero-in/
zero-out single-supply operation with an input voltage noise of
3.1 nV/√Hzat 100 Hz. R5 and R6 set the gain of 1,000, making
this circuit ideal for maximizing dynamic range when amplifying
low level signals in single-supply applications. The OP295/OP495
provide rail-to-rail output swings, allowing this circuit to operate
with 0 V to 5 V outputs. Only half of the OP295/OP495 is used,
leaving the other uncommitted op amp for use elsewhere.
0.1F
R1LED
Q2
2N3906
35
V
IN
2
R2
27k⍀
Q1Q2
MAT-03
R7
510⍀
C1
R3
1500pF
R8
100⍀
6
71
2
3
R4
10F
R5
10k⍀
8
1
4
OP295/OP495
10⍀
C2
10F
R6
V
OUT
Figure 2. Low Noise Single-Supply Preamplifier
The input noise is controlled by the MAT03 transistor pair
and the collector current level. Increasing the collector current
reduces the voltage noise. This particular circuit was tested
with 1.85 mA and 0.5 mA of current. Under these two cases,
the input voltage noise was 3.1 nV/√Hzand 10 nV/√Hz, respectively. The high collector currents do lead to a tradeoff in supply
current, bias current, and current noise. All of these parameters
increase with increasing collector current. For example, typically the MAT03 has an h
= 165. This leads to bias currents
FE
of 11 µA and 3 µA, respectively. Based on the high bias cur-
rents, this circuit is best suited for applications with low source
impedance such as magnetic pickups or low impedance strain
gages. Furthermore, a high source impedance degrades the noise
performance. For example, a 1 kΩ resistor generates 4 nV/√Hz
of broadband noise, which is already greater than the noise of
the preamp.
The collector current is set by R1 in combination with the LED
and Q2. The LED is a 1.6 V Zener diode that has a temperature
coefficient close to that of Q2’s base-emitter junction, which
provides a constant 1.0 V drop across R1. With R1 equal to 270 Ω,
the tail current is 3.7 mA and the collector current is half that,
or 1.85 mA. The value of R1 can be altered to adjust the collector
current. Whenever R1 is changed, R3 and R4 should also be
adjusted. To maintain a common-mode input range that includes
ground, the collectors of the Q1 and Q2 should not go above
0.5 V—otherwise they could saturate. Thus, R3 and R4 must
be small enough to prevent this condition. Their values and the
overall performance for two different values of R1 are summarized in Table I. Lastly, the potentiometer, R8, is needed to
adjust the offset voltage to null it to zero. Similar performance
can be obtained using an OP90 as the output amplifier with a
savings of about 185 µA of supply current. However, the output
swing will not include the positive rail, and the bandwidth will
reduce to approximately 250 Hz.
REV. D
–7–
Page 8
OP295/OP495
Table I. Single-Supply Low Noise Preamp Performance
IC = 1.85 mAIC = 0.5 mA
R1270 Ω1.0 kΩ
R3, R4200 Ω910 Ω
@ 100 Hz3.15 nV/√Hz8.6 nV/√Hz
e
n
en @ 10 Hz4.2 nV/√Hz10.2 nV/√Hz
I
SY
I
B
4.0 mA1.3 mA
11 µA3 µA
Bandwidth1 kHz1 kHz
Closed-Loop Gain1,0001,000
Driving Heavy Loads
The OP295/OP495 is well suited to drive loads by using a power
transistor, Darlington, or FET to increase the current to the load.
The ability to swing to either rail can assure that the device is
turned on hard. This results in more power to the load and an
increase in efficiency over using standard op amps with their
limited output swing. Driving power FETs is also possible with
the OP295/OP495 because of its ability to drive capacitive loads
of several hundred picofarads without oscillating.
Without the addition of external transistors, the OP295/OP495
can drive loads in excess of ±15 mA with ±15 V or +30 V
supplies. This drive capability is somewhat decreased at lower
supply voltages. At ±5 V supplies, the drive current is ±11 mA.
Driving motors or actuators in two directions in a single-supply
application is often accomplished using an “H” bridge. The
principle is demonstrated in Figure 3a. From a single 5 V supply, this driver is capable of driving loads from 0.8 V to 4.2 V in
both directions. Figure 3b shows the voltages at the inverting
and non- inverting outputs of the driver. There is a small crossover glitch that is frequency dependent and would not cause
problems unless this was a low distortion application such as
audio. If this is used to drive inductive loads, be sure to add
diode clamps to protect the bridge from inductive kickback.
5V
2N2222
2N2907
0
VIN 2.5V
5k⍀
1.67V
10k⍀ 10k⍀
10k⍀
2N2222
OUTPUTS
2N2907
Figure 3a. “H” Bridge
100
90
Direct Access Arrangement
OP295/OP495 can be used in a single-supply direct access
arrangement (DAA) as is shown in Figure 4. This figure shows a
portion of a typical DM capable of operating from a single 5 V
supply and it may also work on 3 V supplies with minor modifications. Amplifiers A2 and A3 are configured so that the transmit
signal TxA is inverted by A2 and is not inverted by A3. This
arrangement drives the transformer differentially so that the drive
to the transformer is effectively doubled over a single amplifier
arrangement. This application takes advantage of the OP295/
OP495’s ability to drive capacitive loads, and to save power in
single-supply applications.
390pF
37.4k⍀
0.1F
RXA
0.0047F
OP295/
A1
3.3k⍀
A2
OP295/
OP495
20k⍀
20k⍀
475⍀
OP495
22.1k⍀
0.1F
TXA
2.5V REF
20k⍀
OP295/
OP495
750pF
20k⍀
20k⍀
A3
0.033F
1:1
Figure 4. Direct Access Arrangement
A Single-Supply Instrumentation Amplifier
The OP295/OP495 can be configured as a single-supply instrumentation amplifier as in Figure 5. For our example, V
equal to V+/2 and V
is measured with respect to V
O
REF
is set
REF
. The
input common-mode voltage range includes ground and the
output swings to both rails.
1/2
V+
OP295/
5
8
OP495
V
IN
R1
100k⍀
V
REF
1/2
OP295/
OP495
3
2
1
R2
20k⍀20k⍀100k⍀
VO = 5 +
R
G
200k⍀
R
G
R3
VIN + V
4
6
R4
REF
V
7
O
10
0%
2V
2V
1ms
Figure 3b. “H” Bridge Outputs
Figure 5. Single-Supply Instrumentation Amplifier
Resistor RG sets the gain of the instrumentation amplifier. Minimum gain is 6 (with no R
). All resistors should be matched in
G
absolute value as well as temperature coefficient to maximize
common-mode rejection performance and minimize drift. This
instrumentation amplifier can operate from a supply voltage as
low as 3 V.
REV. D–8–
Page 9
OP295/OP495
5V5V
R1
17.8k⍀
1.23V
AD589
3
4765
GND CLK SRI
LD
V
DD
V
REF
R
FB
2
1
3
2
4
1
8
DIGITAL
CONTROL
5V
OP295/
OP495
R3
5k⍀
R2
41.2k⍀
R4
100k⍀
D
4096
V
O
=(4.096V)
TOTA L POWER DISSIPATION = 1.6mW
I
OUT
8
DAC8043
A Single-Supply RTD Thermometer Amplifier
This RTD amplifier takes advantage of the rail-to-rail swing of
the OP295/OP495 to achieve a high bridge voltage in spite of a
low 5 V supply. The OP295/OP495 amplifier servos a constant
200 µA current to the bridge. The return current drops across
the parallel resistors 6.19 kΩ and the 2.55 MΩ, developing a
voltage that is servoed to 1.235 V, which is established by the
AD589 band gap reference. The 3-wire RTD provides an
equal line resistance drop in both 100 Ω legs of the bridge,
thus improving the accuracy.
The AMP04 amplifies the differential bridge signal and converts
it to a single-ended output. The gain is set by the series resistance of the 332 Ω resistor plus the 50 Ω potentiometer. The
gain scales the output to produce a 4.5 V full scale. The 0.22 µF
capacitor to the output provides a 7 Hz low-pass filter to keep
noise at a minimum.
ZERO ADJ
10-TURNS
26.7k⍀
2.55M⍀
1%
0.5%
100⍀
RTD
200⍀
100⍀
0.5%
6.19k⍀
1%
26.7k⍀
0.5%
2
AD589
1
OP295/
OP495
3
1.235
5V
7
3
2
1/2
37.4k⍀
1
AMP04
4
8
5
5V
50⍀
332⍀
0.22F
6
4.5V = 450ⴗC
0V = 0ⴗC
V
O
bath or oven and adjust the scale-adjust potentiometer for an
output voltage of 2.50 V, which is equivalent to 250°C. Within
this temperature range, the K-type thermocouple is quite accurate and produces a fairly linear transfer characteristic. Accuracy
of ±3°C is achievable without linearization.
Even if the battery voltage is allowed to decay to as low as 7 V,
the rail-to-rail swing allows temperature measurements to 700°C.
However, linearization may be necessary for temperatures above
250°C where the thermocouple becomes rather nonlinear. The
circuit draws just under 500 µA supply current from a 9 V battery.
A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V
Figure 8 shows a complete voltage output DAC with wide output voltage swing operating off a single 5 V supply. The serial
input 12-bit DAC is configured as a voltage output device with
the 1.235 V reference feeding the current output pin (I
the DAC. The V
, which is normally the input now becomes
REF
OUT
) of
the output.
The output voltage from the DAC is the binary weighted voltage of the reference, which is gained up by the output amplifier
such that the DAC has a 1 mV per bit transfer function.
A Cold Junction Compensated, Battery-Powered
Thermocouple Amplifier
The OP295/OP495’s 150 µA quiescent current per amplifier
consumption makes it useful for battery-powered temperature
measuring instruments. The K-type thermocouple terminates
into an isothermal block where the terminated junctions’ ambient temperatures can be continuously monitored and corrected
by summing an equal but opposite thermal EMF to the amplifier,
thereby canceling the error introduced by the cold junctions.
To calibrate, immerse the thermocouple measuring junction in a
0°C ice bath, adjust the 500 Ω zero-adjust potentiometer to 0 V
out. Then immerse the thermocouple in a 250°C temperature
REV. D
Figure 6. Low Power RTD Amplifier
1.235V
24.9k⍀
1N914
COLD
JUNCTIONS
AD589
7.15k⍀
1.5M⍀1%24.9k⍀
475⍀
1%
1%
1%
24.3k⍀
1%
4.99k⍀
1%
500⍀
10-TURN
ZERO
ADJUST
2.1k⍀
1%
9V
1.33M⍀
8
2
3
4
SCALE
ADJUST
20k⍀
1
OP295/
OP495
V
O
5V = 500ⴗC
0V = 0ⴗC
Figure 8. A 5 V 12-Bit DAC with 0 V to 4.095 Output Swing
4 mA to 20 mA Current Loop Transmitter
Figure 9 shows a self-powered 4 to 20 mA current loop transmitter. The entire circuit floats up from the single-supply (12 V
to 36 V) return. The supply current carries the signal within the
4 to 20 mA range. Thus the 4 mA establishes the baseline current
budget with which the circuit must operate. This circuit consumes
only 1.4 mA maximum quiescent current, making 2.6 mA of
current available to power additional signal conditioning circuitry
or to power a bridge circuit.
SPAN ADJ
V
IN
0 + 3V
10k⍀
10-TURN
10-TURN
182k⍀
1%
100k⍀
1.21M
1%
HP
5082-2800
NULL ADJ
3
2
220pF
100k⍀
1%
62
REF02
GND
4
5V
8
4
1/2
100⍀
220⍀
1
2N1711
OP295/
OP495
100⍀
1%
4 TO
20mA
12V
TO
36V
Figure 9. 4 to 20 mA Current Loop Transmitter
–9–
R
L
100⍀
Page 10
OP295/OP495
A 3 V Low-Dropout Linear Voltage Regulator
Figure 10 shows a simple 3 V voltage regulator design. The
regulator can deliver 50 mA load current while allowing a 0.2 V
dropout voltage. The OP295/OP495’s rail-to-rail output swing
handily drives the MJE350 pass transistor without requiring
special drive circuitry. At no load, its output can swing less than
the pass transistor’s base-emitter voltage, turning the device
nearly off. At full load, and at low emitter-collector voltages, the
transistor beta tends to decrease. The additional base current is
easily handled by the OP295/OP495 output.
The amplifier servos the output to a constant voltage, which
feeds a portion of the signal to the error amplifier.
Higher output current, to 100 mA, is achievable at a higher
dropout voltage of 3.8 V.
< 50mA
I
1.235V
L
44.2k⍀
1%
30.9k⍀
1%
OP295/
OP495
1/2
100F
V
O
V
5V TO 3.2V
MJE 350
IN
43k⍀
1
1000pF
8
4
AD589
3
2
Figure 10. 3 V Low Dropout Voltage Regulator
Figure 11 shows the regulator’s recovery characteristic when its
output underwent a 20 mA to 50 mA step current change.
2V
100
50mA
20mA
OUTPUT
90
10
0%
20mV
1ms
STEP
CURRENT
CONTROL
WAVEFORM
Figure 11. Output Step Load Current Recovery
Low-Dropout, 500 mA Voltage Regulator with Fold-Back
Current Limiting
Adding a second amplifier in the regulation loop, as shown in
Figure 12, provides an output current monitor as well as foldback current limiting protection.
Amplifier A1 provides error amplification for the normal voltage
regulation loop. As long as the output current is less than 1 A,
amplifier A2’s output swings to ground, reverse biasing the
diode and effectively taking itself out of the circuit. However, as
the output current exceeds 1 A, the voltage that develops across
the 0.1 Ω sense resistor forces the amplifier A2’s output to go
high, forward-biasing the diode, which in turn closes the current
limit loop. At this point A2’s lower output resistance dominates
the drive to the power MOSFET transistor, thereby effectively
removing the A1 voltage regulation loop from the circuit.
If the output current greater than 1 A persists, the current limit
loop forces a reduction of current to the load, which causes a
corresponding drop in output voltage. As the output voltage
drops, the current limit threshold also drops fractionally, resulting
in a decreasing output current as the output voltage decreases, to
the limit of less than 0.2 A at 1 V output. This “fold-back” effect
reduces the power dissipation considerably during a short circuit
condition, thus making the power supply far more forgiving in
terms of the thermal design requirements. Small heat sinking on
the power MOSFET can be tolerated.
The OP295’s rail-to-rail swing exacts higher gate drive to the
power MOSFET, providing a fuller enhancement to the transistor. The regulator exhibits 0.2 V dropout at 500 mA of load
current. At 1 A output, the dropout voltage is typically 5.6 V.
I
SENSE
0.1⍀
1/4W
(NORM) = 0.5A
O
(MAX) = 1A
I
O
205k⍀
1%
45.3k⍀
1%
124k⍀
1%
5V V
O
IRF9531
SD
6V
G
1N4148
7
1/2
OP295/
5%
OP495
0.01F
1
1/2
100k⍀
OP295/
R
210k⍀
1%
8
5
A2
6
45.3k⍀
1%
3
124k⍀
A1
1%
4
2
OP495
REF43
2
6
4
2.500V
Figure 12. Low Dropout, 500 mA Voltage Regulator
with Fold-Back Current Limiting
Square Wave Oscillator
The circuit in Figure 13 is a square wave oscillator (note the
positive feedback). The rail-to-rail swing of the OP295/OP495
helps maintain a constant oscillation frequency even if the supply voltage varies considerably. Consider a battery-powered
system where the voltages are not regulated and drop over time.
The rail-to-rail swing ensures that the noninverting input sees
the full V+/2, rather than only a fraction of it.
The constant frequency comes from the fact that the 58.7 kΩ
feedback sets up Schmitt trigger threshold levels that are directly
proportional to the supply voltage, as are the RC charge voltage
levels. As a result, the RC charge time, and therefore, the frequency, remains constant independent of supply voltage. The
slew rate of the amplifier limits oscillation frequency to a maximum of about 800 Hz at a 5 V supply.
Single-Supply Differential Speaker Driver
Connected as a differential speaker driver, the OP295/OP495 can
deliver a minimum of 10 mA to the load. With a 600 Ω load, the
OP295/OP495 can swing close to 5 V p-p across the load.
REV. D–10–
Page 11
OP295/OP495
V+
100k⍀
100k⍀
58.7k⍀
C
8
3
1
1/2
4
2
OP295/
OP495
R
FREQ OUT
F
OSC
1
=
< 350Hz @ V+ = 5V
RC
Figure 13. Square Wave Oscillator Has Stable Frequency
Regardless of Supply Changes
High Accuracy, Single-Supply, Low Power Comparator
The OP295/OP495 makes an accurate open-loop comparator.
With a single 5 V supply, the offset error is less than 300 µV.
Figure 15 shows the OP295/OP495’s response time when
operating open-loop with 4 mV overdrive. It exhibits a 4 ms
response time at the rising edge and a 1.5 ms response time at
the falling edge.
1V
100
90
INPUT
(5mV OVERDRIVE
@ OP-295 INPUT)
OUTPUT
10
0%
2V
5ms
Figure 15. Open-Loop Comparator Response Time
with 5 mV Overdrive
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-095AA
BSC
5
4
0.295 (7.49)
0.285 (7.24)
0.275 (6.98)
0.015
(0.38)
MIN
SEATING
PLANE
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
8-Lead Standard Small Outline Package [SOIC]
Narrow Body
(R-8)
S-Suffix
Dimensions shown in millimeters and (inches)
5.00 (0.1968)
4.80 (0.1890)
4.00 (0.1574)
3.80 (0.1497)
0.25 (0.0098)
0.10 (0.0040)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
85
1.27 (0.0500)
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MS-012AA
BSC
6.20 (0.2440)
5.80 (0.2284)
41
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
0.31 (0.0122)
0.25 (0.0098)
0.17 (0.0067)
0.50 (0.0196)
0.25 (0.0099)
8ⴗ
1.27 (0.0500)
0ⴗ
0.40 (0.0157)
ⴛ 45ⴗ
14-Lead Plastic Dual In-Line Package [PDIP]
(N-14)
P-Suffix
Dimensions shown in inches and (millimeters)
0.685 (17.40)
0.665 (16.89)
0.645 (16.38)
14
1
0.100 (2.54)
BSC
0.015 (0.38)
0.180 (4.57)
MAX
0.150 (3.81)
0.130 (3.30)
0.110 (2.79)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
0.022 (0.56)
0.018 (0.46)
0.014 (0.36)
COMPLIANT TO JEDEC STANDARDS MO-095-AB
0.060 (1.52)
0.050 (1.27)
0.045 (1.14)
8
7
MIN
0.295 (7.49)
0.285 (7.24)
0.275 (6.99)
SEATING
PLANE
0.325 (8.26)
0.310 (7.87)
0.300 (7.62)
0.015 (0.38)
0.010 (0.25)
0.008 (0.20)
0.150 (3.81)
0.135 (3.43)
0.120 (3.05)
16-Lead Standard Small Outline Package [SOIC]
Wide Body
(RW-16)
S-Suffix
Dimensions shown in millimeters and (inches)
10.50 (0.4134)
10.10 (0.3976)
16
1
1.27 (0.0500)
BSC
0.30 (0.0118)
0.10 (0.0039)
COPLANARITY
0.10
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN