Datasheet OP27AZ, OP27FP, OP27GP, OP27EJ, OP27EP Datasheet (Analog Devices)

Low-Noise, Precision

a

FEATURES

√

Low Noise: 80 nV p-p (0.1 Hz to 10 Hz), 3 nV/
Low Drift: 0.2 V/C
High Speed: 2.8 V/s Slew Rate, 8 MHz Gain

Bandwidth
Low V
Excellent CMRR: 126 dB at V

: 10 V

OS

of ±11 V

CM

High Open-Loop Gain: 1.8 Million
Fits 725, OP07, 5534A Sockets
Available in Die Form

GENERAL DESCRIPTION

The OP27 precision operational amplifier combines the low
offset and drift of the OP07 with both high speed and low noise.
Offsets down to 25 µV and drift of 0.6 µV/°C maximum make
the OP27 ideal for precision instrumentation applications.
Exceptionally low noise, e

= 3.5 nV/√Hz, at 10 Hz, a low 1/f

n

noise corner frequency of 2.7 Hz, and high gain (1.8 million),
allow accurate high-gain amplification of low-level signals. A
gain-bandwidth product of 8 MHz and a 2.8 V/µsec slew rate
provides excellent dynamic accuracy in high-speed, data­acquisition systems.

A low input bias current of ±10 nA is achieved by use of a
bias-current-cancellation circuit. Over the military temperature
range, this circuit typically holds I

and IOS to ±20 nA and 15 nA,

B

respectively.

The output stage has good load driving capability. A guaranteed
swing of ±10 V into 600 Ω and low output distortion make the
OP27 an excellent choice for professional audio applications.

Hz

(Continued on page 7)

Operational Amplifier

OP27

PIN CONNECTIONS

TO-99

(J-Suffix)

BAL

BAL 1

–IN 2

+IN 3

OP27

4V– (CASE)

NC = NO CONNECT

8-Pin Hermetic DIP

(Z-Suffix)

Epoxy Mini-DIP

(P-Suffix)

8-Pin SO

(S-Suffix)

TRIM

OS

–IN

+IN

1

OP27

2

3

4

NC = NO CONNECT

V

V+

OUT

NC

8

V

TRIM

OS

7

V+

6

OUT

5

NCV–

NONINVERTING

INPUT (+)

INVERTING

INPUT (–)

R1 AND R2 ARE PERMANENTLY

*

ADJUSTED AT WAFER TEST FOR
MINIMUM OFFSET VOLTAGE.

Q6

Q3

R1*

R3

18

V

ADJ.

OS

Q2B

R4

R2*

Q2AQ1A Q1B

Q11 Q12

Figure 1. Simplified Schematic

REV. A

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.

V+

C2

Q22

Q21

R23 R24

Q23

Q27 Q28

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] © Analog Devices, Inc., 2002

Q24

R5

C1

R9

R12

C3 C4

Q20 Q19

Q26

Q46

OUTPUT

Q45

V–

OP27

–SPECIFICATIONS

ELECTRICAL CHARACTERISTICS

(@ VS = ±15 V, TA = 25C, unless otherwise noted.)

OP27A/E OP27F OP27C/G

Parameter Symbol Conditions Min Typ Max Min Typ Max Min Typ Max Unit

INPUT OFFSET
VOLTAGE

LONG-TERM V

STABILITY

1

OS

2, 3

V

OS

10 25 20 60 30 100 µV

VOS/Time 0.2 1.0 0.3 1.5 0.4 2.0 µV/M

INPUT OFFSET
CURRENT I

OS

7 35 9 50 12 75 nA

INPUT BIAS
CURRENT I

INPUT NOISE

VOLTAGE

INPUT NOISE e

Voltage Density

3, 4

3

e

B

n p-p

n

0.1 Hz to 10 Hz 0.08 0.18 0.08 0.18 0.09 0.25 µV p-p

fO = 10 Hz 3.5 5.5 3.5 5.5 3.8 8.0 nV/√Hz
fO = 30 Hz 3.1 4.5 3.1 4.5 3.3 5.6 nV/√Hz

±10 ±40 ±12 ±55 ±15 ±80 nA

fO = 1000 Hz 3.0 3.8 3.0 3.8 3.2 4.5 nV/√Hz

INPUT NOISE i

Current Density

3, 5

n

fO = 10 Hz 1.7 4.0 1.7 4.0 1.7 pA/√Hz
fO = 30 Hz 1.0 2.3 1.0 2.3 1.0 pA/√Hz
fO = 1000 Hz 0.4 0.6 0.4 0.6 0.4 0.6 pA/√Hz

INPUT
RESISTANCE

Differential-Mode
Common-Mode R

6

R

IN

INCM

1.3 6 0.94 5 0.7 4 MΩ
3 2.5 2 GΩ

INPUT VOLTAGE
RANGE IVR ±11.0 ±12.3 ±11.0 ±12.3 ±11.0 ±12.3 V

COMMON-MODE

REJECTION RATIO CMRR VCM = ±11 V 114 126 106 123 100 120 dB

POWER SUPPLY PSRR VS = ±4 V

REJECTION RATIO to ±18 V 1 10 1 10 2 20 µV/V

LARGE-SIGNAL A

VO

VOLTAGE GAIN V

RL ≥ 2 kΩ,

= ±10 V 1000 1800 1000 1800 700 1500 V/mV

O

≥ 600 Ω,

R

L

VO = ±10 V 800 1500 800 1500 600 1500 V/mV

OUTPUT
VOLTAGE SWING V

O

RL ≥ 2 kΩ±12.0 ± 13.8 ± 12.0 ± 13.8 ±11.5 ± 13.5 V
RL ≥ 600 Ω±10.0 ± 11.5 ± 10.0 ± 11.5 ±10.0 ± 11.5 V

SLEW RATE

7

SR RL ≥ 2 kΩ 1.7 2.8 1.7 2.8 1.7 2.8 V/µs

O

GAIN
BANDWIDTH

PRODUCT

7

GBW 5.0 8.0 5.0 8.0 5.0 8.0 MHz

OPEN-LOOP
OUTPUT

RESISTANCE R

O

VO = 0, IO = 07070 70Ω

POWER
CONSUMPTION P

d

V

O

90 140 90 140 100 170 mW

OFFSET
ADJUSTMENT

RANGE RP = 10 kΩ±4.0 ±4.0 ± 4.0 mV

NOTES

1

Input offset voltage measurements are performed ~ 0.5 seconds after application of power. A/E grades guaranteed fully warmed up.

2

Long-term input offset voltage stability refers to the average trend line of VOS versus. Time over extended periods after the first 30 days of operation. Excluding the
initial hour of operation, changes in VOS during the first 30 days are typically 2.5 µV. Refer to typical performance curve.

3

Sample tested.

4

See test circuit and frequency response curve for 0.1 Hz to 10 Hz tester.

5

See test circuit for current noise measurement.

6

Guaranteed by input bias current.

7

Guaranteed by design.

–2–

REV. A

OP27

ELECTRICAL CHARACTERISTICS

(@ VS = ±15 V, –55C ≤ TA ≤ 125C, unless otherwise noted.)

OP27A OP27C

Parameter Symbol Conditions Min Typ Max Min Typ Max Unit

INPUT OFFSET
VOLTAGE

AVERAGE INPUT
OFFSET DRIFT TCV

1

V

OS

TCV

OS

OSn

30 60 70 300 µV

2

3

0.2 0.6 4 1.8 µV/°C

INPUT OFFSET
CURRENT I

OS

15 50 30 135 nA

INPUT BIAS
CURRENT I

B

±20 ±60 ±35 ±150 nA

INPUT VOLTAGE
RANGE IVR ±10.3 ± 11.5 ±10.2 ± 11.5 V

COMMON-MODE

REJECTION RATIO CMRR VCM = ±10 V 108 122 94 118 dB

POWER SUPPLY

REJECTION RATIO PSRR VS = ±4.5 V to ±18 V 2 16 4 51 µV/V

LARGE-SIGNAL
VOLTAGE GAIN A

VO

RL ≥ 2 kΩ, VO = ±10 V 600 1200 300 800 V/mV

OUTPUT
VOLTAGE SWING V

NOTES

1

Input offset voltage measurements are performed by automated test equipment approximately 0.5 seconds after application of power. A/E grades guaranteed fully
warmed up.

2

The TCVOS performance is within the specifications unnulled or when nulled with RP = 8 kΩ to 20 kΩ. TCVOS is 100% tested for A/E grades, sample tested for
C/F/G grades.

3

Guaranteed by design.

O

RL ≥ 2 kΩ±11.5 ± 13.5 ±10.5 ±13.0 V

REV. A

–3–

OP27

(@ VS = ±15 V, –25C¯≤ TA ≤ 85C for OP27J, OP27Z, 0C ≤ TA ≤ 70C for OP27EP,

ELECTRICAL CHARACTERISTICS

Parameter Symbol Conditions Min Typ Max Min Typ Max Min Typ Max Unit

INPUT ONSET
VOLTAGE V

AVERAGE INPUT
OFFSET DRIFT TCV

INPUT OFFSET
CURRENT I

INPUT BIAS
CURRENT I

INPUT VOLTAGE
RANGE IVR ±10.5 ±11.8 ±10.5 ±11.8 ±10.5 ±11.8 V

COMMON-MODE

REJECTION RATIO CMRR VCM = ±10 V 110 124 102 121 96 118 dB

POWER SUPPLY

REJECTION RATIO PSRR VS = ±4.5 V 2 15 2 16 2 32 µV/V

LARGE-SIGNAL

VOLTAGE GAIN A

OUTPUT
VOLTAGE SWING V

NOTES

1

The TCVOS performance is within the specifications unnulled or when nulled with RP = 8 kΩ to 20 kΩ. TCVOS is 100% tested for A/E grades, sample tested for
C/F/G grades.

2

Guaranteed by design.

OS

TCV

OS

B

VO

O

1

OS

2

OSn

to ±18 V

R

≥ 2 kΩ,

L

VO = ±10 V 750 1500 700 1300 450 1000 V/mV

RL ≥ 2 kΩ±11.7 ±13.6 ±11.4 ±13.5 ± 11.0 ± 13.3 V

OP27FP, and –40C ≤ TA ≤ 85C for OP27GP, OP27GS, unless otherwise noted.)

OP27E OP27F OP27G

20 50 40 140 55 220 µV

0.2 0.6 0.3 1.3 0 4 1.8 µV/°C

0.2 0.6 0.3 1.3 0 4 1.8 µV/°C

10 50 14 85 20 135 nA

±14 ±60 ±18 ±95 ±25 ± 150 nA

–4–

REV. A

DICE CHARACTERISTICS

DIE SIZE 0.109  0.055 INCH, 5995 SQ. MILS

(2.77  1.40mm, 3.88 SQ. mm)

1. NULL

2. (–) INPUT

3. (+) INPUT

4. V–

6. OUTPUT

7. V+

8. NULL

OP27

WAFER TEST LIMITS

(@ VS = ±15 V, TA = 25C unless otherwise noted.)

OP27N OP27G OP27GR

Parameter Symbol Conditions Limit Limit Limit Unit

INPUT OFFSET VOLTAGE* V

INPUT OFFSET CURRENT I

OS

OS

35 60 100 µV Max

35 50 75 nA Max

INPUT BIAS CURRENT IB ±40 ±55 ± 80 nA Max
INPUT VOLTAGE RANGE IVR ±11 ±11 ± 11 V Min

COMMON-MODE REJECTION
RATIO CMRR V

= IVR 114 106 100 dB Min

CM

POWER SUPPLY PSRR VS = ±4 V to ±18 V 10 10 20 µV/V Max

LARGE-SIGNAL VOLTAGE
GAIN A

OUTPUT VOLTAGE SWING V

POWER CONSUMPTION P

NOTE
*Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed

for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.

VO

A

VO

O

V

O

d

RL ≥ 2 kΩ, VO = ±10 V 1000 1000 700 V/mV Min
RL ≥ 600 Ω, VO = ±10 V 800 800 600 V/mV Min

RL ≥ 2 kΩ±12.0 ± 12.0 +11.5 V Min
RL2600n ±10.0 ±10.0 ± 10.0 V Min

VO = 0 140 140 170 mW Max

REV. A

–5–

OP27

TYPICAL ELECTRICAL CHARACTERISTICS

(@ VS = ±15 V, TA = 25C unless otherwise noted.)

OP27N OP27G OP27GR

Parameter Symbol Conditions Typical Typical Typical Unit

AVERAGE INPUT OFFSET
VOLTAGE DRIFT* TCVOS or Nulled or Unnulled 0.2 0.3 0.4 µV/°C

TCV

OSn

RP = 8 kΩ to 20 kΩ

AVERAGE INPUT OFFSET
CURRENT DRIFT TCI

OS

80 130 180 pA/°C

AVERAGE INPUT BIAS
CURRENT DRIFT TCI

B

100 160 200 pA/°C

INPUT NOISE VOLTAGE
DENSITY e

n

e

n

e

n

fO = 10 Hz 3.5 3.5 3.8 nV/√Hz
fO = 30 Hz 3.1 3.1 3.3 nV/√Hz
fO = 1000 Hz 3.0 3.0 3.2 nV/√Hz

INPUT NOISE CURRENT
DENSITY i

INPUT NOISE VOLTAGE e

i
i

n

n

n

np-p

SLEW RATE SR R

fO = 10 Hz 1.7 1.7 1.7 pA/√Hz
fO = 30 Hz 1.0 1.0 1.0 pA/√Hz
fO = 1000 Hz 0.4 0.4 0.4 pA/√Hz

0.1 Hz to 10 Hz 0.08 0.08 0.09 µV p-p
≥ 2 kΩ 2.8 2.8 2.8 V/µs

L

GAIN BANDWIDTH
PRODUCT GBW 8 8 8 MHz

NOTE

*Input offset voltage measurements are performed by automated test equipment approximately 0.5 seconds after application of power.

–6–

REV. A

OP27

WARNING!

ESD SENSITIVE DEVICE

(Continued from page 1)

PSRR and CMRR exceed 120 dB. These characteristics, coupled
with long-term drift of 0.2 µV/month, allow the circuit designer
to achieve performance levels previously attained only by dis­crete designs.

Low-cost, high-volume production of OP27 is achieved by
using an on-chip Zener zap-trimming network. This reliable
and stable offset trimming scheme has proved its effectiveness
over many years of production history.

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±22 V

Input Voltage

Output Short-Circuit Duration . . . . . . . . . . . . . . . . Indefinite

Differential Input Voltage
Differential Input Current

1

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±22 V

2

. . . . . . . . . . . . . . . . . . . . . . ±0.7 V

2

. . . . . . . . . . . . . . . . . . . . ±25 mA

4

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

Operating Temperature Range

OP27A, OP27C (J, Z) . . . . . . . . . . . . . . . . –55°C to +125°C

OP27E, OP27F (J, Z) . . . . . . . . . . . . . . . . . –25°C to +85°C

OP27E, OP27F (P) . . . . . . . . . . . . . . . . . . . . . . 0°C to 70°C

OP27G (P, S, J, Z) . . . . . . . . . . . . . . . . . . –40°C to +85°C

Lead Temperature Range (Soldering, 60 sec) . . . . . . . 300°C

Junction Temperature . . . . . . . . . . . . . . . . . –65°C to +150°C

The OP27 provides excellent performance in low-noise, high­accuracy amplification of low-level signals. Applications include
stable integrators, precision summing amplifiers, precision voltage­threshold detectors, comparators, and professional audio circuits
such as tape-head and microphone preamplifiers.

The OP27 is a direct replacement for 725, OP06, OP07, and
OP45 amplifiers; 741 types may be directly replaced by remov­ing the 741’s nulling potentiometer.

Package Type 

3

JA



JC

Unit

TO 99 (J) 150 18 °C/W
8-Lead Hermetic DlP (Z) 148 16 °C/W
8-Lead Plastic DIP (P) 103 43 °C/W
20-Contact LCC (RC) 98 38 °C/W
8-Lead SO (S) 158 43 °C/W

NOTES

1

For supply voltages less than ± 22 V, the absolute maximum input voltage is
equal to the supply voltage.

2

The OP27’s inputs are protected by back-to-back diodes. Current limiting
resistors are not used in order to achieve low noise. If differential input voltage
exceeds ± 0.7 V, the input current should be limited to 25 mA.

3

␪

is specified for worst-case mounting conditions, i.e., ␪JA is specified for

JA

device in socket for TO, CERDIP, and P-DIP packages; ␪JA is specified for
device soldered to printed circuit board for SO package.

4

Absolute Maximum Ratings apply to both DICE and packaged parts, unless
otherwise noted.

ORDERING INFORMATION

1

Package

= 25°C Operating

T

A

V

Max CERDIP Plastic Temperature

OS

(µV) TO-99 8-Lead 8-Lead Range

25 OP27AJ
25 OP27EJ

2, 3

2, 3

OP27AZ

OP27EZ OP27EP IND/COM
60 OP27FP
100 OP27CZ
100 OP27GJ OP27GZ OP27GP XIND
100 OP27GS

NOTES

1

Burn-in is available on commercial and industrial temperature range parts in CERDIP, plastic
DIP, and TO-can packages.

2

For devices processed in total compliance to MIL-STD-883, add /883 after part number.
Consult factory for 883 data sheet.

3

Not for new design; obsolete April 2002.

4

For availability and burn-in information on SO and PLCC packages, contact your local
sales office.

2

3

3

4

MIL

IND/COM
MIL

XIND

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the OP27 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

REV. A

–7–

OP27

–Typical Performance Characteristics

100

90

80

70

60

GAIN – dB

50

TEST TIME OF 10sec FURTHER
LIMITS LOW FREQUENCY

40

(<0.1Hz) GAIN

30

0.01

TPC 1. 0.1 Hz to 10 Hz

0.1 1 10 100
FREQUENCY – Hz

Noise Tester

p-p

Frequency Response

10

TA = 25C

= 15V

V

S

1

0.1

RMS VOLTAGE NOISE – V

0.01
100 1k 10k

BANDWIDTH – Hz

TPC 4. Input Wideband Voltage
Noise vs. Bandwidth (0.1 Hz to
Frequency Indicated)

100k

10

9
8

7
6

5

4

3

I/F CORNER = 2.7Hz

2

VOLTAGE NOISE – nV/ Hz

1

1

10 100 1k

FREQUENCY – Hz

TA = 25C

= 15V

V

S

TPC 2. Voltage Noise Density vs.
Frequency

100

10

TOTAL NOISE – nV/ Hz

1

TA = 25C

= 15V

V

S

AT 10Hz

AT 1kHz

RESISTOR NOISE ONLY

SOURCE RESISTANCE – 

R

R1

R2

S

– 2R1

TPC 5. Total Noise vs. Sourced
Resistance

100

741

10

VOLTAGE NOISE – nV/ Hz

1

1

I/F CORNER =

2.7Hz

INSTRUMENTATION

I/F CORNER

LOW NOISE

OP27

I/F CORNER

RANGE TO DC

10 100 1k

FREQUENCY – Hz

AUDIO OP AMP

AUDIO RANGE

TO 20kHz

TPC 3. A Comparison of Op Amp
Voltage Noise Spectra

5

4

3

2

VOLTAGE NOISE – nV/ Hz

1

10k100 1k

–50 –25 0 25 50 75 100 125

AT 10Hz

AT 1kHz

TEMPERATURE – C

VS = 15V

TPC 6. Voltage Noise Density vs.
Temperature

5

T

= 25C

A

4

3

2

VOLTAGE NOISE – nV/ Hz

1

010 40

TOTAL SUPPLY VOLTAGE (V+ – V–) – V

AT 10Hz

AT 1kHz

20 30

TPC 7. Voltage Noise Density vs.
Supply Voltage

10.0

1.0

CURRENT NOISE – pA/ Hz

0.1

I/F CORNER = 140Hz

10 10k

100 1k
FREQUENCY – Hz

TPC 8. Current Noise Density vs.
Frequency

–8–

5.0

4.0

TA = +125C

3.0

TA = –55C

2.0

SUPPLY CURRENT – mA

1.0

TA = +25C

15 25 35 45

5

TOTAL SUPPLY VOLTAGE – V

TPC 9. Supply Current vs. Supply
Voltage

REV. A

OP27

60

50

40

30
20

10

0

–10

–20

–30

OFFSET VOLTAGE – V

–40

TRIMMING WITH
10k POT DOES

–50

NOT CHANGE

–60

TCV

OS

–70

–75

–50 –25 0 25 50 75 100 125 150 175

TEMPERATURE – C

OP27C

OP27A

OP27A

OP27A

OP27C

TPC 10. Offset Voltage Drift of
Five Representative Units vs.
Temperature

OPEN-LOOP GAIN – dB

30

25

TA =

25C

20

15

10

5

0

–20

= 70C

T

A

THERMAL
SHOCK
RESPONSE
BAND

DEVICE IMMERSED
IN 70C OIL BATH

02040

TIME – Sec

VS = 15V

60 80

100

TPC 13. Offset Voltage Change Due
to Thermal Shock

6

4

2

0

–2

–4

–6

6

4

2

0

–2

CHANGE IN OFFSET VOLTAGE – V

–4

–6

0

1234567

TIME – Months

TPC 11. Long-Term Offset Voltage
Drift of Six Representative Units

INPUT BIAS CURRENT – nA

50

40

30

20

10

0

–50

–25 0 25 50 75 100 125 150

OP27C

OP27A

TEMPERATURE – C

VS = 15V

TPC 14. Input Bias Current vs.
Temperature

TA = 25C

= 15V

V

S

10

OP27 C/G

OP27 F

5

CHANGE IN INPUT OFFSET VOLTAGE – V

1

01 4

TIME AFTER POWER ON – Min

23

OP27 A/E

TPC 12. Warm-Up Offset Voltage
Drift

50

40

30

20

OP27C

10

INPUT OFFSET CURRENT – nA

0

–75

–50 –25 0 25 50 75 100 125

OP27A

TEMPERATURE – C

VS = 15V

TPC 15. Input Offset Current vs.
Temperature

5

130

110

90

70

50

VOLTAGE GAIN – dB

30

10

–10

1

10 100 1k 10k 100k 1M 10M 100M

FREQUENCY – Hz

TPC 16. Open-Loop Gain vs.
Frequency

REV. A

70

60

50

PHASE MARGIN – Degrees

4

3

2

SLEW RATE – V/s

–75

M

GBW

SLEW

–50

–25 0 25 50 75 100 125

TEMPERATURE – C

VS = 15V

10

9

8

7

6

TPC 17. Slew Rate, Gain-Bandwidth
Product, Phase Margin vs.
Temperature

–9–

25

GAIN



PHASE

MARGIN

= 70

FREQUENCY – Hz

GAIN BANDWIDTH PRODUCT – MHz

20

15

10

5

GAIN – dB

0

–5

–10

1M 10M 100M

TPC 18. Gain, Phase Shift vs.
Frequency

TA = 25C

= 15V

V

S

80

100

120

140

160

180

200

220

PHASE SHIFT – Degrees

OP27

2.5
TA = 25C

2.0

RL = 2k

1.5

RL = 1k

1.0

OPEN-LOOP GAIN – V/V

0.5

0

010 40

20 30

TOTAL SUPPLY VOLTAGE – V

50

TPC 19. Open-Loop Voltage Gain vs.
Supply Voltage

100

% OVERSHOOT

80

60

40

20

V

S

V

IN

A

V

= 15V

= +1

= 100mV

28

24

20

16

12

8

PEAK-TO-PEAK AMPLITUDE – V

4

0

1k 10k 100k 1M

FREQUENCY – Hz

T

= 25C

A

= 15V

V

S

10M

TPC 20. Maximum Output Swing vs.
Frequency

20mV

50mV

0V

500ns

A

VCL

= 15pF

C

L

= 15V

V

S

= 25C

T

A

= +1

–50mV

18

16

POSITIVE

14

12

10

8

6

4

MAXIMUM OUTPUT – V

2

0

–2

100

SWING

NEGATIVE
SWING

LOAD RESISTANCE –

1k 10k

TA = 25C

= 15V

V

S



TPC 21. Maximum Output Voltage
vs. Load Resistance

2V

+5V

0V

–5V

2s

A

VCL

= 15V

V

S

= 25C

T

A

= +1

0

0 500 2000

1000 1500

CAPACITIVE LOAD – pF

2500

TPC 22. Small-Signal Overshoot vs.
Capacitive Load

60

TA = 25C

= 15V

V

S

50

40

30

20

SHORT-CIRCUIT CURRENT – mA

10

01 4

TIME FROM OUTPUT SHORTED TO

I

(+)

SC

I

(–)

SC

23 5

GROUND – Min

TPC 25. Short-Circuit Current vs.
Time

TPC 23. Small-Signal Transient
Response

140

120

100

CMRR – dB

80

60

1k

FREQUENCY – Hz

VS = 15V
T

A

V

CM

10k 100k 1M100

TPC 26. CMRR vs. Frequency

= 25C

= 10V

TPC 24. Large-Signal Transient
Response

16

12

8

4

0

–4

–8

COMMON-MODE RANGE – V

–12

–16

0 5

TA = +25C

TA = –55C

TA = +125C

TA = –55C

TA = +25C

TA = +125C

10 15 20

SUPPLY VOLTAGE – V

TPC 27. Common-Mode Input Range
vs. Supply Voltage

–10–

REV. A

OP27

0.1F

100k

OP27

10

D.U.T.

VO LTAG E

GAIN

= 50,000

4.7F

2k

OP12

100k

24.3k

0.1F

4.3k

2.2F

22F

SCOPE  1
RIN = 1M

110k

TPC 28. Voltage Noise Test Circuit
(0.1 Hz to 10 Hz)

2.4
TA = 25C

2.2

= 15V

V

S

2.0

1.8

1.6

1.4

1.2

1.0

0.8

OPEN-LOOP VOLTAGE GAIN – V/V

0.6

0.4

100 1k 10k 100k

LOAD RESISTANCE – 

TPC 29. Open-Loop Voltage Gain vs.
Load Resistance

POWER SUPPLY REJECTION RATIO – dB

160

140

120

100

80

60

40

20

0

1

10 100 1k 10k 100k 1M 10M 100M

NEGATIVE
SWING

POSITIVE

SWING

FREQUENCY – Hz

TA = 25C

TPC 31. PSRR vs. Frequency

1 SEC/DIV

80

40

0

0.1Hz to 10Hz p-p NOISE

VOLTAGE NOISE – nV

120

–40

–90

–120

TPC 30. Low-Frequency Noise

APPLICATION INFORMATION

OP27 series units may be inserted directly into 725 and OP07
sockets with or without removal of external compensation or
nulling components. Additionally, the OP27 may be fitted to
unnulled 741-type sockets; however, if conventional 741 nulling
circuitry is in use, it should be modified or removed to ensure
correct OP27 operation. OP27 offset voltage may be nulled to
zero (or another desired setting) using a potentiometer (see
Offset Nulling Circuit).

The OP27 provides stable operation with load capacitances of
up to 2000 pF and ±10 V swings; larger capacitances should be
decoupled with a 50 Ω resistor inside the feedback loop. The
OP27 is unity-gain stable.

Thermoelectric voltages generated by dissimilar metals at the
input terminal contacts can degrade the drift performance. Best
operation will be obtained when both input contacts are main­tained at the same temperature.

OFFSET VOLTAGE ADJUSTMENT

The input offset voltage of the OP27 is trimmed at wafer level.
However, if further adjustment of V
potentiometer can be used. TCV

is necessary, a 10 kΩ trim

OS

is not degraded (see Offset

OS

Nulling Circuit). Other potentiometer values from 1 kΩ to 1 MΩ
can be used with a slight degradation (0.1 µV/°C to 0.2 µV/°C)
of TCV

. Trimming to a value other than zero creates a drift of

OS

approximately (V
TCV

will be 0.33 µV/°C if VOS is adjusted to 100 µV. The

OS

/300) µV/°C. For example, the change in

OS

offset voltage adjustment range with a 10 kΩ potentiometer is
±4 mV. If smaller adjustment range is required, the nulling
sensitivity can be reduced by using a smaller pot in conjuction
with fixed resistors. For example, the network below will have a
±280 µV adjustment range.

1

V+

84.7k4.7k 1k POT

Figure 2.

NOISE MEASUREMENTS

To measure the 80 nV peak-to-peak noise specification of the
OP27 in the 0.1 Hz to 10 Hz range, the following precautions
must be observed:

1. The device must be warmed up for at least five minutes.
As shown in the warm-up drift curve, the offset voltage
typically changes 4 µV due to increasing chip temperature
after power-up. In the 10-second measurement interval,
these temperature-induced effects can exceed tens-of­nanovolts.

2. For similar reasons, the device has to be well-shielded from
air currents. Shielding minimizes thermocouple effects.

REV. A

–11–

OP27

3. Sudden motion in the vicinity of the device can also
“feedthrough” to increase the observed noise.

4. The test time to measure 0.1 Hz to 10 Hz noise should not
exceed 10 seconds. As shown in the noise-tester frequency
response curve, the 0.1 Hz corner is defined by only one
zero. The test time of 10 seconds acts as an additional zero
to eliminate noise contributions from the frequency band
below 0.1 Hz.

5. A noise-voltage-density test is recommended when measuring
noise on a large number of units. A 10 Hz noise-voltage­density measurement will correlate well with a 0.1 Hz to 10 Hz
peak-to-peak noise reading, since both results are determined
by the white noise and the location of the 1/f corner frequency.

UNITY-GAIN BUFFER APPLICATIONS

When Rf ≤ 100 Ω and the input is driven with a fast, large signal
pulse (>1 V), the output waveform will look as shown in the
pulsed operation diagram (Figure 3).

During the fast feedthrough-like portion of the output, the input
protection diodes effectively short the output to the input and a
current, limited only by the output short-circuit protection, will
be drawn by the signal generator. With R
capable of handling the current requirements (I
the amplifier will stay in its active mode and a smooth transition
will occur.

When R

> 2 kΩ, a pole will be created with Rf and the amplifier’s

f

input capacitance (8 pF) that creates additional phase shift and
reduces phase margin. A small capacitor (20 pF to 50 pF) in
parallel with R

will eliminate this problem.

f

R

f

–

OP27

+

Figure 3. Pulsed Operation

COMMENTS ON NOISE

The OP27 is a very low-noise monolithic op amp. The outstanding
input voltage noise characteristics of the OP27 are achieved mainly
by operating the input stage at a high quiescent current. The input
bias and offset currents, which would normally increase, are held
to reasonable values by the input bias-current cancellation circuit.
The OP27A/E has I

and IOS of only ±40 nA and 35 nA at 25°C

B

respectively. This is particularly important when the input has a
high source resistance. In addition, many audio amplifier design­ers prefer to use direct coupling. The high I
of previous designs have made direct coupling difficult, if not
impossible, to use.

Voltage noise is inversely proportional to the square root of bias
current, but current noise is proportional to the square root of
bias current. The OP27’s noise advantage disappears when high
source-resistors are used. Figures 4, 5, and 6 compare OP27’s
observed total noise with the noise performance of other devices
in different circuit applications.

≥ 500 Ω, the output is

f

≤ 20 mA at 10 V);

L

2.8V/s

, VOS, and TCV

B

OS

12

/




2



+

S







Total Noise



Voltage Noise

()




Current Noise R

=

()




sistor Noise

Re

()




2

+

×

2

Figure 4 shows noise versus source-resistance at 1000 Hz. The
same plot applies to wideband noise. To use this plot, multiply
the vertical scale by the square root of the bandwidth.

100

50

1

OP08/108

OP07

10

5

5534

TOTAL NOISE – nV/ Hz

OP27/37

REGISTER

1

50 10k

NOISE ONLY

100 50k

500 1k 5k

RS – SOURCE RESISTANCE – 

1 RS UNMATCHED

= RS1 = 10k, RS2 = 0

e.g. R

S

2 RS MATCHED

= 10k, RS1 = RS2 = 5k

e.g. R

S

R

S1

R

S2

2

Figure 4. Noise vs. Source Resistance (Including Resistor
Noise) at 1000 Hz

At RS <1 kΩ, the OP27’s low voltage noise is maintained. With
R

<1 kΩ, total noise increases, but is dominated by the resis-

S

tor noise rather than current or voltage noise. lt is only beyond

of 20 kΩ that current noise starts to dominate. The argument

R

S

can be made that current noise is not important for applica­tions with low to moderate source resistances. The crossover
between the OP27, OP07, and OP08 noise occurs in the 15 kΩ to
40 kΩ region.

Figure 5 shows the 0.1 Hz to 10 Hz peak-to-peak noise. Here
the picture is less favorable; resistor noise is negligible and current
noise becomes important because it is inversely proportional to
the square root of frequency. The crossover with the OP07
occurs in the 3 kΩ to 5 kΩ range depending on whether bal­anced or unbalanced source resistors are used (at 3 kΩ the I

B

and IOS error also can be three times the VOS spec.).

1k

OP08/108

500

5534

OP07

100

OP27/37

p-p NOISE – nV

50

REGISTER

10

50 10k

NOISE ONLY

100 50k

RS – SOURCE RESISTANCE – 

1

2

1 RS UNMATCHED
e.g. RS = RS1 = 10k, RS2 = 0

MATCHED

2 R

S

= 10k, RS1 = RS2 = 5k

e.g. R

S

500 1k 5k

R

S1

R

S2

Figure 5. Peak-to-Peak Noise (0.1 Hz to 10 Hz) as Source
Resistance (Includes Resistor Noise)

–12–

REV. A

OP27

Therefore, for low-frequency applications, the OP07 is better
than the OP27/OP37 when R

> 3 kΩ. The only exception is

S

when gain error is important. Figure 6 illustrates the 10 Hz
noise. As expected, the results are between the previous two
figures.

For reference, typical source resistances of some signal sources
are listed in Table I.

Table I.

Source

Device Impedance Comments

Strain Gauge <500 Ω Typically used in low-

frequency applications.

Magnetic <1500 Ω Low is very important to
Tapehead reduce self-magnetization

problems when direct coupling
is used. OP27 I

can be

B

neglected.

Magnetic <1500 Ω Similar need for low I

in

B

Phonograph direct coupled applications.
Cartridges OP27 will not introduce any

self-magnetization problem.

Linear Variable <1500 Ω Used in rugged servo-feedback
Differential applications. Bandwidth of
Transformer interest is 400 Hz to 5 kHz.

Open-Loop Gain

Frequency at OP07 OP27 OP37

3 Hz 100 dB 124 dB 125 dB
10 Hz 100 dB 120 dB 125 dB
30 Hz 90 dB 110 dB 124 dB

For further information regarding noise calculations, see “Minimization of Noise
in Op Amp Applications,” Application Note AN-15.

100

50

OP08/108

1

2

Figure 7 is an example of a phono pre-amplifier circuit using the
OP27 for A1; R1-R2-C1-C2 form a very accurate RIAA net­work with standard component values. The popular method to
accomplish RIAA phono equalization is to employ frequency­dependent feedback around a high-quality gain block. Properly
chosen, an RC network can provide the three necessary time
constants of 3180, 318, and 75 µs.

1

For initial equalization accuracy and stability, precision metal
film resistors and film capacitors of polystyrene or polypropy­lene are recommended since they have low voltage coefficients,
dissipation factors, and dielectric absorption.

4

(High-K ceramic
capacitors should be avoided here, though low-K ceramics—
such as NPO types, which have excellent dissipation factors
and somewhat lower dielectric absorption—can be considered
for small values.)

MOVING MAGNET

CARTRIDGE INPUT

Ra

47.5k

Ca
150pF

A1

OP27

C4 (2)
220F

++

LF ROLLOFF

C3

0.47F

R1

97.6k

R2

7.87k

R3
100

G = 1kHz GAIN

C1

0.03F

C2

0.01F

R1

1 +

= 0.101 ( )

= 98.677 (39.9dB) AS SHOWN

R3

OUT IN

R4

75k

R5

100k

OUTPUT

Figure 7.

The OP27 brings a 3.2 nV/√Hz voltage noise and 0.45 pA/√Hz
current noise to this circuit. To minimize noise from other
sources, R3 is set to a value of 100 Ω, which generates a voltage
noise of 1.3 nV/√Hz. The noise increases the 3.2 nV/√Hz of the
amplifier by only 0.7 dB. With a 1 kΩ source, the circuit noise
measures 63 dB below a 1 mV reference level, unweighted, in a
20 kHz noise bandwidth.

Gain (G) of the circuit at 1 kHz can be calculated by the
expression:

OP07

10

5534

5

TOTAL NOISE – nV/ Hz

OP27/37

REGISTER

1

50 10k

NOISE ONLY

100 50k

500 1k 5k

RS – SOURCE RESISTANCE – 

1 RS UNMATCHED

= RS1 = 10k, RS2 = 0

e.g. R

S

2 RS MATCHED

= 10k, RS1 = RS2 = 5k

e.g. R

S

R

S1

R

S2

Figure 6. 10 Hz Noise vs. Source Resistance (Includes
Resistor Noise)

AUDIO APPLICATIONS

The following applications information has been abstracted
from a PMI article in the 12/20/80 issue of Electronic De­sign magazine and updated.

REV. A

–13–





R

G

=+

0 101 1

.

1









R

3

For the values shown, the gain is just under 100 (or 40 dB).
Lower gains can be accommodated by increasing R3, but gains
higher than 40 dB will show more equalization errors because of
the 8 MHz gain-bandwidth of the OP27.

This circuit is capable of very low distortion over its entire range,
generally below 0.01% at levels up to 7 V rms. At 3 V output
levels, it will produce less than 0.03% total harmonic distortion
at frequencies up to 20 kHz.

Capacitor C3 and resistor R4 form a simple –6 dB-per-octave
rumble filter, with a corner at 22 Hz. As an option, the switch­selected shunt capacitor C4, a nonpolarized electrolytic, bypasses
the low-frequency rolloff. Placing the rumble filter’s high-pass
action after the preamp has the desirable result of discriminating

OP27

against the RlAA-amplified low-frequency noise components and
pickup-produced low-frequency disturbances.

A preamplifier for NAB tape playback is similar to an RIAA
phono preamp, though more gain is typically demanded, along
with equalization requiring a heavy low-frequency boost. The
circuit in Figure 7 can be readily modified for tape use, as shown
by Figure 8.

TA P E

HEAD

–

OP27

Ca

Ra

+

R2

5k

100k

R1

33k

0.01F

0.47F

T1 = 3180s
T2 = 50s

15k

Figure 8.

While the tape-equalization requirement has a flat high-frequency
gain above 3 kHz (T

= 50 µs), the amplifier need not be stabilized

2

for unity gain. The decompensated OP37 provides a greater
bandwidth and slew rate. For many applications, the idealized
time constants shown may require trimming of R1 and R2 to
optimize frequency response for nonideal tapehead performance
and other factors.

5

The network values of the configuration yield a 50 dB gain at
1 kHz, and the dc gain is greater than 70 dB. Thus, the worst-case
output offset is just over 500 mV. A single 0.47 µF output capaci-
tor can block this level without affecting the dynamic range.

The tapehead can be coupled directly to the amplifier input,
since the worst-case bias current of 80 nA with a 400 mH, 100
µ inch head (such as the PRB2H7K) will not be troublesome.

One potential tapehead problem is presented by amplifier bias­current transients which can magnetize a head. The OP27 and
OP37 are free of bias-current transients upon power-up or power­down. However, it is always advantageous to control the speed
of power supply rise and fall, to eliminate transients.

In addition, the dc resistance of the head should be carefully
controlled, and preferably below 1 kS2. For this configuration,
the bias-current-induced offset voltage can be greater than the
100pV maximum offset if the head resistance is not sufficiently
controlled.

A simple, but effective, fixed-gain transformerless microphone
preamp ( Figure 9) amplifies differential signals from low imped­ance microphones by 50 dB, and has an input impedance of 2 kΩ.
Because of the high working gain of the circuit, an OP37 helps
to preserve bandwidth, which will be 110 kHz. As the OP37
is a decompensated device (minimum stable gain of 5), a dummy
resistor, Rp, may be necessary, if the microphone is to be
unplugged. Otherwise the 100% feedback from the open input
may cause the amplifier to oscillate.

Common-mode input-noise rejection will depend upon the
match of the bridge-resistor ratios. Either close-tolerance (0.1%)
types should be used, or R4 should be trimmed for best CMRR.
All resistors should be metal film types for best stability and
low noise.

Noise performance of this circuit is limited more by the input
resistors R1 and R2 than by the op amp, as R1 and R2 each gen­erate a 4 nV/√Hz noise, while the op amp generates a 3.2 nV/√Hz

–14–

noise. The rms sum of these predominant noise sources will be
about 6 nV/√Hz, equivalent to 0.9 µV in a 20 kHz noise band-
width, or nearly 61 dB below a 1 mV input signal. Measurements
confirm this predicted performance.

C1

5F

R6

100

R7
10k

OUTPUT

LOW IMPEDANCE

MICROPHONE INPUT

(Z = 50 TO 200 )

R3

R4

=

R1

R2

R1

1k

R2

1k

Rp
30k

R3

316k

–

OP27/

OP37

+

R4

316k

Figure 9.

For applications demanding appreciably lower noise, a high
quality microphone transformer-coupled preamp (Figure 10)
incorporates the internally compensated OP27. T1 is a JE-115K-E
150 Ω/15 kΩ transformer which provides an optimum source
resistance for the OP27 device. The circuit has an overall gain of
40 dB, the product of the transformer’s voltage setup and the op
amp’s voltage gain.

C2

1800pF

150

SOURCE

T1*

R1

121

R3

100

R2

1100

A1

OP27

*

T1 – JENSEN JE – 115K – E

JENSEN TRANSFORMERS
10735 BURBANK BLVD.
N. HOLLYWOOD, CA 91601

OUTPUT

Figure 10.

Gain may be trimmed to other levels, if desired, by adjusting R2
or R1. Because of the low offset voltage of the OP27, the output
offset of this circuit will be very low, 1.7 mV or less, for a 40 dB
gain. The typical output blocking capacitor can be eliminated in
such cases, but is desirable for higher gains to eliminate switch­ing transients.

+18V

OP27

–18V

Figure 11. Burn-In Circuit

Capacitor C2 and resistor R2 form a 2 µs time constant in this
circuit, as recommended for optimum transient response by the
transformer manufacturer. With C2 in use, A1 must have unity­gain stability. For situations where the 2 µs time constant is not
necessary, C2 can be deleted, allowing the faster OP37 to be
employed.

REV. A

OP27

0.0098 (0.25)

0.0075 (0.19)

0.0500 (1.27)

0.0160 (0.41)

8
0

0.0196 (0.50)

0.0099 (0.25)

 45

85

41

0.1968 (5.00)

0.1890 (4.80)

0.2440 (6.20)

0.2284 (5.80)

PIN 1

0.1574 (4.00)

0.1497 (3.80)

0.0500 (1.27)
BSC

0.0688 (1.75)

0.0532 (1.35)

SEATING

PLANE

0.0098 (0.25)

0.0040 (0.10)

0.0192 (0.49)

0.0138 (0.35)

Some comment on noise is appropriate to understand the
capability of this circuit. A 150 Ω resistor and R1 and R2
gain resistors connected to a noiseless amplifier will generate
220 nV of noise in a 20 kHz bandwidth, or 73 dB below a 1 mV
reference level. Any practical amplifier can only approach this noise
level; it can never exceed it. With the OP27 and T1 specified, the
additional noise degradation will be close to 3.6 dB (or –69.5 refer­enced to 1 mV).

R

P

INPUT

10k

OP27

V–

V+

OUTPUT

Figure 12. Offset Nulling Circuit

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

8-Lead PDIP Package (P-Suffix)

(N-8)

References

1. Lipshitz, S.R, “On RIAA Equalization Networks,” JAES,
Vol. 27, June 1979, p. 458–481.

2. Jung, W.G., IC Op Amp Cookbook, 2nd. Ed., H.W. Sams and
Company, 1980.

3. Jung, W.G., Audio IC Op Amp Applications, 2nd. Ed., H.W.
Sams and Company, 1978.

4. Jung, W.G., and Marsh, R.M., “Picking Capacitors,” Audio,
February and March, 1980.

5. Otala, M., “Feedback-Generated Phase Nonlinearity in
Audio Amplifiers,” London AES Convention, March 1980,
preprint 1976.

6. Stout, D.F., and Kautman, M., Handbook of Operational
Amplifier Circuit Design, New York, McGraw-Hill, 1976.

8-Lead SOIC Package (S-Suffix)

(R-8)

REV. A

0.210

(5.33)

0.160 (4.06)

0.115 (2.93)

0.200 (5.08)

0.200 (5.08)

0.125 (3.18)

0.430 (10.92)

0.348 (8.84)

PIN 1

MAX

0.022 (0.558)

0.014 (0.356)

8

0.100 (2.54)

1

BSC

5

4

0.070 (1.77)

0.045 (1.15)

0.280 (7.11)

0.240 (6.10)

0.060 (1.52)

0.015 (0.38)

0.130
(3.30)
MIN

SEATING
PLANE

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

8-Lead CERDIP Package (Z-Suffix)

(Q-8)

0.005 (0.13)
MIN

PIN 1

MAX

0.023 (0.58)

0.014 (0.36)

0.055 (1.4)
MAX

85

1

4

0.100 (2.54)
BSC

0.405 (10.29) MAX

0.070 (1.78)

0.030 (0.76)

0.310 (7.87)

0.220 (5.59)

0.060 (1.52)

0.015 (0.38)

0.150
(3.81)
MIN

SEATING
PLANE

15
0

0.320 (8.13)

0.290 (7.37)

0.195 (4.95)

0.115 (2.93)

0.015 (0.38)

0.008 (0.20)

–15–

8-Pin (TO-99) Header Package (J-Suffix)

0.185 (4.70)

0.165 (4.19)

0.370 (9.40)

0.335 (8.51)

0.335 (8.51)

0.305 (7.75)

0.040 (1.02) MAX

0.045 (1.14)

0.010 (0.25)

(H-8A)

REFERENCE PLANE

0.750 (19.05)

0.500 (12.70)

0.250 (6.35) MIN

0.050 (1.27) MAX

0.200

(5.08)

BSC

0.019 (0.48)

0.016 (0.41)

0.021 (0.53)

0.016 (0.41)

BASE & SEATING PLANE

0.100 (2.54) BSC

4

3

2

0.100
(2.54)

BSC

5

1

0.034 (0.86)

0.027 (0.69)

0.160 (4.06)

0.110 (2.79)

6

0.045 (1.14)

0.027 (0.69)

7

8

45 BSC

Revision History

Location Page

9/01—Data Sheet changed from REV. 0 to REV. A.

Edits to ORDERING INFORMATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Edits to PIN CONNECTIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Edits to ABSOLUTE MAXIMUM RATINGS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Edits to PACKAGE TYPE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Edits to ELECTRICAL CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2, 3

Edits to WAFER TEST LIMITS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Deleted TYPICAL ELECTRICAL CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Edits to BURN-IN CIRCUIT figure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Edits to APPLICATION INFORMATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

C00317–0–1/02(A)

–16–

PRINTED IN U.S.A.